JP2023050978A - Isolated DC/DC converter - Google Patents

Abstract

To provide an isolated bidirectional DC/DC converter capable of achieving miniaturization or large capacity.SOLUTION: A first switching element S1 is connected between a first end of a first insulating capacitor C1 and a positive terminal P1 on a primary side. A second switching element S2 is connected between a second end of the first insulating capacitor C1 and a negative terminal P4 on a secondary side. A first control current source 110 includes a first inductor L1 and at least one first bridge cell 114 that are serially connected between the first end of the first insulating capacitor C1 and a first end of a second insulating capacitor. Each first bridge cell 114 includes a direct current capacitor. A second control current source 120 includes a second inductor L2 and at least one second bridge cell 124 that are serially connected between the second end of the first insulating capacitor C1 and a second end of the second insulating capacitor C2.SELECTED DRAWING: Figure 1

Description

The present disclosure relates to isolated DC/DC converters.

A DC/DC converter that converts DC voltage to DC voltage is an important elemental technology in moving bodies such as electric railways, electric vehicles, and electric aircraft, photovoltaic power generation, and next-generation DC systems.

Patent Document 1 and Non-Patent Document 1 propose a bidirectional chopper circuit using an auxiliary converter. This bidirectional chopper circuit comprises a main converter corresponding to a conventional bidirectional chopper circuit (Buck converter) and an auxiliary converter connected in series with the inductor of the main converter. The auxiliary converter consists of a cascade connection of single-phase full-bridge converters. Using an auxiliary converter can reduce the inductance to 1/50 or more compared to a conventional bidirectional chopper without it. Also, since the auxiliary converter functions as a semiconductor DC circuit breaker, the low-voltage side DC circuit breaker can be eliminated.

On the other hand, DC/DC converters for electric railways may require electrical isolation between DC voltages for safety reasons and from the viewpoint of improving reliability. For applications requiring isolation, a bidirectional chopper circuit using an auxiliary converter cannot be used, and a flyback converter or forward converter using a transformer is used.

WO2017/038122

Haruna Onishi, Makoto Hagiwara: "Control Method and Operation Verification of Bidirectional Chopper with Auxiliary Converter", The Institute of Electrical Engineers of Japan, Industry Application Division, vol.138, No.7

However, in an insulated DC/DC converter using a transformer, since electric power is transferred via magnetic energy, the magnetic energy inevitably increases as the capacity increases. Since the weight and volume of a transformer are generally proportional to the magnetic energy, there is a problem that the transformer becomes large and heavy. Therefore, the application of the flyback converter is limited to several hundred watts or less.

Further, in the flyback converter, it is necessary to provide a gap inside the iron core in order to store a large amount of magnetic energy. By providing a gap, magnetic energy can be stored efficiently, but leakage magnetic flux due to the gap increases. The magnetic energy stored in the leaked magnetic flux causes overvoltage in the switching element, which lowers the reliability of the system.

In summary, the bi-directional chopper circuit, which is a non-isolated DC/DC converter, is easy to increase voltage and capacity. However, there is a problem that electrical isolation between DC voltages cannot be realized. On the other hand, a flyback converter, which is a representative example of an insulated DC/DC converter, has a problem in that it is difficult to increase the capacity because it exchanges power through magnetic energy.

The present disclosure has been made in such circumstances, and one exemplary object of certain aspects thereof is to provide an isolated bidirectional DC/DC converter that can be miniaturized or increased in capacity.

One aspect of the present disclosure relates to an isolated DC/DC converter. The isolated DC/DC converter includes a primary positive terminal and a primary negative terminal, a secondary positive terminal and a secondary negative terminal, a first insulating capacitor having a first end and a second end, a first a second isolating capacitor having an end and a second end and one between the first end and the positive primary terminal of the first isolating capacitor and between the first end and the negative primary terminal of the second isolating capacitor. A second switching element connected between the second terminal and the secondary side negative terminal of the first insulating capacitor and a second switching element connected to one of the second terminal of the second insulating capacitor and the secondary side positive terminal. each first bridge cell including a switching element, a first inductor and at least one first bridge cell connected in series between a first end of the first isolation capacitor and a first end of the second isolation capacitor; includes a DC capacitor, a first controlled current source, a second inductor and at least one second bridge connected in series between a second end of the first isolating capacitor and a second end of the second isolating capacitor. a second controlled current source, each second bridge cell including a DC capacitor; a first state in which the first switching element is on and the second switching element is off; a second state in which the two switching elements are on, and a controller for controlling the at least one first bridge cell and the at least one second bridge cell.

According to an aspect of the present disclosure, it is possible to reduce the size or increase the capacity of the isolated DC/DC converter.

1 is a circuit diagram of an insulated DC/DC converter according to Embodiment 1. FIG. 4 is a circuit diagram showing a configuration example of a bridge cell; FIG. 3A and 3B are diagrams for explaining the operation of the isolated DC/DC converter of FIG. 1. FIG. FIG. 4 is a diagram showing operation waveforms of an isolated DC/DC converter; 3 is a block diagram showing a configuration example of a controller of the isolated DC/DC converter; FIG. It is a block diagram which shows the structural example of a primary side controller. It is a block diagram of a direct-current capacitor voltage collective control part. 4 is a block diagram showing a configuration example of a converter voltage controller; FIG. It is a figure which shows the structural example of a capacitor voltage balancer. FIG. 4 is a block diagram of an inductor current command value generation section incorporating DC component control of an insulating capacitor voltage; It is a figure which shows the circuit constant used for simulation. FIG. 5 is a diagram showing simulation waveforms during power transmission from the primary side to the secondary side; FIG. 5 is a diagram showing simulation waveforms during power transmission from the secondary side to the primary side; FIG. 10 is a diagram showing simulation waveforms when the direction of power flow is reversed at 20 ms; 4 is a circuit diagram of an insulated DC/DC converter according to Modification 1. FIG. FIG. 11 is a circuit diagram of an insulated DC/DC converter according to Modification 2; FIG. 11 is a circuit diagram of an insulated DC/DC converter according to Modification 3; 18A to 18C are diagrams showing a first controlled current source according to Modification 4. FIG. 14 is a circuit diagram of an insulated DC/DC converter according to Modification 5. FIG.

(Overview of embodiment)
SUMMARY OF THE INVENTION Several exemplary embodiments of the disclosure are summarized. This summary presents, in simplified form, some concepts of one or more embodiments, as a prelude to the more detailed description that is presented later, and for the purpose of a basic understanding of the embodiments. The size is not limited. This summary is not a comprehensive overview of all possible embodiments, and it is intended to neither identify key elements of all embodiments nor delineate the scope of some or all aspects. For convenience, "one embodiment" may be used to refer to one embodiment (example or variation) or multiple embodiments (examples or variations) disclosed herein.

An insulated DC/DC converter according to one embodiment includes a first insulating DC/DC converter having a primary side positive terminal and a primary side negative terminal, a secondary side positive terminal and a secondary side negative terminal, a first end and a second end. a capacitor, a second isolating capacitor having a first end and a second end, between the first end and the positive primary terminal of the first isolating capacitor, the first end of the second isolating capacitor and the negative primary terminal. between the second terminal of the first insulating capacitor and the secondary negative terminal, and between the second terminal of the second insulating capacitor and the secondary positive terminal. a connected second switching element, a first inductor and at least one first bridge cell connected in series between the first end of the first isolating capacitor and the first end of the second isolating capacitor; each first bridge cell including a DC capacitor; a first controlled current source; a second inductor connected in series between a second end of the first isolating capacitor and a second end of the second isolating capacitor; a second controlled current source comprising one second bridge cell, each second bridge cell comprising a DC capacitor; a first state in which the first switching element is on and the second switching element is off; and a first switching a controller for alternating between a second state in which the device is off and the second switching device is on, and for controlling the at least one first bridge cell and the at least one second bridge cell.

This isolated DC/DC converter uses an isolation capacitor to achieve electrical isolation. As a result, the size and weight of the converter can be reduced. Furthermore, since the loss of capacitors is smaller than that of transformers, higher efficiency can be expected.

In addition, conventional flyback converters transmit and receive power between DC power sources via magnetic energy, so they are not suitable for large-capacity power conversion. Large-capacity power conversion can be realized because electric power is exchanged via electrical energy.

In one embodiment, phase-shifted PWM (Pulse-Width Modulation) may be applied to each bridge cell. Thereby, miniaturization of the first inductor and the second inductor can be realized. As a result, a compact and lightweight power converter can be realized, which is most suitable as a power converter to be mounted on a moving object.

In one embodiment, the isolated DC/DC converter is connected between the first end of the first isolation capacitor and the positive primary terminal and the other between the first end of the second isolation capacitor and the negative primary terminal. a fourth switching element connected between the second end of the first insulating capacitor and the secondary side negative terminal, and to the other of the second end of the second insulating capacitor and the secondary side positive terminal; and an element. The controller may turn on the third switching element and turn off the fourth switching element in the first state, and turn off the third switching element and turn on the fourth switching element in the second state.

In one embodiment, the controller determines the first state based on a primary voltage between a positive primary terminal and a negative primary terminal and a secondary voltage between a positive secondary terminal and a negative secondary terminal. and a switching controller that controls the duty cycle of the second state.

In one embodiment, the controller may include a primary side controller that feedback-controls the output voltage of each of the at least one first bridge cells such that the first inductor current through the first inductor approaches its command value. Thereby, current control (power control) can be realized.

In one embodiment, the voltage between the primary positive terminal and the primary negative terminal is V _dc1 , the voltage between the secondary positive terminal and the secondary negative terminal is V _dc2 , and the voltage across the first isolating capacitor is v. When _C1 and the voltage of the second insulating capacitor are v _C2 , the command values of the output voltages of the at least one first bridge cell are V _dc1 in the first state and −V _dc2 +v _C1 +v _C2 in the second state. may include a feedforward component where

By adding a feedforward term, the switching frequency component of the frequency contained in the inductor current can be removed. As a result, a significant reduction in ripple current and inductance can be achieved.

In one embodiment, the at least one first bridge cell may be a plurality of first bridge cells. The primary side controller may include multiple capacitor voltage balancers. Each of the plurality of capacitor voltage balancers is configured so that the voltage of one corresponding DC capacitor of the plurality of first bridge cells approaches the average value of the voltages of the DC capacitors of each of the plurality of first bridge cells. The command value of the output voltage of the bridge cell may be corrected.

This makes it possible to equalize the voltages of the plurality of DC capacitors. Note that an "average value" is equivalent to a sum value, and thus controlling the mean value of multiple values includes controlling the sum of multiple values.

In one embodiment, the primary side controller changes the AC component of the command value of the first inductor current such that the average value of the DC capacitor voltage of each of the at least one first bridge cells approaches the predetermined command value. good too. Thereby, the total voltage of the voltages of the plurality of DC capacitors can be stabilized.

In one embodiment, the primary-side controller changes the DC component of the command value of the first inductor current so that the average value of the voltage of the first insulating capacitor and the voltage of the second insulating capacitor approaches zero. good.

In one embodiment, the controller may include a secondary side controller that generates a command value for each output voltage of the at least one second bridge cell such that a second inductor current through the second inductor approaches its command value. good. The secondary controller may have the same configuration as the primary controller.

(embodiment)
Preferred embodiments are described below with reference to the drawings. The same or equivalent constituent elements, members, and processes shown in each drawing are denoted by the same reference numerals, and duplication of description will be omitted as appropriate. Moreover, the embodiments are illustrative rather than limiting of the disclosure and invention, and not all features or combinations thereof described in the embodiments are necessarily essential to the disclosure and invention.

In this specification, "a state in which member A is connected to member B" refers to a case in which member A and member B are physically directly connected, as well as a case in which member A and member B are electrically connected to each other. It also includes the case of being indirectly connected through other members that do not substantially affect the physical connection state or impair the functions and effects achieved by their combination.

Similarly, "the state in which member C is connected (provided) between member A and member B" refers to the case where member A and member C or member B and member C are directly connected. In addition, it also includes the case of being indirectly connected through other members that do not substantially affect their electrical connection state or impair the functions and effects achieved by their combination.

Further, in this specification, the symbols attached to electrical signals such as voltage signals and current signals, or circuit elements such as resistors and capacitors refer to respective voltage values, current values, resistance values, and capacitance values as necessary. shall be represented.

(Embodiment 1)
FIG. 1 is a circuit diagram of an insulated DC/DC converter 100 according to the first embodiment. A primary power supply PS1 and a secondary power supply PS2 are connected to the isolated DC/DC converter 100 . The isolated DC/DC converter 100 can unidirectionally transfer power from the primary side power supply PS1 to the secondary side power supply PS2, or bidirectionally transfer power between the primary side power supply PS1 and the secondary side power supply PS2. can be transmitted. The voltage of the primary side power source PS1 is referred to as primary side DC voltage V _dc1 , and the voltage of the secondary side power source PS2 is referred to as secondary side DC voltage V _dc2 . In this specification, the symbols i _# and v _# beginning with lowercase letters attached to electrical signals (current signals and voltage signals) represent instantaneous values. It means that it fluctuates depending on the As will be described later, the isolated DC/DC converter 100 can operate under the condition of V _dc1 >V _dc2 , can operate under the condition of V _dc1 <V _dc2 , and can operate under the condition of V _dc1 =V _dc2 . It is possible.

Isolated DC/DC converter 100 has a primary side input 102 and a secondary side output 104 . As described above, isolated DC/DC converter 100 can operate as a bi-directional converter, so the notation of "input" and "output" is for convenience. The primary side input 102 has a primary side positive terminal P1 and a primary side negative terminal P2, and a primary side power supply PS1 is connected between the primary side positive terminal P1 and the primary side negative terminal P2. The secondary output 104 has a secondary positive terminal P3 and a secondary negative terminal P4, and a secondary power supply PS2 is connected between the secondary positive terminal P3 and the secondary negative terminal P4. be.

The isolated DC/DC converter 100 includes a first insulating capacitor C1, a second insulating capacitor C2, a plurality of switching elements S1 to S4, a first controlled current source 110, a second controlled current source 120, and a controller 200.

A first insulating capacitor C1 and a second insulating capacitor C2 are provided at the boundary between the primary side region 106 and the secondary side region 108 to be insulated. Each of the first insulating capacitor C1 and the second insulating capacitor C2 may be composed of a single component (capacitor), or may be composed of a plurality of components (capacitors) connected in series and/or in parallel. good too.

The first switching element S1 is connected between the primary side positive terminal P1 and the primary side electrode (terminal) of the first insulating capacitor C1, and the second switching element S2 is connected to the secondary side of the first insulating capacitor C1. and the secondary side negative terminal P4. The first switching element S1 is configured to cut off the current flowing from the primary side positive terminal P1 to the first insulating capacitor C1, and the second switching element S2 is configured to cut off the current flowing from the first insulating capacitor C1 to the secondary side negative terminal P4. configured to cut off the current directed to the

The third switching element S3 is connected between the primary side negative terminal P2 and the electrode (terminal) on the primary side of the second insulating capacitor C2, and the fourth switching element S4 is connected to the secondary side of the second insulating capacitor C2. and the secondary side positive terminal P3. The third switching element S3 is arranged so as to cut off the current from the second insulating capacitor C2 toward the primary side negative terminal P2. The fourth switching element S4 is arranged so as to be able to cut off the current flowing from the secondary side positive terminal P3 to the second insulating capacitor C2.

The first switching element S1 to the fourth switching element S4 are valve devices capable of blocking unidirectional current, and semiconductor switches (power transistors) can be used. For example, the first switching element S1 to the fourth switching element S4 each include an IGBT (Insulated Gate Bipolar Transistor) connected in parallel and a reverse conducting IGBT (RC- IGBT). Alternatively, the first switching element S1 to the fourth switching element S4 may be composed of MOSFETs (Metal Oxide Semiconductor Field Effect Transistors). A MOSFET has its body diode functioning as a freewheeling diode.

Furthermore, a snubber circuit for suppressing overvoltage may be connected in parallel with the first switching element S1 to the fourth switching element S4. However, since the insulated DC/DC converter 100 of FIG. 1 performs power transmission without using magnetic energy, no magnetic energy is accumulated in the leakage inductance. As a result, the size and weight of the snubber circuit can be reduced.

The first controlled current source 110 is connected between the primary side electrode of the first insulating capacitor C1 and the primary side electrode of the second insulating capacitor C2. A first controlled current source 110 includes a first inductor L1 and a first converter 112 . The first converter 112 can be understood as a variable control voltage source because the voltage across the first converter 112 (referred to as converter voltage) _va can be adjusted. This makes it possible to control the inductor current _iL1 flowing through the first inductor L1. The direction of the inductor current _iL1 from the primary side electrode of the first insulating capacitor C1 to the primary side electrode of the second insulating capacitor C2 is positive.

The first converter 112 is a semiconductor power converter, and is composed of a cascade connection of at least one (N, N≧1) bridge cells (unit cells) 114_1 to 114_N.

FIG. 2 is a circuit diagram showing a configuration example of the bridge cell 114_j. Bridge cell 114_j is a single-phase full-bridge converter including a DC capacitor _Cja and a full-bridge circuit 115 consisting of transistors Q1-Q4. The voltage v _Cja across the DC capacitor C _ja is called the DC capacitor voltage. Also, the voltage between outputs of the full bridge circuit 115 is denoted by v _ja and called the cell voltage. The full bridge circuit 115 switches faster than the first switching element S1 to the fourth switching element S4.

The bridge cell 114 includes a gate driver (not shown) and the like, but the illustration is omitted because it is a known technique. Also, although FIG. 2 uses a single-phase full-bridge converter as the bridge cell 114, any semiconductor power converter with similar functionality can be substituted. The transistors Q1 to Q4 of the bridge cell 114 can be composed of RC-IGBTs like the first switching element S1 to the fourth switching element S4. You may

Return to FIG. The second controlled current source 120 is connected between the secondary side electrode of the first insulating capacitor C1 and the secondary side electrode of the second insulating capacitor C2. The second controlled current source 120 includes a second inductor L2 and a second converter 122 and is configured similarly to the first controlled current source 110 to generate an inductor current _iL2 . The direction of the inductor current _iL2 from the secondary side electrode of the first insulating capacitor C1 to the secondary side electrode of the second insulating capacitor C2 is positive. Note that the first inductor L1 and the second inductor L2 may be mounted using a magnetically coupled inductor.

The second converter 122 is a semiconductor power converter, and is configured by a cascade connection of a plurality of N (N≧2) unit cells 124_1 to 124_N. Bridge cell 124_j is configured similarly to bridge cell 114 of FIG _.

Return to FIG. It is assumed that the primary side negative terminal P2 and the secondary side negative terminal P4 are each grounded from the viewpoint of safety. 1 can operate even if both negative terminals P2 and P4 are grounded, and can operate even if only one side is grounded.

The controller 200 controls the first to fourth switching elements S1 to S4, the first converter 112 and the second converter 122 so that the isolated DC/DC converter 100 is in a target state.

Specifically, the controller 200 alternately switches the isolated DC/DC converter 100 between two states φ1 and φ2. In the first state φ1, among the first switching element S1 to the fourth switching element S4, the switch on the primary side is on and the switch on the secondary side is off. Among the four switching elements S4, the secondary side switch is on and the primary side switch is off.

First state φ1:
1st switching element S1 ON
Third switching element S3 ON
Second switching element S2 OFF
Fourth switching element S4 OFF

Second state φ2:
First switching element S1 OFF
Third switching element S3 OFF
Second switching element S2 ON
Fourth switching element S4 ON

Between the first state φ1 and the second state φ2, a dead time in which all the switching elements S1 to S4 are turned off can be inserted.

The controller 200 also controls the first converter 112 to stabilize the current _iL1 through the inductor L1. Specifically, by controlling the switching operation of the full bridge circuit 115 of the bridge cell 114 that constitutes the first converter 112, the inductor current _iL1 is stabilized. Similarly, the controller 200 controls the second converter 122 to stabilize the current _iL2 through the inductor L2.

The above is the configuration of the isolated DC/DC converter 100 . Next, the operation will be explained.

3A and 3B are diagrams for explaining the operation of the isolated DC/DC converter 100 of FIG. 1. FIG. Here, power transfer from primary side input 102 to secondary side output 104 will be described. To simplify the explanation and facilitate understanding, it is assumed that the wiring resistance and wiring inductance are zero and each bridge cell operates as an ideal control voltage source. That is, the harmonic voltage output by each bridge cell is assumed to be zero. At this time, the combination of the variable controlled voltage source and the inductor operates as an ideal controlled current source.

Further, it is assumed that all the switching elements S1 to S4 are ideal switches, and the dead time when all the switching elements S1 to S4 are turned off is zero.

3A shows an equivalent circuit of the first state φ1 of the isolated DC/DC converter 100, and FIG. 3B shows an equivalent circuit of the second state φ2 of the isolated DC/DC converter 100. be

In the first state φ1 of FIG. 3A, the secondary power supply PS2 is disconnected from the first insulating capacitor C1 and the second insulating capacitor C2, and the current _idc2 flowing through the secondary power supply PS2 becomes zero. In the second state φ2 of FIG. 3(b), the primary power supply PS1 is disconnected from the first insulating capacitor C1 and the second insulating capacitor C2, and the current _idc1 flowing through the primary power supply PS1 becomes zero. That is, by alternately repeating the first state φ1 and the second state φ2, electrical insulation between the primary side power supply PS1 and the secondary side power supply PS2 can be ensured.

なお、本動作期間は各ブリッジセルに使用するコンデンサが静電エネルギーを蓄積する都合上、式（１），（２）において、ｖ_１＞０，ｖ_２＞０の関係が成立する必要がある。 In the first state φ1 of FIG. 3(a), the following voltage-current equation holds.

During this operation period, the capacitors used in the bridge cells accumulate electrostatic energy, so the relationships v ₁ >0 and v ₂ >0 must be established in equations (1) and (2). .

なお、本動作期間は各ブリッジセルに使用するコンデンサが静電エネルギーを放出する都合上、式（６），（７）において、ｖ_１＜０，ｖ_２＜０の関係が成立する必要がある。以上より、バルブデバイスの状態に関わらず、絶縁用コンデンサＣ１，Ｃ２には常に共通の電流が流れる（ｉ_Ｃ１＝ｉ_Ｃ２）。 In the second state φ2 of FIG. 3(b), the following voltage-current equation holds.

During this operation period, the capacitors used in the bridge cells emit electrostatic energy, so the relationships v ₁ <0 and v ₂ <0 must be established in equations (6) and (7). . From the above, regardless of the state of the valve device, a common current always flows through the insulating capacitors C1 and C2 (i _C1 =i _C2 ).

ただしＩ_ｄｃ１，Ｉ_ｄｃ２は、ｉ_ｄｃ１，ｉ_ｄｃ２の一周期平均値を表す（Ｉ_ｄｃ１＞０，Ｉ_ｄｃ２＞０）。変換器損失が零の場合、定常時においてＰ_ｄｃ１＝Ｐ_ｄｃ２の関係式が成立する。このとき、式（１１）より式（１２）の関係が導かれる。

The ON time of the switching elements S1 and S3 is T _ON , the OFF time is T _OFF , and the duty cycle (ratio of the ON time to the switching period, also called energization rate) is d=T _ON /(T _ON +T _OFF ). At this time, the energization rate of the switching elements S2 and S4 is 1-d. Let P _dc1 be the one-cycle average value of the primary power, and P _dc2 be the one-cycle average value of the secondary power. At this time, both are represented by the formula (11).

However, I _dc1 and I _dc2 represent one cycle average values of i _dc1 and i _dc2 (I _dc1 >0, I _dc2 >0). When the converter loss is zero, the relational expression of P _dc1 =P _dc2 is established at steady state. At this time, the relationship of Equation (12) is derived from Equation (11).

つまり、この絶縁型ＤＣ／ＤＣコンバータ１００によれば、デューティサイクルｄに応じて、入力電圧Ｖ_ｄｃ１と出力電圧Ｖ_ｄｃ２の比（昇圧比／降圧比）を制御することができる。具体的にはｄ＝０のとき、ｖ_ｄｃ２＝０であり、０＜ｄ＜０．５の範囲において、Ｖ_ｄｃ２＝０～Ｖ_ｄｃ１となり降圧動作が可能であり、０．５＜ｄ＜１の範囲において、Ｖ_ｄｃ２＞Ｖ_ｄｃ１となり、昇圧動作が可能となる。 Transformation of equation (12) yields equation (12').

That is, according to the isolated DC/DC converter 100, the ratio (step-up ratio/step-down ratio) between the input voltage _Vdc1 and the output voltage _Vdc2 can be controlled according to the duty cycle d. Specifically, when d=0, v _dc2 =0, and in the range of 0<d<0.5, V _dc2 =0 to V _dc1 and step-down operation is possible, where 0.5<d<1. In the range of , V _dc2 >V _dc1 is satisfied, and boosting operation is possible.

In order to keep the DC components contained in the capacitor voltages v _C1 and v _C2 constant, the one cycle average value of the capacitor currents i _C1 and i _C2 must be zero. At this time, equation (13) must be established from equations (4), (9), and (12).

From equations (11) to (13), P _dc1 and P _dc2 can be transformed into equation (14).

From Equation (14), power control can be achieved by appropriately controlling i _L1 and i _L2 .

したがって、ｉ_Ｌ２が直流量Ｉ_Ｌ２の場合、傾きＩ_Ｌ２／Ｃ１、もしくはＩ_Ｌ２／Ｃ２で一次関数的に増加する。 Focusing on the capacitor voltages v _C1 and v _C2 , in the first state φ1 in which the switching elements S1 and S3 are on and the switching elements S2 and S4 are off, the following circuit equation holds from equation (4).

Therefore, when i _L2 is the DC quantity I _L2 , it linearly increases with a slope of I _L2 /C1 or I _L2 /C2.

したがって、ｉ_Ｌ１が直流量Ｉ_Ｌ１の場合、傾き－Ｉ_Ｌ１／Ｃ１、もしくは－Ｉ_Ｌ1／Ｃ２で一次関数的に減少する。前述の式（１３）が成立する場合、増加量と減少量は等しくなり、直流コンデンサ電圧ｖ_Ｃ１，ｖ_Ｃ２の直流分は一定に保たれる。 On the other hand, in the second state φ2 in which the switching elements S1 and S3 are off and the switching elements S2 and S4 are on, the following circuit equation is established from the equation (9).

Therefore, when i _L1 is the DC quantity I _L1 , it linearly decreases with a slope of -I _L1 /C1 or -I _L1 /C2. When the above equation (13) holds, the amount of increase and the amount of decrease are equal, and the DC components of the DC capacitor voltages v _C1 and v _C2 are kept constant.

On the other hand, if _iL2 is made larger than the value given by equation (13), the amount of increase will be greater than the amount of decrease. As a result, the DC components included in v _C1 and v _C2 increase. On the other hand, if i _L2 is less than the value given by equation (13), the amount of decrease will be greater than the amount of increase. As a result, the DC components contained in v _C1 and v _C2 are reduced. The above characteristics can be used to control the amount of direct current contained in the arithmetic mean of v _C1 and v _C2 .

Focusing on the converter voltages _va and _vb , in the first state φ1, when the voltage drop across the inductors L1 and L2 is zero, they can be expressed by Equation (17).

The instantaneous powers flowing into the first transducers 112 and 122 are given by v _a i _L1 and v _b i _L2 respectively. Both powers are positive from the above assumptions (v ₁ >0, v ₂ >0, i _L1 >0, i _L2 >0). Therefore, power is transmitted from the primary power supply PS1 (V _dc1 ) to each converter 112, 122, and the transmitted power is converted into electrostatic energy by the DC capacitor C of each bridge cell 114_j, 124_j (j=1 to N). _ja and _Cjb .

Next, in the second state φ2, when the voltage drop across the inductors L1 and L2 is zero, the converter voltages v _a and v _b can be expressed by equation (18).

The instantaneous power flowing into each converter 112, 122 is given by v _a i _L1 and v _b i _L2 , respectively. Both powers are negative from the above assumptions (v ₁ <0, v ₂ <0, i _L1 >0, i _L2 >0). Therefore, by discharging the electrostatic energy accumulated in the DC capacitors C _{ja and} C _jb of each bridge cell 114_j and 124_j (j=1 to N), the secondary power supply PS2 (V _dc2 ) is transferred.

In short, in the first state φ1, electric power flows into the DC capacitors C _ja and C _jb of the first converter 112 and the second converter 122, respectively, and electrostatic energy is accumulated. On the other hand, in the second state φ2, power flows out from the capacitors C _ja and C _jb to release electrostatic energy. That is, the isolated DC/DC converter 100 shown in FIG. 1 differs from conventional converters in that power is transferred via electrostatic energy.

Also, it should be noted that the insulating capacitor in FIG. 1 is intended only for electrical isolation, and that the withstand voltage during circuit operation is ensured by the valve devices (switching elements S1 to S4).

FIG. 4 is a diagram showing operation waveforms of the isolated DC/DC converter 100. As shown in FIG. Here, _Vdc1 = _Vdc2 = _Vdc , _iL1 = _iL2 = _Idc (>0), and C1=C2. The duty cycle d at this time is 0.5. Focusing on _va , _vb , _{iL1 and iL2} , _{iL1 and iL2} are only DC components whereas va and _vb are AC components only. Therefore, the instantaneous electric power (v _a i _L1 , v _b i _L2 ) flowing into the converters 112 and 122 is also only AC power. As a result, only AC power is supplied to each bridge cell 114, 124, and steady DC power is not generated. Therefore, the DC components of the DC capacitor voltages v _Cja and v _Cjb can be kept constant.

The above is the operation of the isolated DC/DC converter 100 . Next, its advantages will be explained.

The isolated DC/DC converter 100 has the following advantages over conventional bidirectional converters. In other words, the conventional bidirectional converter cannot achieve electrical isolation between DC voltages, whereas the present embodiment can achieve electrical isolation. As a result, a safer and more reliable power converter can be realized.

Insulated DC/DC converter 100 also has the following advantages over flyback converters. That is, the flyback converter uses a transformer to achieve electrical isolation, whereas the isolated DC/DC converter 100 uses isolation capacitors C1 and C2 to achieve electrical isolation. As a result, the size and weight of the converter can be reduced. Furthermore, since the loss of capacitors is smaller than that of transformers, higher efficiency can be expected.

In addition, flyback converters are not suitable for large-capacity power conversion because they transmit and receive power between DC power supplies via magnetic energy. On the other hand, in the present embodiment, power transfer is performed via the electrostatic energy stored in the DC capacitor of each cell, so large-capacity power conversion can be realized.

In addition, it is possible to apply phase-shift PWM (Pulse-Width Modulation) to each bridge cell, thereby realizing miniaturization of inductors L1 and L2. As a result, a compact and lightweight power converter can be realized, which is most suitable as a power converter to be mounted on a moving object.

Although it is difficult to control the inductor current by feedback control in the flyback converter, feedback control can be easily applied in this embodiment. As a result, a high-performance and robust power converter can be realized.

Next, specific control of the insulated DC/DC converter 100 by the controller 200 will be described.

FIG. 5 is a block diagram showing a configuration example of the controller 200 of the isolated DC/DC converter 100. As shown in FIG. This controller 200 controls the primary and secondary powers P _dc1 and P _dc2 and brings them closer to the command values by feedback control. If the voltages V _dc1 and V _dc2 of the primary power supply PS1 and the secondary power supply PS2 can be considered constant, controlling the powers P _dc1 and P _dc2 is nothing but controlling the currents i _dc1 and i _dc2 . That is, the controller 200 performs feedback control in the major loop to bring the primary side current _idc1 and the secondary side current _idc2 close to their command values. In the following example, a feedback loop is configured such that inductor current _iL1 approaches its command value i ^* _L1 and inductor current _iL2 approaches its command value i ^* _L2 .

That is, the controller 200 has the following two functions.
・Duty cycle control ・Inductor current control (power control)

Controller 200 mainly includes switching controller 202 , primary side controller 204 , secondary side controller 206 and sensor group 208 . The sensor group 208 detects electrical signals (voltage signals and current signals) required for control and converts them into digital values.

The functions of the controller 200 may be realized by software processing, hardware processing, or a combination of software processing and hardware processing. Specifically, software processing is implemented by combining processors (hardware) such as CPUs (Central Processing Units), MPUs (Micro Processing Units), microcomputers, and software programs executed by the processors (hardware). Specifically, hardware processing is implemented by hardware such as an ASIC (Application Specific Integrated Circuit), a controller IC, and an FPGA (Field Programmable Gate Array). In the case of hardware processing, it can be implemented with a digital circuit, an analog circuit, or an analog-digital mixed circuit.

For duty cycle control, a switching controller 202 is provided. The switching controller 202 controls the first switching element S1 to the fourth switching element S4. As described above, the duty cycle d is determined from equation (12) based on the primary side voltage _Vdc1 and the secondary side voltage _Vdc2 . In this embodiment, the first state φ1 and the second state φ2 are switched at the duty cycle d determined by the equation (12), and the ON/OFF control signals corresponding to each state are applied to the first switching element S1 to the fourth switching element S1 to the fourth switching element S1. Output to switching element S4. For example, the ON/OFF states of the first switching element S1 to the fourth switching element S4 can be determined by a general triangular wave comparison method. That is, it can be determined by comparing the triangular wave or sawtooth wave (minimum value: 0, maximum value: 1) and the duty cycle d.

As noted above, the duty cycle d is given by equation (12). At that time, the detected instantaneous DC voltages V _dc1 and V _dc2 are used. The pair of switching elements S1 and S3 and the pair of S2 and S4 perform complementary operations. That is, when S1 and S3 are on, S2 and S4 are turned off, and when S1 and S3 are off, S2 and S4 are turned on.

In connection with inductor current control (power control), a primary side controller 204 and a secondary side controller 206 are provided.

Primary-side controller 204 controls first converter 112 so that current iL1 flowing through inductor _L1 approaches its command value i ^* _L1 . Specifically, by generating PWM signals PWM _1a to PWM _Na for controlling the full bridge circuits 115 of each of the plurality of bridge cells 114_1 to 114_N constituting the first converter 112, and controlling the switching operation, each The cell voltages v _1a to v _Na of the bridge cell 114 and their sum, the converter voltage v _{a ,} are varied, thereby stabilizing the inductor current i _L1 .

Similarly, the secondary controller 206 controls the second converter 122 so that the current _iL2 flowing through the inductor L2 approaches its command value i ^* _L2 . Specifically, the PWM signals PWM _1b to PWM _Nb for controlling the switching operations of the full bridge circuits 115 of the plurality of bridge cells 124_1 to 124_N constituting the second converter 122 are generated, and the cells of each bridge cell 124 are generated. Vary the voltages v _1b to v _Nb and their sum, the converter voltage v _b , thereby stabilizing the inductor current i _L2 . Secondary controller 206 is configured similarly to primary controller 204 .

FIG. 6 is a block diagram showing a configuration example of the primary side controller 204. As shown in FIG. Primary side controller 204 includes a converter voltage controller 210 and a plurality of bridge controllers 220_1-220_N. Converter voltage controller 210 includes inductor current controller 212 and inductor current command value generator 230 . Inductor current command value generation unit 230 generates inductor current command value i ^* _L1 so as to obtain desired power. The inductor current controller 212 receives the detected value of the current _iL1 flowing through the inductor L1 and its command value i ^* _L1 , and sets the command value v of the plurality of cell voltages _v1a to _vNa so that their error approaches zero. ^* Generates _1a -v ^* _Na .

The j-th bridge controller 220_j receives the detected value _vCja of the DC voltage of the corresponding DC capacitor _Cja and the command value v ^* _ja of the cell voltage. Bridge controller 220_j includes duty cycle calculator 222 and PWM modulator 224 .

The duty cycle calculator 222 normalizes v ^* _ja by _vCja to calculate the duty cycle (phase shift amount or overlap amount) _dja of the phase shift PWM control. PWM modulator 224 generates a PWM signal _{PWM_ja} having carrier frequency _fSA based on duty cycle _{d_ja} to control bridge cell 114_j.

The above is the configuration example of the controller 200 . Next, preferred control and configuration of controller 200 will be described.

Controller 200 preferably performs the following controls in addition to the duty cycle control and inductor current control (power control) described above.

DC Capacitor Voltage Control Focusing on the operation of each bridge cell 114, 124 in FIG. 1, power is absorbed from the primary side power supply PS1 in the first state φ1 in which the switching elements S1, S3 are on. Also, in the second state φ2 in which the switching elements S2 and S4 are on, power is discharged to the secondary side power supply PS2. The one- _cycle average value of the power that flows into and out of the bridge cells 114 and 124 is zero, and steady _DC power is not generated. However, in reality, it fluctuates under the influence of transient fluctuations and losses in the converters 112 and 122, so it is necessary to apply DC capacitor voltage control.

Here, DC capacitor voltage control can be performed by a combination of two processes.
・Batch control of DC capacitor voltage ・Balance control of DC capacitor voltage

The former controls the average value v _{Ca_ave} of the plurality of DC capacitor voltages v _C1a to v _CNa (the average value v _{Cb_ave} of v _C1b to v _CNb on the secondary side), whereas the latter controls the plurality of DC capacitor voltages Dispersion of v _C1a to v _CNa is eliminated.

・DC capacitor voltage batch control In actual converters 112 and 114, the DC capacitor voltages v _C1a to v _CNa and v _C1b to v _CNb fluctuate due to transient fluctuations and converter losses, so DC capacitor voltage batch control is applied. preferably. The main purpose of the DC capacitor voltage collective control is to control the arithmetic mean values v _{Ca — ave} and v _{Cb — ave} of the DC capacitor voltages given by Equation (19).

DC capacitor voltage collective control can be realized by using the AC component of the inductor current. Although v _{a and} v _b shown in FIG. 4 have square voltage waveforms, the fundamental wave frequency is equal to the carrier frequency f _SM of the first switching element S1 to the fourth switching element S4. The DC capacitor voltage control method using the AC components of the inductor currents i _{L1 and} i _L2 superimposes the sinusoidal AC component of the frequency f _SM on the inductor currents i L1 and i _L2 and controls the amplitude to control the DC capacitor Realize voltage batch control.

FIG. 7 is a block diagram of the DC capacitor voltage collective control section 240. As shown in FIG. Although control on the primary side controller 204 side will be described here, the same applies to the secondary side.

DC capacitor voltage batch control section 240 is incorporated in inductor current command value generation section 230 . In the inductor current command value generation unit 230, the command value i ^* _L1 of the inductor current _iL1 is divided into a DC component i ^* _{L1_dc} and an AC component i ^* _{L1_ac} . The DC capacitor voltage collective control is performed by changing the command value i ^* _{L1_ac} .

Inductor current command value generation unit 230 includes DC capacitor voltage collective control unit 240 and adder 232 .

The DC capacitor voltage collective control unit 240 feedback-controls the AC component command value i ^* _{L1_ac} so that the average value _{vCa_ave} of the detected values of the DC capacitor voltages _vC1a to _vCNa approaches the command value v ^* _C . .

The DC capacitor voltage collective control section 240 includes an average value calculation section 242 , an error detector 244 , a PI controller 246 and an amplitude modulator 248 . The average value calculator 242 calculates an arithmetic average value v _{Ca_ave} of the detected values of the plurality of DC capacitor voltages v _C1a to v _CNa .

The error detector 244 calculates the difference between the command value (target value) v ^* _C of the DC capacitor voltage and the arithmetic average value v _{Ca_ave} of the DC voltages of the plurality of DC capacitors to generate an error. The PI controller 246 proportionally integrates the error generated by the error detector 244 . Amplitude modulator 248 amplitude modulates the sine wave sin(2πf _SM t) with the output of PI controller 246 . A sine wave sin(2πf _SM t) represents a fundamental wave component and an in-phase component included in _va . The output of the amplitude modulator 248 is the command value (target value) i ^* _{L1_ac} of the AC component of the current i _L1 of the inductor L1 (the current i L2 of the inductor _L2 ) in inductor current control (power control).

The adder 232 superimposes (adds) the AC component command value (target value) i ^* _{L1_ac} on the DC component command value (target value) ^i* _{L1_dc} . The DC component i ^* _{L1_dc} controls the power and the AC component i ^* _{L1_ac} controls the DC capacitor voltage.

If v ^* _C > _{vCa_ave} , i ^* _{L1_ac} is in phase with the fundamental wave component contained in _va . As a result, v _a and inductor current i _L1 form a positive active power that flows into converter 112 and DC capacitor voltages v _C1a to v _CNa increase. On the other hand, when v ^* _C < _{vCa_ave} , i ^* _{L1_ac} is in phase with the fundamental wave component included in _va . As a result, v _a and the inductor current i _L1 form a negative active power, and the DC capacitor voltages v _C1a to v _CNa decrease. By using the above characteristics, the DC capacitor voltage can be controlled.

The same applies to the second converter 122 side.

A triangular AC voltage (ripple) equal to the switching frequency of the switching elements S1 and S3 (S2 and S4) is generated in the average voltage v _{Ca — ave} (v _{Cb — ave} ). Since the AC voltage becomes a disturbance to the control system, it must be removed when it is taken into the control system. AC voltage removal can be realized by applying a low-pass filter or a moving average filter.

DC Capacitor Voltage Balance Control In the DC capacitor voltage batch control described above, the average value of a plurality of DC capacitor voltages is stabilized, but an imbalance may occur in the DC capacitor voltages. DC capacitor voltage balance control is introduced to eliminate this imbalance.

DC capacitor voltage balance control achieves voltage balance by forming active power between the output voltage (cell voltage) v ^* _ja of each bridge cell and inductor current _iL1 . At that time, it should be noted that the polarity of the voltage command value is changed according to the polarity of the inductor current _iL1 .

FIG. 8 is a block diagram showing a configuration example of the converter voltage controller 210. As shown in FIG. As described above, the converter voltage controller 210 generates the cell voltage command values v ^* _1a to v ^* _Na for the plurality of bridge cells 114 . A DC capacitor voltage balance control can be incorporated into the converter voltage controller 210 .

Converter voltage controller 210 includes inductor current controller 212 as described above. The inductor current controller 212 performs inductor current control (power control) as described above. Inductor current controller 212 includes an error detector 212a and a PI controller 212b.

フィードフォワード項を追加することで、インダクタ電流ｉ_Ｌ１に含まれる周波数ｆ_ＳＭのスイッチング周波数成分を除去できる。その結果、リップル電流とインダクタンスの大幅な低減が実現できる。 Feedforward controller 214 generates a feedforward voltage command value v ^* _FFa given by equation (20) in addition to voltage term v ^* _FBa based on inductor current control. The two voltage command values v ^* _FBa , v ^* _FFa are added by an adder 216 to generate the command value v ^* _a for the converter voltage _va .

By adding the feedforward term, it is possible to remove the switching frequency component of frequency _fSM contained in the inductor current _iL1 . As a result, a significant reduction in ripple current and inductance can be achieved.

Converter voltage command value v ^* _a is multiplied by 1/N in divider 218 . This corresponds to the cell voltage command value for each bridge cell 114 .

Capacitor voltage balancers 250_1 to 250_N are provided for DC capacitor voltage balance control. The j-th capacitor voltage balancer 250_j sets the cell voltage command value so that the error between the corresponding j-th DC capacitor voltage v _Cja and the average value v _{Ca_ave} of the plurality of DC capacitor voltages v _C1a to v _CNa is small. Correct v ^* _ja .

FIG. 9 is a diagram showing a configuration example of the capacitor voltage balancer 250. As shown in FIG. Capacitor voltage balancer 250 — j comprises error detector 252 , PI controller 254 , polarity inverter 256 and adder 258 . The error detector 252 calculates the error between the corresponding j-th DC capacitor voltage v _Cja and the average value v _{Ca_ave} of the DC capacitor voltages v _C1a to v _CNa . The PI controller 254 proportionally and integrally calculates the error generated by the error detector 252 . The polarity inverter 256 outputs the output of the PI controller 254 as it is when i _L1 >0, and inverts the polarity (sign) of the output of the PI controller 254 when i _L1 <0. Adder 258 adds the output of polarity inverter 256 to v ^* _a /N.

DC component control of insulating capacitor voltage The controller 200 can additionally perform DC component control of the insulating capacitor voltages v _C1 and v _C2 in addition to the above-described duty cycle control and inductor current control (power control). preferable. DC component control of the insulating capacitor voltages v _C1 and v _C2 can be incorporated into the inductor current command value generation unit 230 .

FIG. 10 is a block diagram of an inductor current command value generator 230E incorporating DC component control of the insulating capacitor voltage. The inductor current command value generator 230E includes an insulating capacitor voltage controller 260 .

As described above, since the currents flowing through the respective insulating capacitors C1 and C2 are common, v _C1 and v _C2 cannot be controlled independently. Therefore, the insulating capacitor voltage control section 260 controls the arithmetic average value v _{C_ave} given by the equation (21).

As shown by the ideal waveforms in FIG. 4, triangular AC voltages (ripples) equal to the switching frequency f _SM are generated at v _C1 and v _C2 . Since the AC voltage (ripple) is a disturbance to the control system, it must be removed when it is taken into the control system. AC voltage removal can be realized by applying a low-pass filter or a moving average filter.

As described above, the DC component included in v _{C_ave} can be varied by adjusting the DC component included in i _L2 or i _L1 . Therefore, _{vc_ave} can be made zero by determining the DC current command value i ^* _{L1_dc} or i ^* _{L2_dc} in FIG. 8 from the voltage feedback control that makes _{vc_ave} zero.

Although _{vc_ave} can be suppressed to zero by control, _{vc_ave} cannot be reduced to 0 by control if the DC components included in _vC1 and _vC2 are varied. Variations in the DC components of v _C1 and v _C2 are mainly due to individual differences in capacitances C1 and C2, and may occur mainly during transients. On the other hand, a capacitor has dielectric loss, which is generally simulated by placing an insulation resistor in parallel with the capacitor. Due to the above insulation resistance, the variation of the DC component finally becomes zero.

The insulating capacitor voltage controller 260 includes an average value calculator 262 , an error detector 264 and a PI controller 266 . The average value calculator 262 calculates an average value v _{C_ave} of the voltages v _C1 and v _C2 across the two insulating capacitors C1 and C2. The error detector 264 calculates the difference between the average value v _{C_ave} and its target value of zero (0). The PI controller 266 amplifies the error to generate a DC component correction amount _{iL1_dc_cmp} of the inductor current _iL1 . The adder 234 adds the correction amount _{iL1_dc_cmp} to the command value i ^* _{L1_dc} of the DC component of the inductor current _iL1 . Adders 232 and 234 may be combined into one.

The above is the power control by the controller 200 .

Next, the simulation results will be explained. "PSCAD/EMTDC" was used for the simulation. FIG. 11 is a diagram showing circuit constants used in the simulation.

The number of bridge cells N was set to 3. The DC voltages were V _dc1 =1 kV and V _dc2 =1 kV, and the capacitor voltages v _C1 and v _C2 were 0.4 kV. The carrier frequency f _SM of the switching elements S1 to S4 was set to 450 Hz, and the carrier frequency f _SA of each bridge cell was set to 1800 Hz. Since phase-shifted PWM is applied to each bridge cell, the equivalent carrier frequency of converters 112 and 122 is 10.8 kHz (=2*N*f _SA ).

Since the purpose of the simulation is to confirm the principle, an ideal state is assumed. That is, an analog control system with zero control delay is assumed, and an ideal switch with zero dead time is used.

FIG. 12 is a diagram showing simulation waveforms during power transmission from the primary side to the secondary side. However, the command value i ^* _{L1_dc} =i ^* _{L2_dc} =150 [A] for the direct current component of the inductor current. By applying the control system of FIG. 8, the inductor current _iL1 follows the command value of 150A without steady-state deviation. On the other hand, _iL1 has a switching ripple component of 10.8 kHz, which is sufficiently small with respect to the DC component. The duty cycle d is 0.5 by substituting the circuit constants in Table 1 into equation (12).

Focusing on the primary-side current i _dc1 , i _dc1 =i _L1 +i _L2 =300 [A] from equation (3) in the first state φ1 (S1=ON, S2=OFF). On the other hand, in the case of the second state φ2 (S1=OFF, S2=ON), i _dc1 =0 from equation (8). Focusing on the capacitor current i _C1 , in the first state φ1, i _C1 =i _L2 =150 [A] from equation (4). On the other hand, in the second state φ2, i _C1 =-i _L2 =-150 [A] from equation (9). The one-cycle average value of the primary side power is P ₁ =V _dc1 i _L1 =1 kV×150 A=150 [kW] from equation (14). The one cycle average value of the secondary power is P ₂ =V _dc2 i _L2 =1 kV×150 A=150 [kW] from the same equation.

Focusing on the converter voltage v _a , in the first state φ1, v _a =V _dc1 =1 [kV] from equation (17). On the other hand, in the second state φ2, v _a =−V _dc2 +v _C1 +v _C2 from equation (18), and is affected by the capacitor voltages v _C1 and v _C2 . If the effects of the capacitor voltages v _C1 and v _C2 are excluded, v _a can be regarded as a 180-degree square-wave voltage with a frequency of 450 Hz, which is equal to f _SM . A 10.8 kHz component due to switching is also present in _va , but is sufficiently small compared to the 450 Hz component.

The DC capacitor voltages v _C1a , v _C2a , and v _C3a include DC and AC components, and the DC component satisfactorily follows the command value of 400V. The AC component has a triangular AC waveform of 450 Hz. Focusing on the first state φ1, the DC capacitor voltages v _C1a , v _C2a and v _C3a linearly increase. This means that power flows into the bridge cells 114_1-114_3 and the DC capacitors C _1a -C _3a store electrostatic energy. On the other hand, in the second state φ2, the DC capacitor voltages v _C1a , v _C2a and v _C3a linearly decrease. This means that power flows out from the bridge cells 114_1 to 114_3 to the secondary side, and the DC capacitors C _1a to C _3a discharge electrostatic energy. The difference between the electrostatic energy when the capacitor voltages v _C1a , v _C2a , and v _C3a take the maximum value and the electrostatic energy when the capacitor voltage takes the minimum value corresponds to the electrostatic energy amount to be charged and discharged.

FIG. 13 is a diagram showing simulation waveforms during power transmission from the secondary side to the primary side. However, the command value i ^* _{L1_dc} =i ^* _{L2_dc} =-150 [A] is set for the direct current component of the inductor current. The duty cycle d is 0.5 as in FIG.

Focusing on the primary-side current i _dc1 , in the first state φ1, i _dc1 =i _L1 +i _L2 =−300 [A] from equation (3). On the other hand, in the second state φ2, i _dc1 =0 from equation (8). Focusing on the capacitor current i _C1 , in the first state φ1, i _C1 =i _L2 =−150 [A] from equation (4). On the other hand, in the second state φ2, i _C1 =−i _L1 =150 [A] from equation (9). The one cycle average value of the primary side electric power is P ₁ =V _dc1 i _L1 =1 kV×(−150 A)=−150 [kW] from equation (14). From the same equation, the one-period average value of the secondary power is P ₂ =V _dc2 i _L2 =1 kV×(−150 A)=−150 [kW]. Other waveforms are similar to those in FIG.

FIG. 14 is a diagram showing simulation waveforms when the direction of power flow is reversed at 20 ms. The converters 112 and 122 operate well without overvoltage and overcurrent even when the power flow is changed at a high speed of 20ms. This shows that FIG. 1 has a fast current control function.

Those skilled in the art will understand that the above-described embodiments are examples, and that various modifications can be made to combinations of each component and each processing process. Such modifications will be described below.

(Modification 1)
FIG. 15 is a circuit diagram of an insulated DC/DC converter 100A according to Modification 1. As shown in FIG. As described above, the insulated DC/DC converter 100 according to the embodiment can operate even if both negative terminals P2 and P4 are grounded, and can operate even if only one side is grounded. If only one side of the primary side negative terminal P2 and the secondary side negative terminal P4 is grounded, one of the first switching element S1 and the third switching element S3 can be omitted. Similarly, either one of the second switching element S2 and the fourth switching element S4 can be omitted.

In Modification 1, the third switching element S3 and the fourth switching element S4 are omitted.

When both the primary side negative terminal P2 and the secondary side negative terminal P4 are grounded, all of the first switching element S1 to the fourth switching element S4 are arranged in order to make the current flowing through the ground point zero. There is a need.

(Modification 2)
In FIG. 1, a semiconductor valve device capable of unidirectional current interruption is used as the first switching element S1 to the fourth switching element S4, but a semiconductor valve device capable of bidirectional current interruption may be used.

FIG. 16 is a circuit diagram of an insulated DC/DC converter 100B according to Modification 2. As shown in FIG. A semiconductor valve device capable of bidirectional current interruption can be realized by connecting two semiconductor valve devices capable of unidirectional current interruption in reverse series to each other.

(Modification 3)
FIG. 17 is a circuit diagram of an insulated DC/DC converter 100C according to Modification 3. As shown in FIG. In Modification 3, the first switching element S1 and the fourth switching element S4 are omitted, and the second switching element S2 and the third switching element S3 are composed of semiconductor valve devices capable of blocking current in both directions.

(Modification 4)
In first controlled current source 110 and second controlled current source 120 in FIG. 1, the arrangement of inductors L1 and L2 is not limited to that in FIG. 18A to 18C are diagrams showing the first controlled current source 110 according to Modification 4. FIG. In FIG. 18(a), inductor L1 is placed above bridge cell 114_1. In FIG. 18(b), inductor L1 is placed below bridge cell 114_N. In FIG. 18(c), an inductor L1 is placed between the bridge cells.

(Modification 5)
FIG. 19 is a circuit diagram of an insulated DC/DC converter 100D according to Modification 5. As shown in FIG. The isolated DC/DC converter 100D has a configuration in which a plurality of M (M≧2) units are connected in parallel, with the isolated DC/DC converter 100 of FIG. 1 as one unit.

The configuration and control of controller 200 are not limited to those described above. Modified examples of control will be described below.

(Modification 6)
When the number N of bridge cells 114 (124), that is, the number N of DC capacitors is 1, balance control of the DC capacitor voltage can be omitted. Also, the number of bridge cells 114 on the primary side and the number of bridge cells 124 on the secondary side may be different, and the numbers should be determined based on the respective primary side voltage _Vdc1 and secondary side voltage _Vdc2 . Just do it.

(Modification 7)
In the embodiments, current control (power control) has been described as control by the controller 200, but the operation mode of the isolated DC/DC converter 100 is not limited thereto. For example, a feedback loop may be formed so that primary side voltage _Vdc1 or secondary side voltage _Vdc2 of isolated DC/DC converter 100 approaches a predetermined command value. For example, when stabilizing the secondary voltage _Vdc2 to its command value V ^* _dc2 , the DC command value i ^* _{L1_dc} for the inductor current of the first converter 112 and the DC command value i for the inductor current of the second converter 122 ^* _{L2_dc} may be determined by feedback control. Specifically, i ^* _{L1_dc} and i ^* _{L2_dc} are set to a common value, the error between _Vdc2 and V ^* _dc2 is amplified by a PI controller or the like, and a DC command value is given according to the output of the PI controller.

(Modification 7)
Controller 200 may be implemented in hardware (digital or analog circuitry). When implemented in analog circuitry, the error detector and PI controller are replaced with an error amplifier (op-amp).

(Application)
INDUSTRIAL APPLICABILITY The present invention is expected to be applied to high-voltage, large-capacity DC/DC converters that strongly require small size, light weight, high performance, and electrical insulation.
・DC/DC converters for moving bodies such as electric railways, electric vehicles, and electric aircraft ・DC/DC converters for solar power generation ・DC/DC converters for next-generation DC systems

Those skilled in the art will understand that the embodiments are examples, and that there are various modifications in the combination of each component and each processing process, and that such modifications are also included in the scope of the present disclosure or the present invention. It is about

100 Isolated DC/DC converter 102 Primary side input 104 Secondary side output PS1 Primary side power supply PS2 Secondary side power supply C1 First insulating capacitor C2 Second insulating capacitor L1 First inductor L2 Second inductor S1 First switching element S2 second switching element S3 third switching element S4 fourth switching element P1 primary side positive terminal P2 primary side negative terminal P3 secondary side positive terminal P4 secondary side negative terminal 110 first controlled current source 112 first converter 114 bridge Cell 115 Full Bridge Circuit 120 Second Controlled Current Source 122 Second Converter 124 Bridge Cell 200 Controller 202 Switching Controller 204 Primary Side Controller 206 Secondary Side Controller 208 Sensor Group 210 Converter Voltage Controller 212 Inductor Current Controller 212a Error Detector 212b PI controller 220 bridge controller 222 duty cycle calculator 224 PWM modulator 230 inductor current command value generator 232 adder 240 DC capacitor voltage batch controller 242 average value calculator 244 error detector 246 PI controller 248 amplitude modulator 250 capacitor voltage Balancer 252 Error Detector 254 PI Controller 256 Polarity Inverter 258 Adder 260 Insulating Capacitor Voltage Control Section 262 Average Value Calculation Section 264 Error Detector 266 PI Controller

Claims (10)

a primary side positive terminal and a primary side negative terminal;
a secondary-side positive terminal and a secondary-side negative terminal;
a first isolating capacitor having a first end and a second end;
a second isolating capacitor having a first end and a second end;
a first switching element connected to one of between the first terminal of the first insulating capacitor and the primary side positive terminal and between the first end of the second insulating capacitor and the primary side negative terminal; ,
a second switching element connected between the second terminal of the first insulating capacitor and the secondary negative terminal and to one of the second terminal and the secondary positive terminal of the second insulating capacitor; ,
each first bridge cell including a first inductor and at least one first bridge cell connected in series between the first end of the first isolation capacitor and the first end of the second isolation capacitor; includes a DC capacitor; a first controlled current source;
each second bridge cell including a second inductor and at least one second bridge cell connected in series between the second end of the first isolation capacitor and the second end of the second isolation capacitor; includes a DC capacitor; a second controlled current source;
Alternately repeating a first state in which the first switching element is on and the second switching element is off and a second state in which the first switching element is off and the second switching element is on, a controller for controlling the at least one first bridge cell and the at least one second bridge cell;
An isolated DC/DC converter, comprising: a third switching element connected to the other between the first end of the first insulating capacitor and the primary side positive terminal and between the first end of the second insulating capacitor and the primary side negative terminal; ,
a fourth switching element connected between the second end of the first insulating capacitor and the secondary negative terminal and to the other of the second end of the second insulating capacitor and the secondary positive terminal; ,
further comprising
The controller is
turning on the third switching element and turning off the fourth switching element in the first state;
2. The isolated DC/DC converter according to claim 1, wherein in said second state, said third switching element is turned off and said fourth switching element is turned on. The controller is
Based on the primary side voltage between the primary side positive terminal and the primary side negative terminal and the secondary side voltage between the secondary side positive terminal and the secondary side negative terminal, the first state and the second state are determined. 3. An isolated DC/DC converter as claimed in claim 1 or 2, comprising a switching controller for controlling a two-state duty cycle. The controller is
4. A primary side controller that feedback-controls each of the output voltages of the at least one first bridge cell so that the first inductor current flowing through the first inductor approaches its command value. The isolated DC/DC converter according to any one of . The voltage between the primary positive terminal and the primary negative terminal is V _dc1 , the voltage between the secondary positive terminal and the secondary negative terminal is V _dc2 , and the voltage of the first insulating capacitor is v. _C1 , where v _C2 is the voltage of the second insulating capacitor,
The command value for each of the output voltages of the at least one first bridge cell includes a feedforward component that is V _dc1 in the first state and −V _dc2 +v _C1 +v _C2 in the second state. Item 5. The isolated DC/DC converter according to item 4. The primary side controller
5. The AC component of the command value of the first inductor current is varied so that the average value of the DC capacitor voltage of each of the at least one first bridge cells approaches a predetermined command value. 6. Or the isolated DC/DC converter according to 5. the at least one first bridge cell is a plurality of first bridge cells;
the primary side controller includes a plurality of capacitor voltage balancers;
Each of the plurality of capacitor voltage balancers is configured such that the voltage of the corresponding one of the DC capacitors of the plurality of first bridge cells approaches the average value of the voltages of the DC capacitors of the plurality of first bridge cells, 7. The isolated DC/DC converter according to claim 4, wherein the command value of the output voltage of the corresponding first bridge cell is corrected. The primary side controller,
5. A DC component of the command value of the first inductor current is changed so that an average value of the voltage of the first insulating capacitor and the voltage of the second insulating capacitor approaches zero. 8. The isolated DC/DC converter according to any one of 1 to 7. The controller includes a secondary controller that generates a command value for each output voltage of the at least one second bridge cell such that a second inductor current flowing through the second inductor approaches its command value. The isolated DC/DC converter according to any one of claims 4 to 8. 10. The isolated DC/DC converter according to claim 9, wherein said secondary side controller has the same configuration as said primary side controller.

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