JP2023037306A - Coupling resonance wireless power transmission system, and transmission characteristic adjustment method therefor - Google Patents

Abstract

To provide a simply configured coupling resonance wireless power transmission system which can maintain high transmission efficiency even if a coupling coefficient k is changed caused by a positional relationship change between two resonators.SOLUTION: A coupling resonance wireless power transmission system performs non-contact power transmission from a first resonator to a second resonator using electromagnetic resonance. The external Q of each of the first and second resonators is adjusted in advance to have the most flattened characteristic of power transmission characteristics under a predetermined condition. At system operation, when the positional relationship between the first and the second resonators is changed from the predetermined condition, the power transmission is executed after a control unit adjusts the external Q of one of the first and second resonators in a state that the external Q of the other resonator in a manner to have center-point matching at the resonant frequency of the first and second resonators.SELECTED DRAWING: Figure 16

Description

The present disclosure relates to a coupled resonance wireless power transmission system and a method of adjusting transmission characteristics in the coupled resonance wireless power transmission system.

A method using electromagnetic induction between two coils has long been known as a method of supplying power without contact. In general, a transformer that uses a magnetic material to strengthen coupling is used as a transformer.

In recent years, as shown in FIG. 1, a power transmitting device and a power receiving device are each provided with a resonator having the same resonance frequency, which is configured by combining a coil and a capacitor. BACKGROUND ART A coupled resonance type wireless power transmission system (hereinafter abbreviated as “wireless power transmission system”) that enables power transmission has attracted attention (see, for example, Non-Patent Document 1). According to Non-Patent Document 1, this type of wireless power transmission system can efficiently transmit a sine wave signal of about 10 MHz over a distance of about 1 m using a coil with a diameter of 50 cm and several turns.

Note that FIG. 1 is a conceptual diagram of a wireless power transmission system using a resonator. In FIG. 1, the power transmitting device UA is drawn on the left side of the wireless power transmission system, and the power receiving device UB is drawn on the right side. The power transmission device UA has a resonator composed of a coil 1 and a capacitor 3, and a power supply 5 for transmitting AC power to the resonator. The power receiving device UB also has a resonator configured by the coil 2 and the capacitor 4, and a load 6 that acquires AC power from the resonator. Note that the power receiving device UB and the power transmitting device UA are typically arranged in a separated state, and the positional relationship between the resonator of the power receiving device UB and the resonator of the power transmitting device UA is usually such that power transmission is not possible. Alignment is often a challenge because it varies from scene to scene.

Japanese Patent Application Laid-Open No. 2021-027778

Andre Kurs, et.al, “Wireless Power Transfer via Strongly Coupled Magnetic Resonances”, Science Vol.317, 6 July 2007 T. Ohira, “Angular expression of maximum power transfer efficiency in reciprocal two-port systems”, Published in 2014 IEEE Wireless Power Transfer Conference, Date of Conference: 8-9 May 2014, INSPEC Accession Number: 14395324 Y. Zhang and Z. Zhao, “Frequency Splitting Analysis of Two-Coil Resonant Wireless Power Transfer”, in IEEE Antennas and Wireless Propagation Letters, vol. 13, pp. 400-402, 2014 A. P. Sample, D. T. Meyer and J. R. Smith, “Analysis, Experimental Results, and Range Adaptation of Magnetically Coupled Resonators for Wireless Power Transfer”, in IEEE Transactions on Industrial Electronics, vol. 58, no. 2, pp. 544-554, Feb. 2011 Ikuo Awai, et.al, “Superiority of BPF theory for design of coupled resonator WPT systems”, Published in Asia-Pacific Microwave Conference 2011, Date of Conference: 5-8 Dec. 2011, INSPEC Accession Number: 12656013 Takehiro Imura, “Wireless Power Transmission by Magnetic Resonance,” p. 94, Morikita Publishing (Janry 31, 2017) I.Awai and A.K.Saha, “Open Ring Resonators Applicable to Wide-band BPF,” Proceedings of Asia-Pacific Microwave Conference 2006, ISBN:978-4-902339-08-6

By the way, when wireless power feeding is performed using electromagnetic resonance between two resonators, in order to obtain high transmission efficiency, the Q value of the resonator (here, from the internal resistance of the LC circuit that constitutes the resonator, A fixed Q value, hereinafter referred to as “unloaded Q”) is high, and a coupling coefficient k (hereinafter also referred to as “k value”), which indicates the strength of electromagnetic coupling between two resonators, is high. is required (see, for example, Non-Patent Document 2).

In general, the unloaded Q of a resonator can be increased by lowering the parasitic resistance of a coil that constitutes the resonator.

On the other hand, the k value is expressed with 1 as the upper limit, and its value varies according to the positional relationship between the two resonators. In particular, the k value is easily affected by the distance between the coils of the two resonators (hereinafter also simply referred to as "the distance between the two resonators"). The k value generally increases as the distance between the two resonators becomes shorter, and at this time, the transmission efficiency during power transmission also becomes higher. The k value decreases as the distance between the two resonators increases or as the orientation of the two resonators changes.

However, what is characteristic of resonance power transmission is that when the distance between the two resonators approaches, due to electromagnetic interference, the resonance frequency, which was previously set to be the same for the two resonators, changes to two frequencies, one on the high frequency side and the other on the low frequency side. (referred to as a bimodal frequency characteristic). As a result, it is known that the peak of the transmission efficiency is also separated, and the transmission efficiency at the original resonance frequency, which is approximately at the center, is reduced (for example, see Non-Patent Document 3 and Non-Patent Document 4). . This phenomenon is called frequency decoupling, and is noticeable in short distances where high-efficiency power transmission is inherently possible. At this time, high-frequency power that is not transmitted is returned to the power supply as reflection. In low frequency circuits it is possible to convert it back to useful power again, but at high frequencies it must be thrown away as a loss.

Conventionally, as a means of overcoming the reduction in transmission efficiency that accompanies frequency separation, circuit components arranged in the input/output lines connected to these resonators or the resonators concerned have been used in accordance with the positional relationship between the two resonators. Attempts have been made to change the parameters so that the frequency characteristic has a constant peak position, or to adjust the frequency of the high-frequency power to be transmitted to the frequency-separated peak (also referred to as frequency tracking).

However, in the former method, it is necessary to change the parameters of the resonator or the circuit components arranged in the input/output line connected to the resonator, so that a special resonator or a special circuit configuration is adopted. However, there are issues in terms of cost and versatility. In the latter method, when a circuit component with a constant impedance (for example, a high-frequency power supply, a rectifier circuit of a load, or a coaxial cable) is used in a power transmitting device or a power receiving device, the frequency used for power transmission and reception is changed. The change inevitably leads to a decrease in transmission efficiency. In addition, the latter method is not realistic due to radio regulation restrictions.

The present disclosure has been made in view of the above problems, and maintains high transmission efficiency with a simple configuration even when the positional relationship between the two resonators changes and the coupling coefficient k changes. It is an object of the present invention to provide a coupled resonance type wireless power transmission system and a method for adjusting transmission characteristics in the coupled resonance type wireless power transmission system.

The main disclosure that solves the above-mentioned problems is
A coupled resonance type wireless power transmission system that uses electromagnetic resonance to perform contactless power transmission from a first resonator to a second resonator,
The external Q of each of the first and second resonators is adjusted in advance under predetermined conditions so that the transmission characteristics of the power transmission are the flattest characteristics (see paragraph [0020]),
During system operation, if the positional relationship between the first resonator and the second resonator has changed from the predetermined condition, the controller causes the resonance frequencies of the first and second resonators to After adjusting the external Q of one of the first and second resonators while fixing the external Q of the other to achieve center point matching (see paragraphs [0026] [0046]), the power transfer run the
It is a coupled resonance type wireless power transmission system.

Also, in other aspects,
A method for adjusting transmission characteristics in a coupled resonance type wireless power transmission system that performs contactless power transmission from a first resonator to a second resonator using electromagnetic resonance, comprising:
adjusting the external Q of each of the first and second resonators in advance under predetermined conditions so that the transmission characteristics of the power transmission are the flattest characteristics;
When the positional relationship between the first resonator and the second resonator is changed from the predetermined condition during system operation, center point matching is performed at the resonance frequencies of the first and second resonators. Adjusting the external Q of one of the first and second resonators while fixing the external Q of the other,
adjustment method.

According to the coupled resonance type wireless power transmission system according to the present invention, even when the positional relationship between the two resonators changes and the coupling coefficient k changes, only by adjusting the external Q of one resonator , it is possible to suppress reflection during power transmission and maintain high transmission efficiency.

Conceptual diagram of a wireless power transmission system using a resonator Circuit diagram of a general wireless power transmission system using a series resonator Circuit diagram of a general wireless power transmission system using parallel resonators A circuit diagram obtained by extracting only the configuration of the power transmission device from the circuit diagram of FIG. 2A A circuit diagram in which only the configuration of the power transmission device is extracted from the circuit diagram of FIG. 2B FIG. 2 is a diagram showing the configuration of the wireless power transmission system according to the first embodiment when using series resonators; FIG. 2 is a diagram showing the configuration of the wireless power transmission system according to the first embodiment when parallel resonators are used; A circuit diagram in which only the configuration of the power transmission device is extracted from the circuit diagram of FIG. 4A A circuit diagram obtained by extracting only the configuration of the power transmission device from the circuit diagram of FIG. 4B A diagram showing a modification of the wireless power transmission system according to the first embodiment FIG. 5B is a diagram showing an example of the configuration of a capacitor applied to a wireless power transmission system using a series resonator (that is, the capacitance division series resonance circuit of FIG. 5A); FIG. 5B is a diagram showing an example of the configuration of a capacitor applied to a wireless power transmission system using parallel resonators (that is, the capacitance-dividing parallel resonant circuit of FIG. 5B); A diagram showing transmission characteristics obtained in a state of maximum flatness matching in a wireless power transmission system In the wireless power transmission system, when the condition (k = 0.04) that satisfies the flattest matching shown in FIG. 8 is changed to k = 0.16, the external Q of the receiving side resonator is adjusted to Diagram showing transmission characteristics when point-matched In the wireless power transmission system, when the condition (k = 0.04) that satisfies the flattest matching shown in FIG. Diagram showing transmission characteristics when point-matched In the wireless power transmission system, when the k value varies from the condition (k=0.04) that satisfies the flattest matching shown in FIG. Diagram showing change in passband width when matched In the wireless power transmission system, when the k value changes from the condition (k = 0.04) satisfying the flattest matching shown in FIG. 8, the external Q of the receiving side resonator is adjusted to achieve center point matching. Figure showing the peak transmission efficiency when A diagram plotting the relationship between peak transmission efficiency and passband width from the data in FIGS. 11 and 12 FIG. 4 is a diagram showing measurement results of absolute values of reflectance and transmittance obtained when power is transmitted from a power transmitting resonator to a power receiving resonator in a wireless power transmission system using resonators. FIG. 2 is a diagram showing, on a Smith chart, the reflectance obtained when power is transmitted from a power transmitting resonator to a power receiving resonator in a wireless power transmission system using resonators. 4 is a flowchart showing an operation example during system operation of the wireless power transmission system according to the first embodiment; A diagram showing a schematic configuration of a wireless power transmission system according to a second embodiment. FIG. 4 is a plan view of an open ring resonator of a wireless power transmission system according to a second embodiment; FIG. 11 is a plan view of a state in which open ring resonators for power transmission and reception of a wireless power transmission system according to a second embodiment face each other; Diagram showing inter-resonator distance dependence of optimum port angle of open ring resonator Diagram showing the external Q at the optimum port angle of the open ring resonator Diagram showing the passband width at the optimum port angle of the open ring resonator Diagram showing peak transmission efficiency at optimum port angle of open ring resonator

Preferred embodiments of the present disclosure will be described in detail below with reference to the accompanying drawings. In the present specification and drawings, constituent elements having substantially the same functions are denoted by the same reference numerals, thereby omitting redundant description.

(First embodiment)
In a wireless power transmission system according to an embodiment of the present disclosure, each of a power transmitting device and a power receiving device is provided with a resonator configured by combining a coil and a capacitor and having the same resonance frequency, and the two resonators are caused to resonate. This is a coupled resonance type wireless power transmission system that enables long-distance power transmission.

Wireless power transmission using resonant circuit resonance is functionally a bandpass filter (BPF) that propagates electromagnetic energy through space. Among BPFs, a type of BPF that transmits without degrading transmission characteristics near the resonance frequency is called a flattest filter or a Butterworth filter. Here, the characteristic in which the efficiency in the vicinity of the center frequency is high and almost constant is referred to as the flattest characteristic.

The flattest characteristic is a transmission characteristic in which the transmission efficiency is approximately 100% in the frequency band _f1 to _f2 sandwiching the natural resonance frequency (that is, the center frequency) _f0 of the resonator. , the transmission efficiency drops rapidly and approaches zero. It is known that in the flattest characteristic, the relationship between the frequency range f ₁ to f ₂ and the k value is given by equation (1).

It is already known in microwave filter theory that the external Q of the resonator should be set to 1/k in order to achieve this flattest characteristic (see, for example, Non-Patent Document 5). That is, frequency separation can be prevented by adjusting the external Q in response to changes in the k value.

Since the external Q is a value determined by the impedance of the input/output line to the resonator, in order to set the external Q of the resonator to 1/k, the input/output impedance to the resonator should be adjusted to an appropriate value. However, it is not easy to arbitrarily change the impedance of the power supply and the load, which are the input/output impedances for the resonator, because the selection of parts must be reconsidered. In addition, it is difficult to electrically change the impedance of the high-frequency power supply and load in power transmission that is used under conditions where the high-frequency characteristics of electronic components and the power/voltage resistance characteristics are marginal.

Therefore, this embodiment proposes, as one aspect, a method of adjusting the external Q by devising a method of supplying power to the resonator without changing the impedance of the power supply and the load. Resonators using coils and capacitors include series resonators and parallel resonators depending on the difference in resonance circuit. (see FIGS. 4A, 4B and 6 below). Adjustment of the external Q can be realized by changing the capacitance of the isolated capacitor while keeping the resonance frequency unchanged.

Further, in a general wireless power transmission system, in many cases, at least one of the power transmitting device and the power receiving device is a mobile body, and the positional relationship between the power transmitting side resonator and the power receiving side resonator (that is, the k value ) will be different for each power transmission. In such a configuration, it is complicated to adjust the resonator parameters of the power transmitting side resonator and the power receiving side resonator by communicating between the power transmitting device and the power receiving device each time power is transmitted. It is not preferable because it leads to complication and control system complication. In particular, the power receiving devices to which power is transmitted from the power transmitting device are not necessarily the same, and the electrical characteristics of the power receiving devices change according to the type and individual differences of the power receiving devices. In such a case, each time the power is actually transmitted, communication is performed between the power transmitting device and the power receiving device, and the resonator parameters of the power transmitting side resonator and the power receiving side resonator are adjusted to the electrical characteristics of both. Optimizing accordingly leads to complication of the control system, which is not preferable.

Therefore, this embodiment proposes, as another aspect, a method of adjusting the external Q that can achieve good transmission characteristics by adjusting only one of the power transmitting resonator and the power receiving resonator during actual use. Specifically, in the wireless power transmission system according to the present embodiment, as a preparation, based on the resonance frequencies of the power transmission side resonator and the power reception side resonator, which are set to be the same in advance, the power transmission side resonator is set to the power reception side resonance frequency. The external Q of the power transmission side resonator and the external Q of the power reception side resonator are adjusted in advance so that the transmission characteristics when power is transmitted to the device become the flattest. Alternatively, by adjusting only the external Q value of one of the resonators on the power receiving side, a method of satisfying the condition of zero reflection at the resonance frequency (center point matching condition described later) is adopted (see Examples described later). .

[Circuit Configuration of Wireless Power Transmission System]
First, the configuration of a general wireless power transmission system will be described with reference to FIGS. 2A to 4B.

2A and 2B are circuit diagrams of typical wireless power transmission systems. FIG. 2A is a circuit diagram of a wireless power transmission system using a series resonant circuit as a resonant circuit of a resonator, and FIG. 2B is a circuit diagram of a wireless power transmission system using a parallel resonant circuit as a resonant circuit of a resonator. The basic configuration of FIGS. 2A and 2B is as explained with reference to FIG. In each circuit diagram below, the configuration of the power transmission device UA is drawn on the left side, and the configuration of the power reception device UB is drawn on the right side.

In the circuit diagram of FIG. 2A , the series resonator of the power transmitting device UA is composed of the coil 1 and the capacitor 3 , and the series resonator of the power receiving device UB is composed of the coil 2 and the capacitor 4 . The resistance unit 5 of the power transmission device UA corresponds to the impedance of the power supply 5 (that is, the output impedance of the power supply circuit (hereinafter also referred to as “external impedance”). The resistance unit 6 of the power reception device UB is It corresponds to the impedance of the load 6 (hereinafter also referred to as "external impedance").

The power supply 5 is composed of, for example, a battery and an oscillator that converts DC power supplied from the battery into high-frequency power. Also, the load 6 is composed of, for example, a rectifier circuit that rectifies high-frequency power and a battery that stores the DC power rectified by the rectifier circuit.

Also in the circuit diagram of FIG. 2B, the same reference numerals as in the circuit diagram of FIG. 2A are attached to each circuit component. In the circuit diagram of FIG. 2B, the coil 1 and the capacitor 3 constitute a parallel resonance circuit of the power transmission device UA, and the coil 2 and the capacitor 4 constitute a parallel resonance circuit of the power reception device UB.

In the wireless power transmission system of FIGS. 2A and 2B, the coil 1 and the coil 2 are arranged to face each other and are magnetically coupled. Therefore, in the circuit diagrams of FIGS. and the resonator of the power receiving device UB are represented by the mutual inductance between the two coils.

Here, for the external Q of the resonator, the single series resonant circuit shown in FIG. 3A (a circuit diagram obtained by extracting only the configuration of the power transmission device UA from the circuit diagram of FIG. 2A) and the single parallel resonant circuit shown in FIG. 3B (A circuit diagram obtained by extracting only the configuration of the power transmission device UA from the circuit diagram of FIG. 2B) is used for analysis.

Hereinafter, for convenience of explanation, the circuit in FIG. 3A is also referred to as a "single-capacitance series-resonant resonator", and the circuit in FIG. 3B is also referred to as a "single-capacitance parallel-resonant resonator".

In the following description, the inductance of the coil 1 forming the LC resonator is L ₀ , the capacitance of the capacitor 3 is C ₀ , and the impedance of the power source 5 is Z _S . When considering the power receiving device UB side, the circuit parameters of _each part on the power receiving device UB side are similar to the circuit parameters of each part on the power transmitting device UA side. Let C ₀ and the impedance of the load 6 be Z _S .

First, the resonance frequency _f0 of the LC resonator is represented by the following equation (2).

For subsequent analysis, the characteristic impedance _ZC of the resonator is defined as in the following equation (3). Note that this characteristic impedance _ZC is a dimension value of resistance [Ω].

The Q value of a generally used resonator is the Q value when no load is connected, and is called unloaded Q. This unloaded Q is the ratio of the characteristic impedance of the resonator and the internal resistance of the coil. On the other hand, the Q value due to the impedance of the power supply and load connected to input and output signals to and from the resonator is called external Q (hereinafter referred to as Qe).

The external Q in the case of the series type is represented by the following equation (4). Here, _ZS is the impedance of the power supply 5 or the load 6.

On the other hand, the parallel type external Q is represented by the following equation (5).

Conventionally, it is said that the condition for realizing the flattest characteristic is to simultaneously satisfy the following equations (6a) and (6b) (see, for example, Non-Patent Document 5). Here, the suffix T indicates the power transmission side and R indicates the power reception side (the same applies hereinafter), and the equations (6a) and (6b) are calculated according to the variation of the coupling coefficient k on the power transmission side and the power reception side, respectively. means that the extrinsic Q needs to be adjusted by Hereinafter, this condition will be referred to as "flattest matching".

Contactless power supply is performed by using two of these resonators, and the coupling coefficient k at that time can be expressed by the mutual inductance _Lm of the two coils, and is expressed by the following equation (7).

Here, L _T and L _R are the inductances of the power transmission/reception coils. The analysis of these basic circuits is detailed in Non-Patent Document 6. When the two resonators are of the series resonance type (Fig. 3A), the input impedance _Zin of the power transmitting side resonator, the power receiving side resonator, and the load 6 as viewed from the power supply 5 side is equal to the internal resistance of the coils 1 and 2 being zero. Then, the value of the following equation (8) is obtained. where _ZSR is the impedance of load 6;

To eliminate the reflection in this resonator combination circuit, the idea of impedance matching in microwave circuits is applied. According to this, since the impedance _ZST of the power supply 5 should be equal to _ZIN , satisfying the following equation (9) means satisfying the impedance matching condition.

By transforming the equation (9) using the equations (3), (4), and (7), it is expressed as the following equation (10). Note that _ZCR is the impedance of the resonator of the power receiving device UB, and _ZCT is the impedance of the resonator of the power transmitting device UA.

The condition of formula (10) is relaxed from the conditions for realizing the flattest type (that is, formulas (6a) and (6b)). This condition is hereinafter referred to as "central point matching". This condition is strictly a condition for making the reflectance zero at the center frequency f0 (that is, a condition for impedance matching), and reflection occurs at other frequencies. That is, the transmission band becomes narrower than that of the flattest type. However, unlike information communication, power transmission does not require a wide band because it uses a single frequency.

The center point matching condition expressed by Equation (10) indicates that it is sufficient to adjust only the external Q of either the power transmitting side or the power receiving side when the coupling coefficient k changes. This is an important discovery. Equation (10) is the transmission between series resonators. established regardless.

Here, a method for adjusting the external Q will be described.

The external Q can be adjusted by changing the impedance _ZST of the power supply 5 and the impedance _ZSR of the load 6, as shown in equations (4) and (5). However, when using high frequencies, it is generally necessary to use lines of constant impedance (typically 50Ω) to prevent signal reflection. In addition, it is not easy to change the impedance of the power supply 5 and the load 6, and it is extremely difficult from a practical point of view to set the impedance of the power supply 5 and the load 6 to a value that matches the change in the k value. be. Furthermore, if the characteristic impedance is fixed at 50Ω, a widely used power supply or the like can be used as it is, which has the advantage of reducing the product cost.

The wireless power transmission system according to the present embodiment is designed in consideration of such problems, and can achieve the flattest characteristics while keeping the impedance of the power supply 5 and/or the load 6 constant.

FIG. 4A is a diagram showing the configuration of the wireless power transmission system according to this embodiment when using series resonators, and FIG. 4B is a diagram showing the configuration of the wireless power transmission system according to this embodiment when using parallel resonators. is a diagram showing the configuration of.

In the wireless power transmission system according to the present embodiment, when a series resonance circuit is used (see FIG. 4A), the capacitor of the resonator is divided in parallel, and the power supply 5 and the load 6 are placed in series with one capacitor. The configuration is different from the conventional wireless power transmission system (see FIG. 2A). Also, when a parallel resonant circuit is used (see FIG. 4B), the capacitors (capacitors 3 and 4) are divided in series, and the power source 5 and the load 6 are placed in parallel with one capacitor. This is different from the conventional wireless power transmission system (see FIG. 2B).

In FIG. 4A, the capacitors 3a and 3b arranged in parallel when viewed from the coil 1 are the capacitors 3 constituting the resonator of the power transmission device UA, and the capacitors 4a and 4b arranged in parallel when viewed from the coil 1 are the capacitors 3a and 4b. A capacitor 4 constitutes a resonator of the power receiving device UB. In FIG. 4B, the capacitor 3c and the capacitor 3d arranged in series when viewed from the coil 1 are the capacitor 3 constituting the resonator of the power transmission device UA, and the capacitor 4c and the capacitor 3d arranged in series when viewed from the coil 1 4d is a capacitor 4 forming a resonator of the power receiving device UB. These capacitors 3a, 3b, 3c, 3d, 4a, 4b, 4c and 4d are hereinafter referred to as "dividing capacitors".

Here, the control unit 10A of the power transmission device UA drives the movable electrodes of the capacitor 3 when adjusting the capacitance values of the divided capacitors 3a, 3b (3c, 3d) that constitute the capacitor 3 on the side of the power transmission device UA. (see FIGS. 7A and 7B), and the control unit 10B of the power receiving device UB adjusts the capacitance values of the dividing capacitors 4a and 4b (4c and 4d) constituting the capacitor 4 of the power receiving device UB. Second, it is a controller that drives the movable electrode of the capacitor 4 .

Here, regarding the external Q in the wireless power transmission system according to the present embodiment, the single series resonance circuit shown in FIG. Analysis is performed using a single parallel resonant circuit (a circuit diagram obtained by extracting only the configuration of the power transmission device UA from the circuit diagram of FIG. 4B).

In the following description, the resonant circuit shown in FIG. 5A is referred to as "capacitance division series resonant circuit", and the resonant circuit illustrated in FIG. 5B is referred to as "capacitance division parallel resonant circuit".

First, the combined impedance Z of the single-capacitance series resonance circuit of FIG. 3A is a series connection of the coil 1, the capacitor 3, and the resistor 5, so it is represented by the following equation (11). Incidentally, the external Q at that time is given by the above equation (4).

Here, in the capacitance-dividing series resonator of FIG. 5A, the capacitance value of the dividing capacitor 3a to which the power supply 5 is not connected is C _a , and the capacitance value of the dividing capacitor 3b to which the power supply 5 is connected is C _b , let the impedance of the power source 5 be Z _S ; Then, the overall combined impedance Z of this circuit is approximation that 2πfC _a Z _S is sufficiently smaller than 1, and is given by the following equation (12).

Here, if the capacitance values Ca _{and Cb} of the dividing capacitors 3a and 3b are set so as to satisfy the following equation (13), the resonance frequency of the resonator can be matched with the value in FIG. 3A. .

Here, as a parameter relating to the ratio between the capacitance value _Ca of the dividing capacitor 3a and the capacitance value _Cb of the dividing capacitor 3b, a new parameter a value of the following equation (14) is introduced. The a value is a value between 0 and 1, as can be seen from equation (14).

Substituting this value of a into equation (12) reveals that equation (12) is an equation in which Z _S in equation (11) is replaced with a ² Z _S. From this, it can be seen that the external Q of the circuit of FIG. 5A is given by the following equation (15).

Next, the case of a parallel resonant circuit is analyzed. Since the combined admittance Y of the single-capacitance parallel resonant circuit of FIG. 3B is the parallel connection of the coil 1, the capacitor 3, and the resistor 5, it is expressed as in Equation (16). Incidentally, the external Q at that time is given by the above equation (5).

Here, in the capacitance-dividing parallel resonator of FIG _{. d} , let the impedance of the power supply 5 be _Zs . With the approximation that 2πfC _c Z _S is well above 1, the combined admittance Y of this circuit is given by equation (17) below.

Here, when the capacitance values _Cc and _Cd of the dividing capacitors 3c and 3d are set so as to satisfy the following equation (18), the resonance frequency of the resonator can be matched with the resonance frequency of the original parallel resonator. can be done.

Here, as a parameter relating to the ratio between the capacitance value _Cc of the dividing capacitor 3c and the capacitance value _Cd of the dividing capacitor 3d, a new parameter b value of the following equation (19) is introduced. Incidentally, the b value is a value of 1 or more as can be seen from the equation (19).

Substituting this b value into equation (17) reveals that this equation is obtained by replacing Z _S in equation (16) with b ² Z _S. From this it can be seen that the extrinsic Q is given by the following equation (20).

Summarizing the above, the center point matching condition that the Qe product is 1/k ² is obtained from the following equation (21 ) can be converted into a formula.

If the characteristic impedance of the resonator is represented by the resonance frequency and the inductance value using the equation (3), the equation (21) can be further transformed into the following equation (22).

Similarly, the conditions for center point matching can be converted from Equation (20) to Equation (23) below when both the power transmitting side resonator and the power receiving side resonator are of parallel type.

FIG. 6 is a diagram showing a modification of the wireless power transmission system according to this embodiment. In the wireless power transmission system of FIG. The resonator of is composed of a capacitance division parallel resonance circuit.

In the case of the wireless power transmission system of FIG. 6, the conditions for center point matching can be converted from Equations (15) and (20) to Equation (24) below.

As a modification of the wireless power transmission system according to the present embodiment, it is possible to use a series type resonator for the power transmission side and a parallel type resonator for the power reception side. Just swap T and R.

From these equations (22), (23), and (24), when satisfying the center point matching condition, one of the capacitors of the power transmitting side resonator or the power receiving side resonator (that is, capacitor 3 or capacitor 4 ) is sufficient to adjust the capacitance ratio of the dividing capacitors.

[Configuration of Resonator Capacitor]
Here, an example of the configuration of the resonator capacitors 3 and 4 applied to the wireless power transmission system according to this embodiment will be described.

FIG. 7A is a diagram showing an example of the configuration of capacitors 3 and 4 applied to a wireless power transmission system using series resonators (that is, the capacitance division series resonance circuit of FIG. 5A).

As shown in FIG. 5A, the capacitors 3 and 4 applied to the capacitance division series resonance circuit are divided into two in parallel when viewed from the coil, and one of the two divided capacitors formed by dividing into two It has a configuration in which a power source 5 as a power transmission unit or a load 6 as a power reception unit is connected in series only to a dividing capacitor. For convenience of explanation, one of the two divided capacitors of the capacitor 3 (or the capacitor 4) is hereinafter referred to as a "first divided capacitor" and the other is referred to as a "second divided capacitor".

These capacitors 3 and 4 adjust the ratio between the capacitance value of the first dividing capacitor and the capacitance value of the second dividing capacitor so as to adjust the external Q of the resonator as shown in equation (15). It is composed of a three-terminal variable capacitor that enables

Specifically, the capacitors 3 and 4 applied to the capacitance-dividing series resonant circuit are configured by variable capacitors shown in FIG. 7A so that the capacitance can be easily adjusted. The variable capacitors are electrically separated from each other and arranged side by side to face the same direction as the first fixed electrode E1 and the second fixed electrode E2, and the first fixed electrode E1 and the second fixed electrode E2. While maintaining a constant distance between each of the first fixed electrode E1 and the second fixed electrode E2 so as to face each other, the first fixed electrode E1 and the second fixed electrode E2 are arranged so as to be movable along the direction in which the first fixed electrode E1 and the second fixed electrode E2 are arranged. and a movable electrode E3 provided. A first divided capacitor is formed between the first fixed electrode E1 and the movable electrode E3, and a second divided capacitor is formed between the second fixed electrode E2 and the movable electrode E3. In this variable capacitor, when the movable electrode E3 is moved, the sum of the area where the first fixed electrode E1 and the movable electrode E3 face each other and the area where the second fixed electrode E2 and the movable electrode E3 face each other is configured to be constant.

As a result, the overlapping area of the first fixed electrode E1 and the movable electrode E3 is _S1 , and the overlapping area of the second fixed electrode E2 and the movable electrode E3 is _S2 . Assuming that ₁ + _S2 is always constant (for example _{, S1} + _S2 = _S0 ), the parallel combined capacitance C0 of the capacitance value _C1 of the first dividing capacitor and the capacitance value _C2 of the second dividing capacitor becomes constant as in the following equation (25).

That is, by using this variable capacitor, the capacitance value _C1 of the first dividing capacitor and the capacitance value C of the second dividing capacitor can be obtained while keeping the parallel combined capacitance _C0 of the first dividing capacitor and the second dividing capacitor constant. It is possible to adjust the ratio of _two . This makes it possible to easily adjust the external Q of the resonator while keeping the resonance frequency constant.

FIG. 7B is a diagram showing an example of the configuration of capacitors 3 and 4 applied to a wireless power transmission system using parallel resonators (that is, the capacitance-dividing parallel resonant circuit of FIG. 5B).

As shown in FIG. 5B, the capacitors 3 and 4 applied to the capacitance-dividing parallel resonant circuit are divided into two in series when viewed from the coil, and one of the two divided capacitors formed by dividing into two It has a configuration in which a power source 5 as a power transmission unit or a load 6 as a power reception unit is connected in parallel only to the divided capacitors. For convenience of explanation, one of the two divided capacitors of the capacitors 3 and 4 is hereinafter referred to as the "third divided capacitor", and the other is referred to as the "fourth divided capacitor".

These capacitors 3 and 4 are arranged so that the capacitance value of the third divided capacitor and the fourth divided capacitor is composed of a three-terminal variable capacitor that can adjust the ratio of the capacitance values of .

Specifically, the capacitors 3 and 4 according to the present embodiment are both configured by variable capacitors shown in FIG. 7B so that the capacitance can be easily adjusted. This variable capacitor has a first fixed electrode E4 and a second fixed electrode E5 which are electrically separated and opposed to each other, and between the first fixed electrode E4 and the second fixed electrode E5. While keeping the areas facing the fixed electrode E4 and the second fixed electrode E5 constant, they are arranged so as to be movable along the direction in which the first fixed electrode E4 and the second fixed electrode E5 face each other (the arrow direction in FIG. 7B). and a movable electrode E6. A third divided capacitor is formed between the first fixed electrode E4 and the movable electrode E6, and a fourth divided capacitor is formed between the second fixed electrode E5 and the movable electrode E6. When the movable electrode E6 is moved, this variable capacitor has a distance _d3 between the first fixed electrode E4 and the movable electrode E6 and a distance _d4 between the second fixed electrode E5 and the movable electrode E6. is configured so that the sum of

The first fixed electrode E4, the second fixed electrode E5, and the movable electrode E6 are, for example, electrode plates having the same area, and are arranged such that their facing surfaces are parallel to each other. Here, the area of the first fixed electrode E4, the second fixed electrode E5, and the movable electrode E6 is S ₀ , the distance between the first fixed electrode E4 and the movable electrode E6 is d ₃ , the second fixed electrode E5 and the movable electrode Let the distance between E6 be _d4 . Also, assuming that the positions of the first fixed electrode E4 and the second fixed electrode E5 are fixed and the distance between the first fixed electrode E4 and the second fixed electrode E5 is fixed at _d0 , , d ₃ +d ₄ is constant (d ₃ +d ₄ =d ₀ ) even when the movable electrode E6 is moved. Then, assuming that the capacitance value of the third divided capacitor composed of the first fixed electrode E4 and the movable electrode E6 is C ₃ and the capacitance value of the fourth divided capacitor composed of the second fixed electrode E5 and the movable electrode E6 is C ₄ , C ₃ and _C4 are connected in series, the series combined capacitance _C0 is constant as shown in the following equation (26).

That is, by using this variable capacitor, the capacitance value _C3 of the third dividing capacitor and the capacitance value C4 of the second dividing capacitor can be changed while keeping the series combined capacitance of the third dividing capacitor and the _fourth dividing capacitor constant. It is possible to adjust the ratio. This makes it possible to easily adjust the external Q of the resonator while keeping the resonance frequency constant.

In the wireless power transmission system according to the present embodiment, during actual use, only the external Q of one of the power transmitting resonator and the power receiving resonator is adjusted to satisfy the center point matching condition. From this point of view, of the capacitor 3 of the power transmission side resonator and the capacitor 4 of the power reception side resonator, only the capacitor on the side to be adjusted during actual use is configured by the variable capacitor shown in FIG. 7A or 7B. A non-variable capacitor may be used for the capacitor on the side not to be adjusted. This makes it possible to further simplify the device.

However, in the wireless power transmission system according to the present embodiment, at the stage of preparation, the power transmitting side resonator and the power receiving side Both external Qs of the resonator must be adjusted. Therefore, it is preferable to configure both the power transmission side capacitor 3 and the power reception side capacitor 4 with the variable capacitors shown in FIG. 7A or 7B.

On the other hand, although there are many demerits from a practical point of view as described above, it is also possible to configure one or both of the power transmitting side resonator and the power receiving side resonator with a single-capacitor series resonant circuit or a single-capacitor parallel resonant circuit. is. In this case, the external Q of the power transmitting resonator can be adjusted by adjusting the impedance of the power source 5 , and the external Q of the power receiving resonator can be adjusted by adjusting the impedance of the load 6 .

[Method for adjusting transmission characteristics during power transmission in wireless power transmission system]
A method of adjusting transmission characteristics during power transmission in the wireless power transmission system according to the present embodiment will be described below together with results of verification by simulation.

In the wireless power transmission system according to the present embodiment, as a preparation, power is received from the power transmission side resonator based on the resonance frequencies of the power transmission side resonator and the power reception side resonator that are set to be the same under predetermined conditions in advance. The external Q of the power transmission side resonator and the power receiving side so that the transmission characteristics when power is transmitted to the side resonator are the flattest characteristics (that is, so that the equations (6a) and (6b) are satisfied) Each external Q of the resonator is adjusted. In actual use, if the positional relationship (that is, the relative position and/or angle) between the power transmitting resonator and the power receiving resonator changes, and as a result the coupling coefficient k changes from a predetermined condition, By adjusting only the external Q of one of the resonator and the receiver resonator, the center point matching condition (i.e., Eq. (22), Eq. (23), or Eq. (24)) is satisfied, whereby , to enable power transmission with good transmission efficiency with zero reflection.

Below, the usefulness of the method for adjusting transmission characteristics during power transmission in the wireless power transmission system according to the present embodiment will be described with verification results obtained by simulation.

Specifically, in this simulation, first, when the coupling coefficient k is 0.04, the external Q of each of the power transmitting side resonator and the power receiving side resonator is adjusted so as to satisfy the condition of the flattest matching, After that, the coupling coefficient k was variously changed from 0.04, and the external Q of only one of the power transmitting side resonator and the power receiving side resonator was adjusted to satisfy the center point matching condition. In this simulation, assuming that a user exists on the power receiving device UB side, such as an electric vehicle, in actual use, only the external Q of the power receiving side resonator is adjusted, and the center point matching condition is set. It is configured to satisfy

In this simulation, as described above, each of the power transmission side resonator and the power reception side resonator was configured by a capacitance division series resonance circuit or a capacitance division parallel resonance circuit. The external Q of the power transmitting resonator is adjusted by adjusting the capacitance ratio of the divided capacitors 3 a and 3 b (or 3 c and 3 d) forming the capacitor 3 , and the external Q of the power receiving resonator is adjusted by the capacitor 4 was performed by adjusting the capacitance ratio of the dividing capacitors 4a and 4b (or 4c and 4d) constituting the .

More specifically, in this simulation, when both the resonators of the power receiving device UB are of the divided-capacitor series type, when the resonators of the power transmitting device UA and the power receiving device UB are both of the divided-capacitor parallel type, the power transmitting device UA When the resonator of the power receiving device UB is of the divided capacitance parallel type and the resonator of the power receiving device UB is of the divided capacitance series type, and when the resonator of the power transmitting device UA and the resonator of the power receiving device UB are both of the single capacitance series resonance type A simulation was performed for four modes. The simulation for the case where both the resonator of the power transmitting device UA and the resonator of the power receiving device UB are of the single-capacitor series-resonant type was performed by adjusting the external Q of the resonator by the divided-capacitor series-type and the divided-capacitor parallel-type. can work in the same way as the normal external Q adjustment method.

In this simulation, each circuit parameter of the wireless power transmission system was set as follows with application to an electric vehicle in mind. The inductance of the coil 1 of the power transmitting side resonator was set to 4 μH, and the inductance 2 of the coil 2 of the power receiving side resonator was set to 1.2 μH. Also, it is assumed that the wires of the coils 1 and 2 have an internal resistance of 0.02Ω for a coil of 1.2μH. Here, assuming that the resonance frequency is 10 MHz, from equations (2) and (3), the capacitance of the capacitors used in the power transmission side resonator and the power reception side resonator (here, capacitors 3 and 4 before capacity division) is are 63.3 pF and 211 pF, respectively, and the characteristic impedances of the resonators are determined to be 251 Ω and 75.4 Ω, respectively. Also, in this simulation, the impedance of each of the power source 5 and the load 6 was set to 50Ω.

Here, when the coupling coefficient k is 0.04, the condition for satisfying the flattest matching condition is, as shown in Equations (6a) and (6b), The extrinsic Q is Q _eT =1/k (=25) and Q _eR =1/k (=25).

By adjusting the impedance of the power supply 5 or the load 6, in the case of a single capacitor series type, in the case of a single capacitor series type, to satisfy the condition of maximum flatness matching, from equation (4), 10.1 Ω on the transmitting side and 3.02 Ω on the receiving side, In the case of the single-capacitor parallel type, the power transmission side is 6283 Ω and the power reception side is 1885 Ω from equation (5). Without capacitive splitting (i.e. single capacitor in series or single capacitor in parallel), the impedance of source 5 and load 6 would be far from the commonly used 50Ω anyway. there is For this reason as well, a special circuit design is required to satisfy the condition of maximum flatness matching using a single-capacity series resonator and a single-capacity parallel-type resonator. It can be seen that the single-capacitance parallel resonator cannot be said to be a realistic configuration from a practical point of view.

On the other hand, in order to satisfy the condition of maximum flatness matching by adjusting the split capacitor, in the case of the split capacitor series type, the parameter a is 0.448 for the power transmission side capacitor 3, 0.25 for the power reception side capacitor 4, and In the case of the capacitance parallel type, the parameter b may be set to 11.2 for the power transmission side capacitor 3 and 6.14 for the power reception side capacitor 4 . The reason why the values of the power transmission side capacitor 3 and the power reception side capacitor 4 are different here is that the coil inductance is set to a different value on the power transmission side and the power reception side.

FIG. 8 is a diagram showing transmission characteristics obtained in the most flat matching state in the wireless power transmission system.

Each graph in FIG. 8 represents transmission characteristics of the following aspects. Here, it is assumed that the internal resistances of the coils 1 and 2 are zero.
Graph 11: Transmission when the transmitting side resonator and the receiving side resonator are composed of a single-capacity series resonance circuit, and the impedance of the power supply 5 or the load 6 is adjusted to achieve the flattest matching. Characteristic graph 12: In a mode in which the power transmission side resonator and the power reception side resonator are composed of a divided capacitor series resonance circuit, the maximum flatness matching is achieved by adjusting the a value of each of the power transmission side capacitor 3 and the power reception side capacitor 4. Transmission characteristic graph 13 in the case of: In a mode in which the power transmission side resonator and the power reception side resonator are configured by a divided capacitance parallel resonance circuit, by adjusting the b value of each of the power transmission side capacitor 3 and the power reception side capacitor 4, Transmission characteristic graph 14 for maximum flatness matching: In a mode in which the power transmission side resonator is composed of a divided capacitance parallel resonance circuit and the power receiving side resonator is composed of a division capacitance series resonance circuit, the power transmission side capacitor 3 By adjusting the a value of the power receiving side capacitor 4 and the b value of the receiving side capacitor 4, the transmission characteristics when the flattest matching is achieved

Referring to graphs 11 to 14, it can be seen that substantially the same maximum flatness characteristics are obtained in any case. If the passband width is defined at a level of transmission efficiency of 80% in consideration of the state of use in power transmission, the passband widths of the flattest characteristics of graphs 11 to 14 are all about 400 kHz. In the other graphs below, the passband width of the transmission characteristics is calculated based on the transmission efficiency of 80%.

Next, assuming that the positional relationship between the power transmitting device UA and the power receiving device UB has changed from the above condition (that is, the condition that satisfies the flattest matching), the value of the coupling coefficient k is changed from 0.04 to 0.16. , 0.08, 0.05, 0.03, 0.02, 0.01, 0.004, and 0.001, under which conditions the external Q of the transmitting resonator is 25(Q _eT = 1/k), and the external Q of the resonator on the power receiving side is adjusted to satisfy the center point matching (Q _eT *Q _eR = 1/k ² ).

本シミュレーションの演算結果を参照すると、４態様それぞれで中央点整合を充足させた場合の伝送特性そのものには、大きな違いは無かった（後述する図９、図１０を参照）。即ち、いずれの伝送特性においても、ピーク位置での伝送効率（以下、「ピーク伝送効率」とも称する）が得られる周波数は、送電側共振器及び受電側共振器の共振周波数付近（ここでは、１０ＭＨｚ付近）であり、その際のピーク伝送効率の値や通過帯域幅は、略同一の値となった。 Table 1 shows the parameters ( That is, the a value of the divided capacitor series type, the b value of the divided capacitor parallel type, and the impedance of the load 6 in the single capacitor series resonance type are shown.

Referring to the calculation results of this simulation, there was no significant difference in the transmission characteristics themselves when center point matching was satisfied in each of the four modes (see FIGS. 9 and 10 described later). That is, in any transmission characteristic, the frequency at which the transmission efficiency at the peak position (hereinafter also referred to as "peak transmission efficiency") is obtained is near the resonance frequency of the power transmission side resonator and the power reception side resonator (here, 10 MHz ), and the values of the peak transmission efficiency and the passband width at that time were approximately the same.

FIG. 9 shows the external Q of the receiving side resonator when changing from the condition (k=0.04) that satisfies the flattest matching shown in FIG. 8 to k=0.16 in the wireless power transmission system. FIG. 10 is a diagram showing transmission characteristics when adjusted to achieve center point matching;

Each graph in FIG. 9 represents transmission characteristics of the following aspects.
Graph 11a: In a mode in which the power transmitting side resonator and the power receiving side resonator are composed of single-capacitance series resonant circuits, when k=0.16, the impedance of the power source 5 and the load 6 respectively Transmission characteristic graph 12a when maximum flatness matching is performed in the adjustment: when the power transmitting side resonator and the power receiving side resonator are configured with a divided capacitance series resonance circuit and k = 0.16 Second, a transmission characteristic graph 13a when center point matching is performed by adjusting the a value of the power receiving side capacitor 4: A mode in which the power transmitting side resonator and the power receiving side resonator are configured by a divided capacitance parallel resonance circuit, , and k=0.16, the b value of the power receiving side capacitor 4 is adjusted and center point matching is performed. In a mode in which the power receiving side resonator is composed of a divided capacitor series resonance circuit, and when k=0.16, center point matching is performed by adjusting the b value of the power receiving side capacitor 4. transmission characteristics

Here, when the transmission characteristics were adjusted to the flattest characteristics by adjusting the external Q of both the power transmission side resonator and the power reception side resonator, the passband width was 1.64 MHz from the graph 11a. On the other hand, when only the external Q of the power receiving side resonator is adjusted and the transmission characteristics are adjusted to center point matching, the passband width decreases to 380 to 420 kHz from graphs 12a, 13a, and 14a. However, even with center point matching, the peak transmission efficiency is nearly 100%.

FIG. 9 shows only the transmission characteristics (graph 11a) in a mode in which the power transmission side resonator and the power reception side resonator are composed of single-capacity series resonance circuits as an example of the transmission characteristics of the flattest characteristics. The same applies to transmission characteristics in a mode in which the power transmitting side resonator and the power receiving side resonator are configured by a divided capacitor series resonant circuit or a divided capacitor parallel resonant circuit.

FIG. 10 shows the external Q of the receiving side resonator when changing from the condition satisfying the flattest matching shown in FIG. 8 (k=0.04) to k=0.004 in the wireless power transmission system. FIG. 10 is a diagram showing transmission characteristics when adjusted to achieve center point matching;

Each graph in FIG. 10 represents transmission characteristics of the following aspects.
Graph 11b: In a mode in which the power transmitting side resonator and the power receiving side resonator are composed of single-capacitance series resonant circuits, when k=0.004, the impedance of the power source 5 and the load 6 respectively Transmission characteristic graph 12b when maximum flatness matching is performed in the adjustment: when the power transmitting side resonator and the power receiving side resonator are configured with a divided capacitor series resonance circuit and k = 0.004 In addition, a transmission characteristic graph 11c when center point matching is performed by adjusting the a value of the power receiving side capacitor 4: A transmission characteristic graph when the coils 1 and 2 have internal resistance in the same manner as the graph 11b 12c: Transmission characteristics when coils 1 and 2 have internal resistance in the same manner as graph 12b. In graphs 11b and 12b, the internal resistance of coils 1 and 2 is assumed to be zero.

Here, when the transmission characteristics were adjusted to the flattest characteristics by adjusting the external Q of both the power transmission side resonator and the power reception side resonator, the passband width was 40.0 kHz from graph 11b. On the other hand, when only the external Q on the power receiving side is adjusted and the transmission characteristics are adjusted to match the center point, the passband width is reduced to 4.0 kHz from the graph 12b. The reason why the passband width of each bluff in FIG. 10 is narrower than the passband width of each bluff in FIG. 9 is that in FIG. This is because it is set.

Assuming that the internal resistance of the coils 1 and 2 is zero [Ω], the peak transmission efficiency is almost 100% for both the flattest matching and the center point matching. However, if the internal resistance of coils 1 and 2 is assumed to be a realistic value (for example, 0.02Ω per meter of length), the peak transmission efficiency will also decrease. In our simulations, it dropped to 96% for the flattest match (see graph 11c) and 80% for the midpoint match (see graph 12c). A decrease in peak transmission efficiency due to the internal resistance of the coils 1 and 2 is an unavoidable problem in resonator coupling. I understand.

FIG. 11 shows the adjustment of the external Q of the power receiving side resonator when the k value varies from the condition (k=0.04) that satisfies the flattest matching shown in FIG. 8 in the wireless power transmission system. FIG. 10 is a diagram showing a change in passband width when center point matching is performed.

Each graph in FIG. 11 represents the passband width in the following transmission characteristics.
Graph 31a: A mode in which the transmitting-side resonator and the receiving-side resonator are composed of single-capacitance series-resonant circuits. Passband width graph 32a when performing Passband width when adjusting the a value of the power receiving side capacitor 4 and performing center point matching (the impedance of the power supply 5 and the load 6 is set to 50Ω)

For the flattest match at each k value, the passband width varies proportionally to 1/k as per filter theory (see graph 31a). On the other hand, in the case of non-flattest matching, even after center point matching, the passband width is At the very least it is narrower than for the flattest match (see graph 32a). Also, in this case, the more the k value deviates from k=0.04 (that is, the k value for flattest matching), the narrower the passband width is even after center point matching.

That is, from FIG. 11, it can be seen that the condition for the k value for center point matching (that is, Q _eT *Q _eR =1/k ² ) is the condition for the k value for flattest matching (that is, Q _eT =1 /k and Q _eR =1/k), the passband width after center point matching becomes wider.

FIG. 12 shows that in the wireless power transmission system, when the k value changes from the condition (k=0.04) that satisfies the flattest matching shown in FIG. FIG. 10 is a diagram showing peak transmission efficiency with center point matching;

Each graph in FIG. 12 represents the peak transmission efficiency of the transmission characteristics of the following aspects.
Graph 31b: A mode in which the transmitting side resonator and the receiving side resonator are composed of single-capacitance series resonant circuits. Peak transmission efficiency when performing (when coils 1 and 2 have internal resistance)
Graph 32b: In a mode in which the power transmission side resonator and the power reception side resonator are configured by divided capacitance series resonance circuits, when the k value changes from k = 0.04, the a value of the power reception side capacitor 4 Adjustment, peak transmission efficiency with center point matching (impedance of power supply 5 and load 6 is set to 50Ω) (when coils 1 and 2 have internal resistance)
Graph 33b: Peak transmission efficiency when the internal resistance of coils 1 and 2 is set to zero in the same manner as graph 32b.

Assuming zero internal resistance in coils 1 and 2, the peak transmission efficiency is 100% even for k=0.001 (see graph 33b). However, assuming the resistance values of copper and aluminum, which are actual wiring materials, the peak transmission efficiency decreases at small k values even with maximum flatness matching (see graph 31b). Then, in the case of center point matching, the reduction in peak transmission efficiency becomes even more pronounced (see graph 32b).

FIG. 13 is a diagram plotting the relationship between peak transmission efficiency and passband width from the data of FIGS. 11 and 12. In FIG.

Each graph in FIG. 13 represents the relationship between the peak transmission efficiency and the passband width of the transmission characteristics of the following aspects.
Graph 31c: A mode in which the transmitting side resonator and the receiving side resonator are composed of a single capacitance series resonant circuit, and at each k value, by adjusting the impedance of the source 5 and the load 6 respectively, the flattest matching is achieved. Graph 32c of the relationship between the peak transmission efficiency and the passband width when performing is changed, the relationship between the peak transmission efficiency and the passband width when center point matching is performed by adjusting the a value of the power receiving side capacitor 4 (the impedance of the power supply 5 and the load 6 is set to 50Ω )

The peak transmission efficiency when center point matching is performed is determined by the passband width, and in order to obtain high peak transmission efficiency when performing center point matching, it is important to obtain transmission characteristics with a wide passband width. (see graph 32c). Considering this result together with the results of FIG. 11, the external Q (that is, Q _eT * Q _eR = 1/k ² ) when center point matched is the external Q (that is, Q _eT = 1/k and Q _eR = 1/k), the higher the transmission efficiency after center point matching. It should be noted that this phenomenon was found regardless of the configurations of the power transmitting side resonator and the power receiving side resonator (not shown here).

In this way, by adjusting the external Q of the power transmitting side resonator and the external Q of the power receiving side resonator in advance under predetermined conditions (i.e., a predetermined k value), the flattest matching can be achieved. , during system operation, if the k value fluctuates from a preset predetermined condition (that is, a predetermined k value), the external Q of only one of the power transmitting side resonator and the power receiving side resonator is adjusted. High transmission efficiency can be ensured only by performing (that is, center point matching).

In other words, the external Q of the power transmitting side resonator and the external Q of the power receiving side resonator are not adjusted in advance so as to satisfy the condition of the flattest matching, and when the system is operated, the k value is set to a predetermined condition (i.e., a given k value), even if only the external Q of the receiving resonator is adjusted to satisfy the center point matching (Q _eT *Q _eR =1/k ² ), High transmission efficiency cannot be ensured.

This is a condition in which the external Q of the power transmitting side resonator and the external Q of the power receiving side resonator are not adjusted in advance so as to satisfy the flattest matching condition, and the k value is set in advance under a predetermined condition ( That is, if only the external Q of the receiving resonator is adjusted to satisfy the center-point match when the external Q of the receiving resonator varies from a predetermined k value, the external Q of the receiving resonator is This is because the value is extremely biased. That is, when the external Q of the power transmitting resonator or the power receiving resonator becomes a biased value, it becomes susceptible to the influence of the internal resistance of the coils 1 and 2, the reflected power increases, and the transmission efficiency decreases. is thought to be working.

In this regard, the external Q of each of the power transmission side resonator and the power reception side resonator should be adjusted in advance so that the transmission characteristics of power transmission become the flattest characteristics, assuming the k value during system operation. , even if the center point matching is satisfied by adjusting the external Q of one of the resonators according to the change of the k value during actual system operation, the conditions are close to the flattest matching, so It becomes possible to maintain a sufficiently high transmission efficiency.

From this point of view, the conditions related to the k value for the flattest matching (that is, Q _eT =1/k and Q _eR =1/k) are set so as not to be too far from the k value assumed in actual use. preferably For this purpose, it is important that the external Q of the resonator on the fixed side can also be set to an optimum value, and a configuration in which the external Q can be arbitrarily set on both power transmitting and receiving sides is useful. In this respect, both the power transmission side resonator and the power reception side resonator are configured by the capacitance division series resonance circuit or the capacitance division parallel resonance circuit shown in FIGS. A variable capacitor as shown in FIGS. 7A and 7B is preferred.

<Specific adjustment method>
Here, with reference to FIGS. 14 to 16, a specific example of a method of adjusting the external Q of the resonator performed before power transmission in the wireless power transmission system according to the present embodiment will be described.

As described above, in the wireless power transmission system according to the present embodiment, the external Q of the power transmission side resonator and the external Q of the power reception side resonator are each adjusted in advance under predetermined conditions so as to achieve maximum flatness matching. It is important to

In this preliminary preparation stage, first, the fluctuation range of the positional relationship (that is, k value) between the power transmitting resonator and the power receiving resonator during actual use of the wireless power transmission system (that is, the power transmitting resonator and the power receiving resonator) positional deviation tolerance). When the k value in actual use is lower than the k value that satisfies the flattest matching condition, the passband width tends to become narrower and the peak transmission efficiency tends to decrease significantly. Therefore, in the preliminary preparation stage (that is, when satisfying the flattest matching condition), Q _eT =1/k and Q _eR It is preferable to set the extrinsic Q = 1/k. By doing so, in actual use, it is possible to maintain the transmission efficiency at a high value when the center points are matched in a larger range of k values.

Next, the power transmission side capacitor 3 and the power reception side capacitor 4 are divided according to the k value (that is, the external Q) determined according to the tolerance of positional deviation between the power transmission side resonator and the power reception side resonator. By setting the capacitance (a value or b value) and adjusting the electrode positions using variable capacitors as shown in FIGS. .

Whether or not there is maximum flatness matching (Q _eT =1/k and Q _eR =1/k) can be determined, for example, by phase change of reflectance using a network analyzer (see FIG. 15). can.

14 and 15 are diagrams showing measurement results of reflectance and transmittance obtained when power is transmitted from a power transmitting resonator to a power receiving resonator in a wireless power transmission system. 14 shows the frequency dependence of the absolute values of reflectance and transmittance obtained by measurement, and FIG. 15 shows the absolute values of reflectance and phase change obtained by measurement on a Smith chart. there is Such measurement can be performed using, for example, a network analyzer.

14 and 15, the power transmitting side resonator and the power receiving side resonator are adjusted to have the flattest matching when k=0.04, and the k value is 0.02, 0.04, 0. 05 and 0.06, and reflectance and transmittance are measured when the center point is matched by changing the external Q in the resonator on the power receiving side according to the k value.

Each graph in FIG. 14 represents reflection characteristics and transmission characteristics of the following aspects.
Graph 41: When the coupling coefficient k is 0.02, the reflection characteristics seen from the transmitting side when the center point is matched by adjusting the external Q of the resonator on the receiving side Graph 42: When the coupling coefficient k is 0.03 , Reflection characteristic graph 43 seen from the power transmission side when the center point is matched by adjusting the external Q of the resonator on the power receiving side: When the coupling coefficient k is 0.04, the center point is Graph 44 of reflection characteristics seen from the power transmission side in the case of matching: Graph 45 of reflection characteristics seen from the power transmission side when center point matching is achieved by adjusting the external Q of the resonator on the power reception side when the coupling coefficient k is 0.05 : Reflection characteristic graph 51 seen from the power transmission side when the center point is matched by adjusting the external Q of the resonator on the power receiving side when the coupling coefficient k is 0.06: When the coupling coefficient k is 0.02, the power receiving Graph 52 of transmission characteristics viewed from the power transmission side when the center point is matched by adjusting the external Q of the resonator on the power receiving side: When the coupling coefficient k is 0.03, the center point is matched by adjusting the external Q of the resonator on the power receiving side Graph 53 of the transmission characteristics seen from the power transmission side in the case: When the coupling coefficient k is 0.04, the transmission characteristics graph 54 seen from the power transmission side in the case of center point matching by adjusting the external Q of the resonator on the power reception side: Coupling Graph 55 of transmission characteristics seen from the power transmission side when center point matching is achieved by adjusting the external Q of the resonator on the power receiving side when the coefficient k is 0.05: Resonance on the power receiving side when the coupling coefficient k is 0.06 Transmittance characteristics seen from the transmission side when the center point is matched by adjusting the external Q of the device

It is difficult to determine what condition satisfies the flattest matching condition only by looking at the absolute values of reflectance and transmittance shown in FIG. In this respect, as shown in FIG. 15, it is possible to easily determine whether or not the maximum flatness matching condition is satisfied by paying attention to the phase change in addition to the absolute value of the reflectance.

Each graph in FIG. 15 represents phase change of reflectance in the following manner.
Graph 61: Phase change of reflectance seen from the transmitting side when the center point is matched by adjusting the external Q of the resonator on the receiving side when the coupling coefficient k is 0.02 Graph 62: The coupling coefficient k is 0.03 Graph 63 of reflectance phase change seen from the power transmission side when the center point is matched by adjusting the external Q of the power receiving side resonator: When the coupling coefficient k is 0.04, the external Q of the power receiving side resonator is Phase change graph 64 of the reflectance seen from the transmitting side when the center point is matched by adjusting the Q: when the coupling coefficient k is 0.05 and the center point is matched by adjusting the external Q of the resonator on the receiving side Phase change graph 65 of reflectance seen from the power transmission side: Phase change of reflectance seen from the power transmission side when center point matching is achieved by adjusting the external Q of the resonator on the power receiving side when the coupling coefficient k is 0.06

FIG. 15 shows measurement results of phase change of reflectance using a Smith chart. What is shown here is the signal of _S11 as seen from the transmission side.

As shown in FIG. 15, when the k value is less than 0.04, it becomes a loop and its vertex is at the origin. On the other hand, at k=0.04, which satisfies the flattest matching condition, the loop shrinks to a point and has a sharp angular peak just at the origin. Also, when the k value is greater than 0.04, the sharp peak becomes gentle at the position of the origin, resulting in a gentle curve.

S ₂₂ seen from the power receiving side changes in the opposite direction, but it is the same that a sharp peak appears at the origin at the flattest alignment. By forming a small loop near the origin, the condition of the reflectance is close to zero over a wide frequency range, and as a result, the transmittance peak range is widened. As shown in graph 63, the flattest matching condition is that the S parameter of the reflected signal has a sharp peak at the origin on the Smith chart.

Incidentally, the S parameter can be easily measured with a commercially available network analyzer.

In this manner, the distribution of capacitance values of the power transmission side capacitor 3 and the power reception side capacitor 4 is adjusted so as to achieve the flattest matching. Thereafter, when center point matching is performed during actual system operation, the capacitance value distribution of one of the power transmission side capacitor 3 and the power receiving side capacitor 4 is fixed while the other is adjusted.

FIG. 16 is a flow chart showing an operation example during system operation (that is, during actual use) of the wireless power transmission system according to this embodiment. Here, only an operation example of the wireless power transmission system using the series resonator of FIG. 4A is shown.

The flowchart shown in FIG. 16 is for example a control unit (here, a control unit 10B on the side of the power receiving device UB) provided in a target for adjusting the external Q during center point matching, of the power transmitting device UA or the power receiving device UB. is a process executed according to a computer program. In the processing of this flowchart, for example, a relative position detection sensor (not shown) provided in the power transmitting device UA or the power receiving device UB detects that the relative position of the power receiving device UB with respect to the power transmitting device UA is within a predetermined range. It will be executed as a trigger.

In step S1, the control unit 10B provided in the power receiving device UB communicates with the control unit 10A provided in the power transmitting device UA to execute a power transmission test from the power transmitting device UA to the power receiving device UB. Let

Note that this step S1 specifies the a value (that is, the capacitance values C a and _{C b} of the dividing capacitors 3a and 3b and the capacitance values C _a and C _b of the dividing capacitors 4a and 4b) that can ensure high transmission efficiency. This is an experimental power transmission for Therefore, the power value designated in this step S1 is set to a power value lower than the power value in the case of power transmission according to this embodiment in step S5.

In step S2, the control unit 10B measures the transmission efficiency in the power transmission test. At this time, the control unit 10B measures the transmission efficiency, for example, based on the transmitted power specified by the power transmitting device UA and the received power measured by the power receiving device UB.

At step S3, the control unit 10B determines whether the transmission efficiency measured at step S2 is greater than a threshold value (eg, 95%). If the transmission efficiency measured in step S2 is greater than the threshold (step S3: YES), the process proceeds to step S5, and if less than the threshold (step S3: NO), the process proceeds to step S4.

In step S4, the control section 10B changes the a value of the power receiving side capacitor 4 (that is, the capacitance values Ca _{and Cb} of the dividing capacitors 4a and 4b) by a predetermined amount. In this step S4, the capacitance of the capacitance value _Ca of the dividing capacitor 4a and the capacitance value _Cb of the dividing capacitor 4b is kept constant while the parallel combined capacitance of the capacitance value _Ca of the dividing capacitor 4a and the capacitance value _Cb of the dividing capacitor 4b is kept constant. Change the ratio. In this step S4, regarding the power transmission side capacitor 3, the divided capacitances (that is, the capacitance values C _a and C _b of the divided capacitors 3a and 3b) remain fixed without being changed. Then, returning to step S1, power transmission and transmission efficiency are measured again.

In the wireless power transmission system according to the present embodiment, by repeatedly executing the processing of steps S1 to S4, the a value of the power receiving side capacitor 4 (that is, the capacitance of the dividing capacitors 4a and 4b) that can ensure high transmission efficiency Identify the values C _a , C _b ). That is, this allows the capacitance values C a , C _b of the split capacitors 4 a , 4 b to satisfy the center point matching condition, corresponding to the state of _{electromagnetic} coupling between the two resonators (that is, the coupling coefficient k). can be set.

In step S<b>5 , the control unit 10</b>B causes the power transmission device UA to perform the actual power transmission to the power reception device UB.

In step S6, the control unit 10B waits until the condition for terminating power transmission is satisfied (S6: NO). Then, if the termination condition is satisfied (S6: YES), the processing of the flowchart of FIG. 16 is terminated. For example, when the battery, which is the load 6 of the power receiving device UB, reaches a fully charged state, the control unit 10B determines that the end condition is satisfied.

The wireless power transmission system according to the present embodiment can perform power transmission from the power transmission device UA to the power reception device UB with high transmission efficiency through the series of processes described above.

[effect]
As described above, in the wireless power transmission system according to this embodiment,
Adjusting the external Q of each of the power transmission side resonator and the power reception side resonator in advance under predetermined conditions so that the transmission characteristics of power transmission are the flattest characteristics,
When the positional relationship between the power transmission side resonator and the power reception side resonator is changed from the predetermined condition during system operation, the center points are matched at the resonance frequencies of the power transmission side resonator and the power reception side resonator. Secondly, the transmission characteristics of power transmission are adjusted by adjusting the external Q of one of the power transmission side resonator and the power receiving side resonator while fixing the other external Q.

As a result, even if the positional relationship between the power transmitting side resonator and the power receiving side resonator changes from the assumed positional relationship during actual system operation, the external Q of one of the resonators is Just by adjusting, it is possible to suppress reflection during power transmission and maintain high transmission efficiency. As a result, for example, for a transmission power source with fixed characteristics such as power supply to an electric vehicle or power supply to a portable device, the external Q (for example, the a value or b value), high efficiency can be realized, and highly efficient contactless power supply can be achieved by a simple method.

In particular, in the wireless power transmission system according to the present embodiment, the divided capacitances of the capacitors constituting the power transmitting side resonator and the power receiving side resonator (in the case of the series resonator, the capacitance of the first divided capacitor shown in FIG. 7A It corresponds to the value _C1 and the capacitance value _C2 of the second dividing capacitor, and in the case of a parallel resonator, corresponds to the capacitance value _C3 of the third dividing capacitor and the capacitance value _C4 of the fourth dividing capacitor shown in FIG. ), it is possible to appropriately adjust the external Q value of each of the power transmitting side resonator and the power receiving side resonator. This makes it possible to easily perform adjustment for maximum flatness alignment for preparation and/or adjustment for center point alignment during system operation.

In particular, in the wireless power transmission system according to the present embodiment, wireless power transmission can be performed with high transmission efficiency at the resonance frequencies of the power transmission side resonator and the power reception side resonator (that is, the natural resonance frequency of each resonator alone). It is possible. Therefore, even when the power transmission device UA uses a power source with a fixed impedance, or the power receiving device UB uses a load with a fixed impedance (for example, a rectifier circuit), This is preferable in that it does not cause a drop in transmission efficiency due to the positional deviation of the . This makes it possible to use, for example, a widely used high-frequency power source or coaxial cable with a characteristic impedance of 50Ω.

In addition, in the wireless power transmission system according to the present embodiment, the divided capacitance of the capacitor of the resonator (in the case of a series resonator, the capacitance value _C1 of the first divided capacitor and the capacitance value C of the second divided capacitor shown in FIG. 7A _2. In the case of a parallel resonator, it corresponds to the capacitance value _C3 of the third divided capacitor and the capacitance value _C4 of the fourth divided capacitor shown in FIG. and the power receiving device UB have different shapes and inductances of the resonator coils, and even if the impedance is different between the power supply of the power transmitting device UA and the load of the power receiving device UB, center point matching (and maximum It is useful in that it is possible to achieve flat alignment.

(Second embodiment)
As described above, the adjustment method of the coupled resonance type wireless power transmission system according to the present disclosure is performed by adjusting the external Q of the power transmitting side resonator and the external Q of the power receiving side resonator in advance under predetermined conditions (i.e., predetermined k value) is adjusted to achieve the flattest matching. It is characterized in that high transmission efficiency is ensured only by adjusting the external Q of only one of the side resonators (that is, center point matching).

In the first embodiment, a method for adjusting transmission characteristics in a coupled resonance type wireless power transmission system using an LC resonator has been described. A similar adjustment method can also be used in a coupled resonance type wireless power transmission system using an open ring resonator.

The inventors of the present application disclosed in Patent Document 1, which is a prior application, that in a wireless power transmission system using open-ring resonators, when the positional relationship between the resonators fluctuates, the power-transmitting-side resonator and the power-receiving-side resonator By adjusting the port angles of the open ring resonators that make up each of them, the external Q value of each of the power transmission side resonator and the power reception side resonator can be adjusted, and the frequency characteristics during power transmission can be changed from a bimodal type to a maximum flat type. It was shown that it is possible to change to

The wireless power transmission system according to this embodiment uses the wireless power transmission system disclosed in Patent Document 1, for example.

17, 18, and 19 show the schematic configuration of the wireless power transmission system according to this embodiment. Here, only the outline of the configuration for adjusting the port angle of the open ring resonator will be described (for details, see Patent Document 1).

FIG. 17 is a side view of the overall configuration of the wireless power transmission system, FIG. 18 is a plan view of the open ring resonator 111, and FIG. 211 is a plan view of a state in which 211 are opposed to each other.

In FIG. 17, UA is a power transmitting device of the wireless power transmission system, and UB is a power receiving device. The power transmission device UA includes, for example, an open ring resonator 111, an input/output line 112 (for example, a coaxial line) serving as a feeder line for the open ring resonator 111, a ground plate 113, a control section 114, a drive section 115, a power source, and the like. A circuit board 120 is provided. In addition, the power receiving device UB includes, for example, an open ring resonator 211, an input/output line 212 (for example, a coaxial line) serving as a feeder line for the open ring resonator 211, a ground plate 213, a control unit 214, a driving unit 215, and an electric power supply. It has a circuit board 220 on which a load and the like are mounted.

Here, the same structure is adopted as the support structure of the open-ring resonator in the power transmission device UA and the power reception device UB.

In the wireless power transmission system according to the present embodiment, for example, the open ring resonators 111 and 211 are configured separately from the circuit boards 120 and 220 (for example, configured by metal plates), and the circuit boards 120 and 220 The input/output lines 112 and 212 are extended toward the open ring resonators 111 and 211, and input/output lines are connected to the open ring resonators 111 and 211 via electrode portions 112a and 212a at the tips of the open ring resonators 111 and 211. It is configured to be electrically connected to the lines 112 and 212 . The open ring resonators 111 and 211 are rotatably supported at the positions of the central axes of the resonators 111 and 211 by supporting rods 111S and 211S extending in the vertical direction, and the drive units 115 and 215 are driven. By rotating the support rods 111S and 211S, the positional relationship between the open ring resonators 111 and 211 and the electrode portions 112a and 212a is changed. The port angles of the output lines 112 and 212 are adjustable.

As described above, the port angle represents the position at which the open ring resonator and the input/output line are electrically connected in the circumferential direction of the open ring resonator. In the following, for convenience of explanation, the value of the port angle is defined as the line connecting the center point _A0 of the open ring resonator 111 and the position _A2 where the electrode portion 112a is electrically connected to the open ring resonator 111 and the open ring resonance The angle between the center point _A0 of the resonator 111 and the line connecting the center position _A1 in the longitudinal direction of the open ring resonator 111 (that is, the open ring resonator 111 between the positions _A2 and _A1 angle in the circumferential direction) is defined by ∠A ₁ A ₀ A ₂ (see θ in FIG. 18).

In general, the external Q of an open ring resonator is defined by Z _S as the impedance of the input/output line connected to the open ring resonator, Z R as the characteristic impedance of the ring conductor of the open ring resonator, and Z _R as the port angle of the open ring resonator. It is known that θ can be calculated by the following equation (27) (see Non-Patent Document 7).

The wireless power transmission system according to the present embodiment utilizes this theory, and by making the port angle of the open ring resonator adjustable, the impedance of the input/output line connected to the open ring resonator, the load impedance , and the external Q of an open ring resonator can be appropriately adjusted without changing the power supply impedance or the like. That is, in the wireless power transmission system according to the present embodiment, by adjusting the port angles of the open ring resonators that constitute the power transmission side resonator and the power reception side resonator, the power transmission side resonator and the power reception side resonator respectively appropriately adjust the external Q value of .

In the wireless power transmission system according to this embodiment, the same technique as the technique shown in the first embodiment can be used as a specific adjustment method itself for adjusting the transmission characteristics.

First, in the preliminary preparation stage, under predetermined conditions, the transmission characteristics when power is transmitted from the power transmission resonator to the power reception resonator are the flattest characteristics (that is, equation (6a) and equation (6b) is satisfied), the external Q of the power transmitting side resonator and the external Q of the power receiving side resonator are each adjusted.

For example, as in the first embodiment, this preparatory adjustment process is carried out by checking the phase change of the reflectance using a network analyzer while checking the external Q of the power transmission side resonator (that is, the open ring resonance of the power transmission side). by adjusting the port angle of the receiver) and the external Q of the power receiving side resonator (that is, the port angle of the open ring resonator on the power receiving side). At this time, the conditions of the equations (6a) and (6b) (Q _eT =1/k and Q _eR =1/k) typically make the transmission and reception rings of the same structure. , the port angle of the open ring resonator on the power transmitting side and the port angle of the open ring resonator on the power receiving side are the same angle, as can be seen from equation (27).

During system operation, if the positional relationship (that is, relative position and/or angle) between the power transmitting resonator and the power receiving resonator changes, and as a result, the coupling coefficient k changes from a predetermined condition, power transmission By adjusting only the external Q (for example, the port angle of the open ring resonator on the power receiving side) of one of the resonator on the power receiving side and the resonator on the power receiving side, the center point matching condition (Q _eT * Q _eR =1/k ² ).

This adjustment process during system operation can be performed, for example, by a process similar to that shown in the flowchart of FIG. Specifically, in the wireless power transmission system according to the present embodiment, for example, in the flowchart of FIG. Instead of performing the process, perform the process of "changing the port angle of the power receiving side open ring resonator while fixing the port angle of the power transmitting side open ring resonator", and search for the port angle at which the transmission efficiency is equal to or higher than the threshold. , thereby center point matching.

Here, in order to demonstrate that high peak transmission efficiency can be ensured by the above adjustment method also in the wireless power transmission system according to this embodiment, a verification experiment was conducted using an electromagnetic field simulator.

20 to 23 are diagrams showing the results of verification experiments using an electromagnetic field simulator.

In this simulation, the distance between the open ring resonator on the power transmitting side and the open ring resonator on the power receiving side (hereinafter referred to as the "inter-resonator distance") changes, and the coupling coefficient k achieves the flattest matching. We verified whether high transmission efficiency can be maintained when the center point matching condition is satisfied by changing only the port angle of the open ring resonator on the receiving side when the conditions are changed.

In this simulation, the same support structure of the open ring resonator was adopted for the power transmitting device UA and the power receiving device UB, and the design parameters of both were also set to the same values. Specifically, the diameter of the ring center line of the open ring resonator is 27 mm, the ring width of the open ring resonator is 20 mm, and the distance between the open ring resonator and the back ground substrate is set to 920 MHz. It was set to 8 mm, and as a result, the characteristic impedance of the ring conductor portion of the open ring resonator was about 75Ω. Also, the characteristic impedance _ZS of the input/output line for the open ring resonator was set to 50Ω.

First, the port angle of the open ring resonator on the power transmitting side and the port angle of the open ring resonator on the power receiving side were adjusted so as to achieve the flattest matching under the condition that the distance between the resonators was 90 mm. At that time, the port angle θ that satisfied the flattest alignment was 8°.

Next, the inter-resonator distance (that is, the k value) was variously changed from the condition satisfying the flattest matching, and the port angle of the open ring resonator on the power receiving side was adjusted to achieve center point matching. Here, when the distance between the resonators is 75 mm, 62 mm, 50 mm, and 42 mm, the port angle of the open ring resonator on the power receiving side is adjusted while the port angle of the open ring resonator on the power transmitting side is fixed at 8°. center-point matched.

FIG. 20 is a diagram showing the dependence of the optimum port angle on the inter-resonator distance in an open ring resonator.

Each graph in FIG. 20 represents the following port angles.
Graph 71: Port angle of transmitting resonator (fixed at 8°)
Graph 72: Port angle of receiving side resonator that maximizes efficiency Graph 73: Port angle that achieves flattest matching at each distance between resonators

As can be seen from FIG. 20, the transmission efficiency is peaked and became midpoint matched (see graph 72). When the distance between the resonators is 75 mm, 62 mm, 50 mm, and 42 mm, both the port angle of the open ring resonator on the power transmitting side and the port angle of the open ring resonator on the power receiving side are adjusted to obtain the flattest matching. The port angles when satisfied were 11°, 15°, 21°, 26° (see graph 72). From these port angles, the external Q of each resonator can be determined using equation (27) above.

FIG. 21 is a diagram showing the external Q at the optimum port angle in the open ring resonator. Note that the external Q shown in FIG. 21 is obtained based on the port angle calculated in FIG.

Each graph in FIG. 21 represents the extrinsic Q below.
Graph 74: External Q of transmitting resonator with port angle fixed at 8 degrees
Graph 75: External Q of receiving resonator for maximum efficiency
Graph 76: External Q for flattest matching at each cavity distance
Graph 77: Geometric Mean of Graphs 74 and 75

For comparison, FIG. 21 shows the external Q (see graph 75) of the open ring resonator on the receiving side with center point matched conditions (i.e., at port angles of 16°, 32°, 64°, 128°). ) and the external Q of the open ring resonator on the sending side fixed at 8° (see graph 74) is also shown (see graph 77). This geometric mean approximately agrees with the external Q (see graph 76) of the open ring resonator on the receiving side in the condition of satisfying the flattest matching. That is, the following formula (28) holds, and this relationship is the same as formula (10).

FIG. 22 is a diagram showing the passband width at the optimum port angle of the open ring resonator.

Each graph in FIG. 22 represents the following passband width.
Graph 80: Passband with 80% efficiency for center point matching (assuming perfect conductors)
Graph 81: 80% Efficient Pass Bandwidth for Flattest Match (Assuming Perfect Conductors)

Graph 80 of FIG. 22 shows the passband widths for the center point matched conditions (ie, at port angles of 16°, 32°, 64° and 128°) calculated in FIG. A graph 81 in FIG. 22 shows the passband width at the time of flattest matching. Both the graphs 80 and 81 are for a bandwidth with a transmission efficiency of 80%. Both the graphs 80 and 81 are calculated on the assumption that the open ring resonator is a perfect conductor.

From FIG. 22, the passband width at the flattest matching increases rapidly as the inter-cavity distance decreases (see graph 81), while the passband width at the center point matching decreases as the inter-cavity distance decreases. It can be seen that even in the state where the inter-cavity distance is large, the bandwidth remains as narrow as in the case where the inter-cavity distance is large (see graph 80). It should be noted that power transmission is typically performed at a single frequency, so the narrow passband width during center point matching does not pose a problem per se.

FIG. 23 is a diagram showing peak transmission efficiency at the optimum port angle of the open ring resonator.

Each graph in FIG. 23 represents the following peak transmission efficiency.
Graph 82: Peak transmission efficiency for center point matching (assuming aluminum resistivity)
Graph 83: Peak Transmission Efficiency for Flattest Match (Assumes Aluminum Resistivity)
Graph 84: Peak transmission efficiency assuming a perfect conductor with center point matching

Graphs 82 and 84 of FIG. 23 show peak transmission efficiencies in the center point matched conditions calculated in FIG. , the peak transmission efficiency at the flattest match is also shown.

As can be seen from FIG. 23, assuming the open ring resonator to be a perfect conductor, the peak transmission efficiency is constant at nearly 100% for both the flattest and center point matches (see graph 84). On the other hand, assuming the open-ring resonator to be a conductor with the conductivity of ordinary aluminum, the peak transmission efficiency increases as the inter-resonator distance increases, similar to that for the LC resonator, for the flattest match (graph 82) and for center point alignment (see graph 83) both fall.

From this, it is inferred that the peak transmission efficiency of the open ring resonator is not simply determined uniquely by the distance between the resonators, but is greatly affected by the passband width, although it is not as clear as that of the LC resonator. This is because, as discussed in the above embodiment, when the external Q of the power transmitting side resonator or the power receiving side resonator has a biased value, it becomes susceptible to the internal resistance of the open ring resonator and the reflected power increases. , the phenomenon that the transmission efficiency is lowered is considered to be working. Therefore, as discussed in the above embodiment, in order to obtain high peak transmission efficiency when center point matching is performed, the external Q when center point matching is performed must be equal to the external Q when maximally flat matching is performed. It can be said that it is preferable to set the external Q close to .

As described above, as in the wireless power transmission system according to the present embodiment, even in a mode in which each of the power transmission side resonator and the power reception side resonator is configured as an open ring resonator, the external Q and the external Q of the resonator on the power receiving side are adjusted under predetermined conditions so as to achieve maximum flatness matching. A method of adjusting the external Q of only one of the resonator and the receiver resonator is found to be useful. In other words, in the wireless power transmission system according to this embodiment as well, it is possible to achieve optimum transmission corresponding to changes in the k value by the two-stage adjustment method described above.

(Other embodiments)
The present invention is not limited to the above-described embodiment, and various modifications are conceivable.

For example, in the above-described embodiment, when center point matching is performed in the wireless power transmission system, the control unit 10B provided on the power receiving device UB side controls the external Q of the power receiving side resonator. However, the method of adjusting the external Q of the power receiving side resonator does not necessarily have to be automatic control by the control unit 10B, and may be performed manually by an operator.

Further, the target for adjusting the external Q when performing center point matching may be the power transmission side resonator instead of the power reception side resonator.

Further, in the above-described embodiment, a set of a power transmission side resonator and a power reception side resonator to be used is prepared in advance, and power is actually transmitted and received between them, so that the transmission characteristics during power transmission are as follows. A mode of adjusting the external Q of each of the power transmitting side resonator and the power receiving side resonator is shown so as to obtain the flattest characteristics. Such an aspect is useful when the power transmitting side resonator and/or the power receiving side resonator includes individual differences for each product.

However, if the individual difference between the power transmitting side resonator and/or the power receiving side resonator is so small that it can be ignored, the standard value of the external Q for each of the power transmitting side resonator and the power receiving side resonator is set, and the standard value is set. It is also possible to simply design each of the power transmitting side resonator and the power receiving side resonator so as to satisfy the requirements. As a result, the external Q of each of the power-transmitting-side resonator and the power-receiving-side resonator can be adjusted in advance under a predetermined condition without actually adjusting the external Q for satisfying the flattest matching as the initial setting. Then, the transmission characteristic of power transmission is adjusted to be the flattest characteristic. By doing so, by combining the power transmitting side resonator and the power receiving side resonator that are product-designed according to such standards, it is possible to omit the adjustment work of the external Q for satisfying the flattest matching as the initial setting. can. However, in practice, there are many cases where the power transmission side resonator and the power reception side resonator include individual differences. It is preferable to adjust the external Q of each of the power-transmitting-side resonator and the power-receiving-side resonator so that the transmission characteristic becomes the flattest characteristic.

Further, in the above embodiment, the power transmitting device UA and the power receiving device UB each have the control units 10A and 10B. However, the adjustment of the capacitance values of the divided capacitors of the capacitors 3 and 4 forming the resonator may be performed manually by an operator. In that case, the power transmitting device UA and/or the power receiving device UB may be configured without the controller (10A, 10B).

Further, in the above-described embodiments, the power transmitting side resonator and the power receiving side resonator are configured in the divided capacitance series type or the divided capacitance parallel type. However, as the power transmitting side resonator and/or the power receiving side resonator, a single-capacity series resonant circuit or a single-capacity parallel resonant circuit may be used, although this is not a preferable mode from a practical point of view. In this case, the external Q of the resonator is adjusted by adjusting the impedance of the power source 5 and the impedance of the load 6 .

Although specific examples of the present invention have been described in detail above, these are merely examples and do not limit the scope of the claims. The technology described in the claims includes various modifications and changes of the specific examples illustrated above.

According to the coupled resonance type wireless power transmission system according to the present invention, even when the positional relationship between the two resonators changes and the coupling coefficient k changes, only by adjusting the external Q of one resonator , it is possible to suppress reflection during power transmission and maintain high transmission efficiency.

UA power transmission device UB power reception device 1 coil (power transmission side coil)
2 Coil (receiving side coil)
3 Capacitor (capacitor on the power transmission side)
3a, 3b, 3c, 3d dividing capacitor 4 capacitor (power receiving side capacitor)
4a, 4b, 4c, 4d divided capacitor 5 power source 6 load 10A control unit 10B control unit E1 first fixed electrode E2 second fixed electrode E3 movable electrode E4 first fixed electrode E5 second fixed electrode E6 movable electrode 111, 221 open ring Resonators 111S, 211S Support rods 112, 212 Input/output lines 112a, 212a Electrode parts 113, 213 Ground plates 114, 214 Control parts 115, 215 Driving parts 120, 220 Circuit board

Claims (7)

A coupled resonance type wireless power transmission system that uses electromagnetic resonance to perform contactless power transmission from a first resonator to a second resonator,
The external Q of each of the first and second resonators is adjusted in advance under predetermined conditions so that the transmission characteristics of the power transmission are the flattest characteristics,
During system operation, if the positional relationship between the first resonator and the second resonator has changed from the predetermined condition, the controller causes the resonance frequencies of the first and second resonators to performing the power transfer after adjusting the external Q of one of the first and second resonators while fixing the external Q of the other so as to match the center point;
Coupling resonance type wireless power transmission system. The first and second resonators, respectively,
Consists of a capacitor and a coil,
The capacitor is separated into a first divided capacitor and a second divided capacitor that are connected in parallel to the coil, and an external impedance is present in either the first divided capacitor or the second divided capacitor. A structure that is connected in series and is capable of adjusting the capacitance ratio while keeping the parallel combined capacitance of the first divided capacitor and the second divided capacitor constant, or
The capacitor is separated into a third divided capacitor and a fourth divided capacitor that are connected in series with the coil, and either the third divided capacitor or the fourth divided capacitor has an external impedance. It has a structure that is connected in parallel and that allows the capacitance ratio to be adjusted while the series combined capacitance of the third divided capacitor and the fourth divided capacitor is kept constant.
The coupled resonance type wireless power transmission system according to claim 1. When the system is operated, the control unit performs the power transmission on a trial basis between the first and second resonators while changing the capacitance ratio of the other capacitor of the first and second resonators. determining the capacitance ratio of the other capacitor when the power transmission is actually performed, based on the change in transmission efficiency obtained at that time;
The coupled resonance type wireless power transmission system according to claim 2. The first and second resonators, respectively,
Consists of an open ring resonator,
The open ring resonator has a port angle of an input/output line electrically connected thereto (however, the port angle is the angle between the open ring resonator and the input/output line in the circumferential direction of the open ring resonator) (representing the connecting position) is supported in an adjustable state,
The coupled resonance type wireless power transmission system according to claim 1. A method for adjusting transmission characteristics in a coupled resonance type wireless power transmission system that performs contactless power transmission from a first resonator to a second resonator using electromagnetic resonance, comprising:
adjusting the external Q of each of the first and second resonators in advance under predetermined conditions so that the transmission characteristics of the power transmission are the flattest characteristics;
When the positional relationship between the first resonator and the second resonator is changed from the predetermined condition during system operation, center point matching is performed at the resonance frequencies of the first and second resonators. Adjusting the external Q of one of the first and second resonators while fixing the external Q of the other,
adjustment method. The first and second resonators, respectively,
Consists of a capacitor and a coil,
The capacitor is separated into a first divided capacitor and a second divided capacitor that are connected in parallel to the coil, and an external impedance is present in either the first divided capacitor or the second divided capacitor. A structure that is connected in series and is capable of adjusting the capacitance ratio while keeping the parallel combined capacitance of the first divided capacitor and the second divided capacitor constant, or
The capacitor is separated into a third divided capacitor and a fourth divided capacitor that are connected in series with the coil, and either the third divided capacitor or the fourth divided capacitor has an external impedance. It has a structure that is connected in parallel and that allows the capacitance ratio to be adjusted while the series combined capacitance of the third divided capacitor and the fourth divided capacitor is kept constant.
The adjustment method according to claim 5. The first and second resonators, respectively,
Consists of an open ring resonator,
The open ring resonator has a port angle of an input/output line electrically connected thereto (however, the port angle is the angle between the open ring resonator and the input/output line in the circumferential direction of the open ring resonator) (representing the connecting position) is supported in an adjustable state,
The adjustment method according to claim 5.

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