JP2023005243A - magnetic field sensor - Google Patents

Abstract

To provide a magnetic field sensor that has at least the same performance as a fundamental wave type orthogonal flux gate, can be downsized, and can measure a high-frequency magnetic field of several tens of kHz band.SOLUTION: A magnetic field sensor is equipped with a sensor head 14 formed by winding a detection coil 13 around a magnetic core 12, a current supply portion 11 in which superimposes an excitation AC current that is an AC current for excitation and a bias DC current that is a DC current for bias are superimposed to be supplied to a magnetic core 12, and a detection circuit 15 that is connected to the detection coil 13 and detects a high-frequency magnetic field inputted in the magnetic core 12 as a magnetic field to measured. The detection circuit 15 measures the magnetic field to be measured on the basis of a first item and a third item of (9) expression presenting an induced voltage v(t) induced in the detection coil 13.SELECTED DRAWING: Figure 1

Description

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a fundamental wave type orthogonal fluxgate magnetic field sensor, and more particularly to a magnetic field sensor that measures a high frequency magnetic field with a down-conversion function using frequency mixing.

For example, Patent Literatures 1 and 2 disclose techniques related to fundamental mode orthogonal fluxgates. The technique disclosed in Patent Document 1 is a sensor that has a magnetic wire and a detection coil wound around the magnetic wire, supplies an alternating excitation current to the magnetic wire, and uses a voltage induced in the detection coil as a detection output. The magnetic wire may be replaced with a cylindrical magnetic body, or the magnetic wire may be an amorphous magnetic wire with a non-magnetic distorted composition. , the magnetic wire may have a slightly negative magnetostriction.

The technology disclosed in Patent Document 2 includes a sensor head formed by winding a detection coil around a magnetic core, a current supply section that supplies an alternating current for excitation and a direct current for bias to the magnetic core in a superimposed manner, and a bias sensor. and a detection circuit connected to the detection coil for detecting the magnetic field measured by the sensor head with a feedback current, the first switch making the polarity of the DC current positive. Of the positive and negative bias current values, the offset value that occurs when the positive bias current is applied and the offset value that occurs when the negative bias current is applied when switching is equivalent. is supplied to the magnetic core, and a negative bias current value is supplied to the magnetic core when the first switch switches the polarity of the DC current to negative polarity.

In addition, Non-Patent Document 1 discloses that interference is caused between two magnetic cores when two (a plurality of) fundamental-wave orthogonal fluxgates are arranged close to each other. It has been described as a beat phenomenon between fundamental quadrature fluxgates.

JP-A-2002-277522 JP 2019-211450 A

Feng Han, Shoumu Harada, and Ichiro Sasada, “Beat Interferences in Fundamental Mode Orthogonal Fluxgates”, IEEE TRANSACTIONS ON MAGNETICS, VOL. 50, NO. 7, JULY 2014

It is easiest to use a pickup coil based on Faraday's law of electromagnetic induction to detect a magnetic field in the band of several tens of kHz. However, when a high resolution of the pT order is required, the effective cross-sectional area of the coil and the number of turns of the coil cannot be reduced, and there is a problem that miniaturization is difficult.

On the other hand, the fundamental wave type quadrature flux gate as shown in Patent Document 1 has a resolution of pT order and can be miniaturized, but generally detects a magnetic field in a frequency band of about DC to 1 kHz. As it is, it cannot detect magnetic fields in the tens of kHz band. In order to extend the detectable frequency band to several tens of kHz, in the configuration of the conventional fundamental wave type orthogonal fluxgate, it is necessary to increase the excitation frequency to 10 times or more the frequency of the magnetic field to be measured. Since it is necessary to increase the self-resonance frequency of the coil and widen the bandwidth of the circuit, it is essential to review the elements and redesign the circuit, and it is not easy to put it into practical use.

Also, the technique shown in Non-Patent Document 1 is a technique for arranging a plurality of general fundamental-wave type orthogonal fluxgates close to each other, and is not a technique for detecting a high-frequency magnetic field.

SUMMARY OF THE INVENTION The present invention has been made to solve the above-mentioned problems, and provides a magnetic field sensor that has at least the same performance as a conventional fundamental wave type orthogonal fluxgate, can be miniaturized, and can measure a high-frequency magnetic field. offer.

A magnetic field sensor according to the present invention includes a sensor head formed by winding a detection coil around a magnetic core, an exciting alternating current that is an alternating current for excitation, and a bias direct current that is a direct current for biasing. A power supply unit that supplies power to the magnetic core, and a detection circuit that is connected to the detection coil and detects a high-frequency magnetic field input to the magnetic core as a magnetic field to be measured, wherein the detection circuit is induced by the detection coil. induced voltage v(t)

（ただし、計測対象磁界の振幅をh、その角周波数をω_s、励磁交流電流の励磁角周波数をω_ｃ、磁気コアの実行断面積をS、検出コイルの巻き数をn、入力換算オフセット磁界をH_offとし、k₀,k₁及びk₂は磁気コアの透磁率の近似式の係数でH/mの次元を持つ係数とする）における第一項中の第一項及び第三項に基づいて、前記計測対象磁界を計測するものである。

(However, the amplitude of the magnetic field to be measured is h, its angular frequency is ω _s , the exciting angular frequency of the exciting alternating current is ω _c , the effective cross-sectional area of the magnetic core is S, the number of turns of the detection coil is n, and the input conversion offset magnetic field is is H _off , and k ₀ , k ₁ and k ₂ are the coefficients of the approximate expression of the permeability of the magnetic core and have the dimension of H / m). Based on this, the magnetic field to be measured is measured.

Thus, in the magnetic field sensor according to the present invention, the induced voltage in the first term and the third term in the first term (first parenthesis) of the above equation (9) is synchronously detected at the angular frequency _ωc . By extracting this frequency mixture on the low frequency side, a fundamental wave type quadrature fluxgate with a down-conversion function is realized, and a high frequency magnetic field that cannot be measured with a conventional fundamental wave type quadrature fluxgate is measured. It has the effect of being able to

For example, in the field of non-destructive inspection using magnetism, there are cases where it is desired to detect a high-frequency magnetic field with high precision and perform AD conversion with a resolution of 24 bits or higher. Delta-sigma AD converters are relatively inexpensive and easy to use even at 24-bit, but their conversion speed is about 100kS/s (100,000 samples per second), and the signal frequency must be low. Become. Since the magnetic field sensor according to the present invention has a down-conversion function, it can be measured by converting the frequency to a low frequency while preserving the amplitude and phase of the signal of the magnetic field to be measured, thereby solving the problems related to the above-mentioned AD conversion. becomes possible.

1 is a block diagram showing the configuration of a magnetic field sensor according to a first embodiment; FIG. 3A and 3B are diagrams showing a configuration of a power supply unit and clock waveforms in the magnetic field sensor according to the first embodiment; FIG. FIG. 4 is a diagram showing a switch circuit for obtaining a signal whose sign is inverted in synchronization with the H level and L level of the clock in the magnetic field sensor according to the first embodiment; FIG. 10 is a diagram showing a rotating magnetization model in a fundamental quadrature fluxgate; FIG. 10 is a diagram showing analysis results regarding magnetic permeability in a fundamental wave type orthogonal fluxgate; It is a block diagram showing the configuration of a magnetic field sensor according to a second embodiment.

(First embodiment of the present invention)
An outline of the magnetic field sensor according to the present embodiment will be described. The magnetic field sensor 1 according to this embodiment measures a high-frequency magnetic field in a narrow band with high sensitivity using a fundamental wave type orthogonal fluxgate. Assuming that the driving frequency of the fundamental wave type quadrature fluxgate is f _d , the magnetic field in the band of f _d ±Δf is detected. For this purpose, the relationship of frequency mixing expressed by the following equation is used.

In the above equation, if f ₁ = f _d and f ₂ = f _s (f _s : magnetic field frequency to be measured), the first term on the right side is converted to a low frequency, but |f _d -f _s |< By setting f _d to be Δf, it can be detected by a fundamental wave quadrature fluxgate with a detection band of 0 to Δf. In this case, the frequency is converted from f _s to f _s -f _d , but the phase information does not change due to the conversion from the relationship of the above equation, so it can be detected correctly from the first term on the right side. Also the amplitude of the magnetic field can be scaled correctly by calibration. Regarding the frequency, since the first term on the right side of the above equation gives the same result when f1>f2 and when f2>f1, ambiguity remains, but it can be known by the following method. When it is detected as f _o [Hz] in the 0 to Δf band, there are two possibilities of measurement target magnetic field frequency f _s =f _d ±f _o [Hz] from synchronous detection operation, but f _s = f _d + Whether f _s =f _d -f _o is f _s =f _d +f _o if f _d is slightly increased and _fo becomes smaller, and conversely if _{fo increases} It can be identified that f _s =f _{d -fo} . For example, when the magnetic field frequency to be applied is known as in non-destructive inspection, such an operation for obtaining the frequency is unnecessary, but when the magnetic field frequency to be applied is unknown, the frequency can be accurately calculated.

In the above description, the drive frequency of the sensor is assumed to be f _d to avoid ambiguity, but the drive frequency of the fundamental wave type orthogonal fluxgate is the excitation frequency of the sensor itself, f _d =f _c (ω _d = ω _c ).

As described above, the magnetic field sensor 1 according to the present embodiment uses the magnetic permeability increased by the small-angle rotation operation of the magnetization by DC bias excitation when measuring a high-frequency magnetic field using a fundamental wave type orthogonal fluxgate. is used to cause frequency mixing between the frequency of the magnetic field to be detected and the excitation frequency, and the modulated wave appearing on the low frequency side is detected. For this purpose, the difference between the excitation frequency and the frequency of the magnetic field to be detected is set to about several kHz or less so that the modulated wave on the low frequency side falls within the detection band of the fundamental wave type orthogonal fluxgate. The signal to be measured can be output as a down-converted low-frequency signal, the amplitude and phase of which can be easily correlated with the amplitude and phase of the original magnetic field to be measured. Moreover, when the frequency of the high-frequency magnetic field is unknown, the frequency can be known by changing the drive frequency _fd .

A configuration of a magnetic field sensor 1 according to this embodiment will be described with reference to FIGS. 1 to 3. FIG. FIG. 1 is a block diagram showing the configuration of the magnetic field sensor according to the present embodiment, FIG. 2 is a diagram showing the configuration of the power supply unit 11 and clock waveforms, and FIG. FIG. 10 is a diagram showing a switch circuit for obtaining a signal with an inverted sign; In FIG. 1, the magnetic field sensor 1 includes a current supply unit 11 that supplies an exciting current, a magnetic core 12 that is energized by superimposing an AC exciting current and a bias DC current that constitute the exciting current, and a magnetic core 12 that is wound around the magnetic core 12 . It comprises a sensor head 14 consisting of a rotating detection coil 13 and a detection circuit 15 connected to one end of the detection coil 13 that is not grounded.

As shown in FIG. 2, the current supply unit 11 generates a DC bias current _idc and an AC excitation current _iac in the excitation current generator 11b based on _clock1 with a clock cycle of 2π/ωc from the variable frequency clock generator 11a. Generate sinω _c t. The variable frequency clock generator 11a also provides clock2 for switching the switch of the synchronous detection circuit (the configuration of the detection circuit 15 will be described in detail later) in the detection circuit 15 (time from _clock1 to ΔT at a clock cycle of 2π/ωc). A clock with a delay) is generated and input to the synchronous detection circuit.

A square wave with a duty ratio of 50% can be generated and used for the generation of the clock waveform, with p periods of the master clock of several tens of MHz (for example, 48 MHz) from the crystal oscillator as 1/2 intervals. _ωc can be made variable by changing the integer value p. To obtain a two-phase clock, for example, one waveform may be shifted from the other waveform by p1 periods (p1<2p) of the master clock.

The excitation current generator 11b superimposes a bias DC current larger than the amplitude of the AC excitation current to generate an excitation current, which is directly applied to the magnetic core 12. FIG. As a result, the output of the detection coil 13 can be obtained at the same fundamental frequency as the frequency of the AC exciting current.

The magnetic core 12 can be a magnetic wire or an elongated magnetic ribbon, and for example, a Co-based non-magnetostrictive amorphous magnetic wire or a Co-based non-magnetostrictive amorphous magnetic ribbon is suitable. Since an excitation current in which an AC excitation current is superimposed on a bias DC current is passed through the magnetic core 12, the high-frequency magnetic field to be measured (hereinafter referred to as the magnetic field to be measured, and the angular frequency is ω _s ) is applied to the sensor head. 14 , the detection signal is detected as an induced voltage in the detection coil 13 wound around the magnetic core 12 of the sensor head 14 . In the present embodiment, the angular frequency ω _c of the AC exciting current is set to a value close to the angular frequency ω _s of the magnetic field to be measured. At this time, ω _s ≠ω _c .

An induced voltage appearing in the detection coil 13 is detected by a detection circuit 15 . The detection circuit 15 includes an amplifier 16 that amplifies the voltage signal induced in the detection coil 13 by the magnetic field to be measured, and a synchronous detection circuit 17 that synchronously detects the amplified signal. The amplifier 16 is a narrowband amplifier with a central angular frequency ω _c as shown in FIG. The synchronous detection circuit 17 is composed of a synchronous switch 17a and a bandpass filter 17b for inverting the sign of the amplified signal at predetermined intervals. Synchronous switch 17a is a switch whose switching is controlled by clock2 having a time delay of ΔT from _clock1 (clock cycle 2π/ωc). Invert the sign every half cycle. As a result, the same effect as multiplication by _cosωct is obtained, and only the component corresponding to the passband of the bandpass filter 17b is output from the calculation result.

_{The passband} of the bandpass filter 17b is set so that, for example, when _ωc is _{| ωs} - _ωc | ω ₁ on the side can be set, for example, to remove the noise magnetic field from the commercial power line and the 1/f noise inherent in the fundamental quadrature fluxgate. Since the 1/f noise of the fundamental wave type quadrature fluxgate becomes smaller than the white noise around 10 Hz, it is possible to set, for example, ₁₈₀ ×2π<ω1 to remove the noise magnetic field from the commercial power supply. Note that the bandpass filter 17b may be a lowpass filter as described later in the second embodiment. Since the selection of the bandpass filter 17b is for enhancing the signal-to-noise ratio of the magnetic field detection, if it is necessary to monitor the magnitude of the geomagnetism input to the magnetic field sensor 1 as well, it should be a lowpass filter with ω ₁ =0. can be

1 to 3, the magnetic field sensor 1 according to the present embodiment can measure a high-frequency magnetic field to be measured as a down-converted low-frequency signal.

The theory of the magnetic field sensor 1 according to this embodiment will be described in detail below. The magnetic core 12 of the sensor head 14 of the magnetic field sensor 1 is supplied with an excitation current in which an AC excitation current is superimposed on a DC bias current. It changes with the excitation cycle of the AC excitation current. The induced voltage to the detection coil 13 is expressed by the axial magnetic flux density _Bz , which is the product of the input magnetic field H to be measured and the axial magnetic permeability _μz , the number of coil windings n, and the effective cross-sectional area of the magnetic core 12 of the sensor head 14. It is the time derivative of the interlinkage magnetic flux φ _z (=nSμ _z H) obtained by multiplying S. When the magnetic field to be measured is a magnetic field that can be detected by a conventional fundamental wave type orthogonal fluxgate of about DC to 1 kHz, the induced voltage is mainly generated by the time change of the axial permeability μ _z , but the magnetic field to be measured is is as high as the excitation magnetic field (frequency is, for example, about 30 kHz to 90 kHz, but not limited to this), a large induced voltage is also generated from the magnetic flux component due to the constant component of the magnetic permeability.

In the case of the magnetic field sensor 1 of the present embodiment in which the magnetic field to be measured has a high frequency comparable to the excitation magnetic field, frequency mixing occurs between the excitation frequency and the frequency of the magnetic field to be measured, and a modulated wave is generated in a low frequency range of several kHz or less. Appear. The modulated wave on the low frequency side generated by this frequency mixing can be used as a means for detecting with high sensitivity a high frequency magnetic field far higher than the detection frequency band of the conventional fundamental wave type orthogonal fluxgate. Information possessed by the magnetic field to be measured is the amplitude and phase, and if the frequency is unknown, the frequency is also added as information. We will explain how frequency mixing occurs in a fundamental-wave quadrature fluxgate and how to assemble a high-frequency magnetic field detection system that utilizes frequency mixing.

(1) Longitudinal permeability μ _z under circumferential (width) excitation
A detection coil 13 is wound around the entire length of the magnetic core 12 or a shorter range, and response magnetic flux to the input magnetic field to be measured interlinks with the detection coil 13 . The behavior of the magnetization in the surface layer of the magnetic core 12 determines the operation of the fluxgate. Target magnetic field H and excitation magnetic field H _ex . The operation of the fluxgate is explained by the rotating magnetization model of FIG. 4 described by these quantities. In FIG. 4, the longitudinal direction of the magnetic core 12 is taken as the z-axis, the magnetic field H to be measured is input in the z-axis direction, and the magnetic field H _ex (=H _dc +H _ac ) created by the exciting current flowing through the magnetic core 12 is It is in the circumferential direction of the magnetic core 12 . The magnetic anisotropy is expressed by uniaxial magnetic anisotropy.

The magnitude K of the magnetic anisotropy constant, the angle α formed by the axis of easy magnetization from the circumferential direction, and the saturation magnetization J _s are caused by the properties of the magnetic material of the magnetic core 12 . H is an unknown quantity and the excitation field H _ex is a controlled quantity as a function of time. The angle θ in FIG. 4 is determined as the direction that minimizes the energy E expressed by the following equation (1), so from equation (2) obtained as the derivative of equation (1) with respect to θ=0, θ is solve.

FIG. 5(A) is a waveform showing the time change of the excitation magnetic field. As shown in FIG. 5(A), the excitation magnetic field H _ex =H _dc +H _{ac sinωt} (H _ac ≦H _dc ) changes with time. Calculate the time series of θ by substituting each value into the equation (2). The magnetization component in the axial direction of the magnetic core 12 that interlinks with the detection coil 13 is J _z =J _s sin θ, but the normalized J _z /J _s is given by It is FIG. 5(B) that calculated and plotted. The difference between the value of the dashed line and the value of the solid line in FIG. 5(B) is the normalized increase in magnetization with respect to the increase of 0.5 A/m in the magnetic field to be measured.

The magnetic flux density B is generally expressed as B=J+μ ₀ H using the vacuum permeability μ ₀ , but when J>>μ ₀ H holds, it can be regarded as B≈J. In the case of a Co-based non-magnetostrictive composition amorphous magnetic material, J _s =0.5 to 0.7 T, and even if it is 0.02 J _s , J _z ≈0.01 to 0.014 T. For example, H is 50 A/m, which is about the same as geomagnetism. Since μ ₀ H≈62 μT at time, J _z >>μ ₀ H, and can be regarded as B _z ≈J _z .

The figure shows the results of calculations of how the normalized magnetization (J _z /J _s ), which changes periodically with time due to the excitation magnetic field, changes according to the magnetic field H to be measured. 5(C). In this numerical calculation, J _s =0.7 T, K = 7 Joule/m ³ , α = 10° as physical properties, H _dc is a 40 mA direct current through a wire with a diameter of 120 μm, and H _ac is an alternating current with an amplitude of 30 mA. is the value on the wire surface when the current is flowing (H _dc ≈ 106 A/m, H _ac ≈ 80 A/m), and ΔH = 0.1 A/m.

The behavior of the axial (incremental) permeability ΔB _z /ΔH can be known from the curve group interval in FIG. 5(C). In FIG. 5(C), the interval between adjacent curves is large around 0.75 in terms of time normalized to one cycle period, and conversely, the interval is small around 0.25. Looking at this in relation to the excitation magnetic field, it can be seen that ΔB _z /ΔH is large where H _ex /H _dc is small, and ΔB _z /ΔH is small where H _ex /H _dc is large. By reading the interval of the curve group for each time in FIG. 5(C), similarly reading the magnitude of the excitation magnetic field in FIG. 5(A), and eliminating the parameter time t, the axis A plot of directional permeability versus excitation field can be obtained. The relationship is shown in FIG. 5(D).

The curve in FIG. 5(D) is downwardly convex and monotonic. The fine lines are auxiliary lines that connect the endpoints of the curve so that the degree of curvature of the curve can be seen. Since the variation of the curve is monotonic, it can be approximated by terms up to second order. When this is actually calculated, the approximate expression of the (incremental) permeability is as follows.

Return to Hex/Hdc=( ₁ +Hac/Hdcsinωt) and organize by order of _sinωt . If k is set to ₂ , it can be generalized as follows.

In the case of FIG. 3(D), the magnitude of each coefficient is shown as

becomes. Although k ₀ and k ₂ are positive and k ₁ is negative, their absolute values do not differ by an order of magnitude.

(2) Interlinking magnetic flux φ _z to the detection coil
The relational expression for generating the magnetic flux that the magnetic field H to be measured interlinks with the detection coil 13 by the action of the magnetic field sensor 1 is the magnetic permeability of the equation (4), where S is the effective cross-sectional area of the magnetic core 12 and n is the number of coil turns. It is obtained by multiplying the product of H・n・S. Expression (6) is obtained by further expressing (4) in terms of harmonic components.

In addition to φ _z in the equation (6), there are magnetic fluxes interlinking with the detection coil 13 . The component of _Jz represented by the solid line in FIG. becomes. Since the curves generated by taking the difference between the solid line in FIG. 5B and the adjacent curves in FIG. ), H can be expressed as an appropriate constant. This constant value corresponds to the offset input converted magnetic field, and is designated as _Hoff . The expression of the interlinking magnetic flux considering up to this offset H _off is as shown in the expression (7).

(3) Operation of conventional fundamental wave type orthogonal fluxgate and magnetic field sensor 1 from the viewpoint of frequency modulation/demodulation When the input magnetic field to be measured is H = h sin ω _s t and the excitation angular frequency is ω _c , (7 ), the following magnetic flux is linked to the detection coil 13 from the equation.

By differentiating this with respect to time, the voltage induced in the detection coil 13 can be obtained.

In the case of conventional fundamental-wave orthogonal fluxgates shown in Patent Documents 1 and 2 and Non-Patent Document 1, there is a relationship of ω _s <<ω _c , and Equation (9) can be approximated as follows.

The first term in this equation (10) is a waveform (modulated wave) modulated such that the envelope of the carrier wave _cosωct varies with _hsinωst . If this modulated wave is amplified at a high frequency by an amplifier and multiplied by cos ω _c t, the low frequency component sin ω _s t can be separated as shown in the following formula, and can be extracted using a low-pass filter.

The demodulation process for extracting this envelope waveform is synchronous detection. If the carrier waveform of the first term has a phase ψ, it is multiplied by cos(ω _c t+ψ) accordingly. The cut-off angular frequency of the low-pass filter placed after the multiplication is usually less than 1/10 of ω _c . The induced voltage waveform of the _second term ( _{k2ωc sin2ωctsinωs t} ) in the first term in the equation (10) is the _first term ( _{k1ωc cosωc tsinωst} ) in the first term Although it has the same envelope, synchronous detection at the angular frequency ω _c eliminates the low-frequency component as it does not appear. Thus, in the conventional fundamental-wave orthogonal fluxgate, the second term (k ₁ (ω _s cos ω _s t·sin ω _c t + ω _c cos ω _c t • sinω _s t)) determines its output. In terms of magnetic permeability, the second term of equation (4) plays a decisive role. Multiplying the offset term of the second term in equation (10) by _cosωct gives the constant term, _sinωct , cos( _2ωct ), and sin( _3ωct ) components, which represent the DC offset All but the constant term are removed by a low-pass filter.

On the other hand, in the case of the magnetic field sensor 1 according to the present embodiment, ω _s is close to ω _c to the extent that |ω _s &#8211;ω _c |≦ω _c /10, for example. No such approximation is possible. Therefore, the induced voltage in equation (9) is multiplied by cos ω _c t, which is a carrier wave, in order to perform a synchronous detection operation. As a result, the frequency components obtained from each term of the equation (9) are as follows.

The second term is the same as the offset term of the second term in Equation (10) above. Among the components obtained from the above terms, the frequency component (ω _c -ω _s ) that can be extracted from the first term in the first term and the third term in the first term is the cutoff angular frequency of the filter in the subsequent stage, ω _c / If it is set to about 10, it is a frequency that can be sufficiently detected by synchronous detection. Calculation results for the sum of (ω _c -ω _s ) angular frequency components obtained by synchronous detection of the induced voltage in equation (9) are shown below.

Since the output of formula (11) is obtained by actual measurement, the magnetic field to be measured can be measured. When the angular frequency of the output waveform obtained by actual measurement is ω _m and ω _s is calculated from it, it is necessary to pay attention to whether ω _c >ω _s or ω _c <ω _s . In practice, ω _m =ω _c &#8211;ω _s or ω _m =ω _s &#8211;ω _c . In order to identify this, in the present embodiment, the current supply unit 11 is arranged to make ω _c variable. If ω _c is increased and the angular frequency of the output waveform is decreased, ω _m = ω _s &#8211;ω _c . Conversely, if the angular frequency of the output waveform is increased, ω _m = ω _c &#8211 ; ω _s . As a result, ω _s can be obtained by back calculation.

As described above, in the magnetic field sensor according to the present embodiment, the fundamental wave having a down-conversion function by frequency mixing based on the first term in the first term and the third term in the first term in the above equation (9) A quadrature fluxgate can be realized, and a high-frequency magnetic field that cannot be measured by a conventional fundamental-wave quadrature fluxgate can be measured.

In addition, since the frequency can be converted to a low frequency while preserving the amplitude and phase of the signal of the magnetic field to be measured, there is no need to use a high-speed AD converter for digital signal processing. utensils can be used.

(Second embodiment of the present invention)
A magnetic field sensor according to this embodiment will be described with reference to FIG. The magnetic field sensor according to this embodiment is obtained by modifying the configuration of the detection circuit 15 of the magnetic field sensor 1 according to the first embodiment, and has a feedback function to constitute a negative feedback circuit. In this embodiment, explanations that duplicate those of the first embodiment will be omitted.

FIG. 6 is a block diagram showing the configuration of the magnetic field sensor according to this embodiment. The difference from the configuration of FIG. 1 in the first embodiment is that a low-pass filter 17c is provided instead of the band-pass filter 17b, and an error integrator that compares the component passed through the low-pass filter 17c with 0 and integrates the difference. 18, and outputs a voltage signal obtained by removing a DC component from the output voltage of the error integrator 18 by AC coupling. Also, the output of the error integrator 18 is fed back to the detection coil 13 via the resistor 19 . What is fed back is an AC component and a DC component with an angular frequency of ω _c -ω _s .

When the sensor head 14 is exposed to a direct current magnetic field and the operating point of the magnetic core 12 changes, the magnetic permeability of the above equation (3) changes, resulting in fluctuations in sensitivity to high-frequency magnetic fields. The same thing happens when the input amplitude of the magnetic field to be measured is large. In order to eliminate this, the operating point of the sensor head 14 is always kept constant by performing the feedback as described above. The feedback of the DC component is the same as that of the conventional fundamental quadrature fluxgate. However, in the magnetic field sensor 1 according to this embodiment, the AC component to be fed back and the frequency of the magnetic field to be measured are different, and cannot be considered in the same way as the conventional fundamental wave type orthogonal fluxgate.

In the magnetic field sensor 1 according to this embodiment, the AC component of the feedback signal is a current proportional to sin((ω _c -ω _s )t+φ), and the feedback magnetic field given to the magnetic core 12 by this current is the input magnetic field different from hsinω _s t. At first glance, it seems to cancel out by a different amount from the input amount, but it is proportional to the second term k ₁ sinω _c t and sin((ω _c -ω _s )t+φ) in equation (4) where ω = ω _{c The first _ _ _ _} From the term, a magnetic flux component with an angular frequency ω _s is generated in the magnetic core 12, and the magnetic field h sin ω _s t to be measured can be canceled by controlling the phase by adjusting ΔT.

As described above, in the magnetic field sensor according to the present embodiment, feedback can be performed by using the second term component of the equation (4), and a high-quality magnetic field sensor with stable measurement sensitivity is realized. can do.

Reference Signs List 1 magnetic field sensor 11 current supply unit 11a variable frequency clock generator 11b excitation current generator 12 magnetic core 13 detection coil 14 sensor head 15 detection circuit 16 amplifier 17 synchronous detection circuit 17a synchronous switch 17b bandpass filter 17c lowpass filter 18 error amplifier 19 resistance

Claims (5)

a sensor head formed by winding a detection coil around a magnetic core;
a power supply unit that superimposes an exciting alternating current, which is an alternating current for excitation, and a bias direct current, which is a direct current for biasing, and supplies them to the magnetic core;
A detection circuit connected to the detection coil and detecting a high-frequency magnetic field input to the magnetic core as a magnetic field to be measured,
The detection circuit
The following induced voltage v(t) induced in the detection coil

(However, the amplitude of the magnetic field to be measured is h, its angular frequency is ω _s , the exciting angular frequency of the exciting alternating current is ω _c , the effective cross-sectional area of the magnetic core is S, the number of turns of the detection coil is n, and the input conversion offset magnetic field is is H _off , and k ₀ , k ₁ and k ₂ are the coefficients of the approximate expression of the permeability of the magnetic core and have the dimension of H / m). A magnetic field sensor that measures the magnetic field to be measured based on. The magnetic field sensor of claim 1, wherein
A magnetic field sensor in which a difference between the detection angular frequency ω _s and the excitation angular frequency ω _c is set to |ω _s −ω _c |≦ω _c /10. The magnetic field sensor according to claim 1 or 2,
The detection circuit
an amplifier that amplifies the induced voltage;
A synchronous detection circuit that has a synchronous switch and a bandpass filter and synchronously detects the signal amplified by the amplifier,
A magnetic field sensor, wherein the band-pass filter has at least a DC component as a pass-blocking region. The magnetic field sensor according to claim 1 or 2,
The detection circuit
an amplifier connected to one end of the detection coil and amplifying the induced voltage;
a synchronous detection circuit having a synchronous switch and a low-pass filter and synchronously detecting the signal amplified by the amplifier;
forming a negative feedback feedback circuit comprising an error integrator having one end of an input terminal connected to the output of the synchronous detection circuit and an output terminal connected to the one end of the detection coil via a feedback resistor;
A magnetic field sensor that cancels hsinω _s t, which is the object of measurement, with a feedback current proportional to sin((ωc-ωs)t+φ). The magnetic field sensor according to any one of claims 1 to 4,
A magnetic field sensor comprising frequency varying means for varying the frequency of the exciting alternating current.

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