JP2021191070A - Controller of power converter - Google Patents

Abstract

To allow prevention of a power source short circuit and a load release without switching an operation mode of a three-phase single-phase matrix converter (MC) between a current form commutation and a voltage form commutation.SOLUTION: The controller of the present invention is formed to repeat modes in the order from a first, mode MODE1 to a seventh mode, MODE7 and to control a matrix converter in a power converter. The controller executes, by a voltage form commutation, a switch from the first mode, MODE1 to the fourth mode, MODE2 and a switch from the fourth mode, MODE4 to the fifth mode, MODE5, and executes, by a current form commutation, a switch from the second mode, MODE2 to the third mode, MODE3, a switch from the third mode, MODE3 to the fourth mode, MODE4, a switch from the fifth mode, MODE5 to the sixth mode, MODE6, and a switch from the sixth mode, MODE6 to the seventh mode, MODE7.SELECTED DRAWING: Figure 15

Description

The present invention relates to a control device for a power conversion device, and more particularly to a control device for a power conversion device that performs commutation by switching.

In recent years, the applications of isolated AC / DC converters have been expanding in various fields such as bidirectional interfaces for storage batteries. High power densities are generally achieved in this type of converter by introducing high frequency isolated links. Among them, the MC (matrix converter) type DAB (dual active bridge) type bidirectional isolated AC / DC converter described in Patent Document 1 does not require a bulky electrolytic capacitor and has many advantages such as a long life. , Has been attracting attention in recent years.

As a modulation method of the MC type DAB type bidirectional isolated AC / DC converter, as described in Patent Document 1, the output voltage vMC of the _MC is controlled by pulse width modulation (PWM), and It is common to use a modulation method with a period during which the output voltage v _{MC is zero.} Hereinafter, this modulation method is referred to as a ZPWM (Zero voltage Pulse Width Modulation) modulation method.

In the ZPWM modulation method, the output voltage v _MC is changed by switching each switch element constituting the MC. During this switching, interphase short-circuiting the three-phase alternating current is input to MC (hereinafter, referred to as "power short") and the occurrence of an overvoltage due to the flow path loss of the reactor current i _L is the output of MC (hereinafter, In order to prevent (referred to as "load release"), the operation of switching the four switch elements related to one switching in order instead of switching at once (hereinafter referred to as "commutation operation"). Is done.

The commutation operation is roughly classified into two types, voltage type commutation and current type commutation, depending on the switching order of the switch elements. Voltage-commutation, those that perform switching of the switch elements as the flow path of the reactor current i _L is always ensured, can be prevented theoretically open load, depending on the magnitude of the input phase voltage It is necessary to change the switching pattern (hereinafter referred to as "gate pattern"), and if the magnitude determination of the input phase voltage is incorrect, a power short circuit will occur. On the other hand, in the current type commutation, each switch element is switched while preventing current from flowing between one input end and the other input end of the three-phase side power supply by two diodes connected in opposite directions to each other. but, it can be prevented theoretically supply short circuit, it is necessary to change the gate pattern in accordance with the sign of the reactor current i _L, causing the load opening and erroneous code determination of the reactor current i _L.

In this way, voltage-type commutation and current-type commutation have advantages and disadvantages, but from the viewpoint of failure prevention, it is the power short circuit that should be prevented preferentially, so the magnitude relationship between the power supply phases is irrelevant. When stable, the commutation operation of the MC is usually performed by current commutation. 12 and 13 of Patent Document 1 disclose specific examples of gate patterns in current-type commutation.

Non-Patent Document 1 discloses a technique for using a combination of voltage-type commutation and current-type commutation. In this technology, in the critical area where the voltages of two phases of the three-phase AC voltage input to the MC are almost equal and the failure of voltage-type commutation is likely to occur, the commutation operation by current-type commutation is performed. In other periods, the commutation operation by voltage type commutation is performed.

Japanese Unexamined Patent Publication No. 2020-005462

Harusuke Takuma, 1 outside, "Hybrid commutation method with current direction estimation for three-phase-to-single-phase matrix converter" , 2018 IEEE Applied Power Electronics Conference and Exposition (APEC), p. 672-679

However, in the technique of Non-Patent Document 1, the operation mode of the MC is switched between the current type commutation and the voltage type commutation according to the input voltage of the MC. Such switching of the operation mode is not preferable because it complicates the operation of the MC and may make the operation of the MC unstable at the time of switching.

Therefore, one of the objects of the present invention is to provide a control device for a power conversion device capable of preventing both power short circuit and load release without switching the operation mode of MC between current type commutation and voltage type commutation. To do.

The control device of the power conversion device according to the present invention has a transformer having first and second coils that are magnetically coupled to each other, one end of which is connected to the first phase of three-phase AC, and the other end of which is one end of the first coil. A first bidirectional switch element connected to a first node constituting the above, one end connected to the second phase of the three-phase AC, and the other end connected to the first node. A switch element, one end connected to the third phase of the three-phase AC, the other end connected to the first node, one end connected to the first phase, and the other end connected to the first phase. A fourth bidirectional switch element connected to a second node constituting the other end of the first coil, one end connected to the second phase and the other end connected to the second node. A control device for a power conversion device having a bidirectional switch element of 5 and a matrix converter having a sixth bidirectional switch element having one end connected to the third phase and the other end connected to the second node. The first mode in which the third and sixth bidirectional switch elements are turned on when the first phase is larger than the second phase and the second phase is larger than the third phase. , The second mode in which the first and sixth bidirectional switch elements are on, the third mode in which the second and sixth bidirectional switch elements are on, the third and the sixth. The fourth mode in which the bidirectional switch element is on, the fifth mode in which the third and fourth bidirectional switch elements are on, and the third and fifth bidirectional switch elements are on. It is configured to repeatedly control the matrix converter in the order of a sixth mode, the third mode and the seventh mode in which the sixth bidirectional switch element is on, and the first mode to the second mode. Switching to the mode and switching from the fourth mode to the fifth mode are executed by voltage type commutation, switching from the second mode to the third mode, and the third mode. To switch to the fourth mode, switch from the fifth mode to the sixth mode, and switch from the sixth mode to the seventh mode by current-type commutation. It is a control device.

According to the present invention, unstable sign of the reactor current i _L, switching from a first mode in which the load open-prone and using current-commutation to the second mode, and, from the fourth mode Since the switching to the fifth mode is performed by the voltage type commutation and the other switching that does not cause such a problem is performed by the current type current, the operation of the MC between the current type commutation and the voltage type commutation. It is possible to prevent both power short circuit and load release without switching modes.

It is a figure which shows the structure of the power conversion device 1 and the control device 4 by this embodiment. It is a figure which shows the high frequency equivalent circuit of the power conversion apparatus 1 shown in FIG. Voltage _e u during power _{running, e v, e w, v} MC, v INV, current _{i eu, i ev, i ew} , I dc, i L, and the current command values ^{_{I dc I dc * (= P}} * / It is a signal waveform diagram which shows the simulation result of each waveform of V _dc). It is an enlarged view of a part of FIG. 3 (for 0.01 seconds to 0.00014 seconds). It is a signal waveform diagram which shows typically the waveform at the time of power running _{of voltage v MC} , nv _INV and reactor current i _L in the case of nV _dc / e _{M = 0.25.} Voltage _v MC in the case of nV _dc / _e M = _0.5, is a signal waveform diagram showing schematically a power running of the waveform of _{nv INV} and reactor current _{i L.} It is a figure which shows the detail of the control of a matrix converter 10. It is a figure which shows the detail of the control of a matrix converter 10. It is a figure which shows the detail of the control of a matrix converter 10. It is a figure which shows the detail of the control of a matrix converter 10. It is a figure which shows the detail of the control of a matrix converter 10. It is a figure which shows the commutation sequence when the switching from the 1st mode MODE1 to the 2nd mode MODE2 is performed by the current type commutation. It is a figure which shows the commutation sequence in the case of switching from a 1st mode MODE1 to a 2nd mode MODE2 by voltage type commutation. If carried out by switching the current-commutation from the first mode MODE1 to the second mode MODE2 (However, if the reactor current i _L is commutation operation is initiated while a plus) indicates a commutation sequence of It is a figure. It is a figure which shows the commutation method used at the time of mode switching by the control device 4 by embodiment of this invention. It is a signal waveform diagram which shows typically the waveform at the time of regeneration of the voltage v _MC , nv _INV and the reactor current i _L in the case of nV _dc / e _{M = 0.25.} Voltage _v MC in the case of nV _dc / _e M = _0.5, is a signal waveform diagram schematically showing the regeneration time of the waveform of _{nv INV} and reactor current _{i L. (A) is a signal waveform diagram showing the measurement results of the voltage v MC} , v _INV , the reactor current i _L , and the input current i _eu at the time of power running according to the first comparative example, and (b) is the first signal waveform diagram. It is a signal waveform diagram which shows the measurement result of _{the voltage v MC} , v _INV , the reactor current i _L , and the input current i _eu at the time of power running by an Example. (A) is a figure which shows the measurement result of efficiency [%] in each of 1st comparative example and 1st Example, and (b) is a figure which shows the measurement result of 1st comparative example and 1st Example, respectively. It is a figure which shows the THD [%] of the input current in. (A) is a signal waveform diagram showing the measurement results of _{the voltage v MC} , v _INV , the reactor current i _L , and the input current i _eu at the time of power running according to the second comparative example, and (b) is a signal waveform diagram showing the second comparative example. It is a signal waveform diagram which shows the measurement result of _{the voltage v MC} , v _INV , the reactor current i _L , and the input current i _eu at the time of power running by an Example. (A) is a figure which shows the measurement result of the efficiency [%] in each of the 2nd comparative example and 2nd Example, and (b) is a figure which shows the measurement result of the efficiency [%] in each of the 2nd comparative example and the 2nd Example, respectively. It is a figure which shows the THD [%] of the input current in.

Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings.

FIG. 1 is a diagram showing a configuration of a power conversion device 1 and a control device 4 thereof according to the present embodiment. As shown in the figure, the power conversion device 1 according to the present embodiment is an MC type having a three-phase single-phase matrix converter 10, a transformer 20, a full-bridge type AC / DC converter 30, and a protection circuit 35. It is a DAB type bidirectional isolated AC / DC converter. The matrix converter 10, the transformer 20, and the AC / DC converter 30 are connected between the system power supply 2 and the load 3 in this order. Hereinafter, the seventh node n ₇ 3 one end of the matrix converter 10 connected to the system power supply 2, _{respectively,} the node n ₈ of the _eighth, referred to as a node n ₉ ninth matrix that is connected to the transformer 20 node _{n 1} first each of the two ends of the converter 10, referred to as a second node _{n 2,} a third node two ends of the AC / DC converter 30 connected to the transformer 20, respectively _n 3, referred to as a fourth node _{n 4,} the fifth node _{n 5} 2 two ends of the AC / DC converter 30 connected to the load 3, respectively, referred to as node _{n 6} of the sixth. Further, the voltage of the first node n _{1 with respect to the voltage of the second} node n 2 is referred to as a voltage v _MC, and the voltage of the third node n _{3 with respect to the voltage of the fourth} node n 4 is referred to as a voltage v _INV.

The transformer 20 includes two coils 20a and 20b (first and second coils) that are magnetically coupled to each other. Coil 20a, the number of turns of 20b are each _{N P} and _{N S,} turns ratio n of coils 20a, 20b becomes _{n = N} P / N _S. One end of the coil 20a is connected to a matrix converter 10 in the first node n ₁ through the reactor L, the other end of the coil 20a is connected to a matrix converter 10 in the second node n _2. The reactor L may be an inductance element provided separately from the transformer 20, or may be a leakage inductance of the transformer 20. One end of the coil 20b is connected to the AC / DC converter 30 at a third node _{n 3,} the other end of the coil 20b is connected to the AC / DC converter 30 in the fourth node _{n 4.} Hereinafter, the current flowing through the reactor _L is referred to as a reactor current i L. The orientation of the reactor current i _L, the direction from the first node n ₁ to the second node n ₂ is a positive direction, and the opposite negative direction. The reactor current i _L is expressed as a function of time t as in the following equation (1) when the voltage v _MC and nv _{INV are used.}

The AC / DC converter 30 is a device that mutually converts a single-phase AC voltage (coil 20b side) and a DC voltage (load 3 side). The load 3 is, for example, a power converter for driving a storage battery or a motor of a hybrid car, and is operated (charged) by DC power supplied from the power conversion device 1 (power running), and conversely, power conversion. There is a case where DC power is supplied to the device 1 (regeneration). One end of the load 3 is connected to the AC / DC converter 30 at node _{n 5} of the fifth, the other end of the load 3 is connected to the AC / DC converter 30 at node _{n 6} of the sixth. Hereinafter, the voltage of the fifth node n _{5 with respect to the voltage of the sixth} node n 6 is referred to as a voltage V _dc, and the current flowing from the fifth node n ₅ toward the load 3 is referred to as a current I _dc.

AC / DC converter 30 has one end connected to the node n ₅ of the fifth switch elements G _1, which other end is connected to the third node n _{3 (first} uni-directional switch element), one end of the A switch element G ₂ (second one-way switch element) connected to node n ₆ of 6 and the other end connected to a third node n ₃ , and one end connected to a _{fifth node n 5 and others.} a switching element G ₃ which end is connected to the fourth node n _{4 (third} unidirectional switching elements), one end connected to the node n ₆ of the sixth and the other end connected to the fourth node n ₄ It has a switch element G ₄ (fourth one-way switch element) and a capacitor C1 having _{one end connected to the fifth node n 5} and the other end connected to the sixth node n _6. Will be done.

Each of the switch elements G ₁ ~G _4, for example MOSFET (Metal-Oxide-Semiconductor Field -Effect Transistor) or IGBT (Insulated Gate Bipolar Transistor) semiconductor element and, connected diodes and a snubber capacitor in parallel with the semiconductor element such as It is a one-way switch configured to include and. The snubber capacitor may be a parasitic capacitance of a semiconductor element or an independent capacitive element. The switch element G ₁ is incorporated into the circuit so that the diode anode is _{connected to the third node n 3 , and the switch element G 2} is circuit so that the diode cathode is connected to _{the third node n 3.} incorporated in the switch element G ₃ are, incorporated in the circuit as the anode of the diode is connected to the fourth node n _4, switching element G ₄ are cathode of the diode is connected to the fourth node n ₄ It is incorporated in the circuit so as to be.

The respective control electrodes of the semiconductor elements constituting the switching element G ₁ ~G ₄ is a separate control signal is supplied from the control unit 4. Each control signal is a signal that takes either a high value or a low value. The controller 4 controls the values of these control signals individually to control the on-off states of the respective switching elements G ₁ ~G ₄ individually.

The matrix converter 10 is a device that mutually converts a three-phase AC voltage (system power supply 2 side) and a single-phase AC voltage (coil 20a side). System power supply 2 is a three-phase AC power supply for generating an AC voltage e _u, e _v, e _w represented by a sinusoidal signal with shifted 2 [pi / 3 by phases, for example, a commercial power supply. AC voltage _{e u,} e v, expressed the _{e w} in the formula is as the following equation (2). However, E is the effective value (constant) of the voltage between the input lines, and ωt is the phase angle (constant) of the input voltage.

As shown in FIG. 1, the output end of the system power source 2 corresponding to the AC voltage e _u corresponding to the u-phase (first phase) is connected to a matrix converter 10 at node n ₇ of the 7, v-phase (second output end of the system power source 2 corresponding to the AC voltage e _v corresponding to the phase) is connected to the matrix converter 10 at node n ₈ of the eighth, corresponding to an AC voltage e _w corresponding to w-phase (phase 3) output end of the system power source 2 is connected to a matrix converter 10 at node n ₉ ninth. Hereinafter, the seventh to ninth node _n 7 ~n ₉ a current flowing each current _{i eu,} i _ev, referred to as _{i ew.}

Matrix converter 10 has one end connected to the node n ₇ of the seventh and the other end and connected to the bidirectional switch element S _up to the first node n _{1 (first} bidirectional switch element), one end of the is connected to the node n ₈ 8, the other end and connected to the bidirectional switch element S _vp to the first node n _{1 (second} bidirectional switch element), one end connected to the node n ₉ of 9 , The other end is _{connected to the first node n 1} and the bidirectional switch element _Swp (third bidirectional switch element), and one end is _{connected to the seventh node n 7} and the other end is the second node. A bidirectional switch element _Sun (fourth bidirectional switch element) connected to n _{2 and a bidirectional one end connected to the eighth node n 8} and the other end connected to the second node n _2. switching element S _{vn (the} fifth bidirectional switching devices), one end connected to the node n ₉ ninth and the other end of the second node n ₂ to the connected bidirectional switch element S _{wn (sixth} A bidirectional switch element), an AC reactor Lf, a damping resistor Rf, and an input capacitor Cf.

AC reactor Lf, the node _{n 7} and the bidirectional switch element _S up _seventh, both the inserted inductor, the node _{n 8} eighth between tenth node _{n 10} of a connecting point of _{S un} It is the connection point between the inductor inserted between _{the 11th node n 11} which is the connection point of the direction switch elements S _bp and S _{vn , and the 9th node n 9} and the bidirectional switch elements _Swp and _Swn. It constituted by an inductor inserted between the twelfth node n ₁₂ of the. The damping resistor Rf is composed of three resistance elements connected in parallel with each of these inductors. Input capacitor Cf is composed of three capacitors connected to each other by a delta connection or star connection, the three connecting points are connected to the node n ₁₀ ~n ₁₂ of the 10 to 12 respectively. Hereinafter, referred to as the tenth to twelfth node _n 10 _{~n 12} the current flowing the respective input current _i u _of, i v, _{i w.}

Bidirectional switch element _{S up, S vp,} a _{S wp, S un, S vn} , bidirectional switch composed respectively _{S wn,} comprise two switch elements connected in series. Specifically, the bidirectional switch element _{S up,} the switch element _G pu connected in _series, and _{G up,} switching element _{G pu,} the snubber capacitor connected in parallel with the series circuit composed of _{G up} It is composed. Furthermore, the bidirectional switching element _{S vp} is composed switching element _G pv connected in _series, and _{G vp,} switching element _{G pv,} the snubber capacitor connected in parallel with the series circuit comprising the _{G vp,} bidirectional switch element _{S wp} is composed switching element _G pw connected in _series, and _{G wp,} switching element _{G pw,} the snubber capacitor connected in parallel with the series circuit comprising the _{G wp,} bidirectional switching element _{S un} are configured switching elements _G nu connected in _series, and _{G un,} switching element _{G nu,} by the snubber capacitor connected in parallel with the series circuit comprising the _{G un,} bidirectional switch element S _vn, the switch element _G nv connected in _series, and _{G vn,} switching element _{G nv,} is constituted by a snubber capacitor connected in parallel with the series circuit comprising the _{G vn,} bidirectional switch element _{S wn} consists switching element _G nw connected in _series, and _{G wn,} switching element _{G nw,} by the snubber capacitor connected in parallel with the series circuit comprising the _{G wn.} These switch elements G _pu , G _up , G _pv , G _{v p} , G _{p w} , G _pp , G _nu , _Gun , G _nv , G _vn , G _nw , and G _n are semiconductor devices such as MOSFETs and IGBTs, respectively. This is a one-way switch including a semiconductor element and a diode connected in parallel.

The switch elements G _pu and G _up are connected between the first node n ₁ and the tenth node n ₁₀ in this order and in the direction in which the anodes of the respective diodes are connected to each other. The switch elements G _pv and G _vp are connected between the first node n ₁ and the eleventh node n ₁₁ in this order and in the direction in which the anodes of the respective diodes are connected to each other. The switch elements G _pw and G _wp are connected between the first node n ₁ and the twelfth node n ₁₂ in this order and in the direction in which the anodes of the respective diodes are connected to each other. The switch elements G _nu and _Gun are connected between the second node n ₂ and the tenth node n ₁₀ in this order, and in the direction in which the anodes of the respective diodes are connected to each other. The switch elements G _nv and G _vn are connected between the second node n ₂ and the eleventh node n ₁₁ in this order and in the direction in which the anodes of the respective diodes are connected to each other. The switch elements G _nw and G _wn are connected between the second node n ₂ and the twelfth node n ₁₂ in this order and in the direction in which the anodes of the respective diodes are connected to each other. Regarding the connection of the two switch elements that make up the bidirectional switch, here it is assumed that the anodes of the respective diodes are connected in the direction in which they are connected to each other, but the direction in which the cathodes of the respective diodes are connected to each other. It may be connected by.

The control electrodes of the semiconductor elements constituting the switch elements G _pu , G _up , G _pv , G _vp , G _pw , G _wp , G _nu , _Gun , G _nv , G _vn , G _nw , and G _{w n are} Individual control signals are supplied from the control device 4. Each control signal is a signal that takes either a high value or a low value. The controller 4 controls the values of these control signals individually, the switch element _{G pu, G up, G pv} , G vp, G pw, G wp, G nu, G un, G nv, G vn , G _nw , G _wn Each on / off state is controlled individually.

Protection circuit 35 is a circuit for absorbing reactor current i _L during commutation failure, the rectifier circuit comprising a full-bridge-connected four diodes, connected a smoothing capacitor and a load in parallel with the rectifier circuit It is composed of and. The voltage across the load is V _p . Each of the two input ends of the rectifier circuit is connected to the first node n ₁ and second node n _2.

FIG. 2 is a diagram showing a high frequency equivalent circuit of the power conversion device 1. As shown in the figure, when the power conversion device 1 operates at a high frequency, _{a power supply circuit that outputs a voltage v MC} and a power supply circuit that outputs a voltage nv _INV are connected in series with a reactor L interposed therebetween. It is equivalent to the circuit that was made.

Return to FIG. The control device 4, by the above-mentioned ZPWM modulation method, based on the power command value ^{P *} and the power factor angle command value alpha ^* supplied from the outside, and the value of the voltage _{e u, e v, e w} , V dc bidirectional switch element Te _{S up, S vp,} a _{S wp, S un, S vn} , S wn, G 1 ~G 4 device for controlling the respective off states. By performing this control, the control device 4 applies the power ^{specified by the power command value P *} to the load 3 from the system power supply 2 at the power factor angle specified by the power factor angle command value α ^{* (in the case of power running).} ) Or, transmission from the load 3 to the system power supply 2 (in the case of regeneration) is realized. The operation of the control device 4 during power running and regeneration is basically the same except that the signs of the phase difference δ, which will be described later, are exchanged. Therefore, in the following, the explanation will be continued focusing on the time of powering. The regenerative state will be described with reference to FIGS. 16 and 17 after explaining the operation of the control device 4 during power running.

FIG. 3 shows the command values I _dc ^* _{of the voltage e u} , e _v , e _w , v _MC , v _INV , the current i _eu , i _ev , i _ew , I _dc , i _L , and the current I _dc at the time of power running. It is a signal waveform diagram which shows the simulation result of each waveform of = P ^* / V _dc). The detailed simulation conditions are as shown in Table 1. FIG 3, voltage _{e u,} e v, shows one period of the three-phase AC represented by _{e w.} FIG. 4 is an enlarged view of a part of FIG. 3 (for 0.01 to 0.00014 seconds).

1 cycle of the three-phase alternating current, voltage _{e u,} e v, depending on the phase of the _{e w,} can be divided into twelve space I~XII as shown in FIG. 3. Specifically, the space I whose phase of the voltage e _u is 0 or more and less than π / 6, the space II where π / 6 or more and less than π / 3, and the space III and π / where π / 3 or more and less than π / 2 are present. Space IV that is 2 or more and less than 2π / 3, space V that is 2π / 3 or more and less than 5π / 6, space VI that is 5π / 6 or more and less than π, space VII that is π or more and less than 7π / 6, 7π / 6 or more Space VIII that is less than 4π / 3, space IX that is 4π / 3 or more and less than 3π / 2, space X that is 3π / 2 or more and less than 5π / 3, space XI that is 5π / 3 or more and less than 11π / 6, It can be divided into spaces XII which are 6 or more and less than 2π. The waveform shown in FIG. 4 corresponds to a part of the space IV.

The operation of the control device 4 in each space I-XII, the voltage _{e u,} e v, switching element to be controlled in accordance with a magnitude relationship between _{e w} is replaced, and the voltage _{e u,} e v, the _{e w} It is basically the same except that the output voltage may change depending on the sign of the one that takes an intermediate value. Therefore, in the following, the explanation will be continued focusing on _{the space V (eu} > _ev >0> e _w , _iv ^{*> 0).}

5 and 6 respectively, the voltage _{v MC,} is a signal waveform diagram showing schematically a power running of the waveform of _{nv INV} and reactor current _{i L.} FIG. 5 _{(specifically, nV dc / e M = 0.25} ) is the value of the voltage _{V dc} relatively small example shown in FIG. 1 shows the, in FIG. 6 are shown in FIG. 1 _{(specifically, nV dc / e M = 0.5} ) the value of the voltage _{V dc} relatively large example shows. The voltage _{e M} and _{e m} shown in FIGS. 5 and 6 are represented respectively by the following formula (3) (4). However, a command value of the current _i ^{v *} is input current _{i v} shown in Formula (3). The command values i _u ^* , i _v ^* , i _w ^* of the input currents i _u , i _v , i _w are expressed by the equation (5).

Explaining the outline of the operation of the control device 4 with reference to FIG. 5, first, the control device 4 is configured to periodically control the matrix converter 10 with a period T. The switching frequency of the matrix converter 10 is the reciprocal of this period T. _{The time t 1} shown in FIG. 5 corresponds to the beginning of the cycle T, and the time t ₁₀ corresponds to the end. Details of the control of the matrix converter 10 will be described later.

Further, the control device 4 is configured to periodically control the AC / DC converter 30 with the same period T. _{The time t 2} shown in FIG. 5 corresponds to the beginning of this control cycle, and the time t ₁₁ corresponds to the end. Specifically, the control device 4 turns on the switches G1 and G4 and turns off the switches G2 and G3 in the first half of each cycle, turns on the switches G2 and G3 and turns off the switches G1 and G4 in the latter half of each cycle. It is configured to do. However, when switching the switches, a period (dead time) is provided in which all the switches are once turned off. As a result of the control by the control device 4, _{the value of the voltage nv inv becomes nV dc in} the first half of each cycle and −nV _{dc in} the latter half of each cycle, as shown in FIG. In the following description, the elapsed time length from the start of the control cycle of the matrix converter 10 to the start of the control cycle of the AC / DC converter 30 that arrives for the first time after that is referred to as a phase difference δ.

7 to 11 are diagrams showing details of control of the matrix converter 10. Dashed arrowed shown in this drawing represents the path and orientation of the reactor current i _L. Hereinafter, the control of the matrix converter 10 will be described in detail with reference to these figures together with FIG. Actually, the commutation operation is performed when switching the on / off of each switch element, but here, the commutation operation will be ignored. The commutation operation will be described in detail later with reference to FIGS. 12 to 14.

First, as shown in FIG. 5, the control device 4 is configured to control the matrix converter 10 by sequentially executing seven modes from the first mode MODE1 to the seventh mode MODE7 in each cycle. To. The duration of the first mode MODE1 and seventh mode MODE7 (duty) are identical to each other, in the following, referred to the duration and _d z / 2. The duration of the fourth mode MODE4 is twice the duration of the first mode MODE1 and seventh mode MODE7 (= _{d z).} The duration of the third mode MODE3 and sixth mode MODE6 are also identical to each other, in the following, referred to the duration and _{d m.} The duration of the second mode MODE2 and fifth modes MODE5 is also identical to each other _{(= T / 2-d z} -d m). In the example of FIG. 5, δ < _dz / 2, and the start of the control cycle of the AC / DC converter 30 arrives during the execution of the first mode MODE1.

In FIG. 7 (a), the bidirectional switch element _S up at time _{t 1} is the beginning of the first mode _{MODE1, S vp, S wp,} S un, S vn, shows the state of _{S wn.} Further, in FIG. 7 (b), AC / DC converter 30 each bidirectional switch element _S up at time _{t 2} is the start of the control _{cycle, S vp, S wp, S} un, S vn, the _{S wn} state Is shown. As shown in these figures, in the first mode MODE1, the control device 4, the bidirectional switch element _{S wp,} the _{S wn} is turned on, the other bidirectional switch element _{S up, S vp, S un} , S Turn off _vn. As a result, the first node n ₁ and the second node n ₂ are short-circuited, so that the reactor L is in a state where only the _{voltage nv INV is applied.} Since nv _INV = −nV _dc at time t ₁ to time t ₂ , the current i _L changes in an increasing direction as shown in FIG. On the other hand, since nv _INV = nV _{dc at time t 2} to time t ₃ , the current i _L changes in a decreasing direction as shown in FIG.

In FIG. 8 (a), the bidirectional switch element _S up at time _{t 3} is the beginning of the second mode _{MODE2, S vp, S wp,} S un, S vn, shows the state of _{S wn.} As shown in the figure, in the second mode MODE2, the control device 4, the bidirectional switch element _S Stay _up-, the _{S wn} is turned on, the other bidirectional switch element _{S vp, S wp, S un} , S vn Is turned off. As a result, the voltage e _{M is applied between the first node n 1} and the second node n ₂ , so that the reactor L is in a state where the _{voltage e M} −nv _{INV is applied.} Then, from time t ₃ to time t ₄ , nv _INV = nV _dc , but the voltage e _M is generally set to a value larger than the voltage nV _{dc (e M} > nV _dc ), so as shown in FIG. The current i _L changes in the increasing direction.

In FIG. 8 (b), the bidirectional switch element _S up at time _{t 4} is the start of the third mode _{MODE3, S vp, S wp,} S un, S vn, shows the state of _{S wn.} As shown in the figure, in the third mode MODE3, the control device 4, the bidirectional switch element _{S vp,} the _{S wn} is turned on, the other bidirectional switch element _{S up, S wp, S un} , S vn Is turned off. Thus, since the voltage _{e m} is applied between the first node _{n 1} and the second node _{n 2,} reactor L becomes a state where a voltage _e m _{-nv INV} is applied. Then, a time _{t 4} ~ time _{t 5} in _nv INV = _{nV dc,} as shown in FIG. 5, when the voltage _{e m} and a value larger than the voltage _{nV dc (e m> nV dc} ), the current _{i L} Changes in the direction of increasing. However, the voltage _{e m} since the voltage _{e M} smaller value _(e M> e _m), the rate of increase in current _{i L} is compared with the growth rate at time _{t 3} ~ time _{t 4} as shown in FIG. 5 Becomes smaller. The voltage _{e m} is the voltage _{nV dc} value smaller than in the case _(e m _{<nV dc),} the current _{i L} is varied in a decreasing direction.

FIG. 9 (a), the bidirectional switch element _S up at time _{t 5} is a beginning of the fourth mode _{MODE4, S vp, S wp,} S un, S vn, shows the state of _{S wn.} Further, in FIG. 9 (b), the time _t 6 is a switching timing of the switch element _{G 1 ~G 4 (t 6 -t} 2 = T / 2) the bidirectional switch element in _{S up, S vp, S wp} , _Sun , S _vn , _Swn are shown. As shown in these figures, in the fourth mode MODE4, the control device 4, the bidirectional switch element _{S wp,} the _{S wn} is turned on, the other bidirectional switch element _{S up, S vp, S un} , S Turn off _vn. This is the same state as the first mode MODE1. Therefore, since the first node n ₁ and the second node n ₂ are short-circuited, the reactor L is in a state where only the _{voltage nv INV is applied.} Since nv _INV = nV _dc from time t ₅ to time t ₆ , the current i _L changes in a decreasing direction as shown in FIG. On the other hand, since nv _INV = −nV _{dc at time t 6} to time t ₇ , the current i _L changes in an increasing direction as shown in FIG.

In FIG. 10 (a), the bidirectional switch element _S up at time _{t 7} is a beginning of the fifth mode _{MODE5, S vp, S wp,} S un, S vn, shows the state of _{S wn.} As shown in the figure, in the fifth mode MODE5, the control device 4, the bidirectional switch element _{S wp,} the _{S un} is turned on, the other bidirectional switch element _{S up, S vp, S vn} , S wn Is turned off. As a result, the voltage-e _{M is applied between the first node n 1} and the second node n ₂ , so that the reactor L is in a state where the _{voltage-e M-} nv _{INV is applied.} Then, since nv _INV = −nV _{dc from time t 7} to time t ₈ , if e _M > nV _dc holds as described above, the current i _L changes in a decreasing direction as shown in FIG. do.

The FIG. 10 (b), the sixth the bidirectional switch element _S up at time _{t 8} is the beginning of mode MODE6 _{of, S vp, S wp, S} un, S vn, shows the state of _{S wn.} As shown in the figure, in the sixth mode MODE 6, the control device 4, the bidirectional switch element _{S wp,} the _{S vn} is turned on, the other bidirectional switch element _{S up, S vp, S un} , S wn Is turned off. As a result, the voltage _{−m is applied between the first node n 1} and the second node n ₂ , so that the reactor L is in a state where the _{voltage −m} −nv _{INV is applied.} And, since at time _{t 8} ~ time _{t 9} in _nv INV = _{-nV dc,} when an _e m> _{nV dc} as described above is satisfied, as shown in FIG. 5, the change in the direction to decrease the current _{i L} do. However, since e _M> e _m as described above is satisfied, the reduction rate of the current i _L is smaller than the reduction rate at time t _{7 ~} time t ₈ as shown in FIG.

11, each bidirectional switch element _S up at time _{t 9} is the beginning of the seventh mode of _{MODE7, S vp, S wp,} S un, S vn, shows the state of _{S wn.} As shown in the figure, in the seventh mode MODE7, the control device 4, the bidirectional switch element _{S wp,} the _{S wn} is turned on, the other bidirectional switch element _{S up, S vp, S un} , S vn Is turned off. This is the same state as the first mode MODE1. Therefore, since the first node n ₁ and the second node n ₂ are short-circuited, the reactor L is in a state where only the _{voltage nv INV is applied.} Since nv _INV = −nV _dc at time t ₉ to time t ₁₀ , the current i _L changes in an increasing direction as shown in FIG.

As described so far, the control device 4, the bidirectional switch element _{S wp, S wn} is a mode (i.e. _{v MC} = 0 to be turned on at the same time mode. More specifically, first, fourth, 7 modes MODE1, MODE4, MODE7) are carried out at least once in each cycle. _{Therefore, since it is possible to suppress an increase in the circulating current when the output voltage nV dc} on the AC / DC converter 30 side decreases, it is possible to suppress a decrease in the power factor when the power conversion device 1 is operated in a wide voltage range. become.

Next, the bidirectional switching element _{S up, S vp, S wp} , S un, S vn, the commutation operation is carried out when switching off of _{S wn} be described.

Bidirectional switch element _{S up, S vp, S wp} , S un, S vn, switching on and off of the _{S wn} are, in the case of p-side commutation always bidirectional switch element _{S up, S vp, S} one of _wp and commutation source, bidirectional switch element _S Stay _up-, S _vp, is performed one of the other _{S wp} as commutation destination. Further, in the case of n-side commutation always bidirectional switch element _{S un,} S _vn, one of the _{S wn} and commutation source, bidirectional switch element _{S un,} S _vn, the _{S wn} It is executed with any one of them as the commutation destination. Then, the control device 4 constitutes one of the two switch elements constituting one of the bidirectional switch elements of the commutation source and the commutation destination, and the other of the bidirectional switch elements of the commutation source and the commutation destination. One of the two switch elements, the remaining one of the two switch elements constituting one of the bidirectional switch elements of the commutation source and the commutation destination, and the bidirectional switch of the commutation source and the commutation destination. The commutation operation is executed by controlling the on / off of each switch element in the order of the remaining one of the two switch elements constituting the other of the elements.

The commutation operation is roughly classified into the above-mentioned voltage type commutation and current type commutation according to the order of the switch elements subject to the on / off control performed in this way. Hereinafter, each of the voltage type commutation and the current type commutation will be described in detail by taking the case of switching from the first mode MODE1 to the second mode MODE2 as an example.

FIG. 12 is a diagram showing a commutation sequence in the case where switching from the first mode MODE1 to the second mode MODE2 is performed by current-type commutation. Current-commutation, according to the direction of the reactor current i _{L (output} phase current), a commutation method for performing on-off control of each switch element such that the path of the reactor current i _L is always ensured. Therefore, when performing a current-commutation has to the direction of the reactor current i _L is known in advance. Here, as understood from FIG. 5, the direction of the reactor current i _L in the beginning of the second mode MODE2 (time t ₃₎ is in the normal negative direction. Therefore the control unit 4, assuming the direction of the reactor current i _L is negative direction, performs switching control so as to perform from the first mode MODE1 by current-commutation to the second mode MODE2. Figure 12 is a broken line with arrows shows the reactor current i _L flowing in the negative direction.

The commutation sequence of the current type commutation will be specifically described with reference to FIG. First, FIG. 12A shows a state before the start of commutation operation. In this state, the switch element _G wp constituting the bidirectional switch element _{S wp} to be _{Tenryumoto, G pw} are both turned on, the switch element _G up which constitutes the bidirectional switch element _{S up} to the commutation _{destination, G pu} Are both off.

Control device 4 starts the commutation operation by current-commutation, initially as shown in FIG. 12 (b), according to the direction of the reactor current i _L, the switch element G _wp, turns off of G _pw also the route while the can ensure the reactor current i _{L (reactor} flows in the negative direction current i _L) (specifically, switching elements G _wp) turning off. The reactor current i _L after the switch element G _wp is turned off continues to flow through the diode in the switch element G _wp.

Next, the control unit 4, as shown in FIG. 12 (c), according to the direction of the reactor current _{i L,} the switch element _G Stay _up-, newly constructed path of the reactor current _{i L} by turning on of the _{G pu} On the other hand (specifically, the switch element G _pu ) is turned on. After this, the reactor current i _L will flow through the path through the diode in the switch element _Gwp. As for the diode in the _{switch element Gup} , no current flows because _{eM is applied to both ends.}

Next, as shown in FIG. 12 (d), the control device 4 _{turns off the switch element G pw} that has not yet been turned off among the _{switch elements G wp} and G _pw . As a result, the path of the reactor current i _{L passing through the bidirectional switch element Swp} is cut off, and the reactor current i _L flows only through the path passing through the diode in the switch element _Group.

Finally the control unit 4, as shown in FIG. 12 (e), the switch element _{G up,} among the _{G pu,} turning on the switching element _{G up} that have not yet turned on. Thus commutation from the bidirectional switching element S _wp to the bidirectional switch element S _up is completed.

FIG. 13 is a diagram showing a commutation sequence in the case where switching from the first mode MODE1 to the second mode MODE2 is performed by voltage-type commutation. In the voltage type commutation, one end of the bidirectional switch element that is the commutation source (input end on the system power supply 2 side) and one end of the bidirectional switch element that is the commutation destination (input end on the system power supply 2 side). Depending on the sign of the voltage applied between them (the voltage at one end of the bidirectional switch element that is the commutation destination with respect to one end of the bidirectional switch element that is the commutation source, hereinafter referred to as "input phase voltage"). Each switch element so as not to short-circuit one end of the bidirectional switch element that is the flow destination (input end on the system power supply 2 side) and one end of the bidirectional switch element that is the commutation source (input end on the system power supply 2 side). It is a commutation method that controls the on / off of. Therefore, when performing voltage-type commutation, it is necessary to know the sign of the input phase voltage in advance. Here, as understood from FIG. 5, in the switching from the first mode MODE1 to the second mode MODE2, the input phase voltage changes from 0 _e M (> 0). Therefore, the control device 4 is configured to switch from the first mode MODE1 to the second mode MODE2 by voltage-type commutation, assuming that the input phase voltage is positive. Also in FIG. 13, the broken line with arrows shows the reactor current i _L flowing in the negative direction.

The commutation sequence of voltage-type commutation will be specifically described with reference to FIG. First, FIG. 13A shows a state before the start of commutation operation. This is the same state as in FIG. 12 (a).

Control device 4 starts the commutation operation by voltage-commutation, as shown in FIG. 13 (b) First, depending on the sign of the input phase voltage, switching elements G _{Stay up-,} bidirectional switch element of G _pu while not short-circuit between one end of the one end and the bidirectional switch element S _wp of S _{Stay up-(specifically,} switching elements G _pu) turns on. If Assuming that here it is assumed that turning on the switching element _{G pu} instead switching element _{G Stay up-,} since it is e u> _{e w,} over the one end of the bidirectional switch element _{S up} to one end of the bidirectional switch element _{S wp,} A current path is formed through the diode in the switch element _Gpu. Then, since the current flows toward the one end of the bidirectional switch element S _up to one end of the bidirectional switch element S _wp, power short circuit occurs. If the switch element _Gpu is turned on, such a current path is not formed and a power short circuit does not occur.

Next, as shown in FIG. 13 (c), the control device 4 can secure the path of the reactor current i _L regardless of the direction of the reactor current i _{L even if the switch elements G wp} and G _{p w are turned off.} (Specifically, the switch element _GPw ) is turned off. As a result, when the sign of the reactor current i _L is negative, the reactor current i _L flows _{through the bidirectional switch element Sup} as shown in FIG. 13 (c). Although not shown, when the sign of the _{reactor current i L} is positive, the reactor current i _L flows _{through the bidirectional switch element Swp.}

Next, the control unit 4, as shown in FIG. 13 (d), the switch element _{G up,} among the _{G pu,} turning on the switching element _{G up} that have not yet turned on. And further, as shown in FIG. 13 (e), the switch element _{G wp,} among _{G pw,} to turn off the switching element _{G wp} not yet turned off. Thus commutation from the bidirectional switching element S _wp to the bidirectional switch element S _up is completed.

Next, the problem that may occur when switching from the first mode MODE1 to the second mode MODE2 by the current type commutation shown in FIG. 12 will be described in detail.

Turning to FIG. 5, that the sign of the reactor current i _L before and after the time t ₃ when the switching from the first mode MODE1 to the second mode MODE2 is performed is changing is appreciated. It is difficult to predict the timing of the change completely, as a result, the reactor current i _L during commutation operation is sometimes become positive. However, as described above, the commutation operation in this case is _{executed on the premise that the direction of the reactor current i L} is in the negative direction. Therefore, when the reactor current i _L becomes positive, the commutation operation occurs. It will not be done properly.

FIG. 14 is a diagram showing a commutation sequence in the case where switching from the first mode MODE1 to the second mode MODE2 is performed by current-type commutation. However, the figure _{shows an example in which the commutation operation is started while the reactor current i L} is positive, and the direction of the _{reactor current i L} shown in FIGS. 14 (a) to 14 (d) is shown in the figure. It is the opposite of that of 12 (a) to (d).

Figure 14 (a) to commutation operation starts before the state shown, except that the direction of the reactor current i _L are reversed, the same as the state shown in Figure 12 (a).

The control device 4 that has started the commutation operation by the current type commutation first turns off the _{switch element G wp as shown in FIG. 14 (b). This is expected to continue the reactor current i L} through the diode in the _{switch element G wp} , as described with reference to FIG. 12 (b), but the actual reactor current i _L. Since the direction is positive, the reactor current i _L cannot pass through the diode in the switch element G _wp. As a result, the flow path of the reactor current _{i L} is lost, so that the bidirectional switch element _S Stay _{up-, S wp} reactor current _{i L} in the respective snubber capacitor flows.

Reactor current i _L flowing into the snubber capacitor charges the snubber capacitor. The illustrated Δv (> 0) represents the potential difference between the electrode plates of the snubber capacitor of _{the bidirectional switch element Swp thus charged.} With this Delta] v, electrode plates potential of the snubber capacitor of the bidirectional switching element _{S up} is expressed as _{Δv + e M (> 0)} . When a potential difference is generated between the electrode plates of the snubber capacitor, the first node n ₁ of the voltage by that amount, that is, the voltage v _MC decreases. Specific examples of this decrease are shown in FIGS. 18 (a) and 20 (a), which will be described later.

Thereafter, the control unit 4 turns on the switching element _{G pu,} as shown in FIG. 14 (c), but to turn off the switching element _{G pw} as shown in FIG. 14 (d), even reactor current _{i L} during Continue to charge the snubber capacitor. Incidentally, in this way when the voltage v _MC exceeds the threshold voltage of the protection circuit 35 shown in FIG. 1 while the snubber capacitor is being charged, that some of the reactor current i _L commutates to the protection circuit 35 become. This causes a loss of the protection circuit.

In the example of FIG. 14, _{after the switch element GPw} is turned off, the sign of the reactor current i _{L becomes negative.} Then, the control unit 4, as shown in FIG. 14 (e) is the turning on the switching element G _{Stay up-,} snubber capacitor of the bidirectional switch element S _up is discharged. As a result, the turn-on of the switching element _{G up} becomes hard switching, a spike voltage is generated in the voltage _{v MC} and voltage _{v INV.} Specific examples of this spike voltage are also shown in FIGS. 18 (a) and 20 (a), which will be described later.

Thus, in the switching from the first mode MODE1 according to the example of FIG. 5 to the second mode MODE2, since the sign of the reactor current i _L is unstable, to perform the commutation operation by a current commutation When a flow path loss of the reactor current i _L, i.e. load open may occur. When the load is released, as described above, the voltage v _MC drops during the commutation operation, and a spike voltage is generated in _{the voltage v MC} and the voltage v _INV immediately after the commutation ends. Further, with the occurrence of load release, there is a problem that the total harmonic distortion (THD: Total Harmonic Distortion) of _{the input currents ieu} , _iev , and _{iev increases.} These points are the same for the switching from the fourth mode MODE4 to the fifth mode MODE5 according to the example of FIG.

Therefore, in the control device 4 according to the present embodiment, the load release occurs in the switching from the first mode MODE1 to the second mode MODE2 and the switching from the fourth mode MODE4 to the fifth mode MODE5. In order to prevent and prevent the occurrence of a power short circuit, the current commutation and the voltage commutation are used in combination in the control cycle T of the matrix converter 10. Hereinafter, it will be described in detail.

FIG. 15 is a diagram showing a commutation method used by the control device 4 according to the present embodiment when switching modes. As shown in the figure, the control device 4 executes switching from the first mode MODE1 to the second mode MODE2 and switching from the fourth mode MODE4 to the fifth mode MODE5 by voltage type commutation. Then, switching from the second mode MODE2 to the third mode MODE3, switching from the third mode MODE3 to the fourth mode MODE4, switching from the fifth mode MODE5 to the sixth mode MODE6, and the third mode The switching from the mode MODE 6 of the sixth mode to the seventh mode MODE 7 is configured to be executed by the current type commutation.

The voltage-commutation, as described with reference to FIG. 13, the path of the reactor current i _L regardless of the orientation of the reactor current i _L commutation is performed is ensured. That is, when each switching element is in the state of FIG. 13 (c), the reactor current i _L can flow in either the positive and negative directions. Therefore, by configuring the control device 4 so as to perform the commutation operation by the method shown in FIG. 15, it is possible to prevent the occurrence of load release.

Here, as described above, the voltage type commutation has a problem that if the magnitude determination of the input phase voltage is erroneous, a power short circuit is caused. However, as long as the voltage type commutation is used only for switching from the first mode MODE1 to the second mode MODE2 and switching from the fourth mode MODE4 to the fifth mode MODE5, the control device 4 inputs. There is no mistake in determining the magnitude of the phase voltage. Therefore, even if the control device 4 is configured to perform the commutation operation by the method shown in FIG. 15, the power supply short circuit due to the commutation does not occur.

As described above, according to the control apparatus 4 according to this embodiment, the unstable sign of reactor current i _L, using the current-commutation and the load opening is likely from the first mode the second generation Since the switching to the mode and the switching from the fourth mode to the fifth mode are performed by the voltage type commutation, and the other switching that does not cause such a problem is performed by the current type current, the current type commutation is performed. It is possible to prevent both power short circuit and load release without switching the MC operation mode between current and voltage type commutation.

Next, the description will be focused on the time of regeneration. In the following, we will mainly explain the differences from the time of power running.

16 and 17, respectively, a signal waveform diagram showing schematically a power running of the waveform of the voltage _{v MC, nv INV} and reactor current _{i L.} Figure 16 _{(specifically, nV dc / e M = 0.25} ) is the value of the voltage _{V dc} relatively small example shown in FIG. 1 shows the, in FIG. 17 are shown in FIG. 1 _{(specifically, nV dc / e M = 0.5} ) the value of the voltage _{V dc} relatively large example shows. The signal waveforms shown in FIGS. 5 and 6 _{are the point where the voltage nv INV} precedes the voltage v _MC (that is, the point where the sign of the phase difference δ is opposite), and the first in the second mode MODE2. voltage _{e m} that is applied, voltage _{e M} during the third mode MODE3 first node _{n 1} and the second node _{n 2} between the node _{n 1} of the second node _{n 2} point but applied, the point where the voltage -e _m is applied between the fifth mode MODE5 first node _{n 1} and the second node _{n 2,} the first node n in the sixth mode MODE6 voltage -e _M is different in that it is applied between the ₁ and the second node n _2. Each bidirectional switch element _S up for applying these _{voltages, S vp, S wp, S} un, S vn, specific control method for _{S wn} are the same as in the case of power running.

In the example of FIG. 16, when switching from the third mode MODE3 to the fourth mode MODE4 and when switching from the sixth mode MODE6 to the seventh mode MODE7, the first in the example of FIG. code switching time as well as reactor current _{i L} becomes unstable from modes MODE1 to the second mode MODE2. Therefore, if it performed by a current commutation commutation operation when these switching flow path loss of the reactor current i _L, i.e. load open may occur.

Therefore, as shown in FIG. 16, the control device 4 according to the present embodiment switches from the third mode MODE3 to the fourth mode MODE4 and from the sixth mode MODE6 to the seventh mode MODE7. Executed by voltage type commutation, switching from the first mode MODE1 to the second mode MODE2, switching from the second mode MODE2 to the third mode MODE3, and switching from the fourth mode MODE4 to the fifth mode MODE5. And switching from the fifth mode MODE5 to the sixth mode MODE6 is configured to be executed by current type commutation. This makes it possible to prevent both power short circuit and load release without switching the operation mode of the MC between the current type commutation and the voltage type commutation, as in the case of power running.

Hereinafter, the effects produced by this embodiment will be described with reference to the results of performing a commutation operation using the power conversion device 1 satisfying the conditions shown in Table 1 below. Hereinafter, the first and second embodiments and the first and second comparative examples will be described, but in the first and second embodiments, the commutation operation of the matrix converter 10 is based on the example shown in FIG. The first and second comparative examples are examples in which all the commutation operations of the matrix converter 10 are performed by current type commutation. Further, in the first embodiment and the first comparative example, the voltage V _dc = 200 V and the power command value P ^* = 2 kW, and in the second embodiment and the second comparative example, the voltage V _dc = 400 V and the power command value. P ^* = 3 kW.

FIG. 18 (a), the voltage _v MC of power running of the first comparative _{example, v INV,} a signal waveform diagram showing the measurement results of the reactor current _{i L,} and the input current _{i eu,} FIG. 18 (b) _{, Is} a signal waveform diagram showing the measurement results of _{the voltage v MC} , v _INV , the reactor current i _L , and the input current i eu at the time of power running according to the first embodiment. Further, FIG. 19 (a) is a diagram showing the measurement results of efficiency [%] in each of the first comparative example and the first embodiment, and FIG. 19 (b) is a diagram showing the first comparison. It is a figure which shows the THD [%] of the input current in each of an example and 1st Example. The horizontal axis of FIGS. 18 (a) and 18 (b) is the time [ms or μs], and the horizontal axis of FIGS. 19 (a) and 19 (b) is the input power (= V _dc × I _dc ) [W]. It has become. 19 (a) and 19 (b) also include data at the time of regeneration (input power <0).

As shown in FIG. 18A, in the first comparative example, the voltage v _MC once drops significantly immediately before switching from the first mode MODE1 to the second mode MODE2. This is nothing but a decrease in the voltage v _MC due to the charging of the snubber capacitor as described above. _{For the same reason, the voltage v MC} once rises significantly immediately before switching from the fourth mode MODE4 to the fifth mode MODE5.

_{Further, in the first comparative example, the voltage v MC} and the voltage v _INV temporarily greatly increase at the timing of switching from the first mode MODE1 to the second mode MODE2. This corresponds to the generation of the spike voltage described above. _{For the same reason, the voltage v MC} and the voltage v _INV temporarily drop significantly immediately before switching from the fourth mode MODE 4 to the fifth mode MODE 5.

In contrast, in the first embodiment, as shown in FIG. 18 (b), the voltage drop v _MC due to the charging of the snubber capacitor is not observed at all. Further, in the first embodiment, the generation of the spike voltage is greatly suppressed, and in particular, the voltage v _INV is not observed at all. Therefore, according to the control device 4 of this embodiment, it can be said the drop in voltage v _MC due to the charging of the snubber capacitor, and is the generation of spike voltage in the voltage v _MC and voltage v _INV is suppressed.

Further, as shown in FIGS. 19 (a) and 19 (a), in the first comparative example, only in the case of power running corresponding to the positive input power, the vicinity of the specific input power (specifically, the vicinity of 2 kW). Then, the decrease in the total efficiency and the increase in the THD of _{the input currents ieu} , _iev , and _{iew are remarkably shown.} The increase in THD of the input currents i _eu , i _ev , and i _ew also appears in the waveform of _{the input current i u in FIG.} It should be noted that the reason why these phenomena do not appear at the time of regeneration corresponding to the negative input power is that the voltage v _{MC does not decrease as in the case of power running.}

On the other hand, in the first embodiment, as shown in FIGS. 19 (a) and 19 (b), the decrease in total efficiency and the input currents i _eu , i _ev , i _{ew as in the first comparative example.} No increase in THD has been observed. _{In addition, the THD of the input currents i eu} , i _ev , and i _ew is generally improved as compared with the first comparative example in both the power running and the regeneration. The improvement of THD at the time of regeneration is mainly due to the reduction of the protection circuit loss and the inverter loss by suppressing the load release. Therefore, according to the control device 4 according to the present embodiment, it can be said that the decrease in the total efficiency and the increase in the THD of _{the input currents ieu} , _iev , and _{iew are suppressed.}

20 (a) is a voltage _v MC of power running of the second comparative _{example, v INV,} a signal waveform diagram showing the measurement results of the reactor current _{i L,} and the input current _{i eu,} FIG. 20 (b) _{, Is} a signal waveform diagram showing the measurement results of _{the voltage v MC} , v _INV , the reactor current i _L , and the input current i eu at the time of power running according to the second embodiment. 21 (a) is a diagram showing the measurement results of efficiency [%] in each of the second comparative example and the second embodiment, and FIG. 21 (b) is a diagram showing the second comparison. It is a figure which shows the THD [%] of the input current in each of an example and 2nd Example. The horizontal axis of FIGS. 20 (a) and 20 (b) is the time [ms or μs], and the horizontal axis of FIGS. 21 (a) and 21 (b) is the input power (= V _dc × I _dc ) [W]. It has become. 21 (a) and 21 (b) also include data at the time of regeneration (input power <0).

As shown in FIG. 20A, even in the second comparative example, the amount of decrease is smaller than that in the first comparative example, but immediately before switching from the first mode MODE1 to the second mode MODE2. The voltage v _MC has dropped once. Further, although the spike voltage of the _{voltage v INV} is not generated, a large spike voltage is generated in the _{voltage v MC.}

In contrast, in the second embodiment, as shown in FIG. 20 (b), the voltage drop v _MC due to the charging of the snubber capacitor is not observed at all. In addition, the spike voltage of the _{voltage v MC is also slightly smaller.} Therefore, according to the control device 4 of this embodiment, it can be said the drop in voltage v _MC due to the charging of the snubber capacitor, and is the generation of spike voltage in the voltage v _MC is suppressed.

Further, as shown in FIGS. 21 (a) and 21 (b), the total efficiency is almost the same between the second comparative example and the second embodiment, but in the second comparative example, the positive input power is applied. only when the corresponding power running in the vicinity of the first comparative example as well as certain input power (specifically 2kW vicinity), the input current _{i eu, i ev, i ew} of THD is compared with the second embodiment It has become significantly larger. _{Further, in the second embodiment, the THD of the input currents i-eu} , i- _ev , and i- _ew is generally improved as compared with the second comparative example in both the power running time and the regeneration time. Therefore, according to the control device 4 according to the present embodiment, it can be said that the increase in THD of _{the input currents ieu} , _iev , and _{iew is suppressed.}

Although the preferred embodiments of the present invention have been described above, the present invention is not limited to these embodiments, and the present invention can be implemented in various embodiments without departing from the gist thereof. Of course.

1 Power converter 2 System power supply 3 Load 4 Controller 10 Three-phase single-phase matrix converter 20 Transformer 20a, 20b Coil 30 AC / DC converter 35 Protection circuit C1 Capacitor Cf Input capacitor G _{1 to} G _{4 One-} way switch element G _pu , _{G up, G pv, G vp} , G pw, G wp uni-directional switch element _{G nu, G un, G nv} , G vn, G nw, G wn unidirectional switching element _{S up, S vp, S wp} , S un _{, S} vn, _{S wn} bidirectional switch element L reactor Lf AC reactor MODE1~MODE7 first to seventh modes Rf damping resistor

Claims (7)

A transformer with first and second coils that are magnetically coupled to each other,
A first bidirectional switch element in which one end is connected to the first phase of the three-phase alternating current and the other end is connected to the first node constituting the one end of the first coil, and one end is the first of the three-phase alternating current. A second bidirectional switch element connected to two phases, the other end connected to the first node, one end connected to the third phase of the three-phase alternating current, and the other end connected to the first node. A third bidirectional switch element, one end connected to the first phase and the other end connected to a second node constituting the other end of the first coil. A fifth bidirectional switch element with one end connected to the second phase and the other end connected to the second node, and one end connected to the third phase and the other end connected to the second node. A control device for a power converter having a matrix converter with a sixth bidirectional switch element connected to.
A first mode in which the third and sixth bidirectional switch elements are on when the first phase is larger than the second phase and the second phase is larger than the third phase, the first. And a second mode in which the sixth bidirectional switch element is on, a third mode in which the second and sixth bidirectional switch elements are on, the third and sixth bidirectional switches. A fourth mode in which the element is on, a fifth mode in which the third and fourth bidirectional switch elements are on, and a sixth mode in which the third and fifth bidirectional switch elements are on. It is configured to repeatedly control the matrix converter in the order of the mode, the seventh mode in which the third and sixth bidirectional switch elements are on.
Switching from the first mode to the second mode and switching from the fourth mode to the fifth mode are executed by voltage-type commutation.
Switching from the second mode to the third mode, switching from the third mode to the fourth mode, switching from the fifth mode to the sixth mode, and the sixth mode. The switching from the mode to the seventh mode is executed by the current type commutation.
Control device. A first mode in which the third and sixth bidirectional switch elements are on when the first phase is larger than the second phase and the second phase is larger than the third phase, the first. And a second mode in which the sixth bidirectional switch element is on, a third mode in which the second and sixth bidirectional switch elements are on, the third and sixth bidirectional switches. A fourth mode in which the element is on, a fifth mode in which the third and fourth bidirectional switch elements are on, and a sixth mode in which the third and fifth bidirectional switch elements are on. By repeatedly controlling the matrix converter in the order of the mode, the seventh mode in which the third and sixth bidirectional switch elements are on, the power conversion device is made to perform a power running operation.
The control device according to claim 1. A transformer with first and second coils that are magnetically coupled to each other,
A first bidirectional switch element in which one end is connected to the first phase of the three-phase alternating current and the other end is connected to the first node constituting the one end of the first coil, and one end is the first of the three-phase alternating current. A second bidirectional switch element connected to two phases, the other end connected to the first node, one end connected to the third phase of the three-phase alternating current, and the other end connected to the first node. A third bidirectional switch element, one end connected to the first phase and the other end connected to a second node constituting the other end of the first coil. A fifth bidirectional switch element with one end connected to the second phase and the other end connected to the second node, and one end connected to the third phase and the other end connected to the second node. A control device for a power converter having a matrix converter with a sixth bidirectional switch element connected to.
A first mode in which the third and sixth bidirectional switch elements are on when the first phase is larger than the second phase and the second phase is larger than the third phase, the second. And a second mode in which the sixth bidirectional switch element is on, a third mode in which the first and sixth bidirectional switch elements are on, the third and sixth bidirectional switches. A fourth mode in which the element is on, a fifth mode in which the third and fifth bidirectional switch elements are on, and a sixth mode in which the third and fourth bidirectional switch elements are on. It is configured to repeatedly control the matrix converter in the order of the mode, the seventh mode in which the third and sixth bidirectional switch elements are on.
Switching from the third mode to the fourth mode and switching from the sixth mode to the seventh mode are executed by voltage-type commutation.
Switching from the first mode to the second mode, switching from the second mode to the third mode, switching from the fourth mode to the fifth mode, and the fifth mode. The switching from the mode to the sixth mode is executed by the current type commutation.
Control device. A first mode in which the third and sixth bidirectional switch elements are on when the first phase is larger than the second phase and the second phase is larger than the third phase, the second. And a second mode in which the sixth bidirectional switch element is on, a third mode in which the first and sixth bidirectional switch elements are on, the third and sixth bidirectional switches. A fourth mode in which the element is on, a fifth mode in which the third and fifth bidirectional switch elements are on, and a sixth mode in which the third and fourth bidirectional switch elements are on. By repeatedly controlling the matrix converter in the order of the mode, the seventh mode in which the third and sixth bidirectional switch elements are on, the power conversion device is made to perform a regenerative operation.
The control device according to claim 3. The first to sixth bidirectional switch elements are serially composed of first and second semiconductor elements connected in series from the three-phase AC side and the first and second semiconductor elements, respectively. The snubber capacitor connected in parallel with the circuit, the first diode connected in parallel with the first semiconductor element in the direction in which the anode is connected to the connection point of the first and second semiconductor elements, and the said. It has a second diode connected in parallel with the second semiconductor element in a direction in which the anode is connected to the connection points of the first and second semiconductor elements.
The voltage type commutation
Both one end of the commutation destination bidirectional switch element that is the commutation destination of the first to sixth bidirectional switch elements and the commutation source that is the commutation source of the first to sixth bidirectional switch elements. Of the first and second semiconductor elements of the commutation destination bidirectional switch element, of the commutation destination bidirectional switch element, depending on the sign of the input phase voltage applied between one end of the commutation switch element. A step of turning on one end without short-circuiting between one end and one end of the commutation source bidirectional switch element.
A step of turning off one of the first and second semiconductor elements of the commutation source bidirectional switch element, which can secure the path of the output phase current flowing through the reactor even if it is turned off, regardless of the direction of the output phase current. When,
A step of turning on the other of the first and second semiconductor elements of the commutation destination bidirectional switch element, and
A step of turning off the other of the first and second semiconductor elements of the commutation source bidirectional switch element, and
Performed by,
The control device according to any one of claims 1 to 4. The current type commutation
A step of turning off one of the first and second semiconductor elements of the commutation source bidirectional switch element, which can secure the path of the output phase current even if it is turned off, according to the direction of the output phase current.
Of the first and second semiconductor elements of the commutation destination bidirectional switch element, one of the first and second semiconductor elements of the commutation destination bidirectional switch element, which newly constitutes the path of the output phase current by turning on, is used according to the direction of the output phase current. Steps to turn on and
A step of turning off the other of the first and second semiconductor elements of the commutation source bidirectional switch element, and
A step of turning on the other of the first and second semiconductor elements of the commutation destination bidirectional switch element, and
Performed by,
The control device according to claim 5. The power converter is
A first one-way switch element, one end connected to a fifth node constituting one end of a DC power supply, the other end connected to a third node constituting one end of the second coil, and one end connected to the DC. A second one-way switch element connected to a sixth node constituting the other end of the power supply, the other end connected to the third node, one end connected to the fifth node, and the other end connected to the fifth node. A third one-way switch element connected to a fourth node constituting the other end of the second coil, one end connected to the sixth node, and the other end connected to the fourth node. An AC / DC converter with a fourth one-way switch element,
A reactor inserted between the first node and the first coil,
Have,
The control device according to any one of claims 1 to 6.

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