JP2021093901A - Motor control device - Google Patents

Abstract

To provide a control device that controls the drive of an electric motor by vector control, and can suppress the deterioration of the controllability of the drive of the electric motor when the rotation speed and the modulation rate of a rotor of the electric motor are relatively large.SOLUTION: A control device 1 includes an inverter circuit 2 that drives a rotor of an electric motor M on the basis of the comparison result between a carrier wave and a voltage command value, and a control circuit 3 that obtains the voltage command value V* by vector control using currents Iu, Iv, and Iw flowing through the electric motor M and the rotation speed ω^ and position θ^ of the rotor for each control cycle or the position θ, and the control circuit 3 reduces the control cycle as the rotation speed ω^ of the rotor or the modulation rate corresponding to the rotation speed ω^ increases.SELECTED DRAWING: Figure 1

Description

The present invention relates to a control device for an electric motor.

As a control device for the motor, the position of the rotor of the motor is used to convert the three-phase alternating current flowing through the motor into d-axis current and q-axis current, and the d-axis current and q-axis current approach the current command value. As described above, there is a method in which a voltage command value is obtained and the drive of the motor is controlled by a drive signal according to a comparison result between the voltage command value and a carrier, that is, a so-called vector control is used to control the drive of the motor. Patent Document 1 is a related technique.

Japanese Unexamined Patent Publication No. 2001-169590

However, in the above control device, when the rotation speed (rotation speed) of the rotor or the modulation factor according to the rotation speed of the rotor becomes relatively large, the duty ratio of the drive signal does not change according to the voltage command value. , There is a risk that the controllability of the drive of the electric motor will deteriorate.

An object according to one aspect of the present invention is that in a control device that controls the drive of an electric motor by vector control, the controllability of the drive of the electric motor is lowered when the rotation speed and the modulation factor of the rotor of the electric motor are relatively large. It is to suppress.

The motor control device according to the present invention includes an inverter circuit that drives the rotor of the motor based on a comparison result between a voltage command value and a carrier, a current flowing through the motor for each control cycle, and a rotor. It is provided with a control circuit for obtaining a voltage command value by vector control using the rotation speed and position of.

The control circuit reduces the control cycle as the rotation speed of the rotor or the modulation factor according to the rotation speed of the rotor increases.

As a result, even if the rotation speed of the rotor of the motor or the modulation factor becomes relatively large, it is possible to suppress that the duty ratio of the drive signal does not change according to the voltage command value, so that the control of the motor can be controlled. It is possible to suppress the deterioration of the sex.

Further, the control circuit may be configured to estimate the rotation speed and position of the rotor by using the current flowing through the motor for each control cycle.

Further, the motor control device according to the present invention has an inverter circuit that drives the rotor of the motor based on the result of comparison between the voltage command value and the carrier, the current flowing through the motor, and the rotation speed of the rotor. And a control circuit that obtains the voltage command value by vector control using the position.

As the rotation speed or modulation factor increases, the control circuit determines the control cycle of the acquisition process for acquiring the current flowing through the motor and the estimation process for estimating the position using the acquired current among all the processes of the control circuit. In addition to making it smaller, the control cycle of processes other than the acquisition process and the estimation process may be made constant.

As a result, the number of samplings of the current flowing through the motor can be increased as the rotation speed or the modulation factor increases, and the position estimation accuracy can be improved. Therefore, by calculating the voltage command value using that position, it is possible to suppress a decrease in the controllability of the motor. Further, since the control cycle of a part of all the processes of the control circuit is reduced and the control cycle of the other processes is made constant, the processing load of the control circuit can be suppressed.

According to the present invention, in a control device that controls the drive of an electric motor by vector control, it is possible to suppress a decrease in the controllability of the drive of the electric motor when the rotation speed and the modulation rate of the rotor of the electric motor are relatively large. it can.

It is a figure which shows an example of the control device of the electric motor of 1st Embodiment. It is a figure which shows another example of the control device of the electric motor of 1st Embodiment. It is a figure which shows an example of a carrier wave, a voltage command value, and a drive signal.

<First Embodiment>
Hereinafter, embodiments will be described in detail based on the drawings.

FIG. 1 is a diagram showing an example of a control device for an electric motor according to the first embodiment.

The control device 1 shown in FIG. 1 controls the drive of the electric motor M mounted on a vehicle such as an electric forklift or a plug-in hybrid vehicle, and includes an inverter circuit 2, a control circuit 3, and a current sensor Se1. ~ Se3 is provided.

The inverter circuit 2 drives the motor M by the DC power supplied from the DC power supply P, and includes a capacitor C and switching elements SW1 to SW6 (for example, an IGBT (Insulated Gate Bipolar Transistor)). That is, one end of the capacitor C is connected to the positive electrode terminal of the DC power supply P and each collector terminal of the switching elements SW1, SW3, SW5, and the other end of the capacitor C is the negative electrode terminal of the DC power supply P and the switching elements SW2, SW4, SW6. It is connected to each emitter terminal of. The connection point between the emitter terminal of the switching element SW1 and the collector terminal of the switching element SW2 is connected to the U-phase input terminal of the motor M via the current sensor Se1. The connection point between the emitter terminal of the switching element SW3 and the collector terminal of the switching element SW4 is connected to the V-phase input terminal of the motor M via the current sensor Se2. The connection point between the emitter terminal of the switching element SW5 and the collector terminal of the switching element SW6 is connected to the W phase input terminal of the motor M via the current sensor Se3.

The capacitor C smoothes the voltage output from the DC power supply P and input to the inverter circuit 2.

The switching element SW1 is turned on or off based on the drive signal S1 output from the control circuit 3. The switching element SW2 is turned on or off based on the drive signal S2 output from the control circuit 3. The switching element SW3 is turned on or off based on the drive signal S3 output from the control circuit 3. The switching element SW4 is turned on or off based on the drive signal S4 output from the control circuit 3. The switching element SW5 is turned on or off based on the drive signal S5 output from the control circuit 3. The switching element SW6 is turned on or off based on the drive signal S6 output from the control circuit 3. When the switching elements SW1 to SW6 are turned on or off, the DC power output from the DC power supply P is converted into three AC powers whose phases are 120 degrees different from each other, and the AC powers are the U phase of the motor M. The rotor of the motor M is rotated by being input to the input terminals of the V phase and the W phase.

The current sensors Se1 to Se3 are composed of a Hall element, a shunt resistor, and the like. The current sensor Se1 detects the current Iu flowing in the U phase of the motor M and outputs it to the control circuit 3, and the current sensor Se2 detects the current Iv flowing in the V phase of the motor M and outputs it to the control circuit 3 to output the current sensor. Se3 detects the current Iw flowing in the W phase of the motor M and outputs it to the control circuit 3.

The control circuit 3 includes a drive circuit 4 and a calculation unit 5.

The drive circuit 4 is composed of an IC (Integrated Circuit) or the like, and has voltage command values Vu *, Vv *, Vw * and a carrier wave (triangle wave, sawtooth wave, or reverse sawtooth wave) output from the calculation unit 5 for each control cycle. Etc.), and the drive signals S1 to S6 according to the comparison result are output to the respective gate terminals of the switching elements SW1 to SW6. For example, when the voltage command value Vu * is equal to or higher than the carrier wave, the drive circuit 4 outputs a high-level drive signal S1 and outputs a low-level drive signal S2, and the voltage command value Vu * is smaller than the carrier wave. , The low-level drive signal S1 is output, and the high-level drive signal S2 is output. Further, the drive circuit 4 outputs a high-level drive signal S3 and a low-level drive signal S4 when the voltage command value Vv * is equal to or higher than the carrier wave, and the voltage command value Vv * is smaller than the carrier wave. , The low-level drive signal S3 is output, and the high-level drive signal S4 is output. Further, the drive circuit 4 outputs a high-level drive signal S5 and a low-level drive signal S6 when the voltage command value Vw * is equal to or higher than the carrier wave, and the voltage command value Vw * is smaller than the carrier wave. , The low-level drive signal S5 is output, and the high-level drive signal S6 is output.

In the drive circuit 4, when the amplitude values of the voltage command values Vu *, Vv *, and Vw * are smaller than the amplitude value of the carrier wave, the switching elements SW1 to SW6 are used in one cycle of the voltage command values Vu *, Vv *, and Vw *. Is repeatedly turned on and off (PWM (Pulse Width Modulation) control).

Further, when the amplitude value of the voltage command values Vu *, Vv *, and Vw * is larger than the amplitude value of the carrier wave, the drive circuit 4 is a part of one cycle of the voltage command values Vu *, Vv *, and Vw *. It is assumed that the switching elements SW1 to SW6 are repeatedly turned on and off during the period, and the switching elements SW1 to SW6 are constantly turned on or off during the remaining period (overmodulation control).

Further, in the drive circuit 4, when the amplitude values of the voltage command values Vu *, Vv *, and Vw * are further larger than the amplitude values of the carrier wave, the switching elements SW1 to the switching elements SW1 in a half cycle of the voltage command values Vu *, Vv *, and Vw *. It is assumed that the SW6 is always on or always off, and the switching elements SW1 to SW6 are always on or always off in the remaining half cycle (square wave control).

Further, when the voltage command values Vu *, Vv *, and Vw * are not particularly distinguished, the voltage command value V * is simply used. Further, when the drive signals S1 to S6 are not particularly distinguished, it is simply referred to as the drive signal S.

The calculation unit 5 is composed of a microcomputer or the like, and includes an estimation unit 6, a subtraction unit 7, a speed control unit 8, subtraction units 9, 10 and a current control unit 11, a coordinate conversion unit 12, and a coordinate conversion unit. 13 and. For example, by executing a program stored in a storage unit (not shown), the microcomputer executes an estimation unit 6, a subtraction unit 7, a speed control unit 8, a subtraction unit 9, 10, a current control unit 11, and a coordinate conversion unit 12. , And the coordinate conversion unit 13 is realized.

The estimation unit 6 has the d-axis voltage command value Vd * and q-axis voltage command value Vq * output from the current control unit 11 and the d-axis current Id and q-axis current output from the coordinate conversion unit 13 for each control cycle. Using Iq, the rotation speed (rotation speed) ω ^ and the position θ ^ of the rotor of the motor M are estimated.

For example, the estimation unit 6 calculates the counter electromotive force ed ^ and the counter electromotive force eq ^ by the following equations 1 and 2. Note that R indicates the resistance of the motor M, and L indicates the inductance of the coil possessed by the transmitter M.

ed ^ = Vd * -R × Id + ω ^ × L × Id ・ ・ ・ Equation 1
eq ^ = Vq * -R x Iq-ω ^ x L x Iq ... Equation 2

Next, the estimation unit 6 calculates the error θe ^ by the following equation 3.

θe ^ = tan ^-1 (ed ^ / eq ^) ・ ・ ・ Equation 3

Next, the estimation unit 6 obtains the rotation speed ω ^ such that the error θe ^ becomes zero in the following equation 4. Kp indicates the constant of the proportional term of PI (Proportional Integral) control, and Ki indicates the constant of the integral term of PI control.

ω ^ = Kp × θe ^ + Ki × ∫ (θe ^) dt ・ ・ ・ Equation 4

Then, the estimation unit 6 calculates the position θ ^ by the following equation 5. Note that s indicates a Laplace operator.

θ ^ = (1 / s) × ω ^ ・ ・ ・ Equation 5

The subtraction unit 7 calculates the difference Δω between the rotation speed command value ω * input from the outside and the rotation speed ω ^ output from the estimation unit 6 for each control cycle.

The speed control unit 8 converts the difference Δω output from the subtraction unit 7 into the q-axis current command value Iq * for each control cycle.

For example, the speed control unit 8 obtains the q-axis current command value Iq * such that the difference Δω becomes zero in the following equation 6.

IQ * = Kp × Δω ＋ Ki × ∫ (Δω) dt ・ ・ ・ Equation 6

The subtraction unit 9 calculates the difference ΔId between the predetermined d-axis current command value Id * and the d-axis current Id output from the coordinate conversion unit 13 for each control cycle.

The subtraction unit 10 calculates the difference ΔIq between the q-axis current command value Iq * output from the speed control unit 8 and the q-axis current Iq output from the coordinate conversion unit 13 for each control cycle.

The current control unit 11 converts the difference ΔId output from the subtraction unit 9 and the difference ΔIq output from the subtraction unit 10 into the d-axis voltage command value Vd * and the q-axis voltage command value Vq * for each control cycle. ..

For example, the current control unit 11 calculates the d-axis voltage command value Vd * using the following formula 7, and calculates the q-axis voltage command value Vq * using the following formula 8. Lq indicates the q-axis inductance of the coil of the motor M, Ld indicates the d-axis inductance of the coil of the motor M, and Ke indicates the induced voltage constant.

Vd * = Kp × ΔId + Ki × ∫ (ΔId) dt−ωLqIq ・ ・ ・ Equation 7
Vq * = Kp × ΔIq + Ki × ∫ (ΔIq) dt + ωLdId + ωKe ・ ・ ・ Equation 8

The coordinate conversion unit 12 sets the d-axis voltage command value Vd * and the q-axis voltage command value Vq * to the voltage command value Vv * and the voltage command by using the position θ ^ output from the estimation unit 6 for each control cycle. Convert to the value Vv * and the voltage command value Vw *.

For example, the coordinate conversion unit 12 uses the transformation matrix C1 shown in the following equation 9 to convert the d-axis voltage command value Vd * and the q-axis voltage command value Vq * into the voltage command value Vu *, the voltage command value Vv *, and the voltage command value Vv *. Convert to voltage command value Vw *.

For example, the coordinate conversion unit 12 sets the calculation result of the following equation 10 as the phase angle δ.

δ = tan ^-1 (−Vq * / Vd *) ・ ・ ・ Equation 10

Next, the coordinate conversion unit 12 sets the addition result of the phase angle δ and the position θ ^ as the target position θv.

Next, the coordinate conversion unit 12 sets the calculation result of the following equation 11 as the modulation factor'. It should be noted that 0 <modulation rate ′ <1. In _{addition, V in} is the voltage of the DC power supply P.

Next, the coordinate conversion unit 12 uses the calculation result of the following equation 12 as the modulation factor. It should be noted that -1 <modulation rate <1.

Modulation rate = 2 x modulation rate ′ -1 ・ ・ ・ Equation 12

Then, the coordinate conversion unit 12 provides information indicating the correspondence between the target position θv and the voltage command value Vu *, the voltage command value Vv *, and the voltage command value Vw *, which are stored in advance in a storage unit (not shown). The voltage command value Vu *, the voltage command value Vv *, and the voltage command value Vw * corresponding to the target position θv are obtained with reference to.

The coordinate conversion unit 13 uses the position θ ^ output from the estimation unit 6 for each control cycle to convert the currents Iu, Iv, and Iw detected by the current sensors Se1 to Se3 into the d-axis currents Id and the q-axis currents. Convert to Iq.

For example, the coordinate conversion unit 13 converts the currents Iu, Iv, and Iw into the d-axis current Id and the q-axis current Iq by using the transformation matrix C2 shown in the following equation 13.

FIG. 2 is a diagram showing another example of the control device 1 of the motor M of the first embodiment. The same components as those shown in FIG. 1 are designated by the same reference numerals, and the description thereof will be omitted.

The control device 1 shown in FIG. 2 differs from the control device 1 shown in FIG. 1 in that the position detection unit Sp (which detects the position θ of the rotor of the motor M and outputs the detected position θ to the control circuit 3). It is equipped with a resolver, etc.).

Further, the control device 1 shown in FIG. 2 is different from the control device 1 shown in FIG. 1 in that the calculation unit 5'is provided instead of the calculation unit 5.

The calculation unit 5'is composed of a microcomputer or the like, and includes an estimation unit 6', a subtraction unit 7, a speed control unit 8, subtraction units 9, 10 and a current control unit 11 and a coordinate conversion unit 12'. It is provided with a coordinate conversion unit 13'. For example, by executing a program stored in a storage unit (not shown), the microcomputer executes an estimation unit 6', a subtraction unit 7, a speed control unit 8, a subtraction unit 9, 10, a current control unit 11, and a coordinate conversion unit. 12'and the coordinate conversion unit 13'are realized.

The estimation unit 6'estimates the rotation speed ω ^ of the rotor of the motor M by using the position θ detected by the position detection unit Sp for each control cycle.

For example, the estimation unit 6'estimates the rotation speed ω ^ by dividing the position θ by the control cycle of the control circuit 3.

Further, the coordinate conversion unit 12'sets the d-axis voltage command value Vd * and the q-axis voltage command value Vq * by using the position θ detected by the position detection unit Sp for each control cycle, and sets the voltage command value Vu *. , Voltage command value Vu *, and voltage command value Vw *.

For example, the coordinate conversion unit 12'uses the transformation matrix C1 shown in the above equation 9 to convert the d-axis voltage command value Vd * and the q-axis voltage command value Vq * into the voltage command value Vu *, the voltage command value Vv *, and the voltage command value Vv *. And convert to the voltage command value Vw *. In the above equation 9, the position θ ^ is replaced with the position θ.

For example, the coordinate conversion unit 12'sets the d-axis voltage command value Vd * and the q-axis voltage command value Vq * by using the information stored in advance in the above equations 10 to 12 and the storage unit (not shown). It is converted into a command value Vu *, a voltage command value Vv *, and a voltage command value Vw *. When obtaining the target position θv, the position θ ^ is replaced with the position θ.

The coordinate conversion unit 13'uses the position θ detected by the position detection unit Sp for each control cycle to convert the currents Iu, Iv, and Iw detected by the current sensors Se1 to Se3 into the d-axis currents Id and q-axis. Convert to current Iq.

For example, the coordinate conversion unit 13 converts the currents Iu, Iv, and Iw into the d-axis current Id and the q-axis current Iq by using the transformation matrix C2 shown in the above equation 13. In the above equation 13, the position θ ^ is replaced with the position θ.

In the control circuit 3 shown in FIG. 1 or 2, when the rotation speed ω ^ is equal to or less than the threshold value ωth or the modulation factor is equal to or less than the threshold value Mth, the control cycle of the control circuit 3 is set to the control cycle T1 and the rotation speed ω ^ When is larger than the threshold value ωth or the modulation factor is larger than the threshold value Mth, the control cycle of the control circuit 3 is set to the control cycle T2 smaller than the control cycle T1. The threshold value ωth is the maximum value of the rotation speed ω ^ when the estimation accuracy of the rotation speed ω ^ is not lowered. Further, the threshold value Mth is the maximum value of the modulation factor when the estimation accuracy of the rotation speed ω ^ is not lowered.

In the control circuit 3 shown in FIG. 1 or 2, when the rotation speed ω ^ is the threshold value ωth1 or less or the modulation factor is the threshold value Mth1 or less, the control cycle of all the processes of the control circuit 3 is controlled by the control cycle T1. When the rotation speed ω ^ is larger than the threshold value ωth1 or the modulation factor is larger than the threshold value Mth1, the control cycle of all the processes of the control circuit 3 is set to the control cycle T2, and the rotation speed ω ^ is equal to or more than the threshold value ωth2. When the modulation factor is the threshold value Mth2 or more, the control cycle of all the processes of the control circuit 3 may be set to the control cycle T3. The threshold value ωth1 <threshold value ωth2. Further, the threshold value Mth1 <threshold value Mth2 is set. Further, the control cycle T1> the control cycle T2> the control cycle T3. Further, the threshold value ωth1 is set to the maximum value of the rotation speed ω ^ when the estimation accuracy of the rotation speed ω ^ is not lowered. Further, the threshold value Mth1 is set to the maximum value of the modulation factor when the estimation accuracy of the rotation speed ω ^ is not lowered. That is, the control circuit 3 shown in FIG. 1 or 2 may be configured to reduce the control cycle of all the processes of the control circuit 3 as the rotation speed ω ^ or the modulation factor increases.

3A and 3B are diagrams showing an example of a carrier wave, a voltage command value Vu *, and a drive signal S1. The horizontal axis of the two-dimensional coordinates shown in FIGS. 3A and 3B indicates the target position θv, and the vertical axis indicates the voltage. Further, it is assumed that the frequency of the voltage command value Vu * at the positions θ2 to θ5 is higher than the frequency of the voltage command value Vu * at the positions θ1 to θ2. That is, it is assumed that the rotation speed ω ^ at the positions θ1 to θ2 is equal to or less than the threshold value ωth, and the rotation speed ω ^ at the positions θ2 to θ5 is larger than the threshold value ωth. Alternatively, it is assumed that the modulation factor at positions θ1 to θ2 is equal to or less than the threshold value Mth, and the modulation factor at positions θ2 to θ5 is larger than the threshold value Mth. Further, the control cycle T1 of the control circuit 3 shown in FIG. 3A is constant at the positions θ1 to θ5. Further, in FIG. 3B, the control cycle T2 of the control circuit 3 at the positions θ2 to θ5 is smaller than the control cycle T1 of the control circuit 3 at the positions θ1 to θ2. Further, the amplitude value and frequency of the carrier wave shown in FIGS. 3 (a) and 3 (b) are constant at positions θ1 to θ5.

At positions θ1 to θ2 shown in FIG. 3A, the duty ratio of the drive signal S1 (the ratio of the high level period of the drive signal S1 to one cycle of the carrier wave) following the change in the amplitude value of the voltage command value Vu *). Is changing. That is, at the positions θ1 to θ2 shown in FIG. 3A, when the amplitude value of the voltage command value Vu * increases to the positive side, the duty ratio of the drive signal S1 increases, and the amplitude value of the voltage command value Vu * becomes negative. As it increases to the side, the duty ratio of the drive signal S1 decreases.

On the other hand, at positions θ2 to θ5 shown in FIG. 3A, the rotation speed ω ^ or the modulation factor is larger than that at positions θ1 to θ2, and the duty ratio of the drive signal S1 is the amplitude value of the voltage command value Vu *. It may not be the value according to the change. That is, in the example shown in FIG. 3A, it is desirable that the drive signal S1 is at a low level during the period of positions θ3 to θ4, but since the voltage command value Vu * is equal to or higher than the carrier wave at position θ3, positions θ3 to θ3 to The drive signal S1 is at a high level during the period of θ4. As described above, when the rotation speed ω ^ or the modulation factor becomes relatively large, the duty ratio of the drive signal S1 may not change according to the change in the amplitude value of the voltage command value Vu *.

Therefore, in the control device 1 of the first embodiment, as shown in FIG. 3B, the control cycle T2 at the positions θ2 to θ5 is made smaller than the control cycle T1 at the positions θ1 to θ2. Therefore, the number of samples taken per unit time of the currents Iu, Iv, Iw and the position θ ^ or the position θ at the positions θ2 to θ5 is the unit time of the currents Iu, Iv, Iw and the position θ ^ or the position θ at the positions θ1 to θ2. The number of comparisons between the carrier wave and the voltage command value Vu * per unit time at positions θ2 to θ5 is the number of comparisons between the carrier wave and the voltage command value Vu * per unit time at positions θ1 to θ2. More. As a result, it is possible to prevent the duty ratio of the drive signal S1 from changing according to the change in the amplitude value of the voltage command value Vu *. That is, in the example shown in FIG. 3B, it is desirable that the drive signal S1 is at a low level during the period of positions θ3 to θ4, but the drive signal S1 is at a low level at a part of positions θ3 to θ4. ..

As described above, since the control device 1 of the first embodiment has a configuration in which the control cycle is reduced as the rotation speed ω ^ or the modulation factor of the rotor of the motor M increases, the rotation speed of the rotor of the motor M Even if ω ^ or the modulation speed becomes relatively large, it is possible to suppress that the duty ratio of the drive signal S does not change according to the voltage command value V *, so that the controllability of the motor M deteriorates. It can be suppressed.

Further, in the control device 1 of the first embodiment, when the rotation speed ω ^ or the modulation factor of the rotor of the motor M is relatively small, or when the rotation speed ω ^ or the modulation factor of the rotor of the motor M is relatively large. In comparison, since the control cycle is large, the number of processes of the control circuit 3 per unit time is reduced, and the load applied to the control circuit 3 can be reduced.

<Second Embodiment>
In the control device of the second embodiment, as the rotation speed ω ^ or the modulation factor of the rotor of the motor M increases, among all the processes of the control circuit 3, the process of acquiring the current flowing through the motor M and the current thereof are performed. The control cycle of the process of estimating the position θ ^ is reduced, and the control cycle of other processes is made constant. The configuration of the control device of the second embodiment is the same as the configuration of the control device 1 shown in FIG.

That is, the coordinate conversion unit 13 acquires the currents Iu, Iv, and Iw flowing in each phase of the motor M for each first control cycle, and uses the position θ ^ output from the estimation unit 6 to obtain the current Iu. , Iv, Iw are converted into d-axis current Id and q-axis current Iq.

Further, the estimation unit 6 has a d-axis voltage command value Vd * and a q-axis voltage command value Vq * output from the current control unit 11 and a d-axis current output from the coordinate conversion unit 13 for each first control cycle. Using Id and the q-axis current Iq, the rotation speed ω ^ and position θ ^ of the rotor are estimated.

Further, the subtractor 7 calculates the difference Δω between the rotation speed command value ω * input from the outside and the rotation speed ω ^ output from the estimation unit 6 for each second control cycle.

Further, the speed control unit 8 converts the difference Δω into the q-axis current command value Iq * for each second control cycle.

Further, the subtraction unit 9 calculates the difference ΔId between the predetermined d-axis current command value Id * and the d-axis current Id output from the coordinate conversion unit 13 for each second control cycle.

Further, the subtraction unit 10 calculates the difference ΔIq between the q-axis current command value Iq * output from the speed control unit 8 and the q-axis current Iq output from the coordinate conversion unit 13 for each second control cycle. To do.

Further, the current control unit 11 converts the difference ΔId and the difference ΔIq into the d-axis voltage command value Vd * and the q-axis voltage command value Vq * for each second control cycle.

Further, the coordinate conversion unit 12 uses the position θ ^ output from the estimation unit 6 for each second control cycle to set the d-axis voltage command value Vd * and the q-axis voltage command value Vq * of the motor M. It is converted into voltage command values Vu *, Vv *, and Vw * corresponding to each phase.

Further, the drive circuit 4 compares the voltage command values Vu *, Vv *, Vw * output from the calculation unit 5 with the carrier wave for each second control cycle, and drives signals S1 to corresponding to the comparison result. S6 is output to the respective gate terminals of the switching elements SW1 to SW6.

Then, the control circuit 3 reduces the first control cycle and keeps the second control cycle constant as the rotation speed ω ^ or the modulation factor increases.

For example, when the rotation speed ω ^ is equal to or less than the threshold value ωth, the first and second control cycles are set to the control cycle T1, and when the rotation speed ω ^ is larger than the threshold value ωth, the first control cycle is set to the control cycle T2. At the same time, it is assumed that the second control cycle is left as the control cycle T1. The control cycle T2 is smaller than the control cycle T1.

In this case, when the rotation speed ω ^ is larger than the threshold value ωth, the unit time of the currents Iu, Iv, Iw flowing through the motor M (for example, the currents Iu, Iv, Since the number of samples per unit time of Iw) increases, the number of samples of the d-axis current Id and the q-axis current Iq per unit time also increases. As a result, the error included in the d-axis current Id and the q-axis current Iq can be calculated by calculating the moving average of the d-axis current Id and the q-axis current Iq using the increase in the d-axis current Id and the q-axis current Iq. Can be reduced. Therefore, as the error included in the d-axis current Id and the q-axis current Iq decreases, the estimation accuracy of the position θ ^ estimated by using the d-axis current Id and the q-axis current Iq can be improved.

As described above, in the control device of the second embodiment, as the rotation speed ω ^ or the modulation factor increases, the acquisition process for acquiring the current flowing through the motor M and the estimation for estimating the position θ ^ using the acquired current. The configuration is such that the control cycle of processing is reduced. As a result, the number of samplings of the current flowing through the motor M can be increased as the rotation speed ω ^ or the modulation factor increases, and the estimation accuracy of the position θ ^ can be improved. Therefore, since the voltage command values Vu *, Vv *, and Vw * can be calculated with high accuracy using the position θ ^, it is possible to suppress a decrease in the controllability of the motor M. That is, according to the control device of the second embodiment, the calculation accuracy of the voltage command value V * can be improved even if the rotation speed ω ^ or the modulation factor of the rotor of the motor M becomes relatively large, so that the motor It is possible to suppress a decrease in the controllability of M.

Further, in the control device of the second embodiment, in order to reduce the control cycle of some of the processes of the control circuit 3 and keep the control cycle of the other processes constant, the control circuit 3 of the control circuit 3 is used. The processing load can be suppressed.

Further, the present invention is not limited to the above embodiments, and various improvements and changes can be made without departing from the gist of the present invention.

1 Control device 2 Inverter circuit 3 Control circuit 4 Drive circuit 5, 5'Calculation unit 6, 6'Estimation unit 7 Subtraction unit 8 Speed control unit 9 Subtraction unit 10 Subtraction unit 11 Current control unit 12, 12'Coordinate conversion unit 13, 13'Coordinate conversion unit

Claims (3)

An inverter circuit that drives the rotor of a motor based on the comparison result between the carrier wave and the voltage command value,
A control circuit that obtains the voltage command value by vector control using the current flowing through the motor and the rotation speed and position of the rotor for each control cycle.
With
The control circuit is a control device for an electric motor, wherein the control cycle is reduced as the rotation speed of the rotor or the modulation factor according to the rotation speed of the rotor increases. The motor control device according to claim 1.
The control circuit is a control device for an electric motor, which estimates the rotation speed and the position of the rotor by using the current flowing through the electric motor for each control cycle. An inverter circuit that drives the rotor of a motor based on the comparison result between the carrier wave and the voltage command value,
A control circuit that obtains the voltage command value by vector control using the current flowing through the motor and the rotation speed and position of the rotor.
With
The control circuit estimates the position by using the acquisition process of acquiring the current flowing through the motor and the acquired current among all the processes of the control circuit as the rotation speed or the modulation factor increases. A control device for an electric motor, characterized in that the control cycle of the estimation process is reduced and the control cycle of the acquisition process and the processes other than the estimation process is constant.

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