JP2020031507A - Power conversion device and inverter device using the same - Google Patents

Abstract

To output voltage from full-wave rectification voltage to double voltage by a few switching elements in a power conversion device which converts three-phase AC voltage into DC voltage.SOLUTION: A power conversion device is constituted by providing two full-wave rectification circuits which perform full-wave rectification of three-phase AC voltage in parallel, connecting first and second link capacitors in series on a DC output side of one rectification circuit, connecting first and second switching elements in series on a DC output side of the other full-wave rectification circuit, and connecting a midpoint of the first and second switching elements with a midpoint of the first and second link capacitors, and is further provided with: a voltage sensor which detects voltage of the three-phase AC voltage; and a control circuit which controls the first and second switching elements so as to input a detection value of the voltage sensor to conduct a phase in which an absolute value becomes maximum.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a power conversion device that converts an AC voltage into a DC voltage, and an inverter device using the same.

  In an inverter device that receives a three-phase AC power supply and converts the power to a load such as a motor, the inverter needs a load (for example, an AC motor) after the three-phase AC voltage of the power supply is once converted to a DC voltage by a rectifier circuit. It is common to generate an alternating voltage.

  In this inverter device, the DC voltage generated by the rectifier circuit is determined by the amplitude of the three-phase AC voltage of the power supply. Therefore, the withstand voltage of the semiconductor component and the passive component constituting the inverter device is determined based on this voltage.

  By the way, the three-phase AC voltage of the power supply differs in countries and regions, for example, three-phase 200 V in Japan and three-phase 380 V in China. Therefore, when an inverter device is introduced in a plurality of countries and regions, the withstand voltage of each component is different, so that another inverter device with the same power capacity is required. For this reason, there is a problem that the required number of parts increases and costs such as parts management costs increase.

  As a solution to this problem, use of a power converter that performs three-phase voltage doubler rectification can be considered. With a power converter that performs voltage doubler rectification, for example, in the case of three-phase 200V power reception, the DC voltage of the inverter can be boosted to three-phase 400V power reception. Therefore, the inverter device can be shared for different three-phase AC voltages, and low cost of the entire device can be expected.

  As such a power converter, a circuit system described in Patent Document 1 has been proposed. In this power converter, one of three sets of bidirectional switch circuits is connected in parallel with a diode bridge (rectifier circuit), and the other is connected to the midpoint of two series-connected link capacitors (smoothing capacitors) of the inverter DC section. are doing. By switching the bidirectional switch circuit based on the three-phase AC voltage, the DC voltage can be doubled.

JP 2013-247789 A

  The device described in Patent Document 1 requires a bidirectional switch circuit. Therefore, many switching elements are required. Further, since a driver is required to drive the switching elements forming the bidirectional switch circuit, if the number of switching elements increases, the number of drivers also increases, and the configuration of the entire device becomes complicated. In general, switching elements are expensive, and using many switching elements increases the cost of the entire apparatus.

  The present invention has been made in view of the above circumstances, and has as its object to provide a power converter capable of reducing the number of switching elements and supplying a DC voltage from a three-phase AC voltage to a doubled voltage.

  In order to solve the above-described problem, according to one embodiment of the present invention, a first rectifier circuit that rectifies an AC voltage of a three-phase AC power supply into a DC voltage, and a DC voltage between the DC output side and the load side of the first rectifier circuit. A first link capacitor and a second link capacitor connected to the first rectifier circuit, a second rectifier circuit connected to the three-phase AC power supply in parallel with the first rectifier circuit, and a first switching terminal connected to a DC output side of the second rectifier circuit. A half bridge in which an element and a second switching element are connected in series, a midpoint between the first switching element and the second switching element, and a midpoint between the first link capacitor and the second link capacitor; An AC voltage sensor for detecting a three-phase AC voltage of a three-phase AC power supply; and a switch for the first switching element and the second switching element based on a detection value of the AC voltage sensor. And a control circuit for grayed control.

  According to the present invention, it is possible to provide a power converter that doubles a three-phase AC voltage with a small number of switching elements.

1 is a diagram illustrating a circuit configuration of a power conversion device according to a first embodiment of the present invention. FIG. 3 is a diagram illustrating a configuration of a control unit according to the first embodiment. FIG. 3 is a diagram illustrating a control flowchart according to the first embodiment. FIG. 3 is a diagram illustrating examples of voltage / current waveforms of the power converter according to the first embodiment. FIG. 2 is a diagram illustrating an example of a circuit configuration when a part of the power conversion device in the first embodiment has failed. FIG. 6 is a diagram illustrating an example of each voltage / current waveform of the power converter in FIG. 5. FIG. 6 is a diagram illustrating a circuit configuration of a power conversion device according to a second embodiment of the present invention. FIG. 9 is a diagram illustrating a configuration of a DC voltage control unit according to a second embodiment. FIG. 13 is a diagram illustrating an example of a rotation speed characteristic of a DC voltage command value in the second embodiment. FIG. 9 is a diagram illustrating a control flowchart according to the second embodiment. FIG. 9 is a diagram illustrating an example of each voltage / current waveform of the power converter according to the second embodiment.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings.

  FIG. 1 shows a circuit configuration of a power conversion device 100 according to the first embodiment of the present invention. The power converter 100 includes a first diode bridge 101a that is a first rectifier circuit that rectifies an AC voltage to a DC voltage, a first link capacitor 102a, a second link capacitor 102b, and a second rectifier circuit that is a second rectifier circuit. A two-diode bridge 101b, a half bridge 103, an AC voltage sensor 105 for detecting an AC voltage of the three-phase AC power supply 110, and a control circuit 104 for receiving an output of the AC voltage sensor 105 and controlling a switching element of the half bridge 103. Prepare. The half bridge 103 has a configuration in which a first switching element 103a and a second switching element 103b are connected in series.

  The power converter 100 receives power from the three-phase AC power supply 110 and supplies DC power to both ends (DC link unit 111) of the first link capacitor 102a and the second link capacitor 102b.

  The input sides of the first diode bridge 101a and the second diode bridge 101b are connected to the three-phase AC power supply 110. The output side of the first diode bridge 101a is connected to the DC link unit 111 of the power converter 100, and the first link capacitor 102a and the second link capacitor 102b connected to the DC link unit 111 are connected in parallel. The half bridge 103 is connected to the output side of the second diode bridge 101b. The midpoint of the half bridge 103 (that is, the midpoint of the first switching element 103a and the second switching element 103b) and the midpoint of the first link capacitor 102a and the second link capacitor 102b are connected by a conductor.

  With this configuration, charging of the first link capacitor 102a and the second link capacitor 102b can be controlled by switching control of the first switching element 103a and the second switching element 103b in the half bridge 103. This control is performed by the control circuit 104. At the subsequent stage of the DC link unit 111, another power conversion device such as an inverter or a load is connected.

  The control circuit 104 controls the half bridge 103 (the first switching element 103a and the second switching element 103b) based on each phase voltage information of the three-phase AC power supply 110 detected by the AC voltage sensor 105 input via the signal line 106. Is generated and transmitted to the half bridge 103 via the signal line 107.

  Although a diode bridge is used as the rectifier circuit in FIG. 1, the rectifier circuit is not limited to this. That is, any rectifier circuit that rectifies a three-phase AC voltage into a DC voltage can be employed. Further, the switching element may be a semiconductor element which can control conduction (ON) and non-conduction (OFF) by a gate signal. For example, a semiconductor such as a power transistor (MOSFET, IGBT, or the like) or a thyristor can be used as the switching element. As the signal lines 106 and 107, any communication means such as wired or wireless communication may be used. Further, in the embodiment of FIG. 1, the AC voltage sensor 105 for detecting each phase voltage is used, but the present invention is not limited to this, and the line voltage may be detected.

  Next, a specific configuration of the control circuit 104 in the embodiment of FIG. 1 will be described with reference to FIG.

  Each phase of the three-phase AC power supply 110 is a U-phase, a V-phase, and a W-phase. 1 detects U-phase voltage 201u, V-phase voltage 201v, and W-phase voltage 201w of three-phase AC power supply 110. These detected values are input to the line voltage calculator 202 of FIG. The line voltage calculator 202 generates a line voltage 203uv of the UV phase, a line voltage 203vw of the VW phase, and a line voltage 203wu of the WU phase from the information of each phase voltage. Information on each generated line voltage is input to the line voltage comparator 204. The line voltage comparator 204 compares the absolute values of the line voltages, and selects a phase at which the line voltage becomes maximum at that time. Here, the line voltage comparator 204 outputs a phase transition signal 205 to the gate signal generator 206 when the phase in which the line voltage becomes the maximum transits to another phase. The gate signal generator 206 determines each of the switching elements (the first switching element 103a and the second switching element 103a) based on the phase transition signal 205 indicating that the phase in which the absolute value of the line voltage becomes the maximum has transitioned to another phase. The gate signal 207 of 103b) is generated and output.

  The charging of the first link capacitor 102a and the second link capacitor 102b is performed during a period when the line voltage is equal to or higher than the voltage of each link capacitor. In the steady state, charging occurs in a period near the time when the absolute value of each line voltage takes the maximum value. Therefore, since the absolute value of each line voltage becomes the maximum (positive maximum and negative maximum) twice in one electrical angle cycle, there are a total of six charging modes. In the present embodiment, the control circuit 104 controls each switching element forming the half bridge 103 corresponding to these six modes. Details of this control will be described later.

  According to such control, the charging of the first link capacitor 102a and the charging of the second link capacitor 102b are performed while switching when the phase at which the line voltage becomes maximum transitions. Therefore, each link capacitor can be charged alternately. As a result, a DC voltage that is a total voltage can be obtained at both ends (DC link unit 111) of the two link capacitors. In the control of the control circuit 104, when both the switching elements of the half bridge 103 are controlled to be non-conductive (off), a full-wave rectified voltage by the first diode bridge 101a is obtained. Therefore, in the embodiment of FIG. 1, it is possible to supply a two-stage voltage of a full-wave rectified voltage and a DC voltage that is twice the rectified voltage.

  In the example of FIG. 2, the line voltage is obtained from the phase voltage. However, the present invention is not limited to this. The AC voltage sensor 105 directly detects the line voltage, and the detection result is used as the line voltage comparator of the control circuit 104. It may be input to 204. Although the gate signal 207 is generated by comparing the line voltages, the gate signal 207 may be generated by comparing the phase voltages.

  Next, the processing procedure in the embodiment of FIG. 1 will be described with reference to the flowchart of FIG. The operation based on the flowchart of FIG. 3 is as follows.

  In step 301, the calculation is started based on the calculation cycle set in the control circuit 104 for controlling the power conversion device 100.

  With the start of this process, in step 302, the switching state (ie, on / off state) of each switching element of the half bridge 103 in the previous calculation cycle is called. Next, the routine proceeds to step 303, where it is determined whether or not the phase transition signal 205 has been input.

  If the phase transition signal 205 has been input in step 303 (YES), the process proceeds to step 304. In step 304, the switching state of the two switching elements is changed. That is, the on / off state of the first switching element 103a and the second switching element 103b of the half bridge 103 is alternated.

  If the phase transition signal 205 has not been input in step 303 (NO), the process proceeds to step 306. In step 306, the switching state called in step 302 is maintained.

  In this way, a series of processes is completed in the procedure up to Step 305 (Step 306). This series of processing is performed at every predetermined calculation cycle.

  Next, the operation of each switching element in each mode executed by the control circuit in the embodiment of FIG. 1 and the current path will be described in detail with reference to FIGS.

  FIG. 4 shows each voltage / current waveform in the first embodiment. The waveforms shown in FIG. 4 are the UV-phase line voltage 401uv of the three-phase AC power supply 110, the VW-phase line voltage 401vw, the WU-phase line voltage 401wu, and the gate voltage 402a of the first switching element 103a of the half bridge 103. , A gate voltage 402b of the second switching element 103b, a first link capacitor voltage 403a, a second link capacitor voltage 403b, a DC voltage 404, and a U-phase current 405. M1 to M6 at the top of FIG. 4 indicate the first to sixth modes, respectively.

  First, in the first mode (M1), of the three line voltages (UV-phase line voltage 401uv, VW-phase line voltage 401vw, and WU-phase line voltage 401wu), the UV-phase line voltage 401uv is used. Is a period in which the absolute value of is maximum (positive maximum). In the first mode, the control circuit 104 controls the first switching element 103a to be on (conducting), and controls the second switching element 103b to be off (non-conducting). As a result, in the period of the first mode (M1), the current flows from the U-phase through the second diode bridge 101b, the first switching element 103a, the second link capacitor 102b, and the V through the first diode bridge 101a. It flows from the phase to the three-phase AC power supply 110. Therefore, during the first mode period, the second link capacitor 102b is charged, and the second link capacitor voltage 403b increases.

  The second mode (M2) is a period in which the absolute value of the WU-phase line voltage 401wu is maximum (negative maximum). When transitioning from the first mode to the second mode, the control circuit 104 controls the first switching element 103a of the half bridge 103 to be off and the second switching element 103b to be on. As a result, in the period of the second mode (M2), the current flows from the U phase through the first diode bridge 101a to the first link capacitor 102a, passes through the second switching element 103b, the second diode bridge 101b, and W It flows from the phase to the three-phase AC power supply 110. Therefore, during the second mode, the first link capacitor 102a is charged, and the first link capacitor voltage 403a increases.

  The third mode (M3) is a period in which the absolute value of the VW-phase line voltage 401vw is maximum (positive maximum). When transitioning from the second mode to the third mode, the control circuit 104 controls the first switching element 103a of the half bridge 103 to be on and the second switching element 103b to be off. As a result, in the period of the third mode (M3), the current flows from the V phase to the second diode capacitor 101b through the second diode bridge 101b, the first switching element 103a of the half bridge 103, and the first diode bridge 101a. Flows from the W-phase to the three-phase AC power supply 110. Therefore, during the third mode, the second link capacitor 102b is charged, and the second link capacitor voltage 403b increases.

  The fourth mode (M4) is a period in which the absolute value of the UV-phase line voltage 401uv is maximum (negative maximum). When transitioning from the third mode to the fourth mode, the control circuit 104 controls the first switching element 103a of the half bridge 103 to be off and the second switching element 103b to be on. As a result, in the period of the fourth mode (M4), the current flows from the V phase through the first diode bridge 101a to the first link capacitor 102a, passes through the second switching element 103b, the second diode bridge 101b, and U It flows from the phase to the three-phase AC power supply 110. Therefore, during the period of the fourth mode, the first link capacitor 102a is charged, and the first link capacitor voltage 403a increases.

  The fifth mode (M5) is a period in which the absolute value of the line voltage 401wu of the WU phase is maximum (positive maximum). When transitioning from the fourth mode to the fifth mode, the control circuit 104 controls the first switching element 103a of the half bridge 103 to be on and the second switching element 103b to be off. As a result, in the period of the fifth mode (M5), the current flows from the W phase through the second diode bridge 101b and the first switching element 103a to the second link capacitor 102b, passes through the first diode bridge 101a, and passes through the first diode bridge 101a. It flows from the phase to the three-phase AC power supply 110. Therefore, during the period of the fifth mode, the second link capacitor 102b is charged, and the second link capacitor voltage 403b increases.

  The sixth mode (M6) is a period in which the absolute value of the VW-phase line voltage 401vw is maximum (negative maximum). When transitioning from the fifth mode to the sixth mode, the control circuit 104 controls the first switching element 103a of the half bridge 103 to be off and the second switching element 103b to be on. As a result, during the period of the sixth mode (M6), current flows from the W phase through the first diode bridge 101a to the first link capacitor 102a, passes through the second switching element 103b, the second diode bridge 101b, and V It flows from the phase to the three-phase AC power supply 110. Therefore, during the period of the sixth mode, the first link capacitor 102a is charged, and the first link capacitor voltage 403a increases.

  After the sixth mode, the process returns to the first mode again. The control circuit 104 repeatedly performs on / off control of each switching element corresponding to each of the first to sixth modes.

  By the above operation, the switching of the line voltage for charging the link capacitor and the switching state of the half bridge 103 are synchronized to alternately charge the first link capacitor 102a and the second link capacitor 102b connected in series. be able to. Therefore, the DC voltage 404 comprising the total voltage of the two capacitors can be doubled from the conventional full-wave rectified voltage.

  In this series of operations, the U-phase current 405 has the same waveform as the current at the time of general diode rectification, and the power converter 100 does not generate harmonics in the power supply. Further, a current flows through the first switching element 103a and the second switching element 103b constituting the half bridge 103 in only one direction. Therefore, the bidirectional switch circuit can be eliminated.

  In this embodiment, a DC double voltage can be obtained only by adding a diode bridge and two switching elements, so that the device configuration is simplified. In addition, since there are few components such as switching elements, cost reduction can be realized.

  In the above embodiment, the switching state of the half bridge 103 is determined only by the magnitude relation of the absolute value of the line voltage. However, the turn-off timing may be determined based on the current flowing through the switching element. Charging of the two link capacitors ends when the capacitor voltage becomes substantially equal to the line voltage, and no current flows through the switching element. By detecting this and turning off the switching element, zero current switching becomes possible and the loss can be reduced.

(Operation when some diodes in the rectifier circuit break down)
In the first embodiment (FIGS. 1 and 2), the operation can be continued even when a failure occurs in which a part of the diode of the diode bridge becomes open. Next, this operation will be described with reference to FIGS.

  FIG. 5 shows a state in which the diode 1001 connected to the U-phase of the first diode bridge 101a has an open failure in the embodiment of FIG. FIG. 6 shows operation waveforms in the case of FIG.

  In the case of the failure in FIG. 5, the diode 1001 included in the first diode bridge 101a has an open failure, so that the charging of the first link capacitor 102a is not performed in the second mode among the first to sixth modes. The diode 1001 is connected to the U-phase. In the second mode, a current flows from the U-phase through the diode 1001 to the first link capacitor 102a, so that the U-phase current 405 does not flow. However, similarly, in the first mode in which current flows from the U-phase, the U-phase current 405 flows. In the first mode, a current flows from the U phase to the second link capacitor 102b through the second diode bridge 101b and the half bridge 103. Therefore, the second link capacitor 102b can be charged without passing through the diode 1001. In addition, even when another diode of the first diode bridge fails, the operation can be continued similarly.

  As described above, in the embodiment of FIG. 1, even when one of the diodes in the diode bridge has an open fault, a path through which a current flows through the link capacitor is formed. It becomes possible.

  Next, a configuration of the power conversion device according to the second embodiment will be described with reference to FIGS. In the following description, portions corresponding to the respective portions of the first embodiment are denoted by the same reference numerals, and description thereof may be omitted. In the above-described first embodiment, the power converter capable of supplying the full-wave rectified voltage and the DC voltage twice as much as the full-wave rectified voltage has been described. This is a power conversion device improved to be able to supply an arbitrary DC voltage.

  FIG. 7 illustrates a circuit configuration of the power conversion device 100 according to the second embodiment. 7, a DC voltage sensor 501 for detecting a DC voltage is provided in addition to the configuration of the power conversion device 100 of FIG. The detected DC voltage information is input to the control circuit 104 via the signal line 502. The control circuit 104 controls the conduction period (duty ratio) of each switching element included in the half bridge 103 so as to eliminate the difference between the target DC voltage command and the detected DC voltage, thereby reducing the full-wave DC voltage. It is possible to control to supply an arbitrary DC voltage from the rectified voltage to the doubled voltage. In other words, in each of the first to sixth modes determined from the above-described three-phase AC voltage waveform, the conduction period in each mode is controlled so that the difference between the detected DC voltage and the target DC voltage command is eliminated. Thereby, a desired DC voltage can be obtained.

  FIG. 8 shows a specific configuration of the control circuit 104 of FIG. 8, a half bridge 103, phase voltages 201u to 201w, a line voltage calculator 202, line voltages 203uv to 203wu, a line voltage comparator 204, a phase transition signal 205, a gate signal generator 206, and a gate signal 207 is the same as that described in the first embodiment (FIG. 2).

  8, a DC voltage difference 603 between a DC voltage command 601 input to the control circuit 104 and a DC voltage 602 detected by the DC voltage sensor 501 is input to a duty ratio calculator 604. The duty ratio calculator 604 calculates a duty ratio command value 605 for determining the duration of the ON state of each switching element included in the half bridge 103. This command value is input to the gate signal generator 206. The gate signal generator 206 determines the turn-on time of each switching element based on the phase transition signal 205, and determines the turn-off time of each switching element based on the duty ratio command value 605. Thereby, the charging current of each link capacitor can be controlled, and the DC voltage can be controlled to a target value.

  FIG. 9 is an example of a method of determining a DC voltage command value 701 assuming an inverter device in which a three-phase PWM inverter is connected to a stage subsequent to each link capacitor that supplies a DC voltage of a power converter and a motor is connected as a load. FIG.

  In FIG. 9, the horizontal axis represents the motor rotation speed (hereinafter referred to as the rotation speed), and the vertical axis represents the DC voltage. In a region where the rotation speed is equal to or less than the first predetermined value 702, the DC voltage command value 701 is set to the full-wave rectified voltage 703. That is, each switching element of the half bridge 103 is turned off (disconnected). In this region, since the induced voltage of the motor is lower than that in the high rotation region, the required output voltage of the inverter is also lower. Accordingly, the switching of the half bridge 103 is stopped, and the required voltage can be output even if the DC voltage on the output side of the power converter is changed to the full-wave rectified voltage 703. As a result, a reduction in the switching loss of the half bridge 103 and a reduction in the conduction loss due to a reduction in the number of semiconductor elements (switching elements and diodes) on the current path can be expected, and the power converter 100 can be made more efficient.

  Next, in a region where the rotation speed is larger than the first predetermined value 702 and equal to or smaller than the second predetermined value 704, the DC voltage command value 701 is changed according to the rotation speed. As the rotational speed increases, the induced voltage also increases accordingly. Therefore, it is necessary to increase the output voltage of the inverter. The upper limit of the output voltage is determined by the DC voltage and the modulation factor. When a higher voltage is required, the rotation speed is generally increased by field weakening control. In the field-weakening control, the induced voltage is canceled by injecting a reactive current into the motor, and high rotation can be realized with a low DC voltage. However, the conduction loss increases due to the flow of the reactive current, so that the efficiency decreases. Therefore, as described above, by increasing the DC voltage command value 701 of the power conversion device 100 as the rotation speed increases, the rotation speed can be increased without field-weakening control, and the power conversion device 100 can be made more efficient.

  In a region where the rotation speed is larger than the second predetermined value 704, the DC voltage command value 701 is set to the doubled voltage value 705. In this region, the duty ratio of the half bridge 103 is fixed at 0.5, and the DC voltage is fixed at the doubled voltage value 705.

  With the above operation, power conversion device 100 can be driven with high efficiency in a wide rotation speed range. In the region where the number of revolutions is equal to or less than the second predetermined value 704, the amplitude of the pulse voltage applied to the motor can be reduced, so that deterioration of the motor can be suppressed.

  In FIG. 9, the DC voltage command value 701 is changed in proportion to the rotation speed in the region from the first predetermined value 702 to the second predetermined value 704, but the invention is not limited to this. . Further, in FIG. 9, the DC voltage command value 701 is determined according to the number of revolutions, but may be determined according to the load.

  FIG. 10 is a diagram illustrating a control flowchart in the second embodiment (FIG. 7). The operation based on FIG. 10 is as follows.

  The processing operations in steps 801 to 804 are the same as those in steps 301 to 304 in FIG.

  If a transition signal has not been input in step 803 (NO), the process proceeds to step 805. In this step 805, it is determined whether or not there is a switching element to be turned on in the half bridge 103.

  If there is no switching element to be turned on in step 805 (NO), the process proceeds to step 807, where the switching state called in step 802 is maintained.

  If there is a switching element to be turned on in step 805 (YES), the process proceeds to step 806. In step 806, it is determined whether or not the on-time since the transition to the current switching state is less than the on-time determined by the duty ratio command value 605. In the case of YES, the process proceeds to step 808. In step 808, all the switching elements are turned off.

If there is no switching element to be turned on in step 806 (NO), the process proceeds to step 807.
If “YES” in the step 806, the two switching elements of the half bridge 103 are turned off (step 808).
A series of processing ends in the procedure up to step 808 (step 809).

  Next, the operation of the power converter 100 in the embodiment of FIG. 7 will be described in detail with reference to FIG. FIG. 11 shows each voltage / current waveform in the second embodiment. In the waveform of FIG. 11, the current paths flowing through the respective link capacitors in the first mode (M1) to the sixth mode (M6) are the same as those in the first embodiment (FIG. 1). Therefore, the description of the operation in each mode in FIG. 11 is the same as that in FIG.

  In the second embodiment (FIG. 7), the control circuit 104 sets the period (duty ratio) in which the gate voltage 402a of the first switching element 103a and the gate voltage 402b of the second switching element 103b of the half bridge 103 are on (FIG. 3). Control is performed, for example, so as to be smaller than that shown. Specifically, control circuit 104 controls the ON period (duty ratio) of first switching element 103a and second switching element 103b such that the output of DC voltage sensor 501 matches the target DC voltage command. .

  By such control, the DC voltage is controlled to the target value by controlling the charging current flowing through each link capacitor. For example, when looking at the U-phase current 405 in FIG. 11, it can be seen that the current conduction period in each mode is shorter due to switching than in the case of FIG. In the case of the second embodiment, the DC voltage can be a full-wave rectified voltage at a duty ratio of 0 and a voltage twice the full-wave rectified voltage at a duty ratio of 0.5.

  In the waveform of FIG. 11, the gate voltage 402a of the first switching element 103a and the gate voltage 402b of the second switching element 103b are given equal. However, the present invention is not limited to this. For example, since the capacitances of the first link capacitor 102a and the second link capacitor 102b are not always equal due to individual differences and deterioration, a method of individually controlling the charging current to each link capacitor and separately applying a gate voltage can be considered. As a result, a stable operation can be realized even if the capacity of the link capacitor varies.

(Mounting the power converter on an integrated inverter)
The power converter 100 according to the embodiment of the present invention can be mounted on a mechanical-electrically integrated inverter device. The mechanical and electric integrated type inverter device is a device in which a three-phase inverter is connected to the subsequent stage of the DC link unit shown in FIG. 1 and a motor is connected to the output of the three-phase inverter, and these housings are integrated. is there. By applying the above-described power conversion device 100 to a mechanical and electric integrated inverter device, not only the inverter but also the motor specifications can be shared for different power supply voltage specifications, so that the cost reduction of the entire device is expected. it can.

110 three-phase AC power supply 101a first diode bridge 101b second diode bridge 102a first link capacitor 102b second link capacitor 103 half bridge 103a first switching element 103b second switching element 104 control circuit 105 ... AC voltage sensor

Claims (10)

A first rectifier circuit for rectifying an AC voltage of the three-phase AC power supply into a DC voltage;
A first link capacitor and a second link capacitor connected between a DC output side and a load side of the first rectifier circuit;
A second rectifier circuit connected to the three-phase AC power supply in parallel with the first rectifier circuit;
A half bridge in which a first switching element and a second switching element are connected in series on a DC output side of the second rectifier circuit;
Connecting a middle point of the first switching element and the second switching element, a middle point of the first link capacitor and a middle point of the second link capacitor,
Further, an AC voltage sensor for detecting a three-phase AC voltage of the three-phase AC power supply,
A power converter, comprising: a control circuit that performs switching control of the first switching element and the second switching element based on a detection value of the AC voltage sensor. The power converter according to claim 1,
The control circuit is configured to conduct the first switching element and the third switching element so as to conduct the phase during a period in which the absolute value of the voltage of any phase of the three-phase AC power detected by the AC voltage sensor is maximum. (2) A power converter for controlling a switching element. The power converter according to claim 1,
A power converter in which the first rectifier circuit and the second rectifier circuit are configured by a diode bridge. The power converter according to claim 3,
When any one of the diodes constituting the diode bridge fails, any one of the first link capacitor and the second link capacitor continues from the phase of the three-phase AC power supply to which the failed diode is connected. Power converter to charge the battery. The power converter according to claim 1,
The power conversion device, wherein the AC voltage sensor detects a line voltage of the three-phase AC voltage, and the control circuit controls the first switching element and the second switching element based on the line voltage. The power converter according to claim 1,
A DC voltage detection sensor for detecting a DC voltage between both ends of the first link capacitor and the second link capacitor; and the control circuit detects a DC voltage based on a detection value of the DC voltage detection sensor. A power converter that controls the first switching element and the second switching element so as to match a voltage command. The power converter according to claim 6,
The power conversion device, wherein the control circuit controls a duty ratio of the first switching element and the second switching element so as to match the DC voltage command. The power converter according to claim 7,
The power converter, wherein the control circuit individually controls the duty ratio to be different between the first switching element and the second switching element. An inverter device comprising: a power conversion device that converts a three-phase AC power supply into a DC voltage; and an inverter that converts the DC voltage into an AC voltage and supplies the AC voltage to a load.
An inverter device employing the power converter according to claim 1 as the power converter. The inverter device according to claim 9, wherein the load is a motor, and the power conversion device, the inverter, and the motor are mounted in one housing.

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