JP2019174130A - Radar system and radar system target detection method - Google Patents

Abstract

To respond to persistent interference wave.SOLUTION: The radar system includes a transmission means for transmitting a transmission signal via a transmission antenna, a reception means for receiving a reflected wave reflected by a target via a reception antenna, an orthogonal demodulation means for generating an in-phase component and an orthogonal component of the transmission signal by orthogonally demodulating a received reception signal based on the frequency of the transmission signal, a spectrum generating means for generating a first spectrum having at least one of a first distance function and a first angle function having a maximum intensity according to the position of the target and a second spectrum having at least one of a second distance function and a second angle function orthogonal to the first distance function and the first angle function as well as having a minimum intensity according to the position of the target, and an estimation means for estimating the position of the target based on the first spectrum and the second spectrum generated by the spectrum generating means.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a radar apparatus and a target detection method for the radar apparatus.

  The radar device transmits a transmission signal, receives a reflected wave from the target, and performs signal processing on the received signal to estimate the distance and azimuth of the target, but simultaneously reflects the reflected wave from a plurality of targets. When a plurality of reflected waves are received, a plurality of reflected waves interfere with each other, and an incorrect position may be estimated. In particular, interference is likely to occur when the distance difference between a plurality of targets viewed from the radar apparatus is close or when the difference in azimuth is small.

  As a countermeasure against such an interference problem, for example, there is a technique disclosed in Patent Document 1. The technique disclosed in Patent Document 1 is characterized in that the presence of multipath interference waves is detected from the non-uniformity of the amplitudes of received signals from a plurality of receiving antennas, and the azimuth misestimation result is discarded.

  Further, in Patent Document 2, when a FMCW (Frequency Modulated Continuous Wave) radar apparatus receives a transmission signal of another pulse radar apparatus, a center value is calculated from a plurality of sweep data and used. A method for removing the influence of an interference wave from a pulse radar that occurs instantaneously is described.

  Further, in Patent Document 3, in order to prevent a real image and a mirror ghost from being separated and to estimate a target in a wrong direction, the amplitude of a received wave having a peak reception intensity is used. For each antenna, detects the azimuth of the target based on the phase of the received wave received by each of the multiple antennas, and if the target exists in the detected azimuth, it is estimated that the antenna will receive A technique for outputting the amplitude and phase of a wave for each antenna and comparing the obtained phase and amplitude for each antenna is disclosed.

Japanese Patent Laid-Open No. 9-211114 JP 2010-25901 A International Publication WO2013 / 175558

  However, the interference wave countermeasures by the techniques disclosed in Patent Literature 1 and Patent Literature 2 can deal with an interference wave that occurs instantaneously and ends in a short time, but it can deal with a persistent interference wave. There is a problem that cannot be done. Further, the technique disclosed in Patent Document 3 can discard an erroneous estimation result, but has a problem that the radar function is lost during that time.

  An object of the present invention is to provide a radar apparatus and a target detection method for a radar apparatus that can deal with a continuous interference wave without stopping the radar function.

In order to solve the above-described problems, the present invention provides a radar device that detects a target, a transmission unit that transmits a transmission signal via a transmission antenna, and a reflection that is transmitted by the transmission unit and reflected by the target. Receiving means for receiving a wave via a receiving antenna; and generating a in-phase component and a quadrature component of the transmission signal by performing quadrature demodulation on the reception signal received by the reception means according to the frequency of the transmission signal A quadrature demodulating means; a first spectrum having at least one of a first distance function and a first angle function having a maximum intensity according to the position of the target by the in-phase component and the quadrature component; and At least a second distance function and a second angle function that have a minimum intensity according to the position and are orthogonal to the first distance function and the first angle function. Spectrum generating means for generating a second spectrum having one; and estimation means for estimating the position of the target based on the first spectrum and the second spectrum generated by the spectrum generating means. It is characterized by.
According to such a configuration, it is possible to cope with a continuous interference wave without stopping the radar function.

In the invention, it is preferable that the spectrum generating unit has at least one of a first distance function and a first angle function that has a maximum intensity at the position of the target when the single target is present. One spectrum and a second spectrum having at least one of a second distance function and a second angle function having a minimum intensity at the position of the target are generated.
According to such a configuration, the position of the target can be accurately detected.

Further, in the present invention, the transmission unit transmits the transmission signal a plurality of times, the spectrum generation unit generates a plurality of the second spectrums for the transmission signal transmitted a plurality of times, and the estimation unit includes: A plurality of the second spectra are compared, and a position where the intensity change is minimal is estimated as the position of the target.
According to such a configuration, the position of the target can be accurately detected.

Further, in the present invention, the transmission unit transmits the transmission signal a plurality of times, the spectrum generation unit generates a plurality of the second spectrums for the transmission signal transmitted a plurality of times, and the estimation unit includes: A position where the maximum value at each position of the plurality of second spectra is minimized is estimated as the position of the target.
According to such a configuration, the position of the target can be accurately detected.

Further, in the present invention, the transmission unit generates the transmission signal by modulating a local oscillation signal having a predetermined frequency with a pulsed baseband signal, and the orthogonal demodulation unit converts the reception signal into the station. The I-component which is the in-phase component and the Q-component which is the quadrature component are generated by performing quadrature demodulation by the emitted signal, and the spectrum generating means sets j as an imaginary number unit and sets (I + jQ) as the first distance function. A function having a proportional value and a time axis converted to a distance axis is used, a function indicating the intensity of the first distance function is used as the first distance spectrum, and (I + jQ) is Hilbert as the second distance function. When the value obtained by the conversion is (I ′ + jQ ′), a function having a value proportional to (I ′ + jQ ′) and converting the time axis to the distance axis is used. A function indicating the intensity of the second distance function is used as a tram.
According to such a configuration, the pulse-type radar apparatus can cope with a continuous interference wave without stopping the radar function.

Further, in the present invention, the transmission unit generates an FM modulated signal whose frequency continuously changes as the transmission signal, and the orthogonal demodulation unit generates a beat signal having a frequency difference between the reception signal and the transmission signal. And generating the I component that is the in-phase component and the Q component that is the quadrature component by transforming the beat signal into a frequency domain by Fourier transform, and the spectrum generating means uses j as an imaginary unit, A function having a value proportional to (I + jQ) as the first distance function and a function in which a frequency axis is converted into a distance axis is used, and a function indicating the intensity of the first distance function is used as a first distance spectrum; As a 2-distance function, when the value obtained by Hilbert transform of (I + jQ) is (I ′ + jQ ′), it has a value proportional to (I ′ + jQ ′) and the frequency axis is A function converted to an off-axis is used, and a function indicating the intensity of the second distance function is used as the second distance spectrum.
According to such a configuration, an FM modulation type radar apparatus can cope with a continuous interference wave without stopping the radar function.

Also, the present invention has a plurality of at least one of the transmission antenna and the reception antenna, the spectrum generation means, j is an imaginary unit, (I + jQ) by the combination of the transmission antenna and the reception antenna is an array, By performing the Fourier transform, the first angle function in the angle axis is generated, and when the real part and the imaginary part of the first angle function are Re_A and Im_A, the result of Hilbert transform of Re_A and Im_A is represented by Re_B and When Im_B, the second angle function is a function of an angle axis proportional to (Re_B + jIm_B), and the second angle spectrum is an angle axis signal reflecting the intensity of the second angle function. And
According to such a configuration, the azimuth angle of the target can be accurately detected.

Further, the present invention is characterized in that an odd function window function is used when calculating the second angle function from an array of (I + jQ).
According to such a configuration, the second angle function can be obtained by simple calculation.

Further, the present invention has a plurality of the reception antennas and is arranged side by side in the first direction, and the transmission antenna is arranged at a position shifted from the center of the arrangement of the plurality of reception antennas. .
According to such a configuration, it is possible to prevent unnecessary direct waves from being detected in a direction with an azimuth angle of 0 degrees.

Further, the present invention provides a target detection method for a radar apparatus that detects a target, a transmission step of transmitting a transmission signal via a transmission antenna, and a reflected wave transmitted by the transmission step and reflected by the target A reception step of receiving the received signal via a receiving antenna, and a quadrature for generating an in-phase component and a quadrature component of the transmission signal by orthogonally demodulating the reception signal received by the reception step according to the frequency of the transmission signal A demodulating step, a first spectrum having at least one of a first distance function and a first angle function having a maximum intensity according to the position of the target by the in-phase component and the quadrature component; and the position of the target And the second distance function and the second angle that are minimal in intensity and orthogonal to the first distance function and the first angle function. A spectrum generation step for generating a second spectrum having at least one of the numbers; an estimation step for estimating a position of the target based on the first spectrum and the second spectrum generated by the spectrum generation step; It is characterized by having.
According to such a method, it is possible to cope with a continuous interference wave without stopping the radar function.

  ADVANTAGE OF THE INVENTION According to this invention, it becomes possible to provide the radar apparatus and the target detection method of a radar apparatus which can respond to a continuous interference wave, without stopping a radar function.

It is a figure which shows the structural example of the radar apparatus which concerns on 1st Embodiment of this invention. It is a figure which shows the detailed structural example of the distance estimation part shown in FIG. It is a figure for demonstrating operation | movement of the 1st distance function calculating part and the 1st distance spectrum calculating part which are shown in FIG. It is a figure for demonstrating operation | movement of the 2nd distance function calculating part and 2nd distance spectrum calculating part which are shown in FIG. It is a figure for demonstrating the operation | movement of a 1st distance function calculating part and a 1st distance spectrum calculating part when the positional relationship of a 1st target-a 3rd target, and a radar apparatus changes within the range comparable as a wavelength. is there. It is a figure for demonstrating operation | movement of the 2nd distance function calculating part and the 2nd distance spectrum calculating part when the positional relationship of a 1st target-a 3rd target, and a radar apparatus changes within the range comparable as a wavelength. is there. It is a figure which shows the output of a back | latter stage rather than the 2nd distance spectrum calculating part shown in FIG. It is a figure which shows the structural example of the radar apparatus which concerns on 2nd Embodiment of this invention. It is a figure which shows the example of arrangement | positioning of the transmission antenna of 2nd Embodiment shown in FIG. 8, and a receiving antenna. It is a figure which shows the detailed structural example of the azimuth angle estimation part shown in FIG. It is a figure which shows the 1st angle spectrum and 2nd angle spectrum in case a target exists in azimuth | direction angle 0 degree. It is a figure which shows the angle spectrum at the time of receiving a reflected wave from a single direction. It is a figure which shows the angle spectrum at the time of receiving a reflected wave from a some direction. An angle spectrum is shown when the target is present at azimuth angles of +30 degrees and −30 degrees. It is a figure for demonstrating the specific example of an azimuth angle estimation in the environment where the interference wave using the data of several shots exists. It is a figure which shows operation | movement when a target exists in the position of a phase angle of 0 degree | times and -20 degree | times. It is a figure which shows G ((theta)) and H ((theta)). It is a figure which shows a detection result in case a target exists in the position where a phase angle is 0 degree | times and -20 degree | times. It is a figure for demonstrating a direct wave. It is a figure which shows C ((theta)), D ((theta)), E ((theta)), and F ((theta)). It is a figure which shows G ((theta)) and H ((theta)). It is a figure which shows the detection result in case a target exists in the position of phase angle -20 degrees. It is a figure which shows C ((theta)), D ((theta)), E ((theta)), and F ((theta)). It is a figure which shows G ((theta)) and H ((theta)). It is a figure which shows C ((theta)), D ((theta)), E ((theta)), and F ((theta)). It is a figure which shows G ((theta)) and H ((theta)). It is a figure which shows a detection result in case a target exists in the position of a phase angle of 20 degree | times and -45 degree | times. It is a figure which shows the structural example of the radar apparatus which concerns on 3rd Embodiment of this invention. It is a figure for demonstrating operation | movement of 3rd Embodiment shown in FIG. It is a figure for demonstrating operation | movement of 3rd Embodiment shown in FIG. It is a figure which shows the structural example of the radar apparatus used as the comparison for demonstrating the effect of the offset of the transmission antenna of 3rd Embodiment of this invention. It is a figure for demonstrating operation | movement of the conventional radar apparatus shown in FIG. It is a figure for demonstrating operation | movement of the conventional radar apparatus shown in FIG.

  Next, an embodiment of the present invention will be described.

(A) Description of Configuration of First Embodiment of the Present Invention FIG. 1 is a diagram illustrating a configuration example of a radar apparatus according to the first embodiment of the present invention. The radar apparatus 10 according to the first embodiment of the present invention is mounted on a vehicle such as an automobile, for example, and detects targets such as other vehicles, pedestrians, and obstacles existing around the vehicle.

  Here, the radar apparatus 10 includes a control unit 11, a modulation signal generation unit 12, a VCO (Voltage Controlled Oscillator) 13, a modulation unit 14, a delay unit 15, an amplification unit 16, a transmission antenna 17, a reception antenna 18, an amplification unit 19, Mixers 20 and 21, LPFs (Low Pass Filters) 22 and 23, ADCs (Analog to Digital Converters) 24 and 25, and a received signal processing unit 26 are included.

  Here, the control unit 11 controls the modulation signal generation unit 12, the VCO 13, the reception signal processing unit 26, and the like. In FIG. 1, a broken line indicates a signal line for transmitting a control signal.

  For example, the modulation signal generation unit 12 generates a modulation signal such as a pulse signal and supplies the modulation signal to the modulation unit 14. The VCO 13 generates a local oscillation signal having a predetermined frequency under the control of the control unit 11 and supplies the local oscillation signal to the modulation unit 14, the delay unit 15, and the mixer 20.

  The modulation unit 14 modulates the local oscillation signal supplied from the VCO 13 with the modulation signal supplied from the modulation signal generation unit 12 and outputs the modulated signal as an RF (Radio Frequency) signal.

  The delay unit 15 delays the local oscillation signal supplied from the VCO 13 by π / 2 and supplies the delayed signal to the mixer 21.

  The amplification unit 16 amplifies the RF signal supplied from the modulation unit 14 and supplies the amplified RF signal to the transmission antenna 17 as a transmission signal.

  The transmission antenna 17 transmits the transmission signal supplied from the amplification unit 16 toward the target as an electromagnetic wave.

  The receiving antenna 18 receives the reflected wave transmitted from the transmitting antenna 17 and reflected by the target, and supplies it to the amplifier 19 as a received signal.

  The amplifying unit 19 amplifies the received signal supplied from the receiving antenna 18 with a predetermined gain and outputs the amplified signal as an RF signal.

  The mixer 20 down-converts the RF signal supplied from the amplifying unit 19 with the local signal supplied from the VCO 13 and outputs it as an IF (Intermediate Frequency) signal. The mixer 21 down-converts the RF signal output from the amplifying unit 19 with the local signal output from the delay unit 15 and outputs the result as an IF signal.

  The LPF 22 attenuates the harmonic component from the IF signal output from the mixer 20 and outputs the attenuated component. The LPF 23 attenuates the harmonic component from the IF signal output from the mixer 21 and outputs it.

  The ADC 24 converts the analog signal output from the LPF 22 into a digital signal and supplies the digital signal to the reception signal processing unit 26. The ADC 25 converts the analog signal output from the LPF 23 into a digital signal and supplies the digital signal to the reception signal processing unit 26.

  The received signal processing unit 26 receives the digitized IF signals supplied from the ADCs 24 and 25 and executes target distance estimation processing, velocity estimation processing, azimuth angle estimation processing, clustering processing, tracking processing, and the like. Thus, the target is detected, and the detected result is supplied to an upper ECU (Electric Control Unit) not shown. 1 illustrates a distance estimation unit 27 (described later with reference to FIG. 2) that estimates the distance of the target among the processing units that perform the plurality of processes described above.

  FIG. 2 is a diagram illustrating a configuration example of the distance estimation unit 27 illustrated in FIG. As shown in FIG. 2, the distance estimation unit 27 includes a first distance function calculation unit 271, a first distance spectrum calculation unit 272, a maximum value extraction unit 273, a Hilbert filter 274, a second distance function calculation unit 275, and a second distance. A spectrum calculation unit 276, a maximum value / minimum value extraction unit 277, a maximum value-minimum value calculation unit 278, and an estimation processing unit 279 are included.

  Here, the first distance function calculator 271 generates and outputs a first distance function having a value of (I + jQ) with respect to the distance from the I and Q components supplied from the ADCs 24 and 25.

  The first distance spectrum calculation unit 272 generates and outputs a first distance spectrum having a value of | I + jQ | from the first distance function supplied from the first distance function calculation unit 271. || indicates an absolute value in parentheses.

  The maximum value extraction unit 273 extracts the maximum value from the first distance spectrum supplied from the first distance spectrum calculation unit 272 and supplies the maximum value to the estimation processing unit 279.

  The Hilbert filter 274 performs a Hilbert transform on the first distance function supplied from the first distance function calculation unit 271 and outputs the result. The Hilbert filter 274 is a filter that functions to shift the phase component of the signal by 90 degrees. For example, cos θ is converted into sin θ, and sin θ is converted into −cos θ. Such a filter can be realized by digital signal processing such as an FIR (Finite Impulse Response) filter.

  The second distance function calculation unit 275 generates and outputs a second distance function (I ′ + jQ ′) for the distance from the first distance function subjected to the Hilbert transform supplied from the Hilbert filter 274.

  The second distance spectrum calculation unit 276 generates and outputs a second distance spectrum having a value of | I ′ + jQ ′ | from the second distance function supplied from the second distance function calculation unit 275.

  The maximum value / minimum value extraction unit 277 extracts the maximum value and the minimum value from the second distance spectrum supplied from the second distance spectrum calculation unit 276 and supplies the maximum value and the minimum value calculation unit 278.

  The maximum value-minimum value calculation unit 278 supplies a value obtained by subtracting the minimum value from the maximum value to the estimation processing unit 279.

  The estimation processing unit 279 performs processing for estimating the distance of the target based on information supplied from the maximum value extraction unit 273 and the maximum value-minimum value calculation unit 278.

(B) Description of Operation of First Embodiment of the Invention Next, operation of the first embodiment of the present invention will be described. FIG. 3 is a diagram illustrating an operation when the first target is present at a distance of 0 m from the radar apparatus 10 and the second target is present at a distance of 4 m from the radar apparatus 10. Note that the first target existing at 0 m is actually generated by electromagnetic waves that directly circulate from the transmitting antenna 17 to the receiving antenna 18 of the radar apparatus 10, and is not due to a reflected wave from an actual target. However, since this is a suitable example of a continuous interference wave, which is a problem in the prior art, a sneak signal is also treated as a target here.

  FIGS. 3A and 3B are diagrams illustrating the I and Q components output from the first distance function calculation unit 271 illustrated in FIG. 2. FIG. 3C is a diagram showing a first distance spectrum output from the first distance spectrum calculation unit 272 shown in FIG. In FIG. 3, the horizontal axis indicates the distance [m], and the vertical axis indicates the output value.

  The example in FIG. 3 assumes a 24 GHz band radar device that is a quasi-millimeter wave, and the width of the peak shape in the first distance spectrum output from the first distance spectrum calculation unit 272 is about 2 m. This is due to the limitation of the usable frequency bandwidth defined by the Radio Law.

  Since the transmitting antenna 18 is arranged in the same housing as the receiving antenna 18, the distance between the antennas is short, and the sneak signal tends to be large. Therefore, the peak of the first target in the first distance spectrum is the second object. It is larger than the mark. In the example of FIG. 3, the first target and the second target are separated by 4 m, and the interference between the reflected waves is small. Therefore, the peak search of the first distance spectrum, which is the distance estimation method of the conventional radar apparatus, is used. It becomes possible to correctly estimate the positions of these two targets.

  FIGS. 4A and 4B are diagrams illustrating the second distance component I ′ + jQ ′ output from the second distance function calculation unit 275 illustrated in FIG. 2. FIG. 4C is a diagram illustrating a second distance spectrum | I ′ + jQ ′ | output from the second distance spectrum calculation unit 276 illustrated in FIG. 2. As shown in FIG. 4C, at the distance (0 m and 4 m) at which the first distance spectrum is maximum, the second distance spectrum is minimum.

  The above is an explanation of a case where two first targets and second targets are arranged at a distance of 4 m. Next, a case where a third target exists at a position 1 m in addition to the first target and the second target present at positions 0 m and 4 m from the radar apparatus 10 will be described.

  When the third target is present at a distance of 1 m from the radar apparatus 10, the position is within the peak of the first target, and the reflected waves from the first target and the third target strongly interfere with each other. Each other. Further, because of wave interference, the interference result shows various states depending on the phase of the reflected wave. FIG. 5 shows the first distance function when the positional relationship between the first to third targets and the radar apparatus 10 changes within the range of the same degree as the wavelength (for example, about 1.2 cm in the case of 24 GHz). It is a figure which shows the output from the calculating part 271 and the 1st distance spectrum calculating part 272. FIG. 6 shows the second distance function calculation unit 275 and the second distance spectrum calculation unit when the positional relationship between the first to third targets and the radar apparatus 10 changes within the same range as the wavelength. 276 is a diagram showing an output from 276. FIG.

  In the first distance spectrum shown in FIG. 5C, the peak of the third target cannot be recognized because it is buried in the large peak of the first target. In addition, the first target has no peak at the correct position due to interference with the third target. As a result, it is difficult for the conventional distance estimation method to estimate the correct distance for both the first target and the second target.

  In FIG. 5C, the waveform related to the first target is an interference wave that is continuously generated because it wraps around from the transmitting antenna 17 to the receiving antenna 18.

  On the other hand, in the second distance spectrum shown in FIG. 6C, when attention is paid to the distance position where the target is present, the variation in values is small at the points 0 m, 1 m, and 4 m. This is because the effect of the Hilbert filter 274 reduces the influence of the reflected wave from the target distance in the second distance spectrum because the amplitude of the second distance function becomes zero at the position of the target (that is, This is because no interference occurs only at the target distance).

  FIG. 7 is a diagram showing an output subsequent to the second distance spectrum calculation unit 276. A curve C1 shown in FIG. 7A indicates the maximum value at each distance in the second distance spectrum extracted by the maximum value and minimum value extraction unit 277 shown in FIG. 2, and a curve C2 indicates the second distance spectrum. The minimum value at each distance is shown.

  A curve C3 illustrated in FIG. 7B is a curve indicating a difference value between the curve C1 that is the maximum value and the curve C2 that is the minimum value, which is calculated by the maximum value-minimum value calculation unit 278. In the curve C3, a downward convex bend occurs at the position of each target.

  In the estimation processing unit 279, in the vicinity of the maximum value in the first distance spectrum supplied from the maximum value extraction unit 273, a curve C3 that is a difference value between the maximum value and the minimum value supplied from the maximum value-minimum value calculation unit 278 is used. This is the estimated distance of the distance target that will be a downwardly convex bending point. Thereby, as the curve C4 shown in FIG.7 (C), the distance of a 1st target, a 2nd target, and a 3rd target can be estimated correctly.

  Since the second target is 3 m or more away from the other target, the interference with the reflected wave from the other target is small. For this reason, the variation in the second distance spectrum is small other than the distance of the second target, and the curve C3 that is the difference value between the maximum value and the minimum value does not clearly cause bending. In such a case, the bend in the curve C1 using the maximum value of the second distance spectrum may be used. As a more specific procedure, a clearer one is adopted by comparing the curved bends below the curves C1 and C3. As a quantitative comparison method of the downwardly convex bend, even if the curve C1 and the curve C3 are subjected to second-order differentiation with respect to the distance, a positive value that is a larger value may be adopted as a clear bend. Good.

  As described above, in the first embodiment of the present invention, the above feature is used to estimate the distance that is large in the first distance spectrum and becomes a valley in FIG. 7B obtained from the second distance spectrum. By setting the distance, it is possible to correctly estimate the distances of the first target, the second target, and the third target as shown by a curve C4 in FIG.

  In this way, by using the first distance spectrum and the second distance spectrum measured under a plurality of conditions where the phases of the interference waves are different, the distance of the target can be obtained even in an environment where there is a continuous interference wave. Can be estimated correctly.

  In order to create a plurality of conditions with different phases of interference waves, it is only necessary to slightly change the relative distance between targets. For example, when the radar apparatus 10 uses a 24 GHz band electromagnetic wave (radar wave), since the wavelength of the radar wave is about 12 mm, all phase conditions can be created within a range in which the target moves 12 mm. This is a slight positional displacement with respect to the peak width (about 2 m) in the distance spectrum of the 24 GHz band radar. Further, since the necessary change is a change in the relative position between the radar apparatus 10 and the target, the radar apparatus 10 may move instead of the target. Further, since the phase change is also caused by the wavelength change of the radar wave, the above-described plurality of phase combination conditions may be created by using a plurality of shots in which the center frequency of the radar wave is changed.

  In the example of FIG. 2, the Hilbert filter 274 is used to generate the second distance function. However, when the radar apparatus uses FM modulation, the beat signal is Fourier-transformed to generate the distance function. The window function may be generated by making the window function an odd function. The specific method of using the window function is the same as the azimuth angle estimation method in the second embodiment to be described later.

(C) Description of Configuration of Second Embodiment of Present Invention Next, a second embodiment of the present invention will be described. FIG. 8 is a diagram showing a configuration example of the second embodiment of the present invention. In FIG. 8, portions corresponding to those in FIG. In FIG. 8, compared with FIG. 1, the receiving antenna 18 is replaced with receiving antennas 18-1 to 18-n (n> 1). Further, the amplifying unit 19, the mixers 20 and 21, the LPFs 22 and 23, and the ADCs 24 and 25 are the amplifying units 19-1 to 19-n, the mixers 20-1 to 20-n, 21-1 to 21-n, and the LPFs 22-1 to 21- 22-n, 23-1 to 23-n23, ADC 24-1 to 24-n, 25-1 to 25-n. Since these functions and operations are the same as in the case shown in FIG.

  In the example of FIG. 8, a distance estimation unit 27, a speed estimation unit 28, an azimuth angle estimation unit 29, and a clustering processing unit 30 are illustrated as processing units that constitute the reception signal processing unit 26. Here, the distance estimation unit 27 is a functional block that executes processing for estimating the distance of the target. The speed estimation unit 28 is a functional block that executes a process for estimating the speed of the target. The azimuth angle estimation unit 29 is a functional block that executes processing for estimating the azimuth angle of a target. The clustering processing unit 30 is a functional block that executes processing for specifying a target by clustering a plurality of detection results. Note that the functional blocks shown in FIG. 8 are merely examples, and other processing units (for example, a tracking processing unit that detects a temporal change in the target) may be included.

  FIG. 9 is a diagram illustrating a detailed configuration example of the transmission antenna 17 and the reception antennas 18-1 to 18-4 in the case of n = 4 in FIG. As shown in FIG. 9, the receiving antennas 18-1 to 18-4 are arranged with a plurality of patch antenna elements arranged in the y-axis direction (for example, the vertical direction), and the plurality of patch antenna elements are connected by a feeder line. Configured. The receiving antennas 18-1 to 18-4 are arranged with a predetermined distance d (d ≦ λ / 2) (λ is the wavelength of the electromagnetic wave transmitted from the transmitting antenna 17) in the x-axis direction. Further, the transmission antenna 17 is arranged at a predetermined distance from the reception antenna 18-4 in the x-axis direction. The configuration shown in FIG. 9 is an example, and the present invention is not limited to the case shown in FIG.

  FIG. 10 is a diagram showing a detailed configuration example of the azimuth angle estimation unit 29 shown in FIG. In the example shown in FIG. 10, the first angle spectrum and the second angle spectrum are calculated by multiplying the I and Q reception data of the same distance gate by two types of window functions of W (A) and W (B). Then, the azimuth angle of the target is estimated based on the first angle spectrum and the second angle spectrum.

  More specifically, the example of FIG. 10 includes window function multiplication units 281 to 284, an angle estimation processing unit 285, window function multiplication units 291 to 294, and an angle estimation processing unit 295.

  Here, the window function multipliers 281 to 284 apply WA = (WA 1, WA 2, WA 3, WA 4) = (0.5), which is the first window function, to the input I and Q reception data of the same distance gate. , 1, 1, 0.5) and outputs as the first angular spectrum A (θ, n). The window function multipliers 291 to 294 also receive the second window function WB = (WB1, WB2, WB3, WB4) = (0.5, 1, −1, −0.5) and outputs as the second angular spectrum B (θ, n). The first window function and the second window function are examples, and are not limited to the window functions described above.

Here, in the first angle spectrum A (θ, n) and the second angle spectrum B (θ, n), θ indicates an angle index, and n is a number given to the estimation result of the same distance and the same angle in the past. Indicates. For example, when 10 pieces of data acquired at regular intervals are stored, n = 0 to 9. In addition, it is arbitrary whether it becomes descending order with respect to time. Further, (real part ² + imaginary part ² ) ^0.5 is stored as values of the first angular spectrum A (θ, n) and the second angular spectrum B (θ, n).

  As a method for obtaining the first angle spectrum and the second angle spectrum, for example, (I + jQ) is obtained for each combination of the transmission antenna 17 and the reception antennas 18-1 to 18-4, and Fourier transform is performed. A first angle function (Re_A + jIm_A) in which the part is Re_A and the imaginary part is Im_A is obtained. Then, when Re_B and Im_B are obtained by converting Hilbert transform of Re_A and Im_A, (Re_B + jIm_B) is the second angle function. Then, the intensities of the obtained first angle function (Re_A + jIm_A) and second angle function (Re_B + jIm_B) are calculated, and these can be used as the first angle spectrum | Re_A + jIm_A | and the second angle spectrum | Re_B + jIm_B |. . Of course, other methods may be used.

(D) Description of Operation of Second Embodiment of Present Invention Next, the operation of the second embodiment of the present invention will be described. FIG. 11 is a diagram showing a first angle spectrum and a second angle spectrum when a target is present at an azimuth angle of 0 degrees. As shown in FIG. 11, when there is an incoming wave from a single direction, the first angle spectrum is maximum at the arrival angle of 0 degrees as shown by the solid line, and the second angle spectrum is shown by the broken line. It becomes 0 at the arrival angle of 0 degrees.

  The angle estimation based on the first angle degree spectrum is the same as the conventional Beamformer method in which the angle is estimated with the main lobe. However, in the second embodiment, by suppressing the side lobe by the window function WA, it is possible to solve the problem of the conventional method in which the interference wave is received by the side lobe. In angle estimation using the second angle spectrum, a sharp null (NULL) can be formed in the azimuth angle of the incoming wave by the window function WB. By using such a sharp null, a high angular resolution can be obtained (details will be described later).

  FIG. 12 is a diagram illustrating an angle spectrum when a reflected wave is received from a single direction. More specifically, FIG. 12A shows a case where a single target exists in the direction with an azimuth angle of 0 degrees. In FIG. 12A, the solid line indicates the first angle spectrum, and the broken line indicates the second angle spectrum. In FIG. 12A, the first angle spectrum indicated by the solid line is maximal in the direction of the azimuth angle of 0 degrees where the target exists, and the second angle spectrum indicated by the broken line is the direction of the azimuth angle of 0 degrees where the target exists. Minimal. FIG. 12B shows a case where a single target exists in a direction with an azimuth angle of −20 degrees. In FIG. 12B, the solid line indicates the first angle spectrum, and the broken line indicates the second angle spectrum. In FIG. 12B, the first angle spectrum indicated by the solid line is maximal in the direction of the azimuth angle of −20 degrees where the target exists, and the second angle spectrum indicated by the broken line is that of the azimuth angle −20 degrees where the target exists. Minimal in direction.

  FIG. 13 is a diagram showing an angle spectrum when reflected waves are received from a plurality of directions. More specifically, FIG. 13 (A) shows a case where two targets exist in directions of azimuth angles of 0 degrees and −20 degrees, the solid line shows the first angle spectrum, and the broken line shows the second angle spectrum. Show. FIG. 13A shows a state in which one target is seen due to the low resolution of the first angular spectrum even though there are two targets. At this time, at the angle estimated by the first angle spectrum (−10 degrees in this example), the second angle spectrum is not 0, and the valley portion is floating from the horizontal axis. Since such a float occurs when the number of targets is not one, it is possible to easily detect the occurrence of erroneous estimation by detecting such a float. Since such detection can be realized by the window function WB, the detection can be easily performed as compared with the technique of Patent Document 3.

  FIG. 13B shows a case where two targets are present in directions with azimuth angles of 0 degrees and −20 degrees, and the phase combination is different from that in FIG. In the example of FIG. 13B, compared to FIG. 13A, it seems that two targets can be detected separately, but the azimuth angle estimated from the peak of the first angle spectrum indicated by the solid line Is clearly wrong. Also in this case, since the valley of the second angle spectrum indicated by the broken line is floating, it is possible to detect erroneous estimation due to the interference wave. It is also a feature of the second embodiment of the present invention that the peak angle of the first angle spectrum and the angle that becomes the valley of the second angle spectrum have a deviation. That is, the occurrence of erroneous estimation can be detected by detecting such an angular deviation.

  FIG. 14 shows an angle spectrum when the target exists at azimuth angles of +30 degrees and −30 degrees. More specifically, FIG. 14A shows a case where the target exists at an azimuth angle of +30 degrees. In this example, the peak of the first angle spectrum indicated by the solid line and the valley of the second angle spectrum indicated by the broken line are shown. And are consistent. FIG. 14B shows a case where the target is present at an azimuth angle of −30 degrees. In this example, the peak of the first angle spectrum indicated by the solid line coincides with the valley of the second angle spectrum indicated by the broken line. ing. FIG. 14C shows a case where two targets exist at azimuth angles of +30 degrees and −30 degrees, respectively. In this example, the valley of the second angle spectrum indicated by the broken line is floating (the value of the valley is not 0), but the peak position of the first angle spectrum indicated by the solid line and the valley of the second angle spectrum indicated by the broken line are shown. The positions of match. For this reason, even if the valley of the second angle spectrum is floating, if the position of the peak of the first angle spectrum matches the position of the valley of the second angle spectrum, the target is located at these positions. Can be determined to exist.

  With such a feature, in the second embodiment of the present invention, there are more opportunities to adopt the estimated angle without discarding it even when an interference wave exists. This is not possible with the prior art disclosed in Patent Document 1, and it is not necessary to detect the wave number of the incoming wave unlike the method shown in the prior art disclosed in Patent Document 3. Can be implemented.

  Next, the detailed operation of the second embodiment will be described with a specific example. FIG. 15 is a diagram for describing a specific example of azimuth angle estimation in an environment where interference waves using a plurality of shot data exist. In FIG. 15, the target exists in the direction of azimuth 0 degrees and −20 degrees at a position 5 m away from the radar apparatus 10, and the radar apparatus 10 is set to 0 degrees in order to slightly change the measurement conditions for each shot. 10 shows a simulation result for 10 shots when moved in the direction of 2 mm at intervals of 2 mm. Of a plurality of shot data (n = 0 to 9), the maximum value at each θ in the first angle spectrum A (θ, n) (solid thick line in FIG. 15A) is expressed by the following equation (1). To express. MAX [] is a function for obtaining the maximum value in parentheses.

C (θ) = MAX [A (θ, n)] (1)

  FIG. 15A shows the superposition of the reflected signals from two directions, but takes various values because the reflected waves are strengthened or weakened depending on the combination of the phases of the superposition. In the case of canceling conditions, there may be a peak at an azimuth angle different from the actual one, so the maximum value MAX [] is used here.

  FIG. 15B shows, for the second angle spectrum, among the data for a plurality of hours (n = 0 to 9), the maximum value at each θ is indicated by a solid line, and the minimum value is indicated by a broken line. The maximum value can be expressed by the following formula (2), and the minimum value can be expressed by the following formula (3). Here, MIN [] is a function for obtaining the minimum value in parentheses.

D (θ) = MAX [B (θ, n)] (2)

E (θ) = MIN [B (θ, n)] (3)

  In the second angle spectrum, various values are taken depending on the phase of the received wave to be superimposed. However, since the azimuth angle of the target is a NULL azimuth, one value is always 0, so the value is Does not vary. FIG. 16A shows a case where a single target exists at a position where the phase angle is 0 degrees, and FIG. 16B shows a single target where the phase angle is −20 degrees. Shows when to do. As shown in these figures, in the phase angle of the target, since one is a NULL azimuth, one value is always 0. For this reason, in the azimuth angle of the target, F (θ) in Expression (4) described later becomes a valley.

  FIG. 15C is a diagram illustrating a difference value between the maximum value D (θ) and the minimum value E (θ). The difference value can be expressed by the following formula (4).

F (θ) = D (θ) −E (θ) (4)

  Equation (4) is minimal in the azimuth angle of the target when there are a plurality of targets. One method for detecting the minimum is to use equation (5) obtained by second-order differentiation of equation (4) with θ. However, in equation (5), in order to reduce the influence of the magnitude of the reception intensity, it is normalized by C (θ).

G (θ) = {d ² F (θ) / dθ ² } / C (θ) (5)

  FIG. 17A shows the formula (5). Then, assuming that there is a high possibility that the target exists at an azimuth angle where G (θ) exceeds the threshold Th, H (θ) is obtained by the following equation (6). Here, u [] is a step function, which is 1 when the value in parentheses ≧ 0, and is 0 when the value in parentheses <0.

H (θ) = C (θ) · u [G (θ) −Th] (6)

  FIG. 17B is a diagram of H (θ) shown in Equation (6). It is assumed that Th on the right side of Equation (6) is 0.02. As shown in this figure, the peak of the phase angle +40 degrees in G (θ) of Equation (5) becomes inconspicuous because the value of C (θ) is small, and the phase angle is 0 degrees, which is the azimuth angle of the target. Only the -20 degree peak remains.

  FIG. 18 is a diagram in which the calculation as described above is performed at each distance, converted into luminance information, and drawn. 18A shows an estimation result by C (θ), and FIG. 18B shows an estimation result by H (θ). From comparison of these figures, in FIG. 18 (B), targets are detected at 0 ° and −20 ° which are the phase angles of the targets, but in FIG. 18 (A), these targets are identified. It has not been detected as possible.

  Thus, in the second embodiment of the present invention, the size of the target is reflected to some extent, but C (θ) having a low azimuth resolution and G (θ) that strongly reflects the azimuth information of the target. By calculating the product of the difference between the threshold value and the threshold value, a high azimuth angle resolution can be obtained while reflecting the size of the target.

  In the above-described example, the displacement of the position of the radar device 10 is 18 mm in total (approximately 1.5λ in terms of wavelength), which can be realized with a very small displacement compared to the distance of 5 m to the target. It is.

  As described above, according to the second embodiment of the present invention, the problem that the resolution of the azimuth angle is low in the Beamformer method is solved by combining two window functions even in an environment where an interference wave exists. can do. Further, the main lobe is widened by suppressing the side lobe by the window function. As a result, it is possible to solve the problem that both the desired wave and the interference wave are received by the main lobe when the difference in azimuth angle between the desired wave and the interference wave is small.

  The Capon method, linear prediction method, minimum norm method, and MUSIC method, which are conventional azimuth angle estimation methods that improve the azimuth angle resolution with respect to the Beamformer method, have a number of nulls corresponding to the number of incoming waves. It is necessary to generate a large number of uncorrelated shot data in order to calculate a weight vector for generating the complicated null. However, in the second embodiment of the present invention, the window function WB generates only one null at the center of the main lobe, and angle estimation is performed in an environment where an interference wave exists with such one null. In this method, the angle can be estimated with a smaller number of shots than in the conventional estimation method.

  Further, in the second embodiment of the present invention, since a complicated null is not generated, it is possible to say that it is a stable technique because there is little obvious wrong angle estimation.

  That is, the conventional angle estimation methods Capon method, linear prediction method, minimum norm method, and MUSIC method have a problem that a large number of uncorrelated shot data are required, and Capon method, linear prediction method, minimum The norm method and the MUSIC method have a problem that many erroneous estimations occur. Further, the minimum norm method and the MUSIC method use eigenvalues and eigenvectors, so that they can deal only with fewer incoming waves than the number of antennas of the receiving antenna array. There was a problem. However, according to the second embodiment of the present invention, these problems can be solved.

  In the above, the operation when separation detection is difficult because the reflected waves from the target with a small angle difference work as interference waves is explained. Next, it is difficult to detect because it is buried in strong interference waves. An example of how to solve the problem will be described.

  As shown in FIG. 19, not only the reflected wave from the target but also the direct wave from the transmitting antenna 17 enters the receiving antennas 18-1 to 18-4 of the radar apparatus 10. In recent years, it is desirable that the distance between the transmission antenna 17 and the reception antennas 18-1 to 18-4 be short because of the demand for miniaturization of the radar apparatus 10. However, as these distances are shortened, the amplitude of the direct wave increases, and the reflected wave from the target is buried in the direct wave, which makes it difficult to detect the target.

  When detecting a target that exists far away, an FMCW radar device can cope with the problem by removing components near 0 Hz after down-conversion. Further, in the pulse type radar apparatus, the influence of the direct wave can be reduced by receiving the reflected wave after the reception of the direct wave is completed.

  However, when a target is present in the vicinity of the radar device, it has been difficult for the conventional radar device to detect even if the above-described measures are taken. For example, in the quasi-millimeter wave radar device in the 24 GHz band, the assigned frequency bandwidth is 200 MHz, and thus such a problem becomes serious within a range of about 2 m in terms of distance.

  In the second embodiment of the present invention, a target existing in the vicinity of the radar apparatus 10 can be detected even when a direct wave exists. This will be described below.

  FIG. 20 shows a case where the target is present in an environment where a direct wave exists as shown in FIG. 19 when the target is present at a position where the distance from the radar apparatus 10 is 0.5 m and the azimuth angle is −20 degrees. It is a figure which shows the case where it detects. The interval between the receiving antennas 18-1 to 18-4 is approximately λ / 2.

  FIG. 20A is a diagram showing the first angular spectrum. As shown in this figure, when the interval between the receiving antennas 18-1 to 18-4 is about λ / 2, the sign in the direction of ± 90 degrees cannot be distinguished. Therefore, in A (θ, n), ± 90 Since a reflected wave from a target having a strong peak and an amplitude of about 1/10 is buried in this peak, a peak in the −20 degree direction cannot be confirmed even in C (θ).

  FIG. 20B is a diagram illustrating the second angular spectrum. In this figure, D (θ) indicated by a solid bold line indicates the maximum value of the second angle spectrum indicated by the above-described equation (2), and E (θ) indicated by a bold bold line is the above-described equation (3). The minimum value of the 2nd angle spectrum shown by is shown.

  FIG. 20C is a diagram illustrating F (θ) according to Equation (4) of the second angle spectrum. In this figure, a valley appears around -20 degrees. FIG. 21A is a diagram illustrating G (θ) according to Equation (5), and FIG. 21B is a diagram illustrating G (θ) according to Equation (6). In FIG. 21B, a target having an azimuth angle of -20 degrees and a distance of 0.5 m is accurately detected in spite of an environment where a direct wave exists.

  FIG. 22A is a diagram illustrating an estimation result based on C (θ) related to the first angular spectrum. In the example of this figure, a target existing at an azimuth angle of −20 degrees is masked by a direct wave and cannot be discriminated.

  FIG. 22B is a diagram illustrating an estimation result based on H (θ). In the example of this figure, a target having an azimuth angle of −20 degrees and a distance of 0.5 m is detected in a distinguishable manner.

  As described above, since the direct wave from the transmitting antenna 17 to the receiving antennas 18-1 to 18-4 has a larger amplitude than the reflected wave from the target, in the conventional method, in the vicinity of the radar device 10 The reflected wave from the target existing in was buried directly in the wave and could not be detected. On the other hand, according to the second embodiment of the present invention, the buried reflected wave component can be extracted by the characteristic behavior of the second angular spectrum by the window function WB.

  In the second embodiment of the present invention, the direct wave from the transmitting antenna 17 to the receiving antennas 18-1 to 18-4, which is an unnecessary wave, does not appear in the angle estimation result. 1 to 18-4 is incident from the direction of −90 °, whereas the receiving antennas 18-1 to 18-4 do not have an angular resolution in the ± 90 degrees direction. This is because it is not reflected in (θ).

  That is, by arranging the transmitting antenna 17 at an asymmetrical position with respect to the central axis in the array direction of the receiving antennas 18-1 to 18-4, an unnecessary direct wave having a large amplitude can be transmitted to a target near the radar apparatus 10. It becomes possible to remove from the detection target.

  The above is the azimuth angle estimation method under the influence of the interference wave, which is a particular problem in the conventional Beamformer method. By adopting the angle spectrum D (θ) as the angle spectrum F (θ), the presence or absence of the interference wave can be determined. The azimuth angle can be estimated without consideration. This will be described below.

  FIG. 23 to FIG. 26 are diagrams illustrating the operation when the target is placed at the positions of 1 m and 5 m from the radar apparatus 10. More specifically, FIGS. 23 to 24 show an operation when a target is placed at a position 1 m from the radar apparatus 10 and an azimuth angle of −45 degrees. FIGS. 25 to 26 show the operation when the target is placed at a position 5 m from the radar apparatus 10 and the azimuth angle is +20 degrees.

  FIG. 23A is a diagram illustrating C (θ) related to the first angle spectrum when a target is placed at a position 1 m from the radar apparatus 10 and an azimuth angle of −45 degrees. FIG. 23B is a diagram showing D (θ) related to the second angular spectrum in the same case. FIG. 23C is a diagram showing F (θ) and D (θ) related to the second angular spectrum in the same case. FIG. 24A is a diagram showing G (θ) related to the second angular spectrum in the same case. Further, FIG. 24B is a diagram showing H (θ) related to the second angular spectrum in the same case.

  FIG. 25A is a diagram illustrating C (θ) related to the first angle spectrum when a target is placed at a position 5 m from the radar apparatus 10 and an azimuth angle of +20 degrees. FIG. 25B is a diagram showing D (θ) related to the second angular spectrum in the same case. FIG. 25C is a diagram showing F (θ) and D (θ) related to the second angular spectrum in the same case. FIG. 26A is a diagram showing G (θ) related to the second angular spectrum in the same case. Further, FIG. 26B is a diagram showing H (θ) related to the second angular spectrum in the same case.

  FIG. 27 is a diagram illustrating a detection result when the above-described two targets exist (a first target with a distance of 1 m and an azimuth angle of −45 degrees and a second target with a distance of 5 m and an azimuth angle of +20 degrees). is there. FIG. 27A shows an estimation result by C (θ) related to the first angle spectrum. In this example, the target cannot be detected behind the interference wave. On the other hand, FIG. 27B shows an estimation result by H (θ) related to the second angle spectrum. In this example, two first targets and second targets are detected at +20 degrees and −45 degrees.

  As described above, in the second embodiment of the present invention, the two first targets and the second target have different distances from the radar apparatus 10, and therefore, before the azimuth estimation unit 29 performs processing. In the distance estimation unit 27, separation is possible. For this reason, the first target is under the influence of an interference wave that is a direct wave from the transmission antenna 17 to the reception antennas 18-1 to 18-4, and the second target is an azimuth in a state without the influence of the interference wave. Angle estimation. As shown in FIG. 27, by combining two window functions, it is possible to estimate the azimuth angle of an incoming wave from a target regardless of the presence or absence of an interference wave.

(E) Description of Configuration of Third Embodiment of the Present Invention Next, a third embodiment of the present invention will be described. The circuit configuration of the third embodiment of the present invention is the same as that of FIG. 8, but the arrangement of the transmitting antenna 17 and the receiving antennas 18-1 to 18-4 is different from that of FIG. FIG. 28 is a diagram illustrating an arrangement example of the transmission antenna 17 and the reception antennas 18-1 to 18-4 in the third embodiment. In the example of FIG. 28, the receiving antennas 18-1 to 18-4 are arranged at a predetermined interval d (d ≦ λ / 2) in the x-axis direction, as in FIG. On the other hand, the transmitting antenna 17 is disposed below the receiving antennas 18-1 to 18-4 in the y-axis direction, and is offset from the center of the receiving antennas 18-1 to 18-4 indicated by broken lines. ing.

  In FIG. 9, since the transmission antenna 17 and the reception antennas 18-1 to 18-4 are arranged on the same line, the transmission signals that are strongly radiated from the transmission antenna 17 onto the plane composed of the x axis and the z axis are the same. Since the reception antennas 18-1 to 18-4 on the surface are easy to receive, the influence of the sneak signal from the transmission antenna to the reception antenna was great. However, in the third embodiment of FIG. By offsetting in the direction, wraparound to the receiving antennas 18-1 to 18-4 can be reduced. Furthermore, by arranging the transmission antenna 17 offset from the center of the reception antennas 18-1 to 18-4 in parallel to the x-axis, the influence of the sneak signal can be further reduced.

(F) Description of Operation of Third Embodiment of the Present Invention FIGS. 29 to 30 are diagrams for explaining the operation of the third embodiment with respect to the wraparound from the transmission antenna 17 to the reception antennas 18-1 to 18-4. It is. More specifically, FIG. 29A shows C (θ) related to the first angular spectrum, FIG. 29B shows D (θ) related to the second angular spectrum, and FIG. F (θ) related to the two-angle spectrum is shown. FIG. 30A shows G (θ), and FIG. 30B shows H (θ) when Th = 0.02.

  FIG. 31 shows an arrangement example of the transmission antenna 17 and the reception antennas 18-1 to 18-4 as an example for explaining the effect of the offset of the transmission antenna 17 in the third embodiment. In the example of FIG. 31, the arrangement position of the transmission antenna 17 is different from that of FIG. That is, in the example shown in FIG. 31, the transmission antenna 17 is arranged at the same position as the central axis of the reception antennas 18-1 to 18-4 indicated by a broken line in the drawing.

  FIGS. 32 to 33 are diagrams for explaining the operation with respect to the wraparound from the transmitting antenna 17 to the receiving antennas 18-1 to 18-4 in the example having no offset shown in FIG. More specifically, FIG. 32 (A) shows C (θ) related to the first angular spectrum, FIG. 32 (B) shows D (θ) related to the second angular spectrum, and FIG. F (θ) related to the two-angle spectrum is shown. FIG. 33A shows G (θ), and FIG. 33B shows H (θ) when Th = 0.02.

  29 to FIG. 30 and FIG. 32 to FIG. 33, in the example having no offset shown in FIG. 33, the object is directed in the direction of 0 azimuth by the wraparound from the transmitting antenna 17 to the receiving antennas 18-1 to 18-4. The mark is misdetected. On the other hand, in FIGS. 29 to 30, the target in the direction of the azimuth angle of 0 degrees is not erroneously detected.

  As described above, in the third embodiment of the present invention, the transmitting antenna 18 is offset and arranged at an asymmetric position with respect to the central axis of the receiving antennas 18-1 to 18-4. A direct wave having a large amplitude can be removed from the detection target as a target in the vicinity of the radar apparatus 10.

(G) Description of Modified Embodiment It goes without saying that the above embodiment is merely an example, and the present invention is not limited to the case described above. For example, in each of the above embodiments, one transmission antenna 17 and four reception antennas 18-1 to 18-4 are provided. However, two or more transmission antennas and three or less or five or more reception antennas are used. You may make it have.

  Further, in the second embodiment shown in FIG. 8, the amplification unit 19-1, the mixers 20-1 to 20-n, 21-1 to 21-n, and the LPF 22 are respectively provided for the reception antennas 18-1 to 18-4. -1 to 22-n, 23-1 to 23-n, and ADCs 24-1 to 24-n and 25-1 to 25-n, the amplification unit 19, the mixers 20 and 21, the LPF 22, 23 and ADCs 24 and 25, and any one of the outputs of the receiving antennas 18-1 to 18-4 may be selected and supplied to the amplifying unit 19.

  In each of the above embodiments, the pulse modulation type radar apparatus that transmits a pulse signal has been described as an example. However, the present invention can also be applied to an FM modulation type radar apparatus. In the case of a pulse-type radar device, the I and Q signals output from the ADCs 24, 25, 24-1 to 24-n, 25-1 to 25-n are time-series signals obtained by orthogonal demodulation of the received signal at the local frequency. The first distance function is (I + jQ), which is a signal obtained by converting the time axis into the distance axis, and the first distance spectrum is a distance axis signal having a value of | I + jQ |. On the other hand, in the case of an FM modulation type radar apparatus, the transmission signal is an FM modulation signal whose frequency is continuously modulated, and is output from the ADCs 24, 25, 24-1 to 24-n, 25-1 to 25-n. The I and Q signals are beat signals of the received signal and the frequency-modulated transmission signal, and the beat signal is converted into data in the frequency domain by Fourier transform, so that the I and Q signals corresponding to the distance are newly added. The function having the value of (I + jQ) calculated and further converted from the frequency axis to the distance axis is a first distance function, and the first distance spectrum is a distance axis signal having a value of | I + jQ |. it can.

  In addition, the first spectrum having at least one of the first distance function and the first angle function having the maximum intensity at the target position, and the first distance function and the first angle having the minimum intensity at the target position. Generating a second spectrum having at least one of a second distance function and a second angle function orthogonal to each of the functions, and estimating the position of the object based on the first spectrum and the second spectrum You may do it. More specifically, the second spectrum may be referred to determine whether or not there is an interference wave, determine a position estimation error due to the interference wave, or perform position estimation in an environment where the interference wave exists.

DESCRIPTION OF SYMBOLS 10 Radar apparatus 11 Control part 12 Modulation signal production | generation part 13 Reception antenna 14 Modulation part 15 Delay part 16 Amplification part 17 Transmission antenna 18, 18-1 to 18-4 Reception antenna 19, 19-1 to 19-n Amplification part 20, 21, 20-1 to 20-n, 21-1 to 21-n mixer 22, 23, 22-1 to 22-n, 23-1 to 23-n LPF
26 reception signal processing unit 27 distance estimation unit 28 speed estimation unit 29 azimuth angle estimation unit 30 clustering processing unit 271 first distance function calculation unit 272 first distance spectrum calculation unit 273 maximum value extraction unit 274 Hilbert filter 275 second distance function calculation Unit 276 second distance spectrum calculation unit 277 maximum value / minimum value extraction unit 278 maximum value-minimum value calculation unit 279 estimation processing unit 281-284 window function multiplication unit 291-294 window function multiplication unit 285, 295 angle estimation processing unit

Claims (10)

In a radar device that detects a target,
A transmission means for transmitting a transmission signal via a transmission antenna;
Receiving means for receiving a reflected wave transmitted by the transmitting means and reflected by the target via a receiving antenna;
Quadrature demodulation means for generating an in-phase component and a quadrature component of the transmission signal by quadrature demodulating the reception signal received by the reception means according to the frequency of the transmission signal;
A first spectrum having at least one of a first distance function and a first angle function having a maximum intensity according to the position of the target by the in-phase component and the quadrature component, and an intensity according to the position of the target Spectrum generating means for generating a second spectrum having at least one of a second distance function and a second angle function for which
Estimating means for estimating the position of the target based on the first spectrum and the second spectrum generated by the spectrum generating means;
A radar apparatus comprising:   The spectrum generating means includes a first spectrum having at least one of a first distance function and a first angle function that has a maximum intensity at the position of the target when a single target is present; The radar apparatus according to claim 1, wherein a second spectrum having at least one of a second distance function and a second angle function having a minimum intensity at the position of the target is generated. The transmission means transmits the transmission signal a plurality of times,
The spectrum generating means generates a plurality of the second spectra for the transmission signal transmitted a plurality of times,
The estimating means compares a plurality of the second spectra, and estimates a position where an intensity change is a minimum as a position of the target;
The radar apparatus according to claim 1 or 2, wherein The transmission means transmits the transmission signal a plurality of times,
The spectrum generating means generates a plurality of the second spectra for the transmission signal transmitted a plurality of times,
The estimating means estimates a position where a maximum value at each position of the plurality of second spectra is a minimum as the position of the target;
The radar apparatus according to claim 1 or 2, wherein The transmission means generates the transmission signal by modulating a local oscillation signal having a predetermined frequency with a pulsed baseband signal,
The quadrature demodulation means generates the I component that is the in-phase component and the Q component that is the quadrature component by performing quadrature demodulation on the received signal with the local oscillation signal,
The spectrum generating means includes
A function that has a value proportional to (I + jQ) as the first distance function and uses a function in which the time axis is converted into the distance axis as the first distance function, and indicates the strength of the first distance function as the first distance spectrum. Use
When the value obtained by Hilbert transform of (I + jQ) is (I ′ + jQ ′) as the second distance function, the value is proportional to (I ′ + jQ ′) and the time axis is converted to the distance axis A function is used, and a function indicating the intensity of the second distance function is used as the second distance spectrum.
The radar apparatus according to any one of claims 1 to 4, wherein the radar apparatus is characterized in that: The transmission means generates an FM modulation signal whose frequency continuously changes as the transmission signal,
The quadrature demodulation means generates a beat signal having a frequency difference between the reception signal and the transmission signal, and transforms the beat signal into a frequency domain by Fourier transform, whereby the I component that is the in-phase component and the quadrature component And a Q component that is
The spectrum generating means includes
a function that has a value proportional to (I + jQ) as the first distance function and uses a function in which the frequency axis is converted to the distance axis, and indicates the intensity of the first distance function as the first distance spectrum, where j is an imaginary unit Use
As the second distance function, when the value obtained by Hilbert transform of (I + jQ) is (I ′ + jQ ′), the value is proportional to (I ′ + jQ ′) and the frequency axis is converted to the distance axis. A function is used, and a function indicating the intensity of the second distance function is used as the second distance spectrum.
The radar apparatus according to any one of claims 1 to 4, wherein the radar apparatus is characterized in that: A plurality of at least one of the transmitting antenna and the receiving antenna;
The spectrum generating means includes
j is an imaginary unit, (I + jQ) based on the combination of the transmission antenna and the reception antenna is an array, and Fourier transform is performed to generate the first angle function on the angle axis,
When the real part and the imaginary part of the first angle function are Re_A and Im_A, and the result of Hilbert transform of Re_A and Im_A is Re_B and Im_B, the second angle function is an angle proportional to (Re_B + jIm_B) The second angle spectrum is an angle axis signal reflecting the intensity of the second angle function.
The radar apparatus according to claim 1, wherein   The radar apparatus according to claim 7, wherein an odd function window function is used when calculating the second angle function from an array of (I + jQ). A plurality of receiving antennas and arranged side by side in a first direction;
The transmitting antenna is disposed at a position shifted from the center of the array of the plurality of receiving antennas.
The radar apparatus according to claim 1, wherein In a target detection method of a radar device that detects a target,
A transmission step of transmitting a transmission signal via a transmission antenna;
A receiving step of receiving the reflected wave transmitted by the transmitting step and reflected by the target via a receiving antenna;
A quadrature demodulation step for generating an in-phase component and a quadrature component of the transmission signal by performing quadrature demodulation on the reception signal received by the reception step according to the frequency of the transmission signal;
A first spectrum having at least one of a first distance function and a first angle function having a maximum intensity according to the position of the target by the in-phase component and the quadrature component, and an intensity according to the position of the target Generating a spectrum having a second spectrum having at least one of a second distance function and a second angle function having a minimum;
An estimation step for estimating a position of the target based on the first spectrum and the second spectrum generated by the spectrum generation step;
A target detection method for a radar apparatus, comprising:

Images