JP2019154206A - LLC resonant circuit and power conversion device including the same - Google Patents

Abstract

To provide an LLC resonance circuit capable of controlling an output voltage in a wide range.SOLUTION: An LLC resonance circuit 1 includes: a resonance unit 10 that receives a DC input voltage and outputs a voltage; a conversion unit 20 that converts the voltage output from the resonance unit into a predetermined DC output voltage and outputs the voltage. The conversion unit 20 includes: a boost capacitor C21 that forms a secondary winding of a transformer TR1 and a closed loop circuit 21; and a rectifier diode D21. Furthermore, a first switching element Q21 for controlling a charging voltage of the boost capacitor C21 by turning on and off in response to a predetermined control signal is provided in the closed loop circuit 21.SELECTED DRAWING: Figure 1

Description

  The present invention relates to an LLC resonant circuit and a power conversion device including the same.

  2. Description of the Related Art In recent years, in power converters, in order to reduce the cost of a DC power supply, it is required to use a DC power supply with a low voltage and large current DC power. In the power conversion device, if the specified output power can be obtained at a low input voltage, there is a problem that it becomes difficult to control the output voltage at a high input voltage and a low load. Therefore, Patent Document 1 describes that in a full-bridge composite resonance type DC-DC converter, the full-bridge composite resonance circuit is shifted to burst oscillation at light load.

JP 2011-142765 A

  However, when burst oscillation is performed, ON / OFF control is repeated for a high input voltage, so that noise is generated, which may cause deterioration of EMI (Electromagnetic interference), malfunction of the control system, and the like.

  The present invention has been made in view of the above problems, and an object of the present invention is to provide an LLC resonance circuit capable of controlling an output voltage over a wide range without causing burst oscillation.

  An LLC resonant circuit according to a first aspect of the present invention includes a switching unit, a resonant capacitor, and a transformer, receives a DC input voltage at the switching unit, and outputs an output of the switching unit via the resonant capacitor. A resonance part that is fed to and resonates with the primary winding, a booster capacitor that forms a closed loop circuit with the secondary winding of the transformer, and a rectifier diode, and converts the output voltage of the resonance part into a predetermined DC output voltage. The closed loop circuit is provided with a first switching element that controls a charging voltage of the boost capacitor by turning on and off in response to a predetermined control signal. .

  A power conversion device according to a second aspect of the present invention includes the LLC resonant circuit according to the first aspect and a control circuit that controls on / off of the first switching element.

  In the first and second aspects, since the first switching element is provided in the closed loop circuit of the conversion unit, the charging voltage of the boost capacitor can be controlled by turning on and off the first switching element. Thereby, the output voltage of an LLC resonance circuit and a power converter provided with the same (hereinafter referred to as an LLC resonance circuit (power converter)) can be adjusted to an arbitrary voltage. Specifically, for example, if the first switching element is kept on, a voltage doubled from the secondary winding voltage of the transformer is output. On the other hand, if the first switching element is kept off, the secondary winding voltage of the transformer and the charging voltage of the boost capacitor are canceled, and the output voltage from the LLC resonance circuit (power converter) is almost “0 (zero). )"become. That is, with the configuration as in this aspect, the output voltage of the LLC resonant circuit (power converter) can be controlled in a wide range from a substantially zero state to a desired output voltage. In other words, by turning on and off the first switching element, it is possible to stably output a preset output voltage for a wide range of input voltages and loads.

  According to the present invention, it is possible to control the output voltage while maintaining a stable resonance operation over a wide range of input voltages and loads.

It is a figure which shows the circuit structural example of a power converter device. It is a figure for demonstrating operation | movement of a power converter device. It is a figure for demonstrating operation | movement of a power converter device. It is a figure for demonstrating operation | movement of a power converter device. It is a wave form diagram which shows the operation example of the power converter device of FIG. It is a wave form diagram which shows the other operation example of the power converter device of FIG. It is a figure which shows the relationship between the switching frequency and output voltage of a general LLC resonant circuit. It is a figure which shows the relationship between the timing which the switching element of the power converter device of this embodiment turns on, and an output voltage. It is a figure which shows the other circuit structural example of a power converter device. It is a wave form diagram which shows the operation example of the power converter device of FIG.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. The following description of the preferred embodiments is merely exemplary in nature and is not intended to limit the invention, its scope of application, or its application.

  FIG. 1 is a diagram illustrating a circuit configuration example of the power conversion device A according to the present embodiment.

  The power conversion apparatus A according to the present embodiment receives a DC input voltage Vi (hereinafter referred to as power supply voltage Vi) from the DC power supply E, and converts it into a predetermined DC output voltage Vo (hereinafter simply referred to as output voltage Vo). This is a power conversion device using LLC resonance that is output to a load RL. This power converter A is characterized in that it can output a desired output voltage Vo while maintaining a stable resonance operation with respect to a wide range of input voltage fluctuations and / or load fluctuations.

  Hereinafter, each component will be specifically described.

-Configuration of power converter-
As shown in FIG. 1, the power conversion device A includes an LLC resonance circuit 1 and a control circuit 4 that controls the operation of the LLC resonance circuit 1. In the present embodiment, the LLC resonance circuit refers to a resonance circuit that utilizes the leakage inductance and excitation inductance of the transformer and the resonance of the capacitor.

  The LLC resonant circuit 1 receives a power supply voltage Vi at an input terminal IN, converts it to alternating current and outputs it, and converts the output of the resonant part 10 into a predetermined output voltage Vo and outputs it from the output terminal OUT. The unit 20 is provided. In the following description, when the input terminal IN is described separately on the positive electrode side and the negative electrode side, it may be described with reference numerals INP and INM. Similarly, when the output terminal OUT is described separately on the positive electrode side and the negative electrode side, it may be described with OUTP and OUTM, respectively.

  The resonance unit 10 includes a switching unit 11, resonance capacitors C11 and C12, and a transformer TR1.

  The switching unit 11 includes first and second switching elements Q11 and Q12 provided in series between a positive input terminal INP and a negative input terminal INM. FIG. 1 shows an example in which N-type MOS-FETs are used as the first and second switching elements Q11 and Q12. Further, FIG. 1 also shows a parasitic diode formed in parallel with the MOS-FET. In the parasitic diode, a current from the source to the drain of the MOS-FET passes through the parasitic diode. Note that the configuration of the switching unit 11 is not limited to the configuration of FIG. 1, and other switching circuits having the same function may be applied. For example, a high-speed diode element may be externally attached to the MOS-FET so as to be connected in parallel with the parasitic diode.

  The resonance capacitors C11 and C12 include a current resonance capacitor (hereinafter referred to as a first resonance capacitor C11) and a voltage resonance capacitor (hereinafter referred to as a second resonance capacitor C12). The second resonance capacitor C12 is connected in parallel to the first switching element Q11. Similarly, a series circuit of the first resonance capacitor C11 and the primary winding of the transformer TR1 is connected in parallel to the first switching element Q11.

  In other words, the positive-side input terminal INP is connected to one end (for example, the winding start end) of the primary winding of the transformer TR1 via the first resonance capacitor C11. The other end (for example, the winding end) of the primary winding of the transformer TR1 is connected to the negative-side input terminal INM via the second switching element Q12 to form the first current path Ia (see FIG. 2C). is doing. Further, the first switching element Q11, the first resonance capacitor C11, and the primary winding of the transformer TR1 form a closed loop circuit, which forms the second current path Ib (see FIG. 2A).

  In the present disclosure, the term “connection” includes all electrical connections. In other words, the “connection” is not limited to a direct connection, but includes a concept including an electrical connection via, for example, a resistance element or a semiconductor element.

  The conversion unit 20 is a circuit that converts the voltage output from the resonance unit 10 into a predetermined output voltage Vo and outputs the voltage. Specifically, the converter 20 includes a second capacitor C21 and a diode D22 connected in series between one end (for example, the winding start end) of the secondary winding of the transformer TR1 and the positive output terminal OUTP. The diode D22 is connected in the forward direction from one end of the secondary winding of the transformer TR1 toward the positive output terminal OUTP. The other end (for example, the winding end) of the secondary winding of the transformer TR1 is connected to the negative output terminal OUTM.

  The converter 20 further includes a rectifier diode D21 connected in series between the intermediate node N21 between the second capacitor C21 and the diode D22 and the output terminal OUTM on the negative electrode side, and charge control as the first switching element. Switching element Q21 (hereinafter referred to as charge control switch Q21). The rectifier diode D21 is connected in a direction opposite to the direction from the intermediate node N21 toward the negative output terminal OUTM. An output capacitor C22 is connected between the output terminals OUT.

  In other words, in the converter 20, a closed loop circuit 21 (third current path Ic (see FIG. 2B)) is formed by the secondary winding of the transformer TR1, the second capacitor C21, the rectifier diode D21, and the charge control switch Q21. is doing. Further, the positive output terminal OUTP is connected to the output terminal OUTM via the second capacitor C21 and the secondary winding of the transformer TR1, thereby forming a fourth current path Id (see FIG. 2C). is doing.

  The control circuit 4 controls the operation of the entire power conversion apparatus A, and can be realized by, for example, an IC (Integrated Circuit). The control circuit 4 operates, for example, according to a program or sequence registered in advance in the own device. More specifically, the control circuit 4 includes, for example, control signals Vg11, Vg12, ON / OFF control for the first and second switching elements Q11, 12 of the switching unit 11 and the charge control switch Q21 of the conversion unit 20, respectively. Vg21 is output.

-Operation of power converter-
Next, operation | movement of the power converter device A is demonstrated concretely, referring FIG. 2A-FIG. 2C, FIG. 3, FIG. 2A to 2C schematically show the ON / OFF state of the switching element and the current flowing direction at each timing. 3 and 4 each show an example of a timing chart.

  Each switching element Q11, Q12, Q21 will be described as being turned on when the signal waveform of FIGS. 3 and 4 is “HIGH” and turned off when “LOW”. 2A to 2C, the MOS-FETs that constitute the switching elements Q11, Q12, and Q21 are indicated by simple contact symbols for easy understanding of the description. Further, it is assumed that a series of processes from time Ta to Td is performed before time Ta described below.

  For convenience of explanation, the directions of the primary side current I1 and the secondary side current I2 are defined as follows. The primary-side current I1 is defined as “forward” in the direction from the positive-side input terminal INP to the negative-side input terminal INM (the direction from the winding start side to the winding end side of the primary winding of the transformer TR1). The opposite is the “reverse direction”. Similarly, in the conversion unit 20, the direction from the positive output terminal OUTP to the negative output terminal OUTM (the direction from the winding start side to the winding end side of the secondary winding of the transformer TR1) is “forward direction”, The opposite is the “reverse direction”.

  Furthermore, the current flowing through the switching element Q11 is described as IDH, and the current flowing through the switching element Q12 is described as IDL. The primary side current I1 is a current that flows in the primary side circuit of the power conversion device A including the resonance unit 10, and the secondary side current I2 is the secondary of the power conversion device A including the conversion unit 20. This is the current flowing in the side circuit.

(Operation during period Ta)
First, operation | movement of the power converter device A in the period Ta of FIG. 3 is demonstrated.

  At time T31, which is the start point of the period Ta in FIG. 3, the control circuit 4 turns on the first switching element Q11 (Vg11) and turns off the second switching element Q12 (Vg12).

  Specifically, in the period Td before the period Ta, in the resonance unit 10, a current flows from the source side to the drain side of the first switching element Q11 due to a resonance phenomenon. That is, the current IDH flowing through the first switching element Q11 is negative. The control circuit 4 turns on the first switching element Q11 while the current IDH is negative. Here, the time when the first switching element Q11 is turned on is T31.

  Thereafter, the current IDH changes to a plus, and the primary side current I1 in the direction indicated by the arrow in FIG. 2A flows through the second side current path Ib in the primary side circuit of the power converter A. The primary current I1 at this time is a resonance current between the resonance capacitors C11 and C12 and the transformer TR1, and is a current that changes in a sine wave shape (see FIG. 3). As a result, a primary winding voltage V1 in the forward direction (downward in FIG. 2) is induced in the primary winding of the transformer TR1.

  In the converter 20, the secondary winding voltage V2 in the reverse direction (downward in FIG. 2) is induced in the secondary winding of the transformer TR1. However, in the period Ta, since the charge control switch Q21 of the conversion unit 20 is controlled to be off, no current flows through the third current path Ic (closed loop circuit 21). Further, since the current is blocked by the diode D22, no current flows through the fourth current path Id. Therefore, the secondary side current I2 in this period is “0 (zero)”.

(Operation during period Tb)
Next, operation | movement of the power converter device A in the period Tb of FIG. 3 is demonstrated.

  The control circuit 4 turns on the charge control switch Q21 (Vg21) at time T32 when a predetermined time Ta has elapsed from time T31 when the first switching element Q11 is turned on (when Vg11 changes from Low to High). At this time, the control circuit 4 continues the on control of the first switching element Q11 and the off control of the second switching element Q12. Then, the current IDH flowing through the first switching element Q11 further increases, and accordingly, the current flowing through the second current path Ib (see the arrow in FIG. 2A) increases. Thereby, in the converter 20, the secondary winding voltage V2 is continuously induced.

  In the converter 20, a reverse current (secondary current I2) flows through the third current path Ic by the secondary winding voltage V2 (see the secondary arrow in FIG. 2B).

  Due to the reverse current, the second capacitor C21 is charged with electric charge according to the ON period of the charge control switch Q21. That is, the second capacitor C21 is charged to a charging voltage corresponding to the ON period of the charging control switch Q21. At this time, the secondary current I2 continues to rise until the first switching element Q11 is turned off. That is, the secondary current I2 continues to increase during the period Tb. On the other hand, the primary current I1 gradually decreases toward “0 (zero)” during the period Tb.

(Operation during period Tc)
Next, operation | movement of the power converter device A in the period Tc of FIG. 3 is demonstrated.

  The control circuit 4 turns off the first switching element Q11 at time T33 when the period Tb has elapsed since turning on the charging control switch Q21. Then, the secondary current I2 flowing through the third current path Ic gradually decreases and becomes “0 (zero)” at time T34 when the period Tc has elapsed from time T33.

  In summary, the period during which the secondary current I2 flows is defined by the first timing when the charge control switch Q21 is turned on and the second timing when the first switching element Q11 is turned off. Therefore, by controlling the first and second timings, it is possible to control the time (Tb + Tc) during which the secondary current I2 flows, and as a result, the charging voltage of the second capacitor C21 is controlled. be able to.

(Operation during period Td)
Next, operation | movement of the power converter device A in the period Td of FIG. 3 is demonstrated.

  At time T34, the control circuit 4 turns on the second switching element Q12. In addition, the control circuit 4 maintains the off state of the first switching element Q1 of the resonance unit 10 and maintains the on state of the charge control switch Q21. In FIG. 3, the time when the secondary current I2 flowing through the third current path Ic becomes “0 (zero)” and the time when the second switching element Q12 is turned on are both T34, which will be described later. As shown in FIG. 4, both times may be different from each other.

  The timing at which the control circuit 4 turns on the second switching element Q12 is, for example, before the current in the reverse direction is in the direction indicated by the arrow in FIG. 2C, that is, the current flowing through the first current path Ia (primary current I1 ) Before turning in the forward direction.

  In FIG. 3, the third timing at which the secondary current I2 becomes “0 (zero)” and the fourth timing at which the control circuit 4 turns on the second switching element are aligned. However, the present invention is not limited to this. . For example, in the example of FIG. 4 described later, an example in which the third timing and the fourth timing are different from each other is shown.

  Subsequently, in the resonating unit 10, a reverse current (see the arrow on the primary side in FIG. 2B) flows through the first current path Ia. The primary current I1 is a resonance current of the resonance capacitor C11, the boost capacitor C21, and the transformer TR1, and is a current that changes in a sine wave shape (see FIG. 3).

  When the secondary-side current I2 becomes “0 (zero)”, the primary winding voltage V1 in the reverse direction (upward in FIG. 2B) is continuously induced in the primary winding of the transformer TR1, and 2 of the transformer TR1. A secondary winding voltage V2 in the reverse direction (upward in FIG. 2B) is induced on the secondary side.

  Then, in the conversion unit 20, a forward current (see the secondary arrow in FIG. 2C) flows through the fourth current path Id. At this time, since the charge previously charged in the boost capacitor C21 in the period Tb is also discharged, the output voltage Vo rises by an amount corresponding to the charged charge (charge voltage).

  Next, the control circuit 4 turns off the second switching element Q12 and the charging control switch Q21 at time T36. As a result, in the resonance unit 10, due to the resonance action of the resonance capacitor C11, the boost capacitor C21, and the transformer TR1, the arrow shown in FIG. 2C is passed through the parasitic diode of the first switching element Q11 to the first current path Ia. And reverse current flows.

  At this time, in the conversion unit 20, the current in the fourth current path Id shown in FIG. 2C starts to flow from the time (time T34) when the secondary current I2 becomes “0 (zero)” in the period Tc. Then, when the second switching element Q12 is turned off at time T36, the secondary current I2 becomes “0 (zero)” in a short time (see between time T36 and time T37 in FIG. 3).

  In summary, in this embodiment, in the converter 20, the charge control switch Q21 is provided in the closed loop circuit 21 formed by the secondary winding of the transformer TR1, the boost capacitor C21, and the rectifier diode D21. By providing such a charging control switch Q21, the charging voltage of the boost capacitor C21 can be controlled. Specifically, by changing the time Ta until the charge control switch Q21 is turned on and the time (Tb + Tc) during which the charge control switch Q21 is turned on, the amount of charge charged in the boost capacitor C21 and thus the charge voltage of the boost capacitor C21 is controlled. Can do. Since the boost capacitor C21 is a capacitor that is discharged when the output voltage Vo is boosted, the output voltage Vo of the power converter A can be controlled by controlling the charge amount of the boost capacitor C21.

  In the example of FIG. 3, since the secondary current I2 does not turn positive within the period of Td, the ON period of the second switching element Q12 is longer than the time (Tb + Tc), and the time when the next cycle starts It can be arbitrarily set in the period up to T37 (T31). When the secondary current I2 turns to positive, a counter electromotive force may be generated, but this is not the case in the present embodiment.

-Control of output voltage Vo-
Hereinafter, the control of the output voltage Vo of the power conversion device A will be described more specifically.

  FIG. 5 is a graph showing the relationship between the switching frequency and the output voltage of a general LLC resonant circuit. Specifically, the LLC resonance circuit whose characteristics are illustrated in FIG. 5 is one in which the charge control switch Q21 is not provided in the closed loop circuit 21 of the conversion unit 20 as in this embodiment.

  In FIG. 5, what is plotted with a triangle mark is the characteristic when “low input voltage and heavy load”, and what is plotted with a circle mark is the characteristic when “high input voltage and light load”. As can be seen from FIG. 5, when the input voltage increases and the load decreases, the characteristic curve moves upward.

  For example, FIG. 5 shows an example in which the specified output voltage is V21 and the output voltage V21 is designed to be stably output at “high input voltage and light load (circle)”. In the case of such a design, control cannot be performed so that the output voltage V21 becomes a specified value at “low input voltage and heavy load” indicated by a triangle mark in FIG.

  On the other hand, it is conceivable that the output voltage V21 can be output stably with “low input voltage and heavy load (triangle mark)”, but in that case, it is defined by “high input voltage and light load (circle mark)”. The output voltage V21 cannot be controlled. That is, in a general LLC resonance circuit, it is very difficult to control so that a specified output voltage Vo can be output with respect to a wide range of input voltage fluctuations and load fluctuations.

  For example, in a power generation device using natural energy such as solar power generation, there is a possibility that the input voltage input to the power conversion device may greatly come from a low voltage to a high voltage due to environmental factors such as solar radiation. Moreover, there is a possibility that the load state may vary greatly. For example, when it is assumed that the load is installed in a consumer, there is a possibility of a wide change from a light load state that hardly consumes power to a heavy load state that consumes a large amount of power. Then, a problem as shown in FIG. 5 occurs in a general LLC circuit. On the other hand, in the power conversion device A according to the present embodiment, by changing the length of the predetermined time (Tb + Tc) that is a period during which the charge control switch Q21 is on-controlled, by controlling the charging voltage of the boost capacitor C21, The output voltage value can be controlled.

  FIG. 6 is a diagram illustrating the relationship between the output voltage and the timing at which the charging control switch Q21 of the conversion unit 20 is turned on (indicated as phase in FIG. 6) in the power conversion device A according to the present embodiment. FIG. 6 shows an example in which the switching frequency is fixed to 60 [kHz] and the timing at which the charging control switch Q21 is turned on is changed. The duty ratio between the first switching element Q11 and the second switching element Q12 is set to 0.5.

  In FIG. 6, the phase 0 [degree] indicates the timing when the charging control switch Q <b> 21 is turned on simultaneously with the first switching element Q <b> 11 of the resonance unit 10. The phase 180 [degrees] indicates the timing at which the secondary winding voltage V2 switches from the reverse direction to the forward direction. As shown in FIG. 6, the output voltage Vo can be changed from V21 to almost “0 (zero)” by changing the phase, that is, the timing when the charging control switch Q21 is turned on (in FIG. 5, the range D1). Equivalent). As described above, since the output voltage can be controlled over a wide range at the timing when the charging control switch Q21 is turned on, it is adjusted so that a desired output voltage can be supplied to the load even when an input voltage fluctuation or a load fluctuation occurs. It becomes possible to do.

  The case where the output voltage Vo is substantially “0 (zero)” is when the charge control switch Q21 is always set to OFF. This will be specifically described below.

  First, in the period Td in FIG. 3, the control circuit 4 turns off the first switching element Q11 and turns on the second switching element Q12. Then, as shown in FIG. 2C, the secondary winding voltage V2 in the forward direction (upward in FIG. 2) is induced, and the boost capacitor C21 is charged in the reverse direction (leftward in FIG. 2).

  Next, in the period Ta to the period Tc, the control circuit 4 turns on the first switching element Q11 and turns off the second switching element Q12. Then, as shown in FIG. 2A, the secondary winding voltage V2 in the reverse direction (downward in FIG. 2) is induced. However, since the charge control switch Q21 is off-controlled, no current flows through the third current path Ic, and no current flows through the fourth current path Id since it is blocked by the diode D22. State is maintained.

  Thereafter, when the period Td in FIG. 3 is reached, the secondary winding voltage V2 in the forward direction (upward in FIG. 2) and the charging voltage of the boost capacitor C21 cancel each other and are canceled, and the output voltage Vo is substantially “ 0 (zero) ".

-Operation under no load-
The LLC resonant circuit 1 (power converter A) in the present embodiment has a feature that it can control the output voltage while maintaining a stable and stable resonant operation even in a no-load state.

  Hereinafter, a specific description will be given with reference to FIG. In FIG. 4, the time from when the control circuit 4 turns on the first switching element Q11 to when the charge control switch Q21 is turned on is T1, the period during which the secondary current I2 flows is T2, and the charge control switch Q21 The on period will be described as T3.

  As shown in FIG. 4, when there is no load, the time T1 that defines the timing for turning on the charging control switch Q21 is different from that in FIG. Specifically, the time T1 is longer than the ON period of the first switching element Q11, or the time when the second half of the ON period of the first switching element Q11 and the first half of the ON period of the charge control switch Q21 slightly overlap. It is preferable to set to.

  Moreover, it is preferable to set the ON period T3 of the charging control switch Q21 to be longer than the period T2 in which the secondary current I2 turns to plus. This is because if the ON period T3 of the charging control switch Q21 is shorter than the above-described period T2, back electromotive force may be generated in the secondary winding voltage V2. Furthermore, it is preferable to set the timing for turning off the charging control switch Q21 before turning the secondary current I2 from the reverse direction to the forward direction. Thereby, it is possible to avoid the generation of the counter electromotive force in the secondary winding voltage V2.

(Other embodiments)
Although the embodiments of the present invention have been described above, various modifications can be made.

  For example, in the example of FIG. 3, the second switching element Q12 and the charging control switch Q21 are simultaneously turned off at time T36, but the present invention is not limited to this. Specifically, the timing for turning off the charging control switch Q21 is after time T34, and can be arbitrarily set until the current flowing in the third current path Ic (secondary current I2) turns in the forward direction. Is possible. This is because a counter electromotive force may be generated in the secondary winding voltage if it is turned off after turning to plus. However, the timing for turning off the charging control switch Q21 may be set after the secondary current I2 turns in the forward direction. In this case, a counter electromotive force is generated in the switching element Q21 of the closed loop circuit 21, but for example, by increasing the breakdown voltage of the switching element Q21, it is possible to prevent the influence of the counter electromotive force.

  Moreover, you may provide the regeneration part 3 in the power converter device A so that it may illustrate in FIG. The regenerative unit 3 is a circuit for regenerating the back electromotive force generated in the secondary winding of the transformer TR1 to the primary side input unit, and a high voltage resulting from the back electromotive force is applied to the charge control switch Q21. Can be prevented.

  In the power conversion device A of FIG. 7, the regenerative unit 3 is between the intermediate node N22 between one end of the secondary winding of the transformer TR1 and the boost capacitor C21 and the other end of the secondary winding of the transformer TR1. Connected.

  The regenerative unit 3 includes a transformer TR3 having one end of a primary winding connected to the intermediate node N22 via a capacitor C31, and a switching element Q32 connected between the other end of the transformer TR3 and the intermediate node N22. It has. Furthermore, between one end of the primary winding of the transformer TR3 and the other end of the secondary winding of the transformer TR1, a diode D31 connected in the reverse direction and a switching element Q31 as a second switching element are connected in series. It is connected.

  One end of the secondary winding of the transformer TR3 of the regenerative unit 3 is connected to the positive input terminal INP via the diode D32 connected in the forward direction, and the other end is connected to the negative input terminal INM. Yes. The transformer TR3, the switching element Q32, and the diode D32 constitute a flyback power source.

  In addition to the control signals Vg11, Vg12, and Vg21, the control circuit 4 outputs control signals Vg31 and Vg32 for on / off control of the switching elements Q31 and Q32, respectively.

  Other configurations are the same as those in FIG. 1, and detailed description thereof is omitted here.

  Next, the operation of the power conversion device A of FIG. 7 will be specifically described with reference to FIG. In FIG. 8, times Th to Tk correspond to times Ta to Td in FIG. 3, respectively. In the following description, differences from FIG. 3 will be mainly described.

  In the period Th of FIG. 8, “HIGH” is given to the switching element Q31 as the control signal Vg31 at a time just before the charge control switch Q21 is turned off ((start time of the period Th) + (T11: on period of Q21) −T13). To turn on the switching element Q31. Thereby, even if the charging control switch Q21 is turned off, the secondary current I2 flows to the capacitor C31 without interruption. At this time, the switching element Q31 is turned on until charging of the capacitor C31 is completed, that is, until the charging current becomes zero. As a result, no back electromotive force is generated in the secondary winding of the transformer TR1.

  This time, it is necessary to discharge the charge charged in the capacitor C31 before the start of charging of the capacitor C31 in the next cycle. Therefore, the control circuit 4 causes the charging voltage of the capacitor C31 to be given as the input voltage of the flyback power supply (transformer TR3, switching element Q32 and diode D32). Specifically, the control circuit 4 determines that after the switching element Q31 is turned off, that is, after the on period T12 of the control signal Vg31 has elapsed, until the next switching element Q11 is turned on (until the next period Th starts). In the period, the switching element Q32 is turned on, and the electric charge accumulated in the capacitor C31 is regenerated at the input part of the primary side circuit of the LLC resonant circuit 1.

  As described above, since the regenerative unit 3 is provided, the back electromotive force generated by turning off the charge control switch Q21 is regenerated to the input terminal IN of the power conversion device A via the transformer TR3. It is possible to prevent back electromotive force from being applied to Q21.

  According to the present invention, the output voltage of the LLC resonant circuit can be controlled in response to a wide range of input voltage fluctuations and load fluctuations, which is extremely useful.

DESCRIPTION OF SYMBOLS 10 Resonant part 11 Switching circuit 20 Rectifier 21 Closed loop circuit C11, C12 Resonance capacitor TR1 Transformer Q21 Switching element (first switching element)
C21 Boost capacitor D21 Rectifier diode

Claims (9)

A resonance unit having a switching unit, a resonance capacitor, and a transformer, receiving a DC input voltage at the switching unit, and applying an output of the switching unit to the primary winding of the transformer through the resonance capacitor to resonate; ,
Including a step-up capacitor and a rectifier diode that form a closed loop circuit with the secondary winding of the transformer, and a conversion unit that converts the output voltage of the resonance unit into a predetermined DC output voltage and outputs the converted voltage.
The LLC resonant circuit, wherein the closed loop circuit is provided with a first switching element that controls a charging voltage of the boost capacitor by being turned on and off in response to a predetermined control signal. The LLC resonant circuit according to claim 1,
The boost capacitor is provided between one end of the secondary winding of the transformer and one output terminal of the LLC resonant circuit,
The LLC resonant circuit, wherein the rectifier diode and the first switching element are connected in series between a node on the output side of the boost capacitor and the other end of the secondary winding. The LLC resonant circuit according to claim 1,
A regenerative unit for regenerating the voltage of the closed loop circuit to the switching unit;
An LLC resonance circuit, comprising: a second switching element that controls whether or not to operate the regeneration unit and that is turned on in accordance with a timing at which the first switching element is turned off. The LLC resonant circuit according to claim 3,
The LLC resonance circuit, wherein the regeneration unit is connected between an input side node of the boost capacitor and the other end of the secondary winding. The LLC resonant circuit according to any one of claims 1 to 4,
And a control circuit that controls on / off of the first switching element. The power conversion device according to claim 5, wherein
The control circuit controls the amount of charge to the boost capacitor by adjusting a time from the discharge start time of the resonant capacitor to the on-control of the first switching element. The power conversion device according to claim 6, wherein
The power converter according to claim 1, wherein the control circuit controls the first switching element to be turned off after the current of the closed loop circuit is stopped. In claim 5,
The control circuit adjusts a period during which the first switching element is on-controlled according to the DC input voltage and / or a load connected to the converter. In claim 5,
The power converter according to claim 1, wherein the control circuit turns on the first switching element when an output voltage of the LLC resonant circuit reaches a predetermined control voltage.

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