JP2018014841A - Voltage converter, step-down control method of power conversion circuit, step-up control method of power conversion circuit and computer program - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a power converter capable of reducing the size of a core around which two inductors in a voltage conversion circuit connected in parallel and being driven with phases different from each other, and to provide a step-down control method of the voltage conversion circuit, step-up control method of the voltage conversion circuit and a computer program.SOLUTION: In a voltage converter, SW elements Q11, Q21, inductors L11, L21, and SW elements Q12, Q22 connected with capacitors C1, C2 at the opposite ends are connected, in this order, with voltage conversion circuits 1, 2 connected in parallel, and one ends of inductors L12, L22 are connected with the junction point of the inductors L11, L21, and SW elements Q12, Q22. Two inductors L12, L22 are wound around the core 3 to be cancelled magnetically each other, and when a control section 5 turns the SW elements Q11, Q21 on/off alternately, the voltages inputted to terminals H, G are stepped down, and outputted in parallel to terminals L, G.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a voltage conversion device in which two voltage conversion circuits for converting a DC voltage into voltage are connected in parallel, a step-down control method for the voltage conversion circuit, a step-up control method for the voltage conversion circuit, and a computer program.

  A DC-DC converter (hereinafter simply referred to as a converter) that steps up and down a DC voltage is widely used as a power source for in-vehicle devices and industrial devices. In response to the demand for miniaturization of power supplies, the operating frequency of converters tends to be increased in order to make it possible to use passive components such as inductors and capacitors having a small volume. On the other hand, another problem that the switching loss of the switching element that switches the current flowing through the inductor increases as the operating frequency increases.

  On the other hand, Patent Document 1 discloses a converter that performs zero-current switching that switches a semiconductor switching element (switching element) from on to off when a resonance current flowing in a resonance circuit including a reactor (inductor) and a capacitor is inverted. Is disclosed. By performing such zero current switching, the switching loss of the switching element that switches the current flowing through the inductor is reduced.

  On the other hand, in Patent Document 2, two switching circuits for controlling on / off of input currents are operated in multiphase at phases different from each other by 180 degrees, and output currents of the respective switching circuits are multiplexed and synthesized. A technique is disclosed in which a ripple suppression effect equivalent to that obtained when switching is performed at a frequency substantially higher than the switching frequency of the switching circuit is obtained.

JP 2002-262552 A JP 2002-44941 A

  However, even when the techniques disclosed in Patent Documents 1 and 2 are combined, an inductor having a size corresponding to the operating frequency is still required, and the size of the inductor is still different from that of other components. There was a problem that it had to be large in comparison.

  The present invention has been made in view of such circumstances, and an object thereof is to provide a size of a core around which two inductors are wound in a voltage conversion circuit that is driven in different phases and connected in parallel. Converter, voltage conversion circuit step-down control method, voltage conversion circuit step-up control method, and computer program are provided.

  The voltage conversion device according to an aspect of the present invention has one end at a connection point between the high-voltage side switching element and the low-voltage side switching element connected in series via the first inductor, and the first inductor and the low-voltage side switching element. Two voltage conversion circuits having a connected second inductor and a capacitor connected in parallel to the low-voltage side switching element, and a controller for alternately turning on and off the high-voltage side switching element, A voltage conversion device having a voltage conversion circuit connected in parallel, wherein the second inductor is wound around a core so as to cancel each other magnetically, and the control unit causes the voltage conversion circuit to perform step-down conversion. In this case, the high-voltage side switching element is turned on to allow a resonance current to flow through the first inductor, the second inductor, and the capacitor. When the current of the high-voltage side switching element is smaller than a predetermined current threshold value in the first mode, the high-voltage side switching element is turned off and the return current of the two inductors is supplied to the low-voltage side switching element or the capacitor. The high-voltage side switching element is turned on after the operation in the second mode to shift to the first mode.

  The voltage conversion device according to an aspect of the present invention has one end at a connection point between the high-voltage side switching element and the low-voltage side switching element connected in series via the first inductor, and the first inductor and the low-voltage side switching element. Two voltage conversion circuits having a connected second inductor and a capacitor connected in parallel to the low-voltage side switching element, and a controller for alternately turning on and off the low-voltage side switching element, A voltage conversion device having a voltage conversion circuit connected in parallel, wherein the second inductor is wound around a core so as to cancel each other magnetically, and the control unit causes the voltage conversion circuit to perform step-up conversion. In this case, the low-voltage switching element is turned on to operate in the first mode in which current flows through the second inductor, and after the operation in the first mode The low-voltage side switching element is turned off to operate in a second mode in which a resonance current from the first inductor, the second inductor, and the capacitor flows, and in the second mode, the voltage of the low-voltage side switching element is higher than a predetermined voltage threshold value. If low, the low-voltage side switching element is turned on to shift to the first mode.

  A voltage step-down control method for a voltage conversion circuit according to one aspect of the present invention includes a high-voltage side switching element and a low-voltage side switching element connected in series via a first inductor, and a connection between the first inductor and the low-voltage side switching element. Two voltage conversion circuits having a second inductor having one end connected to the point, and a capacitor connected in parallel to the low-voltage side switching element, and a control unit for alternately turning on / off the high-voltage side switching element A control method in which the voltage conversion circuit is stepped down by the control unit in the voltage conversion device in which the second inductor is wound around a core so as to magnetically cancel each other and the two voltage conversion circuits are connected in parallel. The high-voltage side switching element is turned on to resonate by the first inductor, the second inductor, and the capacitor. When operating in a first mode for flowing a current, and the current of the high-voltage side switching element is less than a predetermined current threshold in the first mode, the high-voltage side switching element is turned off and the low-voltage side switching element or the capacitor is turned on 2. Operate in a second mode in which the return current of the inductor flows. After the operation in the second mode, the high-voltage side switching element is turned on to shift to the first mode.

  According to one aspect of the present invention, there is provided a voltage control circuit boost control method comprising: a high-voltage side switching element and a low-voltage side switching element connected in series via a first inductor; and a connection between the first inductor and the low-voltage side switching element. Two voltage conversion circuits having a second inductor having one end connected to the point, and a capacitor connected in parallel to the low-voltage side switching element, and a control unit for alternately turning on / off the low-voltage side switching element A control method in which the voltage conversion circuit in the voltage conversion device in which the second inductor is wound around a core so as to magnetically cancel each other and the two voltage conversion circuits are connected in parallel to the voltage conversion circuit. The low-voltage side switching element is turned on to operate in a first mode in which current flows through the second inductor, and in the first mode, After the operation, the low-voltage side switching element is turned off to operate in a second mode in which a resonance current is flown by the first inductor, the second inductor, and the capacitor. In the second mode, the voltage of the low-voltage side switching element is a predetermined value When it is lower than the voltage threshold, the low-voltage side switching element is turned on to shift to the first mode.

  A computer program according to an aspect of the present invention has one end connected to a connection point between a high-voltage side switching element and a low-voltage side switching element connected in series via a first inductor, and the first inductor and the low-voltage side switching element. A second voltage conversion circuit having a second inductor and a capacitor connected in parallel to the low-voltage side switching element; and a control unit that alternately turns on and off the high-voltage side switching element, the second inductor Is a computer program for causing the voltage conversion circuit to perform step-down conversion in the control unit in the voltage conversion device in which the two voltage conversion circuits are connected in parallel so as to magnetically cancel each other. The controller turns on the high-voltage side switching element to turn on the first inductor and the second inductor. And a step of operating in a first mode in which a resonance current is caused to flow by the capacitor, and when the current of the high-voltage side switching element is less than a predetermined current threshold in the first mode, the high-voltage side switching element is turned off and the low-voltage side A step of operating in a second mode in which a return current of the two inductors is caused to flow through the switching element or the capacitor; and a step of turning on the high-voltage side switching element and shifting to the first mode after the operation in the second mode. Let it run.

  A computer program according to an aspect of the present invention has one end connected to a connection point between a high-voltage side switching element and a low-voltage side switching element connected in series via a first inductor, and the first inductor and the low-voltage side switching element. A second voltage conversion circuit having a second inductor and a capacitor connected in parallel to the low-voltage side switching element; and a control unit that alternately turns on and off the low-voltage side switching element. Is a computer program for causing the voltage conversion circuit to perform step-up conversion in the control unit in the voltage conversion device in which the two voltage conversion circuits are connected in parallel so as to magnetically cancel each other. The control unit turns on the low-voltage side switching element to flow a current through the second inductor. And operating in a second mode in which after the operation in the first mode, the low-voltage side switching element is turned off and a resonance current is generated by the first inductor, the second inductor, and the capacitor, When the voltage of the low-voltage side switching element is lower than a predetermined voltage threshold in the second mode, the step of turning on the low-voltage side switching element and shifting to the first mode is executed.

  The present application can be realized as a voltage conversion device, a voltage conversion circuit step-down control method, and a voltage conversion circuit step-up control method each including such a characteristic processing unit and step, and corresponds to such a characteristic processing unit. Not only can be realized as a computer program for causing a computer to execute the step, but also part or all of the voltage converter can be realized as a semiconductor integrated circuit, or can be realized as another system including the voltage converter. Can be.

  According to the above, it is possible to reduce the size of the core around which the two inductors are wound in the voltage conversion circuit that is driven in different phases and connected in parallel.

1 is a block diagram illustrating a configuration example of a voltage conversion device according to a first embodiment. 3 is a timing chart illustrating an example of waveforms of respective units during a step-down operation of the voltage conversion circuit according to the first embodiment. 6 is an explanatory diagram illustrating an example of a state D1 in a step-down operation of the voltage conversion circuit according to the first embodiment. FIG. 6 is an explanatory diagram illustrating an example of a state D2 in the step-down operation of the voltage conversion circuit according to the first embodiment. FIG. 6 is an explanatory diagram illustrating an example of a state D3 in the step-down operation of the voltage conversion circuit according to the first embodiment. FIG. 6 is an explanatory diagram illustrating an example of a state D4 in the step-down operation of the voltage conversion circuit according to the first embodiment. FIG. 4 is a flowchart illustrating a processing procedure of a CPU that performs step-down conversion in the voltage conversion circuit according to the first embodiment. It is a circuit diagram which shows the equivalent circuit of the inductor wound around the core. It is a graph which shows the relationship between a duty ratio in case leakage inductance is 2 microH, and a ripple current. It is a graph which shows the relationship between a duty ratio in case leakage inductance is 4 microH, and a ripple current. It is explanatory drawing for demonstrating the magnetic flux which links with an inductor through the inside of a core. It is a graph which shows the relationship between a duty ratio in case leakage inductance is 2 microH, and a ripple current. It is a graph which shows the relationship between the duty ratio before and behind reduction of leakage inductance, and a ripple current. It is a graph which compares and shows the waveform of the ripple current before and behind reduction of leakage inductance. FIG. 10 is an explanatory diagram illustrating an example of a state U1 in a boost operation of the voltage conversion circuit according to the second embodiment. FIG. 10 is an explanatory diagram illustrating an example of a state U2 in the boosting operation of the voltage conversion circuit according to the second embodiment. FIG. 10 is an explanatory diagram illustrating an example of a state U3 in the boosting operation of the voltage conversion circuit according to the second embodiment. FIG. 10 is an explanatory diagram illustrating an example of a state U4 in the boosting operation of the voltage conversion circuit according to the second embodiment. FIG. 10 is an explanatory diagram illustrating an example of a state U5 in the boosting operation of the voltage conversion circuit according to the second embodiment. FIG. 10 is an explanatory diagram illustrating an example of a state U6 in the boosting operation of the voltage conversion circuit according to the second embodiment. 6 is a flowchart illustrating a processing procedure of a CPU that causes a voltage conversion circuit according to a second embodiment to perform step-up conversion.

[Description of Embodiment of the Present Invention]
First, embodiments of the present invention will be listed and described. Moreover, you may combine arbitrarily at least one part of embodiment described below.

(1) A voltage converter according to an aspect of the present invention includes a high-voltage switching element and a low-voltage switching element connected in series via a first inductor, and a connection point between the first inductor and the low-voltage switching element. Two voltage conversion circuits having a second inductor having one end connected to the low-voltage side switching element and a capacitor connected in parallel to the low-voltage side switching element, and a control unit that alternately turns on and off the high-voltage side switching element, In the voltage conversion device in which the two voltage conversion circuits are connected in parallel, the second inductor is wound around a core so as to magnetically cancel each other, and the control unit is connected to the voltage conversion circuit. In the case of step-down conversion, the high-voltage side switching element is turned on, and the resonance current generated by the first inductor, the second inductor, and the capacitor is changed. In the first mode, when the current of the high-voltage side switching element is smaller than a predetermined current threshold in the first mode, the high-voltage side switching element is turned off and the two inductors are connected to the low-voltage side switching element or the capacitor. The high-voltage side switching element is turned on after the operation in the second mode to shift to the first mode.

(7) A step-down control method for a voltage conversion circuit according to an aspect of the present invention includes a high-voltage side switching element and a low-voltage side switching element connected in series via a first inductor, the first inductor, and the low-voltage side switching. Two voltage conversion circuits having a second inductor having one end connected to the connection point of the elements, and a capacitor connected in parallel to the low-voltage side switching element, and control for alternately turning on / off the high-voltage side switching element A control unit in the voltage conversion device in which the two voltage conversion circuits are connected in parallel by winding the second inductor around the core so as to magnetically cancel each other. The high-voltage side switching element is turned on by the first inductor, the second inductor, and the capacitor. When operating in a first mode in which a resonance current flows, and when the current of the high-voltage side switching element is less than a predetermined current threshold in the first mode, the high-voltage side switching element is turned off and the low-voltage side switching element or the capacitor is turned off. The operation is performed in the second mode in which the reflux current of the two inductors flows, and after the operation in the second mode, the high-voltage side switching element is turned on to shift to the first mode.

(9) A computer program according to an aspect of the present invention is provided at a connection point of a high-voltage side switching element and a low-voltage side switching element connected in series via a first inductor, and the first inductor and the low-voltage side switching element. Two voltage conversion circuits having a second inductor having one end connected thereto, and a capacitor connected in parallel to the low-voltage side switching element, and a control unit for alternately turning on / off the high-voltage side switching element, A computer program for causing the voltage conversion circuit to step down the voltage conversion circuit in the voltage conversion device in which the second inductor is wound around a core so as to magnetically cancel each other and the two voltage conversion circuits are connected in parallel. Then, the control unit turns on the high-voltage side switching element to turn on the first inductor and the second inductor. A step of operating in a first mode in which a resonant current is flown by the inductor and the capacitor; and when the current of the high-voltage side switching element is less than a predetermined current threshold in the first mode, the high-voltage side switching element is turned off and the low voltage side Operating in a second mode in which a return current of the two inductors flows to the side switching element or the capacitor; and turning on the high-voltage side switching element after the operation in the second mode to shift to the first mode; Is executed.

  In this aspect, in each of the voltage conversion circuits connected in parallel, the high-voltage side switching element, the first inductor, and the low-voltage side switching element having capacitors connected to both ends are connected in this order. In addition, one end of the second inductor is connected to a connection point between the first inductor and the low-voltage side switching element. The two second inductors are wound around the core so as to magnetically cancel each other, and the control unit alternately turns on / off the high-voltage side switching elements, whereby the voltage input to the high-voltage side of the two voltage conversion circuits Is stepped down and output in parallel to the low pressure side.

  The control unit causes one of the voltage conversion circuits to operate in a first mode in which the high-voltage side switching element is turned on while the low-voltage side switching element is turned off and a resonance current is generated by the first inductor, the second inductor, and the capacitor. In the first mode, the resonance current flows into the capacitor, the capacitor voltage starts to rise, and energy is stored in the second inductor.

  Thereafter, the resonance current is reversed, the voltage of the capacitor starts to decrease, and the current of the high-voltage side switching element decreases. In the first mode, when the current of the high-voltage side switching element becomes smaller than a predetermined current threshold, the control unit turns off the high-voltage side switching element and causes the return current of the second inductor to flow through the low-voltage side switching element or the capacitor. Operate in 2 mode. Then, a control part turns on a high voltage | pressure side switching element in 2nd mode, and makes it transfer to 1st mode.

  The current threshold value may be, for example, 0 A or a current close to 0 A. Since the high voltage side switching element is turned off when the current of the high voltage side switching element is relatively small, the switching loss represented by the voltage × current when the high voltage side switching element is turned off is reduced.

  While the control unit periodically performs the above-described series of step-down control for one voltage conversion circuit, the control unit performs the same step-down control for the other voltage conversion circuit at a different phase in the same cycle. As a result, the high-voltage side switching elements of the two voltage conversion circuits are alternately turned on / off.

(2) In the voltage conversion device according to an aspect of the present invention, the control unit detects a voltage of the low-voltage side switching element, and the voltage of the low-voltage side switching element is a predetermined voltage in the second mode. When the voltage is lower than the voltage threshold, it is preferable to turn on the low-voltage side switching element, turn off the low-voltage side switching element, and then turn on the high-voltage side switching element.

  In this aspect, the control unit turns on the low-voltage side switching element for an appropriate time when the voltage of the low-voltage side switching element in the second mode becomes lower than the predetermined voltage threshold, and turns off the high-voltage side switching element after turning it off. The side switching element is turned on to shift to the first mode. The voltage threshold may be set to 0V or a voltage close to 0V, for example. Since the low voltage side switching element is turned on when the voltage of the low voltage side switching element is relatively low, the switching loss represented by the voltage × current when the low voltage side switching element is turned on is reduced.

(3) A voltage converter according to an aspect of the present invention includes a high-voltage switching element and a low-voltage switching element connected in series via a first inductor, and a connection point between the first inductor and the low-voltage switching element. Two voltage conversion circuits having a second inductor having one end connected to the low-voltage side switching element and a capacitor connected in parallel to the low-voltage side switching element, and a control unit that alternately turns on and off the low-voltage side switching element, In the voltage conversion device in which the two voltage conversion circuits are connected in parallel, the second inductor is wound around a core so as to magnetically cancel each other, and the control unit is connected to the voltage conversion circuit. When performing step-up conversion, the low-voltage side switching element is turned on to operate in the first mode in which current flows through the second inductor, and the operation in the first mode is performed. The low-voltage side switching element is turned off to operate in a second mode in which a resonance current is flown by the first inductor, the second inductor, and the capacitor, and the voltage of the low-voltage side switching element is a predetermined voltage in the second mode. When lower than the threshold value, the low-voltage side switching element is turned on to shift to the first mode.

(8) A step-up control method for a voltage conversion circuit according to an aspect of the present invention includes a high-voltage side switching element and a low-voltage side switching element connected in series via a first inductor, the first inductor and the low-voltage side switching. Two voltage conversion circuits having a second inductor having one end connected to the connection point of the elements, and a capacitor connected in parallel to the low-voltage side switching element, and control for alternately turning on / off the low-voltage side switching element A control unit in the voltage conversion device in which the two voltage conversion circuits are connected in parallel by winding the second inductor around the core so as to magnetically cancel each other. A first mode in which the low-voltage side switching element is turned on to pass a current through the second inductor; After the operation in the switch, the low-voltage side switching element is turned off to operate in a second mode in which a resonance current from the first inductor, the second inductor, and the capacitor flows, and the voltage of the low-voltage side switching element in the second mode Is lower than a predetermined voltage threshold, the low-voltage side switching element is turned on to shift to the first mode.

(10) A computer program according to an aspect of the present invention provides a connection point between a high-voltage side switching element and a low-voltage side switching element connected in series via a first inductor, and the first inductor and the low-voltage side switching element. Two voltage conversion circuits having a second inductor having one end connected thereto and a capacitor connected in parallel to the low-voltage side switching element, and a controller for alternately turning on and off the low-voltage side switching element, A computer program for causing the voltage conversion circuit to perform step-up conversion by the control unit in the voltage conversion device in which the second voltage converter is wound around a core so as to magnetically cancel each other and the two voltage conversion circuits are connected in parallel. Then, the control unit turns on the low-voltage side switching element to pass a current through the second inductor. A step of operating in the first mode; and a step of operating in the second mode in which the low-voltage side switching element is turned off after the operation in the first mode and a resonance current is generated by the first inductor, the second inductor, and the capacitor. And, when the voltage of the low-voltage side switching element is lower than a predetermined voltage threshold in the second mode, the step of turning on the low-voltage side switching element and shifting to the first mode is executed.

  In this aspect, in each of the voltage conversion circuits connected in parallel, the high-voltage side switching element, the first inductor, and the low-voltage side switching element having capacitors connected to both ends are connected in this order. In addition, one end of the second inductor is connected to a connection point between the first inductor and the low-voltage side switching element. The two second inductors are wound around the core so as to magnetically cancel each other, and the control unit alternately turns on / off the low-voltage side switching elements, whereby the voltage input to the low-voltage side of the two voltage conversion circuits Is boosted and output in parallel to the high voltage side.

  The control unit causes the one voltage conversion circuit to operate in the first mode in which the low-voltage side switching element is turned on while the high-voltage side switching element is turned off and current is supplied to the second inductor. Thereby, energy is accumulated in the second inductor.

  Thereafter, the control unit turns off the low-voltage side switching element and operates in the second mode in which the resonance current is generated by the first inductor, the second inductor, and the capacitor. In the second mode, the current flowing through the low-voltage side switching element first flows through a capacitor connected in parallel to the low-voltage side switching element, and a resonance current between the capacitor and the second inductor flows. Furthermore, when the voltage of the capacitor rises and the first switching element is appropriately turned on, a resonance current flows through the first inductor, the second inductor, and the capacitor.

  In the second mode, the voltage of the capacitor, that is, the voltage of the low-voltage side switching element once increases due to resonance and then decreases. When the voltage of the low-voltage side switching element becomes lower than the predetermined voltage threshold in the second mode, the control unit turns on the low-voltage side switching element to shift to the first mode. The voltage threshold may be set to 0V or a voltage close to 0V, for example. Since the low voltage side switching element is turned on when the voltage of the low voltage side switching element is relatively low, the switching loss represented by the voltage × current when the low voltage side switching element is turned on is reduced.

  While the above-described series of boost control is periodically performed for one voltage conversion circuit, the control unit performs similar boost control for the other voltage conversion circuit at a different phase in the same cycle. As a result, the low-voltage side switching elements of the two voltage conversion circuits are alternately turned on / off.

(4) In the voltage conversion device according to one aspect of the present invention, the control unit detects the voltage of the capacitor, and the voltage of the capacitor is higher than a second voltage threshold in the second mode. It is preferable to turn on the low voltage side switching element after turning on the high voltage side switching element and turning off the high voltage side switching element.

  In this aspect, when the voltage of the capacitor in the second mode is higher than the second voltage threshold, the control unit turns on the high voltage side switching element and turns off the low voltage side switching element after turning it off. Transition to the first mode. The second voltage threshold may be a voltage to be output to the high voltage side, for example. Thereby, a current is output to the high voltage side through the high voltage side switching element having a low on-resistance.

(5) In the voltage converter according to one aspect of the present invention, it is preferable that the second inductor has a coupling coefficient in the range of −0.99 or more and −0.78 or less.

In this embodiment, the second inductor has a coupling coefficient k closely related to the leakage inductance in the range of −0.99 ≦ k ≦ −0.78.
As a result, even when the duty ratio for turning on the high-voltage side or low-voltage side switching element changes in a relatively wide range, it is smaller than the self-inductance when k is 0 in a range where the ripple current of the second inductor is not increased. A second inductor having a leakage inductance can be applied.

(6) A voltage conversion device according to an aspect of the present invention includes N sets (N is a natural number) of the two voltage conversion circuits, the voltage conversion circuits of each set are connected in parallel, and the control The unit preferably turns on / off one or the other low-voltage side switching element of each set of voltage conversion circuits with a phase different by π / N.

In this aspect, the control unit alternately turns on / off the two low-voltage side switching elements included in each of the N sets of voltage conversion circuits with a phase difference of π, and π for any of the two low-voltage side switching elements. / N Turns on / off at different phases.
As a result, the switching loss is evenly distributed to the N sets of voltage conversion circuits, and the ripple current included in the output is reduced to 1 / N.

[Details of the embodiment of the present invention]
Specific examples of a voltage conversion device, a step-down control method for a voltage conversion circuit, a step-up control method for a voltage conversion circuit, and a computer program according to embodiments of the present invention will be described below with reference to the drawings. In addition, this invention is not limited to these illustrations, is shown by the claim, and intends that all the changes within the meaning and range equivalent to a claim are included. In addition, the technical features described in each embodiment can be combined with each other.

(Embodiment 1)
FIG. 1 is a block diagram illustrating a configuration example of the voltage conversion apparatus according to the first embodiment. In the figure, reference numerals 1 and 2 denote voltage conversion circuits that step down the voltage supplied from the high-voltage side terminals H and G and output the voltage from the low-voltage side terminals L and G in parallel. And 2 and a control unit 5 that controls voltage conversion by the voltage conversion circuits 1 and 2. A capacitor 41 is connected between the terminals H and G, and a relatively high voltage battery such as a lithium ion battery is connected to the outside. A capacitor 42 is connected between the terminals L and G, and a relatively low voltage battery such as a lead storage battery is connected to the outside. The voltage converter can perform a boosting operation that boosts the voltage supplied from the terminals L and G and outputs the boosted voltage in parallel from the terminals H and G. The boosting operation will be described in a second embodiment to be described later.

The voltage conversion circuit 1 (or 2) includes a high-voltage side switching element Q11 (or Q21) and a low-voltage side SW switching element (hereinafter referred to as “the low-voltage side SW switching element”) connected in series between the terminals H and G via the first inductor L11 (or L21). The high-voltage side and low-voltage side switching elements are also simply referred to as SW elements) Q12 (or Q22), and a second inductor having one end connected to the connection point of the first inductor L11 (or L21) and SW element Q12 (or Q22) ( Hereinafter, the first and second inductors are simply referred to as inductors) L12 (or L22), and a capacitor C1 (or C2) connected to both ends of the SW element Q12 (or Q22). The inductors L12 and L22 are wound around the core 3 so as to cancel each other magnetically, and the other end is connected to the terminal L. In the first embodiment, the constants of the inductor L11 (or L21), the inductor L12 (or L22), and the capacitor C1 are 0.2 μH, 3.4 μH (provided that k described later is 0) and 350 nF, respectively.

  The SW element Q11 (or Q21) has a drain connected to the terminal H and a gate connected to the control unit 5. The SW element Q12 (or Q22) has a source connected to the terminal G and a gate connected to the control unit 5. Here, the SW element is an N-channel MOSFET (Metal Oxide Field Effect Transistor), but is not limited to this, and other switching elements such as a bipolar transistor, an IGBT (Insulated Gate Bipolar Transistor), and the like. It may be.

  The voltage conversion circuit 1 (or 2) further includes a SW element Q13 (or Q23) having a drain connected to a connection point between the SW element Q11 (or Q21) and the inductor L11 (or L21). The SW element Q13 (or Q23) has a source connected to the terminal G and a gate connected to the control unit 5. In the first embodiment, each of the SW elements Q13 and Q23 may be replaced with a diode whose anode is connected to the terminal G.

  The control unit 5 includes a CPU (Central Processing Unit) (not shown), and controls the operation of each unit according to a control program stored in advance in a ROM (Read Only Memory), and performs processing such as input / output and calculation. . A computer program that defines the procedure of each process by the CPU may be loaded in advance into a RAM (Random Access Memory) using means (not shown), and the loaded computer program may be executed by the CPU. 5 may be configured by a dedicated hardware circuit.

  The controller 5 alternately turns on / off the SW elements Q11 and Q21 by giving drive signals having phases different by π to the gates of the SW elements Q11 and Q21. The control unit 5 also detects the voltages of the capacitors C1 and C2 (that is, the voltages at the drains of the SW elements Q12 and Q22), and turns on / off the SW elements Q12 and Q22 as appropriate based on the detection results. . In the first embodiment, the switching frequency of the SW elements Q11 and Q21 is 200 kHz.

  FIG. 2 is a timing chart showing an example of the waveform of each part during the step-down operation of the voltage conversion circuit 1 according to the first embodiment. The five timing charts shown in FIG. 2 all have the same time axis (t) as the horizontal axis, and the voltages of the SW element Q11, the inductor L11, the capacitor C1, the SW element Q12, and the inductor L12 are sequentially shown from the top in the figure. And the current waveform. The thick solid line in the figure indicates the voltage, and the thin solid line indicates the current. In particular, for the SW elements Q11 and Q12, the control voltage applied to the gate is indicated by a broken line. In the first embodiment, the direction from the terminal H to the terminal L is the current direction of the inductor L11, the inductor L12, and the SW element Q11.

Hereinafter, the step-down operation of the voltage conversion circuit 1 in the states D1 to D4 illustrated in FIG. 2 will be described for each state.
FIG. 3 is an explanatory diagram illustrating an example of a state D1 in the step-down operation of the voltage conversion circuit 1 according to the first embodiment. State D1 corresponds to the first mode. The control unit 5 operates the voltage conversion circuit 1 in the first mode in which a sinusoidal resonance current flows through the inductor L11, the inductor L12, and the capacitor C1 by turning on the SW element Q11 with the SW element Q12 turned off. . In the first mode, current flows from the terminal H via the SW element Q11 and the inductor L11, and current flows from the terminal L via the inductor L12.

  In the state D1 shown in FIG. 3, while the voltage of the capacitor C1 that rises in a sine wave is lower than the voltage between the terminals H and G, the current in the inductor L11 continues to increase in a sine wave, and the voltage of the capacitor C1 changes to the terminals H and G. After becoming higher than the voltage between, the current of the inductor L11 starts to decrease. During this time, the voltage of the inductor L11 continues to decrease in a sine wave shape. During the state D1, the voltage of the capacitor C1 continues to rise with time.

  FIG. 4 is an explanatory diagram illustrating an example of a state D2 in the step-down operation of the voltage conversion circuit 1 according to the first embodiment. State D2 also corresponds to the first mode. When the direction of the resonance current flowing in the capacitor C1 is reversed in the state D1 shown in FIG. 3 and the voltage of the capacitor C1 changes from rising to lowering, the voltage conversion circuit 1 becomes the state D2. In the state D2, a sinusoidal discharge current of the capacitor C1 flows to the terminal L side through the inductor L12. While the voltage of the capacitor C1 is higher than the voltage between the terminals H and G, the current of the inductor L11 continues to decrease.

  FIG. 5 is an explanatory diagram illustrating an example of a state D3 in the step-down operation of the voltage conversion circuit 1 according to the first embodiment. State D3 corresponds to the second mode. In the state D2 shown in FIG. 4, when the sinusoidal current flowing in the SW element Q11 and the inductor L11 becomes less than 0A or a current threshold value close to 0A, the control unit 5 turns off the SW element Q11, thereby causing the voltage conversion circuit. 1 becomes state D3. Thereby, since the SW element Q11 is turned off while the current of the SW element Q11 is relatively small, the switching loss of the SW element Q11 is reduced. In the state D3, initially, the return current of the inductor L12 flows through the capacitor C1.

  The time point at which the current of the inductor L11 becomes 0A or close to 0A in the state D2 can be estimated from constants such as the inductors L11 and L12 and the capacitor C1, but the current sensor detects the current of the inductor L11. It may be.

  If the control unit 5 turns off the SW element Q11 before the current of the inductor L11 becomes close to 0 A in the state D2, a surge voltage may be generated in the inductor L11. On the other hand, in the first embodiment, the SW element Q13 is connected between the connection point of the SW element Q11 and the inductor L11 and the terminal G. For this reason, the current flowing through the inductor L11 before the SW element Q11 is turned off continues to flow through the parasitic diode of the SW element Q13, and the surge voltage is suppressed. The control unit 5 may positively turn on the SW element Q13 from when the SW element Q11 is turned off to an appropriate time.

  FIG. 6 is an explanatory diagram illustrating an example of a state D4 in the step-down operation of the voltage conversion circuit 1 according to the first embodiment. In the state D3 shown in FIG. 5, after the voltage of the capacitor C1 and the SW element Q12 becomes lower than a predetermined voltage threshold, the control unit 5 turns on the SW element Q12, so that the voltage conversion circuit 1 becomes the state D4. Thereby, the return current of the inductor L12 flows through the SW element Q12 having a low on-resistance. Further, since the SW element Q12 is turned on while the voltage of the SW element Q12 is relatively low, the switching loss of the SW element Q12 is reduced.

  In the state D4, when an appropriate time corresponding to the step-down ratio of the output voltage to the terminals L and G with respect to the input voltage from the terminals H and G has elapsed, the control unit 5 turns on the SW element Q11 after turning off the SW element Q12. As a result, the voltage conversion circuit 1 enters the state D1 shown in FIG. In this way, the state transition from state D1 to state D4 is repeated.

Below, operation | movement of the control part 5 mentioned above is demonstrated using the flowchart which shows it. The processing shown below is executed by a CPU (not shown) according to a control program stored in advance in a ROM (not shown).
FIG. 7 is a flowchart illustrating a processing procedure of the CPU that causes the voltage conversion circuit 1 according to the first embodiment to perform step-down conversion. The process whose procedure is shown in FIG. 7 is activated when the voltage converter starts step-down conversion.

  When the process of FIG. 7 is started, the CPU turns off the low-voltage side switching element Q12 (S11) and turns on the high-voltage side switching element Q11 (S12) to resonate between the inductors L11 and L12 and the capacitor C1. (S13). As a result, the voltage conversion circuit 1 is operated in the first mode. Next, the CPU estimates the current of the high-voltage side switching element Q11 and determines whether or not the estimated current is less than a predetermined current threshold (S14). If not less (S14: NO), the CPU is less than the current threshold. Wait until

  When the current of the high-voltage side switching element Q11 is smaller than the predetermined current threshold (S14: YES), the CPU turns off the high-voltage side switching element Q11 (S15) and operates the voltage conversion circuit 1 in the second mode. This causes the return current of the inductor L12 to flow through the capacitor C1 or the low-voltage side switching element Q12 (S16). Next, the CPU detects the voltage of the low-voltage side switching element Q12 and determines whether or not the detected voltage is lower than a predetermined voltage threshold (S17). If not lower (S17: NO), it is lower than the voltage threshold. Wait until

  When the voltage of the low voltage side switching element Q12 is lower than the predetermined voltage threshold (S17: YES), the CPU turns on the low voltage side switching element Q12 (S18). Next, after waiting for a time corresponding to the step-down ratio (S19), the CPU determines whether or not to end the step-down conversion (S20). If not (S20: NO), the voltage conversion circuit 1 is set to the first. In order to shift to the mode, the process proceeds to step S11. On the other hand, when ending the step-down conversion (S20: YES), the CPU ends the process of FIG. 7 after turning off the low-voltage side switching element Q12 (S21).

  Next, a ripple current flowing through the inductors L12 and L22 (hereinafter, simply referred to as a ripple current unless otherwise confused) will be described with reference to FIGS. 1 and 2 and other drawings described later. The fluctuation component of the current flowing through each of the inductors L12 and L22 is a ripple current. When the period of each signal shown in FIG. 2 is T, the control unit 5 alternately executes the step-down control of FIG. 7 for the voltage conversion circuit 1 with a delay of T / 2 for the voltage conversion circuit 2 as well. That is, the drive signals for turning on / off the SW elements Q11 and Q21 are out of phase with each other by π.

  When the SW element Q11 is turned on in the voltage conversion circuit 1, the current in the inductor L12 increases from the middle of the state D1 to the state D2 (hereinafter, this period is referred to as a first period). Since a current is induced in the inductor L22 so as to cancel the current flowing through the inductor L12 during the first period, the current of the inductor L22 also increases during the same first period. The increased current flows back to the inductor L22 via the capacitor 42 and an external battery (not shown) and the capacitor C2 or the SW element Q22.

  Similarly, when the SW element Q21 is turned on in the voltage conversion circuit 2, the current in the inductor L22 increases in the second period delayed by T / 2 from the first period. Since a current is induced in the inductor L12 so as to cancel the current flowing through the inductor L22 in the second period, the current in the inductor L12 also increases in the same second period. For this reason, the current of the inductor L12 shown in FIG. 2 repeatedly increases and decreases in the T / 2 period.

Here, the leakage inductance of the inductors L12 and L22 will be described.
FIG. 8 is a circuit diagram showing an equivalent circuit of the inductors L12 and L22 wound around the core 3. As shown in FIG. The self-inductances of the inductors L12 and L22 are L1 and L2, respectively, and the coupling coefficient between the inductors L12 and L22 is k. When this is developed into an equivalent circuit, the inductors L12 and L22 are represented by a three-terminal circuit in which one end of the inductors 31 and 32 corresponding to the leakage inductances Le1 and Le2 and one end of the inductor 33 corresponding to the mutual inductance M are connected. Is done. Each of Le1 and Le2 is represented by the following formulas (1) and (2).

Le1 = (1-k) L1 (1)
Le2 = (1-k) L2 (2)

  The coupling coefficient k and the mutual inductance M are defined with the same sign, and the difference between k and M being positive or negative depends on the direction of magnetic coupling between the inductors L12 and L22. In general, the sign of k is often defined as positive regardless of the direction of magnetic coupling. Therefore, the following description will be made assuming that k is represented by 0 or a positive number. The fact that k should be represented by 0 or a negative number indicates that the direction of the magnetic coupling with each other is reversed by the dot symbol given to one end of each of the inductors L12 and L22. It becomes clear by doing.

  More specifically, when current flows into one end of each of the two inductors coupled to each other to which the dot symbol is given, when the magnetic flux is strengthened (or weakened), the sign of the mutual inductance M Is defined as positive (or negative). Therefore, when it is determined that the sign of the coupling coefficient k and the mutual inductance M between the inductors L12 and L22 shown in FIG. 1 is positive, the mutual magnetic flux is weakened when current flows into one end of the inductors L12 and L22. Is understood from the figure, it becomes clear that the directions of mutual magnetic coupling in the inductors L12 and L22 are opposite to each other.

  When strictly defining FIG. 1 by defining that the sign of the coupling coefficient k and the mutual inductance M is negative, a dot symbol is given to one end of the inductors L12 and L22 connected to the inductors L11 and L21, respectively. In other words, it is difficult to grasp that the directions of the magnetic couplings are opposite to each other. Therefore, such a description method is not used here.

  When the coupling coefficient k between the inductors L12 and L22 is in the range of 0 <k <1, the component that replaces the self-inductance L1 and functions as the choke coil among the inductance components of the inductor L12 is the leakage inductance Le1 (FIG. 8). Hereinafter, the leakage inductance relating to the inductors L12 and L22 or two inductors instead thereof is simply referred to as leakage inductance. Since the ripple current of the inductor L12 when the self-inductance L1 in the case of k = 0 is replaced with the leakage inductance Le1 having the same magnitude, as shown in FIG. 2, the increase and decrease repeats in smaller increments than in the case of k = 0. It becomes smaller than the case of k = 0.

Next, the result of simulating the relationship between the duty ratio for turning on each of the SW elements Q11 and Q12 and the ripple current will be described.
FIG. 9 is a graph showing the relationship between the duty ratio and the ripple current when the leakage inductance is 2 μH, and FIG. 10 is a graph showing the relationship between the duty ratio and the ripple current when the leakage inductance is 4 μH. 9 and 10, the horizontal axis represents the duty ratio, and the vertical axis represents the ripple current (App: Ampere peak-to-peak).

  9 and 10, the solid line and the broken line respectively show the change characteristics of the ripple current with respect to the duty ratio when the coupling coefficient k is 0 and 0.99. When k = 0, the leakage inductance should be called self-inductance, but here, the leakage inductance is unified. The ripple current in the case of k = 0 becomes a maximum when the duty ratio is 0.5, and draws a convex curve upward. Further, the ripple current in the case of k = 0.99 becomes a minimum of about 0 when the duty ratio is 0.5, and becomes a maximum between 0 and 0.5 and between 0.5 and 1. Draw a curve.

  Comparing the cases shown by the solid lines in FIGS. 9 and 10, it is understood that when k = 0, the ripple current is halved over the entire range of the duty ratio when the leakage inductance is doubled from 2 μH to 4 μH. The Similarly, when the cases indicated by the broken lines are compared, it is understood that when k = 0.99, the ripple current is halved over the entire range of the duty ratio when the leakage inductance is doubled from 2 μH to 4 μH. That is, it can be said that the ripple current and the leakage inductance are in an inversely proportional relationship.

Next, the relationship between the core 3 and the inductors L12 and L22 will be described in detail.
FIG. 11 is an explanatory diagram for explaining the magnetic flux interlinking with the inductors L12 and / or L22 through the core 3. The core 3 is a so-called EI core, and a gap is formed at a central leg 3b between a part of the core 3 having an E shape and another part of the core 3 having an I shape. Yes. The core 3 is not limited to the EI core, and may be, for example, an EE core. The inductors L12 and L22 are wound around the leg 3a on one side of the core 3 and the leg 3c on the other side, respectively.

  When a voltage is applied from the inductor L11 to one end of the inductor L12, the exciting current im flows through the inductor L12, and the main magnetic flux φm is generated in the core 3 by the exciting inductance corresponding to the mutual inductance M. The main magnetic flux φm is linked to the inductor L22, the load current i2 flows by electromagnetic induction, and the load current i1 flows to the inductor L12 according to the load current i2.

  In this case, a leakage flux φ1 not linked to the inductor L22 is generated by the winding current (i1 + im) of the inductor L12, and a leakage flux φ2 not linked to the inductor L12 is generated by the winding current (i2) of the inductor L22. . Leakage magnetic fluxes φ1 and φ2 pass through the central leg 3b of the core 3. When the leakage inductances Le1 and Le2 described above are used, the leakage magnetic fluxes φ1 and φ2 are expressed by the following equations (3) and (4).

φ1 = Le1 (i1 + im) (3)
φ2 = Le2 · i2 (4)

  The main magnetic flux φm does not increase due to the load currents i1 and i2, whereas the leakage magnetic fluxes φ1 and φ2 increase in proportion to the load currents i1 and i2. Therefore, when the load currents i1 and i2 are constant, the equations (3) and (4) indicate that the magnetic flux (magnetic flux amount) Φ in the core 3 is proportional to the leakage inductances Le1 and Le2. Hereinafter, it is assumed that the load currents i1 and i2 are constant and compared. Since the magnetic flux Φ is represented by the product of the magnetic flux density B in the core 3 and the effective sectional area Ae of the core 3, when the magnetic flux density B in the core 3 is constant, the effective sectional area Ae and the leakage inductance Le1. And Le2 are in a proportional relationship.

  In general, however, the magnetic flux density in the core cannot exceed the maximum magnetic flux density Bmax. Considering this point, for example, when reducing the size without changing the shape of the core, to reduce the effective cross-sectional area of the core without changing the magnetic flux density in the core, the effective disconnection is based on the above proportional relationship. What is necessary is just to reduce a leakage inductance by the ratio which reduces an area. However, as described above, since the ripple current and the leakage inductance are in an inversely proportional relationship, there is a dilemma that if the leakage inductance is reduced in order to reduce the size of the core, the ripple current increases.

  Therefore, in order to find a way to reduce the core size in a range where the ripple current of the inductor is not increased compared to when the coupling coefficient k is 0, two inductors having a constant leakage inductance and varying the coupling coefficient were used. In this case, a graph showing how the ripple current changes was drawn.

  FIG. 12 is a graph showing the relationship between the duty ratio and the ripple current when the leakage inductance is 2 μH. In the figure, the horizontal axis represents the duty ratio, and the vertical axis represents the ripple current (App). Each of the solid line, the broken line, and the alternate long and short dash line in the figure indicates a change characteristic of the ripple current with respect to the duty ratio when the coupling coefficient k is 0.0, 0.9, and 0.99. As described above, since it is clear that the ripple current tends to decrease as the coupling coefficient k increases, no drawing is performed when the coupling coefficient k is in the range of more than 0.0 and less than 0.9.

  From another point of view, the solid line shown in FIG. 12 is obtained using two inductors α having a leakage inductance (actually equivalent to a self-inductance) of 2 μH and a coupling coefficient k of 0, from 0 to 1. It is a change characteristic α of the ripple current with respect to the duty ratio in the range. Further, the broken line or the alternate long and short dash line shown in FIG. 12 is a range from 0 to 1 obtained by using two inductors β having a leakage inductance of the same magnitude as the self-inductance of the inductor α and having k close to −1. The ripple current change characteristic β with respect to the duty ratio. These change characteristics are sequentially compared over a certain range of duty ratios, and the lowest reduction rate of the ripple current in the change characteristic bar and the like with respect to the ripple current in the change characteristic α can be calculated based on the value read from FIG.

  In this case, as described above, the ripple current and the leakage inductance are in an inversely proportional relationship, and the effective cross-sectional area of the core is reduced by the reduced ratio when the magnetic flux density in the core is constant and the leakage inductance is reduced. It has been found to be reduced. From these facts, the ripple current is increased by reducing the leakage inductance of the inductor β by the reduction rate calculated by reading from FIG. 12, and the ripple current is reduced by changing k from 0 to 0.99. The ratio can be offset. In this way, the core size can be reduced without changing the magnetic flux density in the core in a range where the ripple current of the inductor is not increased compared to when the coupling coefficient k is 0.

  More specifically, for example, when the duty ratio varies from 0.1 to 0.7 (or from 0.15 to 0.7) in FIG. 12, the coupling coefficient k is changed from 0 to 0.99 (or 0.90 and The ratio at which the ripple current is reduced when approaching 0.95) interpolated from the case of 0.99 is the smallest when the duty ratio is 0.1 (or 0.15). The reduction rate is 0.55 (or 0.59).

  In other words, for example, when the duty ratio is 0.1, the ripple current is reduced only 0.45 times by changing k from 0 to 0.99, and the ripple current is greatly reduced at other duty ratios. And becomes smaller than 0.45 times. Therefore, by reducing the leakage inductance to 0.9 μH, which is 0.45 times 2 μH, and setting the coupling coefficient k to 0.99, for a duty ratio in the range of 0.1 to 0.7, k The ripple current can be prevented from increasing compared to when 0 is zero. By reducing the leakage inductance to 0.45 times (the reduction rate is 0.55), the core size (effective cross-sectional area) can be increased to 0.45 times without changing the magnetic flux density in the core. .

Next, the influence on the ripple current by reducing the leakage inductance and changing k from 0 to 0.99 will be described.
FIG. 13 is a graph showing the relationship between the duty ratio and the ripple current before and after the leakage inductance is reduced. In the figure, the horizontal axis represents the duty ratio, and the vertical axis represents the ripple current (App). The solid line in the figure shows the change characteristic of the ripple current with respect to the duty ratio when the leakage inductance (leakage L) is 2 μH and k = 0. This solid line is exactly the same as the solid line in FIG. Also, the broken line indicates the change characteristic of the ripple current with respect to the duty ratio when the leakage inductance is 0.9 μH and k = 0.99.

  The ripple current indicated by the broken line draws a curve having a minimum of about 0 when the duty ratio is 0.5, and a maximum between 0 and 0.5 and between 0.5 and 1. The same as in the case of FIGS. According to FIG. 13, even when the leakage inductance is reduced from 2 μH to 0.9 μH and k is changed from 0 to 0.99, the ripple is at least over the duty ratio range from 0.1 to 0.7. The increase in current is suppressed.

Here, for the four types of coupling coils A, B, C, and D in which two inductors are coupled through the core and the leakage inductance is 0.9 μH, the coupling coefficients are 0.78, 0.86,. Trials were made to be 96 and 0.98, and the core volume was measured. As a result of the measurement, the volumes of the cores of the coupling coils A, B, C, and D were 22 cm ³ , 24 cm ³ , 23 cm ^3, and 21 cm ³ , respectively. On the other hand, for the reference coupling coil having a leakage inductance of 2.0 μH and k = 0 as a reference for comparison, the core volume was 38 cm ³ . From these results, it can be said that the core size is sufficiently reduced over a wide range of k from 0.78 to 0.98. Obviously, if k is 0.99, the size of the core is further reduced.

  The above is the case where the leakage inductance is reduced to 0.45 times. As shown in FIG. 12, the leakage inductance is reduced to 0.41 times (the reduction rate is 0.59), and k is changed from 0 to 0. In the case of .95, an increase in ripple current is suppressed over a duty ratio range from 0.15 to 0.7. Moreover, even if the leakage inductance (self-inductance) when k = 0 before reduction is other than 2 μH, the leakage inductance is reduced by the same 0.41 reduction ratio so that k becomes 0.95. Thus, an increase in ripple current is suppressed over a range of duty ratios from 0.15 to 0.7.

  For example, the self-inductance when the inductor L12 of the first embodiment is k = 0 is 3.4 μH, but when this is reduced to 0.41 times and the leakage inductance is 1.4 μH, k By setting = 0.95, the core size (effective cross-sectional area) can be increased 0.41 times without changing the magnetic flux density in the core.

Next, the effect when the leakage inductance is reduced will be described.
FIG. 14 is a graph showing a comparison of ripple current waveforms before and after the reduction of leakage inductance. In the figure, the horizontal axis represents time (t), and the vertical axis represents the current (A) of the inductor L12 or L22. A time scale of 5 μs serving as a reference for the horizontal axis and a current scale of 10 A serving as a reference for the vertical axis are shown in appropriate places in the drawing. The upper part of the figure is a graph when k = 0 and the self-inductance of the inductor L12 is 3.4 μH, and the lower part is a case where k = 0.95 and the leakage inductance of the inductor L12 is 1.4 μH. It is a graph. The duty ratio is 0.3 in all cases. It can be confirmed that the ripple current that was 16 App before the reduction of the leakage inductance is suppressed to 15 App after the reduction.

  In the first embodiment, the reduction rate of the ripple current when k is approximated from 0 to 0.99 or 0.95 is calculated using FIG. 12, but k is not necessarily 0.99 or 0. It is not necessary to get close to 95. From the above description, it is clear that the rate at which the ripple current is reduced increases as k is closer to 1 (actually -1), and the size of the core 3 is increased as the rate at which the ripple current is reduced. The ratio that can be reduced increases.

  As described above, according to the first embodiment, the SW elements Q11 and Q21, the inductors L11 and L21, and the capacitors C1 and C2 are connected to both ends in each of the voltage conversion circuits 1 and 2 connected in parallel. The SW elements Q12 and Q22 are connected in this order, and one end of the inductors L12 and L22 is connected to the connection point of the inductors L11 and L21 and the SW elements Q12 and Q22. The two inductors L12 and L22 are wound around the core 3 so as to magnetically cancel each other, and the control unit 5 alternately turns on / off the SW elements Q11 and Q21, thereby causing the two voltage conversion circuits 1 and 2 to The voltages input to the terminals H and G are stepped down and output in parallel to the terminals L and G.

  The control unit 5 causes the voltage conversion circuit 1 to operate in the first mode in which the SW element Q11 is turned on while the SW element Q12 is turned off and the resonance current from the inductor L11, the inductor L12, and the capacitor C1 flows. In the first mode, the resonance current flows into the capacitor C1, the voltage of the capacitor C1 starts to rise, and energy is stored in the inductor L12. After that, when the resonance current is inverted and the voltage of the capacitor C1 starts to decrease, and the current of the SW element Q11 decreases and becomes less than a predetermined current threshold, the control unit 5 turns off the SW element Q11 and switches the SW element Q11. The operation is performed in the second mode in which the return current of the inductor L12 flows through Q12 or the capacitor C1. Thereafter, the control unit 5 turns on the SW element Q11 in the second mode to shift to the first mode. The control unit 5 performs the same step-down control for the voltage conversion circuit 2 at different phases in the same cycle while periodically performing the above-described series of step-down control for the voltage conversion circuit 1. As a result, the SW elements Q11 and Q21 of the two voltage conversion circuits 1 and 2 are alternately turned on / off.

  That is, since the inductors L12 and L22 are wound around the core 3 so as to magnetically cancel each other, even if the leakage inductances Le1 and Le2 of the inductors L12 and L22 are reduced within a certain limit, the inductors L12 and L22 The voltage conversion circuits 1 and 2 are step-down controlled within a range in which the ripple current of L22 is not increased. Therefore, the inductors L12 and L22 in the voltage conversion circuits 1 and 2 that are driven in different phases and connected in parallel are wound together with the size of the core 3 being reduced in accordance with the reduction of the leakage inductances Le1 and Le2. It is possible to reduce the size of the rotated core 3.

  Further, according to the first embodiment, when the voltage of the SW element Q12 in the second mode becomes lower than the predetermined voltage threshold, the control unit 5 turns on the SW element Q12 for an appropriate time and turns it off. The SW element Q11 is turned on to shift to the first mode. Therefore, since the SW element Q12 is turned on when the voltage of the SW element Q12 is relatively low, it is possible to reduce the switching loss represented by the voltage × current when the SW element Q12 is turned on.

Further, according to the first embodiment, the inductors L12 and L22 have a coupling coefficient k that is closely related to the leakage inductances Le1 and Le2 of 0.99 ≧ k ≧ 0.78 (actually −0.99 ≦ k ≦ −). 0.78).
Therefore, even when the duty ratio for turning on the SW element Q11 changes in a relatively wide range, the inductor L12 having a leakage inductance smaller than the self-inductance when k is 0 in a range where the ripple current of the inductor L12 is not increased. Can be applied.

(Embodiment 2)
The first embodiment is a mode in which the voltage from the terminals H and G is stepped down by the voltage converter and output from the terminals L and G, whereas the second embodiment is a voltage from the terminals L and G in the voltage converter. Is boosted and output from terminals H and G. Since the configuration of the voltage conversion apparatus in the second embodiment is the same as that in the first embodiment, the same reference numerals are given to the portions corresponding to the first embodiment, and the description thereof is omitted. Further, the timing chart at the time of boosting to be compared with FIG. 2 of the first embodiment is omitted, and the boosting operation of the voltage conversion circuit 1 will be described for each state. In the second embodiment, the direction from the terminal L to the terminal H is the current direction of the inductor L11, the inductor L12, and the SW element Q11.

  FIG. 15 is an explanatory diagram illustrating an example of the state U1 in the boosting operation of the voltage conversion circuit according to the second embodiment. The state U1 corresponds to the first mode. The control unit 5 operates the voltage conversion circuit 1 in the first mode in which a current flows through the inductor L12 by turning on the SW element Q12 with the SW element Q11 turned off. In the first mode, current flows from the terminal L via the inductor L12 and the SW element Q12, and energy is stored in the inductor L12.

  FIG. 16 is an explanatory diagram illustrating an example of a state U2 in the boosting operation of the voltage conversion circuit according to the second embodiment. State U2 corresponds to the second mode. When an appropriate time corresponding to the step-up ratio of the output voltage to the terminals H and G with respect to the input voltage from the terminals L and G has elapsed in the state U1, the control unit 5 turns off the SW element Q12, thereby causing a voltage conversion circuit. 1 becomes the state U2, and a sinusoidal resonance current flows through the inductor L12 and the capacitor C1.

  In the state U2 shown in FIG. 16, while the voltage of the capacitor C1 is lower than the voltage between the terminals L and G, the current of the inductor L12 continues to increase sinusoidally, and the voltage of the capacitor C1 is higher than the voltage between the terminals L and G. After that, the current of the inductor L12 starts to decrease. Even after the voltage of the capacitor C1 exceeds the voltage between the terminals L and G, the voltage continues to rise with time.

  FIG. 17 is an explanatory diagram illustrating an example of a state U3 in the boosting operation of the voltage conversion circuit according to the second embodiment. State U3 also corresponds to the second mode. When the voltage of the capacitor C1 and the SW element Q12 becomes higher than the second voltage threshold value in the state U2 shown in FIG. 16, the control unit 5 turns on the SW element Q11, so that the voltage conversion circuit 1 becomes the state U3. In the state U3, a sinusoidal resonance current flows through the inductors L11 and L12 and the capacitor C1, and the resonance current of the inductor L11 flows to the terminal H side via the SW element Q11 having a low on-resistance.

  FIG. 18 is an explanatory diagram illustrating an example of a state U4 in the boosting operation of the voltage conversion circuit according to the second embodiment. The state U4 also corresponds to the second mode. When the direction of the resonance current flowing in the capacitor C1 is reversed and the voltage of the capacitor C1 changes from rising to lowering, the voltage conversion circuit 1 enters the state U4. In the state U4, while the voltage of the capacitor C1 is higher than the voltage between the terminals H and G, the current of the inductor L11 continues to increase sinusoidally, and after the voltage of the capacitor C1 becomes lower than the voltage between the terminals H and G. Then, the current of the inductor L11 starts to decrease. Even after the voltage of the capacitor C1 falls below the voltage between the terminals H and G, the voltage continues to decrease with time.

  FIG. 19 is an explanatory diagram illustrating an example of a state U5 in the boosting operation of the voltage conversion circuit according to the second embodiment. In the state U4 shown in FIG. 18, when the voltages of the capacitor C1 and the SW element Q12 become lower than a predetermined voltage threshold, the control unit 5 turns on the SW element Q12, so that the voltage conversion circuit 1 becomes the state D4. Thereby, the current of the inductor L12 flows to the SW element Q12 having a low on-resistance. Further, since the SW element Q12 is turned on while the voltage of the SW element Q12 is relatively low, the switching loss of the SW element Q12 is reduced.

  When the voltage of the capacitor C1 does not sufficiently decrease in the state U4, the control unit 5 turns on the SW element Q13, thereby causing the charge of the capacitor C1 to flow to the SW element Q13 through the inductor L11, and the voltage of the capacitor C1 is set. It can be lower than a predetermined voltage threshold.

  FIG. 20 is an explanatory diagram illustrating an example of a state U6 in the boosting operation of the voltage conversion circuit according to the second embodiment. When the current flowing through the inductor L11 decreases in the state U5 shown in FIG. 19, a part of the current flowing through the inductor L12 conducts the SW element Q12, and the direction of the current flowing through the SW element Q12 is reversed to convert the voltage. Circuit 1 enters state U6. Thereafter, when the current flowing through the inductor L11 becomes less than 0A or a current threshold value close to 0A, the control unit 5 turns off the SW element Q11, so that the voltage conversion circuit 1 enters the state U1 shown in FIG. In this way, the state transition from the above-described states U1 to U6 is repeated.

Below, operation | movement of the control part 5 mentioned above is demonstrated using the flowchart which shows it.
FIG. 21 is a flowchart illustrating the processing procedure of the CPU that causes the voltage conversion circuit 1 according to the second embodiment to perform step-up conversion. The process whose procedure is shown in FIG. 21 is activated when the voltage converter starts boost conversion.

  When the process of FIG. 21 is started, the CPU turns off the high-voltage side switching element Q11 (S31), turns on the low-voltage side switching element Q12 (S32), and operates the voltage conversion circuit 1 in the first mode. . Next, after waiting for a time corresponding to the boost ratio (S33), the CPU turns off the low-voltage side switching element Q12 (S34) and operates the voltage conversion circuit 1 in the second mode.

  Thereafter, the CPU detects the voltage of the capacitor C1 and determines whether or not the detected voltage is higher than the second voltage threshold (S35). If not higher (S35: NO), the CPU determines from the second voltage threshold. Wait until it gets high. When the voltage of the capacitor C1 is higher than the second voltage threshold (S35: YES), the CPU turns on the high-voltage side SW element Q11 (S36), and resonates between the inductors L11 and L12 and the capacitor C1 (S37).

  In this state, the CPU determines whether or not the voltage of the low-voltage side SW element Q12 is lower than a predetermined voltage threshold (S38). If not lower (S38: NO), the CPU waits until it becomes lower than the voltage threshold. When the voltage of the low voltage side switching element Q12 is lower than the predetermined voltage threshold (S38: YES), the CPU turns on the low voltage side switching element Q12 (S39).

  Thereafter, the CPU estimates the current flowing through the high voltage side switching element Q11 and waits until the estimated current becomes smaller than a predetermined current threshold (S40). Next, the CPU determines whether or not to end the step-up conversion (S41). If not (S41: NO), the CPU moves the process to step S31 in order to shift the voltage conversion circuit 1 to the first mode. On the other hand, when the step-up conversion is terminated (S41: YES), the CPU ends the processing of FIG. 21 after turning off the high-voltage side switching element Q11 and the low-voltage side switching element Q12 (S42).

  Next, the ripple current flowing through the inductors L12 and L22 will be described. The control unit 5 alternately executes the boost control of FIG. 21 for the voltage conversion circuit 1 with a delay of T / 2 for the voltage conversion circuit 2 as well. That is, the drive signals for turning on / off the SW elements Q12 and Q22 are out of phase with each other by π.

  When the SW element Q12 is turned on in the voltage conversion circuit 1, the current in the inductor L12 increases during the period of the state U1. Since current is induced in the inductor L22 so as to cancel the current flowing through the inductor L12 during this period, the current of the inductor L22 also increases during the same period. The increased current flows back to the inductor L22 via the capacitor 42 and an external battery (not shown) and the capacitor C2 or the SW element Q22.

  Similarly, when the SW element Q21 is turned on in the voltage conversion circuit 2, the current of the inductor L22 increases in a period delayed by T / 2 from the period of the state U1. Since current is induced in the inductor L12 so as to cancel the current flowing through the inductor L22 during this period, the current of the inductor L12 also increases during the same period. For this reason, the current of the inductor L12 repeatedly increases and decreases in a T / 2 cycle.

  In the second embodiment, as in the case of the first embodiment, it is clear that the ripple current and the leakage inductance are in an inversely proportional relationship. The same applies to the case where the effective cross-sectional area of the core is proportional to the leakage inductance when the magnetic flux density in the core is constant. Although illustration is omitted, the ripple current is reduced as the coupling coefficient k approaches 1 while keeping the leakage inductance constant, as in the case of the first embodiment.

  As described above, according to the second embodiment, the control unit 5 alternately turns on / off the SW elements Q12 and Q22, whereby the voltages input to the terminals L and G of the two voltage conversion circuits 1 and 2 are changed. The voltage is boosted and output to terminals H and G in parallel.

  The control unit 5 causes the voltage conversion circuit 1 to operate in the first mode in which the SW element Q12 is turned on while the SW element Q11 is turned off and current is supplied to the inductor L12. Thereby, energy is accumulated in the inductor L12. Thereafter, the control unit 5 turns off the SW element Q12 and operates in the second mode in which the resonance current from the inductor L11, the inductor L12, and the capacitor C1 flows. Thereafter, when the voltage of the capacitor C1, that is, the voltage of the SW element Q12 becomes lower than a predetermined voltage threshold due to resonance, the control unit 5 turns on the SW element Q12 to shift to the first mode. The control unit 5 performs similar boost control on the voltage conversion circuit 2 at different phases in the same cycle while periodically performing the above-described series of boost control on the voltage conversion circuit 1. As a result, the SW elements Q12 and Q22 of the two voltage conversion circuits 1 and 2 are alternately turned on / off.

  That is, since the inductors L12 and L22 are wound around the core 3 so as to magnetically cancel each other, even if the leakage inductances Le1 and Le2 of the inductors L12 and L22 are reduced within a certain limit, the inductors L12 and L22 The voltage conversion circuits 1 and 2 are step-up controlled within a range in which the ripple current of L22 is not increased. Therefore, the inductors L12 and L22 in the voltage conversion circuits 1 and 2 that are driven in different phases and connected in parallel are wound together with the size of the core 3 being reduced in accordance with the reduction of the leakage inductances Le1 and Le2. It is possible to reduce the size of the rotated core 3.

According to the second embodiment, the control unit 5 turns on the SW element Q11 when the voltage of the capacitor C1 in the second mode is higher than the second voltage threshold, and turns on the SW element Q12 after turning it off. To shift to the first mode.
Therefore, it is possible to output current to the terminals H and G via the SW element Q11 having a low on-resistance.

In the first and second embodiments, only one set of voltage conversion circuits 1 and 2 is provided. However, the number of sets of voltage conversion circuits 1 and 2 is not limited to 1, and N sets (N is Natural number) voltage conversion circuits 1 and 2 may be provided and these may be connected in parallel.
When N sets of voltage conversion circuits 1 and 2 are provided, the control unit 5 alternately turns on / off the SW elements Q11 and Q21 included in the N voltage conversion circuits 1 and 2 with a phase difference π. Both the SW elements Q11 and Q21 are turned on / off at a phase different by π / N. That is, all the SW elements Q11 and Q21 included in the N sets of voltage conversion circuits 1 and 2 are turned on / off at phases different by 2π / 2N.
Therefore, the switching loss can be evenly distributed to the N sets of voltage conversion circuits 1 and 2, and the ripple current included in the output can be reduced to 1 / N.

DESCRIPTION OF SYMBOLS 1, 2 Voltage conversion circuit 3 Core 3a, 3b, 3c Leg part 41, 42 Capacitor 5 Control part Q11, Q21 High voltage side switching element (SW element)
Q12, Q22 Low voltage side switching element (SW element)
Q13, Q23 SW element L11, L21 First inductor (inductor)
L12, L22 Second inductor (inductor)
C1, C2 capacitors H, L, G terminals

Claims (10)

A high-voltage side switching element and a low-voltage side switching element connected in series via a first inductor; a second inductor having one end connected to a connection point of the first inductor and the low-voltage side switching element; and the low-voltage side switching element Voltage conversion circuit comprising two voltage conversion circuits having capacitors connected in parallel to the elements, and a control unit for alternately turning on / off the high-voltage side switching elements, wherein the two voltage conversion circuits are connected in parallel A device,
The second inductors are wound around a core so as to cancel each other magnetically,
The controller is
When the voltage conversion circuit performs step-down conversion, the high-voltage side switching element is turned on and operated in a first mode in which a resonance current is flown by the first inductor, the second inductor, and the capacitor,
In the first mode, when the current of the high-voltage side switching element is less than a predetermined current threshold, the high-voltage side switching element is turned off, and the second mode in which the reflux current of the two inductors flows through the low-voltage side switching element or the capacitor To work with
A voltage conversion device that turns on the high-voltage side switching element and shifts to the first mode after the operation in the second mode. The controller is
The voltage of the low-voltage side switching element is detected,
When the voltage of the low-voltage side switching element is lower than a predetermined voltage threshold in the second mode, the low-voltage side switching element is turned on,
The voltage converter according to claim 1, wherein the high-voltage side switching element is turned on after the low-voltage side switching element is turned off. A high-voltage side switching element and a low-voltage side switching element connected in series via a first inductor; a second inductor having one end connected to a connection point of the first inductor and the low-voltage side switching element; and the low-voltage side switching element A voltage conversion circuit comprising: two voltage conversion circuits having capacitors connected in parallel to the elements; and a control unit that alternately turns on and off the low-voltage side switching elements, wherein the two voltage conversion circuits are connected in parallel. A device,
The second inductors are wound around a core so as to cancel each other magnetically,
The controller is
When the voltage conversion circuit performs step-up conversion, the low-voltage side switching element is turned on to operate in a first mode in which a current flows through the second inductor,
After the operation in the first mode, the low-voltage side switching element is turned off to operate in the second mode in which a resonance current is generated by the first inductor, the second inductor, and the capacitor,
A voltage conversion device that turns on the low-voltage side switching element to shift to the first mode when the voltage of the low-voltage side switching element is lower than a predetermined voltage threshold in the second mode. The controller is
A voltage of the capacitor is detected;
When the voltage of the capacitor is higher than a second voltage threshold in the second mode, the high-voltage side switching element is turned on,
The voltage converter according to claim 3, wherein the low-voltage side switching element is turned on after the high-voltage side switching element is turned off.   5. The voltage conversion device according to claim 1, wherein the second inductor has a coupling coefficient in a range of −0.99 or more and −0.78 or less. N sets of the two voltage conversion circuits (N is a natural number),
Each set of voltage conversion circuits are connected in parallel,
6. The voltage converter according to claim 1, wherein the control unit turns on / off one or the other low-voltage side switching element of each set of voltage conversion circuits at a phase different by π / N. 6. A high-voltage side switching element and a low-voltage side switching element connected in series via a first inductor; a second inductor having one end connected to a connection point of the first inductor and the low-voltage side switching element; and the low-voltage side switching element Two voltage conversion circuits having capacitors connected in parallel to the elements, and a control unit for alternately turning on and off the high-voltage side switching elements, and winding the second inductor around the core so as to magnetically cancel each other A control method in which the voltage conversion circuit in the voltage conversion device in which the two voltage conversion circuits are connected in parallel is stepped down to the voltage conversion circuit,
Turning on the high-voltage side switching element to operate in a first mode in which a resonance current is caused to flow by the first inductor, the second inductor, and the capacitor;
In the first mode, when the current of the high-voltage side switching element is smaller than a predetermined current threshold value, the second mode in which the high-voltage side switching element is turned off and the reflux current of the two inductors flows through the low-voltage side switching element or the capacitor To work with
A voltage step-down control method for a voltage conversion circuit in which the high-voltage side switching element is turned on after the operation in the second mode to shift to the first mode. A high-voltage side switching element and a low-voltage side switching element connected in series via a first inductor; a second inductor having one end connected to a connection point of the first inductor and the low-voltage side switching element; and the low-voltage side switching element Two voltage conversion circuits having capacitors connected in parallel to the elements, and a control unit for alternately turning on and off the low-voltage side switching elements, and winding the second inductor around the core so as to magnetically cancel each other A control method in which the control unit in the voltage conversion device in which the two voltage conversion circuits are connected in parallel causes the voltage conversion circuit to perform boost conversion,
Turning on the low-voltage switching element and operating in a first mode in which a current flows through the second inductor;
After the operation in the first mode, the low-voltage side switching element is turned off to operate in the second mode in which a resonance current is generated by the first inductor, the second inductor, and the capacitor,
A step-up control method for a voltage conversion circuit that turns on the low-voltage side switching element and shifts to the first mode when the voltage of the low-voltage side switching element is lower than a predetermined voltage threshold in the second mode. A high-voltage side switching element and a low-voltage side switching element connected in series via a first inductor; a second inductor having one end connected to a connection point of the first inductor and the low-voltage side switching element; and the low-voltage side switching element Two voltage conversion circuits having capacitors connected in parallel to the elements, and a control unit for alternately turning on and off the high-voltage side switching elements, and winding the second inductor around the core so as to magnetically cancel each other A computer program for causing the voltage conversion circuit to perform step-down conversion in the control unit in a voltage conversion device in which the two voltage conversion circuits are connected in parallel,
In the control unit,
Turning on the high-voltage side switching element and operating in a first mode in which a resonance current from the first inductor, the second inductor, and the capacitor flows;
In the first mode, when the current of the high-voltage side switching element is less than a predetermined current threshold, the high-voltage side switching element is turned off, and the second mode in which the reflux current of the two inductors flows through the low-voltage side switching element or the capacitor Steps to work with,
And a step of turning on the high-voltage side switching element after the operation in the second mode to shift to the first mode. A high-voltage side switching element and a low-voltage side switching element connected in series via a first inductor; a second inductor having one end connected to a connection point of the first inductor and the low-voltage side switching element; and the low-voltage side switching element Two voltage conversion circuits having capacitors connected in parallel to the elements, and a control unit for alternately turning on and off the low-voltage side switching elements, and winding the second inductor around the core so as to magnetically cancel each other A computer program for causing the voltage conversion circuit to perform step-up conversion in the control unit in the voltage conversion device in which the two voltage conversion circuits are connected in parallel,
In the control unit,
Turning on the low-voltage side switching element and operating in a first mode in which a current flows through the second inductor;
Turning off the low-voltage side switching element after operation in the first mode and operating in a second mode in which a resonance current is flown by the first inductor, the second inductor, and the capacitor;
When the voltage of the low-voltage side switching element is lower than a predetermined voltage threshold in the second mode, the computer program for executing the step of turning on the low-voltage side switching element and shifting to the first mode.

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