JP2017216764A - Motor controller, and electric system using the same - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a motor controller capable of properly controlling a motor by accurately detecting current off set amount, which is included in a detected value of DC bus current, free from erroneous detection.SOLUTION: The motor controller includes: an electric power converter; a three phase motor which is driven by the electric power converter; and control means that controls the electric power converter by detecting a motor current flowing on the three-phase motor based on the detected value of the DC bus current, creating a command voltage based on the motor current, and using the command voltage. The control means calculates the current phase when the DC bus current does not include the current off set amount, and estimates the current off set amount included in the detected value of the DC bus current based on the current phase.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a motor control device that controls a motor current using a detected value of a DC bus current, and an electric system using the motor control device.

  In a motor control device including a power converter such as an inverter that drives a motor, in order to appropriately control the motor, in addition to flowing the motor current by appropriately controlling the magnitude and phase of the motor applied voltage, In order to reduce the number of current sensors to be detected, it is required to detect the motor current with high accuracy using the detected value of the DC bus current of the inverter.

  On the other hand, the prior art described in Patent Document 1 is known. In the present technology, an instantaneous value of the DC bus current is detected to detect a current offset amount of the DC bus current detection circuit. In particular, the DC bus current in the timing during which one of the high potential side switching element and the low potential side switching element of the inverter is turned off (zero voltage (V0) vector period) is detected as an offset compensation amount.

JP 2014-128087 A

  In the above prior art, when detecting the pulsed DC bus current flowing into the inverter with the PWM pulse, the current detection circuit uses the DC bus current detection value of the V0 vector section in which all three phases are turned off. Detect the offset amount.

  However, when a ground fault of the motor winding (short-circuit between winding and ground (GND)) occurs, the ground fault current flows in the PWM section for generating the V0 vector, and therefore the ground fault current is offset current. And the current detection accuracy is reduced by the offset correction process using the erroneously detected offset detection value. Also, when a motor winding power fault (short-circuit between winding and power supply) occurs, the power fault current flows to the DC bus in the PWM interval that generates the V0 vector, so the power fault current is mistaken for the offset current. The current detection accuracy is lowered by the offset correction processing using the detected offset detection value that is erroneously detected.

  Furthermore, since the offset is detected in the section of the V0 vector in the above prior art, when the PWM duty is close to 100%, the current detected by the V0 vector is buried in the current ringing. Detection accuracy decreases.

  Therefore, the present invention provides a motor control device that appropriately detects a current offset amount included in a detected value of a DC bus current current without erroneous detection and appropriately controls the motor, and an electric system using the same.

  In order to solve the above problems, a motor control device according to the present invention includes a power converter, a three-phase motor driven by the power converter, and a motor current flowing through the three-phase motor based on a detected value of the DC bus current. And a control unit that detects and creates a command voltage based on the motor current and controls the power converter using the command voltage. The control unit includes a DC bus current that does not include a current offset amount. The current phase in the case is calculated, and the current offset amount included in the detected value of the DC bus current is estimated based on the current phase.

  According to the present invention, since the current offset amount is estimated based on the current phase when the current offset amount is not included, the current offset amount included in the detected value of the DC bus current can be detected accurately without erroneous detection. Can do.

  Problems, configurations, and effects other than those described above will become apparent from the following description of embodiments.

It is a block diagram which shows the structure of the motor control apparatus which is 1st Embodiment. The relationship between an output voltage vector, a motor current, and a DC bus current is shown. The PWM phase voltage pulse and the DC bus current are shown. 1 shows a block diagram of an offset detector. An example of a detection current including a current offset is shown. An estimated DC bus current calculated from the voltage phase and the current estimation phase and a current detection value detected based on the DC bus current are shown. The relationship of the d-axis voltage and d-axis current in 1st Embodiment is shown. It is a pulse waveform diagram showing the operation of the pulse shift calculator in the first embodiment. It is a block diagram of the electric power steering apparatus which is 2nd Embodiment. It is a system block diagram which shows the structure of the vehicle brake device which is 3rd Embodiment.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. In each figure, the same reference numerals indicate the same constituent elements or constituent elements having similar functions.

(First embodiment)
FIG. 1 is a block diagram showing a configuration of a motor control device having an inverter device according to the first embodiment of the present invention. In addition, the motor control apparatus of this embodiment is suitable for the use which drives a motor highly efficiently by changing the PWM carrier frequency of an inverter according to a motor output, and improving the detection precision of the DC bus current of an inverter. It is.

  As shown in FIG. 1, the motor control device 500 includes a three-phase motor 300 and an inverter device 100 that drives the three-phase motor 300.

  The inverter device 100 includes an inverter circuit 130 that is a power converter, a shunt resistor Rsh that detects a DC bus current of the inverter circuit, a current detector 120, a pulse shift calculator 230, an offset detector 240, and a three-phase / dq calculator 111. A current controller 110 and a PWM generator 220. The battery 200 is a DC voltage source of the inverter device 100, and the DC voltage (VB) of the battery 200 is converted into a three-phase AC voltage having a variable voltage and a variable frequency by the inverter circuit 130 of the inverter device 100. A voltage is applied to the three-phase motor 300.

  The inverter circuit 130 constitutes a three-phase bridge circuit composed of six semiconductor switching elements (IGBT (Insulated Gate Bipolar Transistor in FIG. 1)). These six semiconductor switching elements are turned on / off by a pulse width modulation signal, whereby the DC power of the battery 200 is converted into three-phase AC power.

  The three-phase motor 300 is a synchronous motor that is driven to rotate by supplying a three-phase AC voltage, for example, a permanent magnet synchronous motor. A rotational position sensor 320 is attached to the three-phase motor 300 in order to control the phase of the three-phase AC applied voltage in accordance with the phase of the induced voltage of the three-phase motor 300. The rotational position detector 150 calculates the detected position θ from the input signal of the rotational position sensor 320. Here, as the rotational position sensor 320, a resolver composed of an iron core and a winding, a magnetoresistive element such as a GMR (Giant Magneto Resistive) sensor, a sensor using a Hall element, or the like can be applied.

  The inverter device 100 has a current control function for controlling the output of the three-phase motor 300, and a pulsed DC bus current flowing into the inverter circuit 130 is inserted between the smoothing capacitor and the inverter circuit 130. It is detected as the voltage (current detection value Idc) across the shunt resistor Rsh. The shunt resistor Rsh is attached to the negative electrode side of the battery 200 in FIG. 1, but may be attached to the positive electrode side of the battery 200.

The current detector 120 detects at least two detection values (I _d1 , I _d2 ) as DC bus current values within the PWM1 period by the trigger timing (Trig) of the pulse shift calculator 230. The current corrector 250 calculates a three-phase motor current from the DC bus current value (I _d1 , I _d2 ), the current offset amount of the DC bus current detected by the offset detector 240, and the PWM pulse pattern (PWM). The value I _uvw (I _u , I _v , I _w ) is calculated.

The three-phase / dq calculator 111 uses the rotational position θ detected by the rotational position detector 150 to _{convert the} three-phase motor current value I _uvw (I _u , I _v , I _w ) into the dq axis in the rotational coordinate system. Conversion to current values (Id, Iq). Current controller 110, dq-axis current value _(I d, I _q) and, dq axis current command value created according to the target torque _{(I d *, I q *} ) so that the match, the rotating coordinate The dq axis voltage command value (V _d ^* , V _q ^* ) in the system is calculated. Further, the current controller 110 converts the dq voltage command value (V _d ^* , V _q ^* ) into the three-phase voltage command value V _uvw ^* (V _u ^* , V _v ^* , V _w ^* ) using the rotational position θ. Convert to and output.

The PWM generator 220 uses a three-phase voltage command value V _uvw ^* (V _u ^* , V _v ^* , V _w ^* ) as a modulation wave signal, a carrier wave signal (triangular wave, sawtooth wave, etc.) and a three-phase voltage command value V _uvw. By comparing with ^* , a so-called pulse width modulation pulse signal (PWM) is output. The PWM pulse signal performs on / off control of the semiconductor switch element in the inverter circuit 130 via a drive circuit (not shown). Thereby, the output voltage of the inverter circuit 130 is adjusted.

When the motor controller 500 controls the rotational speed of the three-phase motor 300, the motor rotational speed (ω _r ) is calculated from the time change of the rotational position θ (ω _r = dθ / dt), and the higher order A voltage command value or a current command value is created so as to coincide with the speed command from the controller. Further, when controlling the motor output torque, a dq axis current command value (I _d ^* , I _q ^* ) is calculated using a mathematical expression or a map showing the relationship between the dq axis current (I _d , I _q ) and the motor torque. Is created.

  Next, the operation for detecting the DC bus current and calculating the three-phase motor current will be described with reference to FIGS.

FIG. 2 shows the relationship between the output voltage vector (PWM pattern) of the inverter device, the three-phase motor currents (I _u , I _v , I _w ), and the DC bus current (I _dc ).

  As shown in FIG. 2, a voltage vector (V0 to V7) is output by turning on and off the semiconductor switching element of the inverter circuit 130 according to the PWM pattern. Each PWM pattern, the three-phase motor currents and the direction in which they flow, and the DC bus current correspond to each other as shown in FIG. 2, and the correspondence relationship shown in FIG. 2 is obtained from the detected value of the pulsed DC bus current. Based on this, a three-phase motor current can be calculated.

FIG. 3 shows PWM phase voltage pulses V _u , V _v , V _w for one cycle of the carrier frequency (PWM 1 cycle) and a pulsed DC bus current.

(A) shows the PWM generation timer operation. In (a), the PWM phase voltage pulse shown in (b) is generated at the timing when the sawtooth wave (which may be a triangular wave) matches the voltage command value. FIG. 3 shows the generation timing of the U-phase voltage pulse V _u as an example. The U-phase voltage pulse V _u rises at the timing T1 when the sawtooth waveform timer count value coincides with the voltage command value V _u 1, the U-phase voltage (V _u ) is output as the inverter output, and the saw with the voltage command value V _u 2 The U-phase voltage pulse V _u falls at the timing when the wavy timer count values match. Although not shown, the same applies to the V phase and the W phase. At this time, the U phase is the maximum phase and the W phase is the minimum phase.

(C) shows the DC bus current Idc flowing in the DC bus according to the relationship shown in FIG. 2 according to the phase voltage pulses V _u , V _v and V _w shown in (b). By performing current sampling twice in one PWM period, the motor current of two phases can be detected, and the remaining one phase can be obtained by calculation from the relationship of I _u + I _v + I _w = 0. As an example, in this view, a negative current of W-phase is the minimum phase (-I _w) is detected as I _d1 when the voltage vector V2 is output when the voltage vector V1 is output A positive current (I _u ) of the U phase that is the maximum phase is detected as I _d2 . As described above, the minimum phase current is detected at the first current detection timing and the maximum phase current is detected at the second current detection timing during one PWM period.

In order to reliably detect the peak of the pulsed DC bus current (I _d1 and I _{d2 in the} figure), a current detection width equal to or greater than the minimum pulse width (TPS) depending on the characteristics of the current detection circuit is required. Factors that determine the minimum pulse width (TPS) include the magnitude of the main circuit inductance of the inverter, the slew rate and response of the detection circuit, and the sampling time of the A / D converter. On the other hand, in this embodiment, in order to improve the detection accuracy with respect to a narrow PWM pulse, the pulse shift calculator 230 (FIG. 1) uses a signal difference (line voltage) between two phases of PWM pulses. ) In advance, and current sampling is executed at an appropriate detection timing (Trig) of the current detector 120.

  Next, the offset detector (240 in FIG. 1) in this embodiment will be described.

  FIG. 4 shows a block diagram of the offset detector 240. The offset detector 240 includes a speed calculation unit 241, a dq axis voltage command conversion unit 242, a voltage phase calculation unit 243, a current phase calculation unit 244, a phase angle estimation unit 245, and an offset derivation unit 246. First, the speed calculation unit 241, the dq axis voltage command conversion unit 242, and the voltage phase calculation unit 243 will be described.

  The speed calculation means 241 derives the electrical angular velocity ω of the three-phase motor by differentiating the electrical angle θ of the three-phase motor detected by the rotational position detector 150.

The dq-axis voltage command conversion unit 242 uses the equation (1) to convert the three-phase voltage commands V _u ^* , V _v ^* , V _w ^* using the electrical angle θ of the three-phase motor (θ _{d in} equation (1)). The dq-axis voltage commands V _d ^* and V _q ^* are converted.

The voltage phase calculation means 243 converts the dq axis voltage commands V _d ^* and V _q ^* into the voltage phase θ _Vdq using the equation (2).

  Next, the current phase calculation unit 244, the phase angle estimation unit 245, and the offset derivation unit 246 in the offset detector 240 will be described.

The current phase calculation unit 244 derives the current phase θ _Idq from the current not including the offset. The current phase θ _Idq ′ calculated from the detected current (I _dc , I _qc ) that can include an offset is expressed by Equation (3), and the actual current phase θ _Idq is expressed by Equation (4). When the current offset amount is 0, the detected dq-axis current (I _dc , I _qc ) matches the dq-axis current (I _d , I _q ) that actually flows through the three-phase motor. I _dc is the detected d-axis current, I _qc is the detected q-axis current, I _d is the d-axis current that actually flows through the three-phase motor, and I _q is the q-axis current that actually flows through the three-phase motor. is there.

However, when the offset is included, the current phase θ _Idq ′ calculated from the detected current is different from the actual current phase θ _Idq . Therefore, in this embodiment, the dq-axis voltage commands V _d ^* and V _q ^* and three motor constants, that is, the d-axis inductance, the q-axis inductance, the winding resistance R, and the induced voltage constant are used, and the current does not include an offset. The phase angle is estimated, and the current offset amount is obtained from the current phase angle.

Equation (5) is a voltage equation of a commonly used three-phase synchronous motor. V _d is a d-axis voltage, V _q is a q-axis voltage, L _d is a d-axis inductance, L _q is a q-axis inductance, K _e is an induced voltage constant, and p is a differential term. If the steady state is assumed, Equation (5) is transformed into Equation (6) with the differential term set to zero.

dq-axis command voltage _V d *, when estimating the d-axis estimated current ^ _{I d} and the q-axis estimated current ^ _{I q} with _{V q} *, the formula (7) is obtained from equation (6).

  When the d-axis estimated current and the q-axis estimated current are obtained from Expression (7), Expression (8) is obtained.

When the inverse matrix in Equation (8) is calculated and the d-axis estimated current is divided by the q-axis estimated current, the current estimated phase (θ _Idq ) is expressed by Equation (9).

In the case of a surface permanent magnet synchronous motor (SPMSM: Surface Permanent Magnet Synchronous Motor) in which the d-axis inductance L _d and the q-axis inductance L _q substantially match, Expression (9) is expressed using the addition theorem of the inverse function of the tangent. It can be simplified as in (10).

In the current phase calculation means 244, in the case of an interior permanent magnet synchronous motor (IPMSM) in which the d-axis inductance L _d and the q-axis inductance L _q are different, the equation (9) is used, and the d-axis inductance In the case of a surface magnet type synchronous motor in which L _d and q-axis inductance L _q substantially match, Expression (10) is used. The current estimation phase that does not include the current offset amount can be derived from Expression (9) and Expression (10). In addition, when Formula (10) is used, since the amount of calculation becomes small compared with Formula (9), calculation time can be shortened.

  Further, when the three-phase motor is stopped (ω = 0), the equation (10) can be simplified as the equation (11), and the calculation time can be shortened.

The phase angle estimation unit 245 uses the voltage phase θ _Vdq , the current phase θ _Idq, and the electrical angle θ, which are outputs of the voltage phase calculation unit 243, as described later, the first phase θ _mod1 and the second phase θ. _mod2 is derived.

FIG. 5 shows an example of a detected current including a current offset. In addition to the current detection value (c) detected based on the DC bus current, the three-phase motor voltage command value (a) and the three-phase motor current (b) are shown. In this example, the current phase is on the q axis that is perpendicular to the direction of the magnet magnetic flux of the three-phase motor, the voltage phase is advanced 30 degrees with respect to the q axis, the voltage phase θ _Vdq is 120 degrees, The estimated phase θ _Idq is 90 degrees. As indicated by the current detection value (c) in the figure, the waveform of the current detection value detected based on the DC bus current changes according to the maximum phase and the minimum phase of the three-phase voltage.

FIG. 6 shows the voltage phase θ _Vdq calculated from the equation (2), the estimated DC bus current calculated from the current estimated phase θ _Idq calculated from any of the equations (9) to (11), and the DC bus current. The current detection value detected based on is shown. Since the estimated minimum phase current and maximum phase current do not include an offset, the current offset I _ofst can be derived using the estimated current and the DC bus current detection value.

Hereinafter, means for deriving the estimated DC bus current from the voltage phase and the current phase will be described. Note that the peak value of the estimated current not including the current offset amount is _Ip . This peak value _Ip can be derived from the mean square sum of the d-axis estimated current ^ _Id and the q-axis estimated current ^ _{Iq in} the equation (8), but the first current detection timing and the second The calculation can be omitted when the current offset amounts substantially coincide with each other at the current detection timing (see (c) in FIG. 3).

When the current offset amounts substantially coincide with each other in the first current detection timing and the second current detection timing, the minimum phase current I _d1 and the maximum phase current I _d2 are expressed by Expression (12) and Expression (13), respectively. .

The first current phase angle θ _mod1 and the second current phase angle θ _mod2 are expressed by Expression (14) and Expression (15), respectively. Here, MOD (A, B) is a kind of so-called mathematical Modular operator, and indicates a remainder obtained by dividing A by B.

  Therefore, Equation (12) and Equation (13) show current waveforms that include the offset amount shown in FIG. 6 and have a partially sinusoidal shape.

From Expression (12) and Expression (13), the current offset amount I _ofst is expressed by Expression (16).

The peak value I _p is not included in the equation (16). Therefore, it can be derived from the DC current I _d1 detected at the first current detection timing, the DC current I _d2 detected at the second current detection timing, and the phases θ _mod1 and θ _mod2 . Therefore, the calculation load is reduced.

In the above description, the current offset amount generated between the minimum phase current I _d1 acquired at the first current detection timing and the maximum phase current I _d2 acquired at the second current detection timing is the same, but the offset amount is If they are different, the offset amount can be calculated individually at the first current detection timing and the second current detection timing using the equations (12) and (13) and the estimated current peak value _Ip. .

  As described above, according to the present embodiment, the current offset amount can be calculated based on the motor constant and the DC bus current detected at a predetermined timing. Is prevented. Therefore, since the detection accuracy of the current offset amount is improved, the accuracy of motor current detection based on the DC bus current is improved. Thereby, a three-phase motor can be controlled appropriately.

  By calculating the current offset amount as described above at a predetermined timing, it is possible to detect a failure in the current detection circuit of the DC bus current. For example, when the detected current offset amount is larger than a predetermined value, for example, 10% of the rated current, it is determined that an abnormality has occurred in the current detection circuit or the like, thereby preventing failure of the inverter device or the power supply. be able to. Further, when the current offset amount fluctuates more than a predetermined amount, it is determined that the motor constant is incompatible if it is determined to be abnormal when the three-phase motor to be driven is changed.

  The derivation of the current offset amount in the present embodiment uses a motor constant of a three-phase motor, so that the sensitivity to changes in the motor constant is high. A measuring means for the winding resistance R having the highest sensitivity among the motor constants will be described with reference to FIG.

  FIG. 7 shows the relationship between the d-axis voltage and the d-axis current in the present embodiment. In the present embodiment, the winding resistance R is measured from the relationship between the d-axis command voltage and the d-axis command current while changing the d-axis command voltage. Here, when there is a current offset amount and when there is no current offset amount, the relationship between the d-axis command voltage and the d-axis command current is expressed by equations (17) and (18), respectively (note that the q-axis current is zero). .

  As shown in equations (17) and (18) and FIG. 7, the relationship between the d-axis command voltage and the d-axis command current is represented by a straight line, the slope of the straight line indicates the winding resistance, and the current offset amount is a straight line. Appears as section size. Therefore, as shown in FIG. 7, when the d-axis voltage is applied to the three-phase motor while changing the operating point of the d-axis voltage, the d-axis command current at that time is plotted, and the slope of the approximate straight line is derived. The winding resistance can be obtained without being affected by the offset amount. If the value of this winding resistance is used, the current offset amount can be obtained with high accuracy even if the magnitude of the winding resistance changes with temperature or the like.

  When calculating the current offset amount using the motor constant as in this embodiment, when the three-phase motor is changed, the accurate current offset amount is not calculated due to variations in the motor constant, and the three-phase motor is excessive. There is a concern that torque pulsation due to a current offset amount may occur.

On the other hand, in the present embodiment, the current offset amount is calculated using the DC bus current detection values (I _d1 , I _d2 ) detected by the current detector 120 while using the motor constant for the calculation of the current phase. Yes. Thereby, the influence of the dispersion | variation in a motor constant is relieve | moderated. For example, when the motor speed is low and the motor can be almost regarded as being locked, the d-axis current command I _d ^* can be regarded as almost zero. At this time, according to the present embodiment, the current offset amount generated by the direct current due to the q-axis current can be detected. Therefore, even if there is a current detection error due to the current offset amount, the current can be controlled to be constant, and torque pulsation can be reduced. .

In the present embodiment, the DC bus current detection value detected at the general DC bus current detection timing shown in FIG. 3 is used, and as described above, the offset detector 240 shown in FIG. The current offset amount I _ofst can be calculated. The current corrector 250 (FIG. 1) calculates a DC bus current value that does not include the current offset amount from the detected current value (I _d1 , I _d2 ) and the current offset amount I _ofst , and further includes this current offset amount. The three-phase motor currents (I _u , I _v , I _w ) are calculated based on the current DC bus current value. Thereby, highly accurate motor control becomes possible.

  Next, the minimum pulse width TPS capable of detecting the DC bus current will be described with reference to FIG.

FIG. 8 is a pulse waveform diagram showing the operation of the pulse shift calculator 230 (FIG. 1) in the present embodiment. (A) is a sawtooth timer value indicating a carrier cycle for generating a PWM pulse, (b) is a PWM pulse of a general inverter, and shows a 1 PWM section with an instantaneous voltage command, (c) is A DC bus current waveform I _dc is shown. The pulse width between each of the U-phase PWM pulse width U _pw , V-phase PWM pulse width V _pw and W-phase PWM pulse width W _pw is smaller than the time Tad required for sampling of the A / D converter and can be detected by a microcomputer or the like. difficult. When a small motor current is applied, a line voltage is applied to the motor according to the signal difference between the PWM pulses of the three phases, and the motor current flows. However, if the minimum pulse width TPS is not ensured, the DC bus current Idc cannot be reliably detected, and appropriate motor current control cannot be performed.

Therefore, as shown in (d), the DC bus current Idc can be detected by generating the minimum pulse width TPS by phase-shifting (pulse-shifting) the inter-phase waveform of the PWM pulse. The pulse width (U _pw , V _pw , W _pw ) of each phase shown in (d) is the same as the pulse width of each phase in (b), and on the falling edge side of the PWM pulse, with reference to the V-phase pulse The phase of the U phase pulse is delayed by the pulse shift amount Tt2, the pulse width is widened so that the interphase pulse width of the U phase pulse and the V phase pulse becomes the minimum pulse width TPS, and A / D sampling is performed.

  On the rising edge side of the PWM pulse, the inter-phase pulse width of the U-phase pulse and the V-phase pulse is reduced, and as shown in FIG. An inversion pulse is generated. As a result, while generating a sufficient A / D sampling time, the average value of the motor applied voltage can be equivalent to the case where the pulse shift shown in (b) is not performed within one PWM interval, and the motor is adjusted by adjusting the motor applied voltage and phase. Can be controlled. At this time, in the DC bus current waveform Idc shown in (e), the area of the current pulse on the rising side of the PWM pulse edge becomes smaller (in the figure, it becomes a negative area) and the PWM pulse edge rises. The area of the current pulse on the falling side increases. The total area of the current pulse in the 1 PWM section shown in (e) is equivalent to the area of the current pulse in the 1 PWM section shown in (c), but the current detection value detected by A / D is the pulse shift equivalent current (Ips). DC bus current value (I1, I2) which is increased only by

  In this way, even when PWM current is shifted, such as when a motor current is detected near zero, the current offset is detected and the current detection value is offset compensated to calculate the motor current with high accuracy. The motor can be controlled appropriately.

  As described above, according to the present embodiment, since the current offset amount is estimated based on the current phase when the current offset amount is not included, the current offset amount included in the detected value of the DC bus current is erroneously detected. And can be detected with high accuracy. Accordingly, since the offset correction amount can be detected even when the motor has a ground fault and a power fault, the motor current detection accuracy can be improved and the motor can be controlled with high accuracy. Further, even when the PWM duty is close to 100%, the current offset amount can be detected.

(Second Embodiment)
Next, an electric power steering apparatus will be described as an example of an electric system to which the motor control apparatus according to the present invention is applied.

  FIG. 9 is a configuration diagram of an electric power steering apparatus according to the second embodiment of the present invention.

  As shown in FIG. 9, the electric actuator includes a torque transmission mechanism 902 and a motor control device 500 that applies torque to the torque transmission mechanism 902. The motor control device 500 includes a motor 300 that outputs torque, an inverter device 100 that drives the motor 300, and the like, and the first embodiment described above is applied.

  The electric power steering apparatus includes an electric actuator, a handle (steering wheel) 900, a steering detector 901, and an operation amount command unit 903, and the operation force of the handle 900 that is steered by the driver is torque-assisted by the electric actuator. .

The torque command τ ^* for the electric actuator is a steering assist torque command for the handle 900 and is generated by the operation amount command unit 903 according to the operation force of the handle 900 detected by the steering detector 901. The operating force (steering force) applied to the steering wheel 900 by the driver is reduced by the motor torque output from the electric actuator in response to the torque command τ ^* .

The motor control device 500 receives the torque command τ *, and uses the motor constant of the motor 300 set in advance to compensate the current offset amount as described above, while the output torque τ _m of the motor 300 becomes the torque command value τ. ^* Control motor current to follow.

The output torque τ _m of the motor 300 output from the output shaft directly connected to the rotor of the motor 300 is transmitted through a torque transmission mechanism 902 using a speed reduction mechanism such as a worm, a wheel, or a planetary gear, or a hydraulic mechanism. 910 is transmitted. As a result, the driver's operating force is reduced (assisted) by the electric force, and the steering angles of the wheels (tires) 920 and 921 are operated.

  At this time, the assist amount, that is, the torque command τ is set to the operation amount detected as the steering angle or the steering torque by the steering detector 901 that detects the steering state incorporated in the steering shaft, and the state amount such as the vehicle speed or the road surface state. Based on the operation amount command unit 903 based on this.

  According to the present embodiment, by applying the motor control device according to the first embodiment, when the vehicle is put in the garage or the like, or when the steering operation at extremely low vehicle speed is repeated for a long time, an excessive motor is used. A current offset amount due to a temperature drift of the current detection circuit when the temperature of an ECU (Electronic Control Unit) rises due to flowing a large motor current to obtain an output can be compensated with high accuracy. Thereby, the motor controllability is prevented from being lowered, and the operation force can be assisted appropriately. Furthermore, even when a variation in pulse shift current due to a change in the di / dt characteristic of the inverter drive circuit occurs, the current can be detected with high accuracy, so that the operating force can be assisted smoothly even during the steering wheel turning operation.

(Third embodiment)
Next, a vehicle brake device will be described as an example of an electric system to which the motor control device according to the present invention is applied.

  FIG. 10 is a system block diagram showing the configuration of the vehicle brake device according to the third embodiment of the present invention.

  The motor control apparatus according to the first embodiment described above is applied to the assist control unit 706 and the motor 731 in FIG. The control microcomputer in the assist control unit 706 having the inverter device and its control unit is programmed so as to perform a vehicle brake operation. Further, the motor 731 is integrally attached to the braking assist device 700 and further has an integral structure with the assist control unit 706 via the casing 712.

  The vehicle brake device includes a brake pedal 701, a braking assist device 700, a booster device 800, and wheel mechanisms 850a to 850d. The braking assist device 700 includes an assist mechanism 720, a master cylinder 721, and a reservoir tank 712. Inside the master cylinder 721, a primary piston 726 and a secondary piston 727 are disposed, thereby forming a primary liquid chamber 721a and a secondary liquid chamber 721b. The amount of operation of the brake pedal 701 that the driver steps on is input to the assist mechanism 720 via the input rod 722, and is transmitted to the primary fluid chamber 721a by moving the primary piston 726 to generate fluid pressure. Further, the secondary piston 727 is moved by the hydraulic pressure generated in the primary hydraulic chamber 721a, so that hydraulic pressure is also generated in the secondary hydraulic chamber 721b. A wheel cylinder 851 is provided in each of the wheel mechanisms 850a to 850d. Each wheel cylinder 851 brakes the vehicle by the hydraulic pressure applied from the master cylinder 721.

  Further, a brake operation amount detected by a stroke sensor 702 attached to the brake pedal 701 or a pedaling force sensor (not shown) is input to an assist control unit 706 that controls the assist mechanism 720. The assist control unit 706 controls the motor 731 so that the rotational position is in accordance with the input brake operation amount. The rotational torque of the motor is transmitted to a rotation-translation conversion device 725 that converts rotational power into translational power via the speed reduction device 723, presses the primary piston 726, and increases the hydraulic pressure in the primary fluid chamber 721a. The secondary piston 727 is pressurized to increase the fluid pressure in the secondary fluid chamber 721b.

  The booster mechanism 800 inputs the hydraulic pressure of the hydraulic fluid pressurized in the liquid chambers 721a and 721b via the master pipes 750a and 750b, and applies the hydraulic pressure to the wheel mechanisms 850a to 850d according to the command of the boost control unit 830. To communicate. Thereby, the braking force of the vehicle is obtained.

  The assist control unit 706 controls the displacement amount of the primary piston 726 in order to adjust the pressing amount of the primary piston 726. Since the displacement amount of the primary piston 726 is not directly detected, the rotation angle of the drive motor 731 is calculated based on a signal from a rotation position sensor (not shown) provided in the motor 731, and rotation-translation. The displacement amount of the primary piston 726 is obtained by calculation from the propulsion amount of the conversion device 725.

  If the drive motor 731 stops due to a failure and the return control of the shaft of the ball screw 725 becomes impossible, the braking force of the driver is hindered by returning the shaft of the ball screw 725 to the initial position by the reaction force of the return spring 728. I try not to. For this reason, instability of the vehicle behavior due to brake drag can be avoided.

  The booster mechanism 800 includes two systems of hydraulic pressure adjustment mechanisms 810a and 810b that adjust hydraulic fluid for each of the two diagonal wheels of the four wheels, and stably stops the vehicle even if one system failure occurs. The braking force of the diagonal two-wheel wheel mechanisms 850a and 850b can be individually adjusted. Since the two systems of hydraulic pressure adjustment mechanisms 810a and 810b operate in the same manner, the hydraulic pressure adjustment mechanism 810a will be described below.

  The hydraulic pressure adjustment mechanism 810a controls the supply of hydraulic fluid to the pump 853 that boosts the master pressure generated by the hydraulic fluid pressure from the master pipe 750a, the pump motor 852 that drives the pump 853, and the wheel cylinder 851. Gate OUT valve 811, gate IN valve 812 that controls supply of hydraulic fluid to pump 853, and IN valve that controls hydraulic fluid pressure from master pipe 750 a or supply of hydraulic fluid from pump 853 to each wheel cylinder 851 814a and 814b and OUT valves 813a and 813b for controlling the pressure reduction of the wheel cylinder 851 are provided.

  For example, when hydraulic control is performed for anti-lock brake control, a signal from the wheel rotation sensor 853 in the wheel mechanisms 850a and 850b is processed by the boost control unit 830 to detect wheel lock during braking. Then, each IN / OUT valve (electromagnetic type) and pump 853 are operated to adjust the hydraulic pressure of the hydraulic fluid to such a hydraulic pressure value that each wheel does not lock. Note that this hydraulic pressure adjustment mechanism can also be applied when hydraulic pressure control is performed for vehicle behavior stabilization control.

  In the vehicle brake device, the assist control unit 706 is configured to include the motor control device described in the first embodiment, and the motor control device is used to control the displacement amount of the primary piston 726, and the driver. The brake pedal depressing force by is stably assisted. For this reason, in a vehicle brake device, it is possible to continue a highly accurate and stable operation and to detect an abnormality accurately.

  Further, when the charge capacity of the battery 200 that is a normal power source is reduced, the assist amount is reduced, so that the brake assist operation can be continued using the auxiliary power source 400 as a power source. In this case, since the auxiliary power supply 400 is for emergency backup, it is required to suppress a large current output.

  On the other hand, the assist control unit 706 in the present embodiment may limit the current from the power source to about 1/10 of the normal when the power source is switched to the auxiliary power source 400. Even when such a current is small, the motor control device can control the power source current with high accuracy by detecting the DC bus current with high accuracy and controlling the motor current. Thereby, stable brake assist control can be continued even when the battery is abnormal.

  According to the present embodiment, when the hydraulic fluid pressure is substantially constant, the current phase angle calculated by the voltage command is used to estimate the current offset amount (current detection error) included in the detected value of the DC bus current. By using it (see the above-described equation (11)), the current offset amount can be estimated with high accuracy and in a short time.

  In addition, this invention is not limited to embodiment mentioned above, Various modifications are included. For example, the above-described embodiments have been described in detail for easy understanding of the present invention, and are not necessarily limited to those having all the configurations described. Further, it is possible to add, delete, and replace other configurations for a part of the configuration of each embodiment.

DESCRIPTION OF SYMBOLS 100 ... Motor control apparatus, 110 ... Current controller, 111 ... Three phase / dq converter,
120 ... current detector, 130 ... inverter circuit, 150 ... rotational position detector,
200 ... battery, 220 ... PWM generator, 230 ... pulse shift calculator,
240 ... offset detector, 241 ... speed calculation means, 242 ... dq axis voltage command converter,
243 ... Voltage phase calculation means, 244 ... Current phase calculation means, 245 ... Phase angle estimation means,
246 ... Offset derivation means, 300 ... Three-phase motor, 320 ... Rotation position sensor,
400 ... Auxiliary power source, 500 ... Motor control device, 700 ... Brake assist device,
701 ... Brake pedal, 702 ... Stroke sensor, 706 ... Assist control unit,
712 ... Reservoir tank, 720 ... Assist mechanism, 721a ... Primary liquid chamber,
721b ... secondary liquid chamber, 722 ... input rod, 723 ... speed reducer,
725 ... Ball screw, 726 ... Primary piston, 727 ... Secondary piston,
728 ... Return spring, 731 ... Motor, 750a, 750b ... Master piping,
800 ... Boost mechanism, 810a, 810b ... Hydraulic pressure adjustment mechanism, 811 ... Gate OUT valve,
812 ... Gate IN valve, 813a, 813b ... OUT valve, 814a, 814b ... IN valve,
830 ... Boost control unit, 850a, 850b, 850c, 850d ... Wheel mechanism,
851 ... Wheel cylinder, 852 ... Pump motor, 853 ... Pump,
900 ... handle, 901 ... steering detector, 902 ... torque transmission mechanism,
903 ... Operation amount command device, 910 ... Rack, 920, 921 ... Wheel

Claims (17)

A power converter;
A three-phase motor driven by the power converter;
Control means for detecting a motor current flowing through the three-phase motor based on a detected value of a DC bus current, creating a command voltage based on the motor current, and controlling the power converter using the command voltage;
In a motor control device comprising:
The control means calculates a current phase when the DC bus current does not include a current offset amount, and estimates the current offset amount included in the detected value of the DC bus current based on the current phase. A motor control device. The motor control device according to claim 1,
The motor control device, wherein the current phase is calculated from the command voltage, a motor constant of the three-phase motor, and a rotation speed of the three-phase motor. The motor control device according to claim 2,
The command voltage is a d-axis command voltage and a q-axis command voltage,
The motor control device characterized in that the motor constants are winding resistance, d-axis inductance, q-axis inductance, and induced voltage constant of the three-phase motor. The motor control device according to claim 3,
The d-axis command voltage and the q-axis command voltage are V _d ^* and V _q ^* , respectively.
The winding resistance, the d-axis inductance, the q-axis inductance and the induced voltage constant, respectively, and _R, and L d, _{L q} and _{K e,}
The rotation speed is ω,
The current phase (θ _Idq ) is given by equation (9)

The motor control device calculated by the following. The motor control device according to claim 1,
The motor control device according to claim 1, wherein when the three-phase motor is stopped, the current phase is calculated from the command voltage. The motor control device according to claim 5,
The motor control device, wherein the command voltage is a d-axis command voltage and a q-axis command voltage. The motor control device according to claim 6,
The d-axis command voltage and the q-axis command voltage are represented as V _d ^* and V _q ^* , respectively.
The current phase (θ _Idq ) is _expressed by equation (11),

The motor control device calculated by the following. The motor control device according to claim 1,
The control means further calculates a voltage phase of the command voltage, and based on the voltage phase, the current phase, the detected value of the DC bus current, and a rotational position of the three-phase motor, A motor control device that estimates a current offset amount. The motor control device according to claim 8, wherein
The motor control device according to claim 1, wherein the detected value of the DC bus current is two detected values detected at different timings. The motor control device according to claim 9,
The current phase is θ _Idq ,
The voltage phase is θ _Vdq ,
The two detection values are I _d1 and I _d2 ,
Assuming that the rotational position of the three-phase motor is θ,
The current offset amount (I _ofst ) is expressed by Equation (14),

Formula (15),

Formula (16)

A motor control device estimated by The motor control device according to claim 3,
The motor control device, wherein the control means derives the winding resistance based on a command current and the command voltage. The motor control device according to claim 1,
The motor control apparatus according to claim 1, wherein the control means determines an abnormality when the current offset amount is larger than a predetermined value. The motor control device according to claim 1,
A motor control device, wherein an abnormality is determined based on the current offset amount when the three-phase motor is changed to another three-phase motor having a different motor constant. A power converter;
A three-phase motor driven by the power converter;
Control means for detecting a motor current flowing through the three-phase motor based on a detected value of a DC bus current, creating a command voltage based on the motor current, and controlling the power converter using the command voltage;
In a motor control device comprising:
The control means includes
An offset detector for detecting a current offset amount included in the detected value of the DC bus current;
The offset detector estimates a current phase of the motor current from a voltage command and a motor constant,
A motor control device that performs current control by correcting the current offset amount from the current phase. A handle,
An electric actuator for torque assisting the operating force of the handle;
In an electric power steering apparatus comprising:
The electric actuator includes the motor control device according to claim 1. A piston that moves by actuation of an electric motor to apply a brake fluid pressure to a wheel mechanism of the vehicle;
A control device that moves the piston by controlling the electric motor based on an operation amount of a brake pedal,
The said control apparatus is comprised including the motor control apparatus described in Claim 1, The brake device for vehicles characterized by the above-mentioned. The vehicle brake device according to claim 16, wherein
A battery that is a normal power source of the control device;
An auxiliary power source serving as a power source for the control device when the battery is abnormal;
Even after switching to the auxiliary power source, the vehicle brake device continues to be controlled by the motor control device according to claim 1.

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