DE112017002259T5 - Motor control device and electrical system using this - Google Patents

Abstract

There is provided a motor control apparatus capable of accurately detecting the amount of a current offset included in a detected value of a DC bus current without error and controlling a motor appropriately.
The engine control device includes a power converter; a three-phase motor configured to be driven by the power converter; and a control unit configured to detect a motor current flowing through the three-phase motor based on a detected value of a DC bus current, to generate a command voltage based on the motor current, and to control the power converter using the command voltage, wherein the control unit is configured to: to calculate a current phase for a case where the DC bus current does not include a current offset, and to estimate the amount of current offset included in the detected value of the DC bus current based on the current phase.

Description

Technical area

The present invention relates to a motor control apparatus which controls a motor current intensity using a detected value of a DC bus current, and to an electric power system using the motor control apparatus.

State of the art

In order to suitably control the motor in a motor control apparatus including a power converter such as an inverter for driving a motor, in addition to properly controlling the magnitude and phase of a voltage applied to the motor, it is necessary to provide motor current with high accuracy to detect, using a detected value of DC bus current strength of the inverter to reduce the number of current intensity sensors for detecting the motor current intensity.

As the prior art in this technical field, there is a disclosure in PTL 1. In this technology, the amount of current offset of a DC bus current-magnitude detection circuit is detected by detecting a DC bus current value instantaneous value. More specifically, a DC bus current at a timing of a period (zero voltage vector period (VO vector period)) to which either a high potential side switching element and a low potential side switching element of an inverter is turned off is detected as an offset correction amount.

Citation List

Patent Document (s)

PTL 1: JP-A-2014-128087

Summary of the invention

Technical problem

In the prior art, in the case of detecting a PWM pulse and a pulsed DC bus current flowing into an inverter, the detected values of a DC bus current in a VO vector period during which all three phases are turned off are used to detect the offset current of a current detection circuit ,

However, when a ground fault of a motor winding (a short between the winding and ground (GND)) occurs, a reverse current of a ground current flows in a PWM section to generate a VO vector. As a result, the ground leakage current is erroneously detected as the offset current, and thus the current detection accuracy deteriorates due to offset correction processing using the erroneously detected offset current. In addition, when a supply short-circuit fault of the motor winding (a short circuit between the winding and a power supply) occurs, a supply short-circuit current flows to a DC bus in the PWM section for generating the VO vector. As a result, the supply short-circuit current is erroneously detected as the offset current, and thus the current-detection accuracy deteriorates due to offset correction processing using the erroneously detected offset current.

Further, in the prior art, since an offset in the VO vector period is detected, when the duty of a PWM becomes almost 100%, a current detected in the VO vector period is concealed in a current overshoot, and thus one deteriorates current detection accuracy.

Accordingly, the present invention provides a motor control apparatus capable of accurately detecting the amount of a current offset included in a detected value of a DC bus current without error and controlling a motor appropriately, and an electric system incorporating the Motor control device used.

Solution to the problem

To solve the above problem, a motor control apparatus includes a power converter; a three-phase motor configured to be driven by the power converter; and a control unit configured to detect a motor current flowing through the three-phase motor based on a detected value of a DC bus current, to generate a command voltage based on the motor current, and to control the power converter using the command voltage, wherein the control unit is configured to: to calculate a current phase for a case where the DC bus current does not include a current offset, and to estimate the amount of current offset included in the detected value of the DC bus current based on the current phase.

Advantageous effect of the invention

According to the present invention, since the amount of current offset in a case where no current offset is included is estimated based on a current phase, the amount of current offset included in a detected value of DC bus current can be accurately detected without error.

Other problems, structures and effects than those described above will become apparent from the following description of the embodiments.

list of figures

• 1 FIG. 10 is a block diagram showing a configuration of a motor control apparatus according to a first embodiment. FIG.
• 2 FIG. 16 shows a relationship between an output voltage vector, a motor current value, and a DC bus current.
• 3 FIG. 12 is a diagram showing a PWM phase voltage pulse and a DC bus current. FIG.
• 4 is a block diagram of an offset detector.
• 5 FIG. 15 is a diagram showing an example of a detected current that includes a current offset.
• 6 FIG. 12 is a graph showing an estimated DC bus current strength calculated from a voltage phase and an estimated current phase and a detected current value that is detected based on a DC bus current. FIG.
• 7 FIG. 15 is a diagram showing a relationship between a d-axis voltage and a d-axis current in the first embodiment.
• 8th FIG. 12 is a pulse waveform diagram showing an operation of a pulse shift calculator in the first embodiment. FIG.
• 9 FIG. 10 is a configuration diagram of an electric power steering apparatus according to a second embodiment. FIG.
• 10 FIG. 10 is a system block diagram showing a configuration of a vehicle brake device according to a third embodiment. FIG.

Description of embodiments

Hereinafter, embodiments of the present invention will be described with reference to the drawings. In each drawing, like reference numerals denote like components or components having similar functions.

First Embodiment

1 FIG. 10 is a block diagram showing a configuration of a motor control device having an inverter device according to a first embodiment of the present invention. Here, the motor control apparatus according to the present embodiment is capable of detecting a detection accuracy To improve DC bus current of an inverter by the PWM carrier frequency is switched in accordance with the motor output signals, whereby it is suitable for driving a motor with high efficiency.

As in 1 shown comprises a motor control device 500 a three-phase motor 300 and an inverter device 100 that the three-phase motor 300 controls.

The inverter device 100 includes an inverter circuit 130 , which is a power converter, a shunt resistor Rsh for detecting a DC bus current of the inverter circuit, a current-magnitude detector 120 , a pulse shift calculator 230 , an offset detector 240 , a three-phase / dq calculator 111 , a current controller 110 and a PWM generator 220 , A battery 200 is a DC power source of the inverter device 100 , A DC voltage (VB) of the battery 200 is through the inverter circuit 130 the inverter device 100 converted into a three-phase alternating voltage with variable voltage / variable frequency and the three-phase AC voltage is applied to the three-phase motor 300 created.

The inverter circuit 130 forms a three-phase bridge circuit with six semiconductor switching elements (IGBT (insulated gate bipolar transistor)) 1 ). The ON / OFF switching of these six semiconductor switching elements is controlled by a pulse width modulation signal, whereby the DC power of the battery 200 is converted into three-phase AC power.

The three-phase engine 300 is a synchronous motor which is rotatively driven by supplying a three-phase AC voltage, for example, a permanent magnet synchronous motor. A rotational position sensor 320 is on the three-phase motor 300 attached to the phase of an applied three-phase AC voltage in accordance with the phase of an induced voltage of the three-phase motor 300 to control. A rotational position detector 150 calculates a detected position θ from a signal input to the rotational position sensor 320 , As a rotational position sensor 320 Here, a magnetoresistive element such as a resolver having an iron core and a wire wound thereon and a GMR sensor (giant magnetoresistance sensor) or a sensor using a Hall element may be used.

The inverter device 100 has a current control function for controlling the output of the three-phase motor 300 and detects one in the inverter circuit 130 flowing, pulsed DC bus current as a voltage (a detected current value Idc) at both ends of the shunt resistor Rsh connected between a smoothing capacitor and the inverter circuit 130 is inserted. Even if the shunt resistor Rsh in 1 on the negative electrode side of the battery 200 is attached, the shunt resistor Rsh on the positive electrode side of the battery 200 to be appropriate.

The amperage detector 120 detects at least two detection values ( I _d1 and I _d2 ) as DC bus current values within a PWM period according to a trip timing (Trig) of the pulse shift calculator 230 , The amperage corrector 250 calculates the three-phase motor current values I _{and many more} ( I _u . I _v and I _w ) from the DC bus current values ( I _d1 and I _d2 ), an amount of current offset of the DC bus current through the offset detector 240 is detected, and a PWM pulse pattern PWM.

The three-phase / dq calculator 111 converts the three-phase motor current values I _{and many others} ( I _u . I _v and I _w ) in dq-axis current values ( I _d and I _q ) in a rotary coordinate system using the rotary position sensor 150 detected rotational position θ order. The amperage controller 150 computes the dq-axis voltage command values ( V _d * and V _q *) in the rotation coordinate system such that the dq-axis current values ( I _d and I _q ) with the dq-axis current command value ( I _d * and I _q *), which is created according to a target torque, are consistent. Furthermore, the amperage controller converts 110 the dq voltage command values ( V _d * and V _q *) using the rotational position in three-phase voltage command values V _uvw * ( V _u * V _v * and V _w *) and outputs the three-phase voltage command values V _uvw * ( V _u * V _v * and V _w *) out.

The PWM generator 220 compares carrier wave signals (triangular wave, sawtooth wave, etc.) with the three-phase voltage command value V _uvw * using the three-phase voltage command values V _uvw * (V _u *, V _v * and V _w *) as modulated wave signals, whereby a so-called pulse width modulation pulse signal (PWM pulse signal) is issued. The PWM pulse signal controls ON / OFF switching of a semiconductor switching element in the inverter circuit 130 via a drive circuit (not shown). As a result, the output voltage of the inverter circuit 130 customized.

In the case of controlling a rotational speed of the three-phase motor 300 in the engine control device 500 Also, an engine speed (ω _r ) is calculated from the time-dependent change of the rotational position θ (ω _r = dθ / dt), and a voltage command value or a current command value is established to be consistent with a speed command from an upper controller. In the case of controlling an engine output torque, the dq-axis current command values ( I _d * and I _q *) using a mathematical expression or a map showing a relationship between the dq-axis currents ( I _d and I _q ) and a motor torque is generated.

Next are with reference to 2 and 3 Operations for detecting a DC bus current and calculating a three-phase motor current are described.

2 FIG. 15 is a diagram showing a relationship between an output voltage vector (PWM pattern) of an inverter device and three-phase motor currents (FIG. I _u . I _v and I _w ) and a DC bus current ( I _dc ) shows.

As in 2 voltage vectors are shown ( V0 to V7 ) when semiconductor switching elements of the inverter circuit 130 switched on and off according to the PWM pattern output. The PWM pattern, the three-phase motor current, flow directions thereof, and the DC bus current are the same as each other in FIG 2 and a three-phase motor current can be calculated from a detection value of a pulsed DC bus current based on the in 2 calculated relationship can be calculated.

3 is a diagram showing the PWM phase voltage pulses V _u . V _v and V _w which corresponds to one period of a carrier frequency (a PWM period) and a pulsed DC bus current.

A section (a) shows a PWM generation timer operation, and in (a) a PWM phase voltage pulse is generated as shown in (b) at a timing where a sawtooth wave (or triangle wave) coincides with a voltage command value. 3 shows by way of example a generation timing of a U-phase voltage pulse V _u , The U-phase voltage pulse V _u rises at a time T1 to which a voltage command value V _u 1 coincides with a timer count with a sawtooth waveform, the U-phase voltage V _u is output as an inverter output and the U-phase voltage pulse V _u rises at a timing to which a voltage command value V _u2 coincides with a timer count with a sawtooth waveform. Although not shown, the same applies to the V phase and the W phase. At this time, the U phase becomes the maximum phase and the W phase becomes the minimum phase.

A section (c) shows the DC bus current I _dc flowing through a DC bus, according to the in 2 in accordance with the phase voltage pulses shown in (b) V _u . V _v and V _w , A motor current of two phases can be detected by performing a current scan twice within a PWM period, and the remaining one phase can be obtained by a calculation based on the relationship I _u + I _v + I _w = 0 based. In 3 For example, if a voltage vector V2 is output, a negative current (- I _w ) with the W phase, which is the minimum phase, as I _d1 detected. If a voltage vector V1 is output, moreover, a positive current ( I _u ) with the U phase, which is the maximum phase, as I _d2 detected. In this regard, during a PWM period, a minimum phase current is detected at a first current detection time limit and a maximum phase current is detected at a second current level detection timing.

To lace ( I _d1 and I _d2 in the figure) of the pulsed DC bus current, a detected current width equal to or greater than the minimum pulse width (TPS) is required depending on the characteristics of a current detection circuit. Among the factors that determine the minimum pulse width (TPS) include the size of the main circuit inductance of an inverter, a slew rate and the response of a detection circuit, the sampling time of an A / D converter, etc. On the other hand, in the present embodiment, to improve the detection accuracy with respect to a narrow PWM pulse in the pulse shift calculator 230 ( 1 ) calculates a signal difference between PWM pulses of two phases (one pulse width of the line voltage) in advance, and a current-magnitude scan becomes a suitable detection timing (Trig) of the current-magnitude detector 120 carried out.

Next is the offset detector ( 240 in 1 ) in the present embodiment.

4 is a block diagram of the offset detector 240 , The offset detector 240 includes a speed calculation unit 241 , a dq-axis voltage command conversion unit 242, a voltage phase calculation unit 243 , a current phase calculation unit 244 , a phase angle estimation unit 245 and an offset derivation unit 246 , First, the speed calculation unit 241 , the dq-axis voltage command conversion unit 242, and the voltage phase calculation unit 243 described.

The speed calculation unit 241 conducts an electrical angular velocity ω of a three-phase motor by differentiating that through the rotational position detector 150 detected electrical angle θ of the three-phase motor.

The dq-axis voltage command conversion unit 242 converts three-phase voltage commands V _u *, V _v *, and V _w * to the dq-axis voltage commands V _d * and _d , using the electrical angle θ (θ _d in equation (1)) of the three-phase motor V _q * um.
[Equation 1] $$\left[\begin{array}{c}{V}_d^{*}\\ V_q^{*}\end{array}\right]=\frac{2}{3}\left[\begin{array}{lll}\text{cos}{\theta }_d\hfill & \text{cos}\left({\theta }_d-\frac{2}{3}\pi \right)\hfill & \text{cos}\left({\theta }_d+\frac{2}{3}\pi \right)\hfill \\ -\text{sin}{\theta }_d\hfill & -\text{sin}\left({\theta }_d-\frac{2}{3}\pi \right)\hfill & -\text{sin}\left({\theta }_d+\frac{2}{3}\pi \right)\hfill \end{array}\right]\left[\begin{array}{c}{V}_u^{*}\\ V_v^{*}\\ V_w^{*}\end{array}\right]$$

The voltage phase calculation unit 243 converts the dq-axis voltage commands V _d * and V _q * to a voltage _phase θ _Vdq using equation (2).
[Equation 2] $${\theta }_{Vdq}={\text{tan}}^{-1}\left(\frac{V_q}{V_d}\right)$$

Next are the current phase calculation unit 244 , the phase angle estimation unit 245 and the offset derivation unit 246 in the offset detector 240 described.

[Gleichung 4] $${\theta }_{Idq}={\text{tan}}^{-1}\left(\frac{I_q}{I_d}\right)$$

The current phase calculation unit 244 derives a current _phase θ _Idq from a current that contains no offset. One from detected currents ( I _dc and I _qc ) which may contain offset, calculated current _phase θ _Idq 'is expressed by Equation (3), while an actual current _phase θ _{Idq is expressed} by Equation (4). If the amount of a current offset 0 is to be detected, the dq-axis currents to be detected (I _dc and I _qc ) agree with the dq axis currents ( I _d and I _q ) that actually flow through the three-phase motor. Here, I _dc denotes a detected d-axis current, I _qc denotes a detected q-axis current, I _d denotes a d-axis current actually flowing through the three-phase motor, and Iq denotes a q-axis current actually flowing through the three-phase motor.
[Equation 3] $${\theta }_{Idq}\text{'}={\text{tan}}^{-1}\left(\frac{I_{qc}}{I_{dc}}\right)$$

[Equation 4] $${\theta }_{Idq}={\text{tan}}^{-1}\left(\frac{I_q}{I_d}\right)$$

However, if an offset is included, the current _phase θ _Idq 'calculated from a detected current _differs from the actual current _phase θ _Idq . Therefore, in the present embodiment, using the dq-axis voltage commands V _d * and V _q * and three motor constants, ie, a d-axis inductance, a q-axis inductance and a winding resistance R, and a Induction voltage constant estimated a current phase angle that does not contain the offset, and the amount of current offset is calculated from the current phase angle.

[Gleichung 6] $$\left[\begin{array}{c}{V}_d\\ V_q\end{array}\right]=\left[\begin{array}{cc}R& -\omega L_q\\ \omega L_d& R\end{array}\right]\left[\begin{array}{c}{I}_d\\ I_q\end{array}\right]+\left[\begin{array}{c}0\\ \omega K_e\end{array}\right]$$

Equation (5) is a voltage equation of a commonly used three-phase synchronous motor. V _d denotes a d-axis voltage, V _q denotes a q-axis voltage, L _d denotes a d-axis inductance, L _q denotes a q-axis inductance, K _e denotes an induced voltage constant and p denotes a differential term. Assuming that there is a stationary state, equation (5) is transformed into equation (6), where the differential term is zero.
[Equation 5] $$\left[\begin{array}{c}{V}_d\\ V_q\end{array}\right]=\left[\begin{array}{cc}R+p{L}_d& -\omega L_q\\ \omega L_d& R+p{L}_d\end{array}\right]\left[\begin{array}{c}{I}_d\\ I_q\end{array}\right]+\left[\begin{array}{c}0\\ \omega K_e\end{array}\right]$$

[Equation 6] $$\left[\begin{array}{c}{V}_d\\ V_q\end{array}\right]=\left[\begin{array}{cc}R& -\omega L_q\\ \omega L_d& R\end{array}\right]\left[\begin{array}{c}{I}_d\\ I_q\end{array}\right]+\left[\begin{array}{c}0\\ \omega K_e\end{array}\right]$$

In case of estimating an estimated d-axis current I _d and an estimated q-axis current I _q using the dq axis command voltages V _d * and V _q * equation (7) is obtained from equation (6).
[Equation 7] $$\left[\begin{array}{c}{V}_d^{*}\\ V_q^{*}\end{array}\right]=\left[\begin{array}{cc}R& -\omega L_q\\ {\omega }_{L_d}& R\end{array}\right]\left[\begin{array}{c}{\widehat{I}}_d\\ {\widehat{I}}_q\end{array}\right]+\left[\begin{array}{c}0\\ \omega K_e\end{array}\right]$$

When the estimated d-axis current and the estimated q-axis current are obtained according to equation (7), equation (8) is obtained.
[Equation 8] $$\left[\begin{array}{c}{\widehat{I}}_d\\ {\widehat{I}}_q\end{array}\right]={\left[\begin{array}{cc}R& -\omega L_q\\ \omega L_d& R\end{array}\right]}^{-\text{I}}\left[\begin{array}{c}{V}_d^{*}\\ V_q^{*}-\omega K_e\end{array}\right]$$

When the inverse matrix in Equation (8) is calculated and the estimated d-axis current is divided by the estimated q-axis current, an estimated current _phase (θ _Idq ) is expressed by Equation (9).
[Equation 9] $${\theta }_{Idq}={\text{tan}}^{-1}\left\{\frac{R{V}_d^{*}+\omega L_q\left(V_q^{*}-\omega K_e\right)}{-\omega L_d{V}_d^{*}+R\left(V_q^{*}-\omega K_e\right)}\right\}$$

In the case of a surface permanent magnet synchronous motor (SPMSM) in which the d-axis inductance L _d and the q-axis inductance L _q substantially coincide with each other, equation (9) as equation (10) can be simplified by using the addition theorem of the arctangent function.
[Equation 10] $${\theta }_{Idq}={\text{tan}}^{-1}\left(\frac{\frac{V_d^{*}}{V_q^{*}-\omega K_e}+\frac{\omega L_q}{R}}{1-\frac{V_d^{*}}{V_q^{*}-\omega K_e}\cdot \frac{\omega L_q}{R}}\right)={\text{tan}}^{-1}\left(\frac{V_d^{*}}{V_q^{*}-\omega K_e}\right)+{\text{tan}}^{-1}\left(\frac{\omega L_q}{R}\right)$$

In the current phase calculation unit 244 in the case of an internal permanent magnet synchronous motor (IPMSM) in which the d-axis inductance Ld and the q-axis inductance Lq are different from each other, equation ( 9 ) used. In the case of an SPMSM in which the d-axis inductance Ld and the q-axis inductance Lq substantially coincide with each other, equation (10) is used. By using equations (9) and (10), an estimated current phase can be derived which contains no contribution of a current offset. In the case of using equation (10), the computation time can be shortened since a computational effort is smaller than that of equation (9).

When a three-phase motor is stopped (ω = 0), equation (10) like equation (11) can be simplified, and the calculation time can be shortened.
[Equation 11] $${\theta }_{Idq}={\text{tan}}^{-1}\left(\frac{V_d*}{V_q*}\right)$$

The phase angle estimation unit 245 derived as described hereinafter a first phase and a second phase θ _mod1 θ _mod2 using a voltage phase θ _Vdq, a current phase θ _Idq and the electrical angle θ, the outputs of the voltage phase calculation unit 243 are off.

5 FIG. 15 is a diagram showing an example of detected current intensity including a current offset. In addition to a detected current value (c) detected based on a DC bus current, a three-phase motor voltage command value (a) and a three-phase motor current value (b) are shown. In this example, the current phase on the q-axis which is perpendicular to the magnetic flux direction of a three-phase _motor , the voltage phase is 30 degrees before the q-axis, the voltage _{phase θVdq} is 120 degrees and the estimated current _phase θ _Idq is 90 degrees. As indicated by the detected current value (c) in 5 2, the waveform of a detected current value detected based on the DC bus current intensity varies according to the maximum phase and the minimum phase of a three-phase voltage.

6 _{FIG. 15} is a graph that calculates the voltage _{phase θVdq} calculated in accordance with Equation (2), an estimated DC _bus current value calculated from the estimated current _phase θ _Idq calculated according to any of Equations (9) to (11), and a detected current value; which is detected based on a DC bus current. Since an estimated current at the minimum phase and an estimated current at the maximum phase contain no offset, a current offset I can be _derived using these estimated currents and a detected value of the DC _bus current.

The following describes a means for deriving an estimated DC bus current from a voltage phase and a current phase. Here, the peak value of an estimated current that does not contain an amount of current offset is included I _p designated. This peak I _p can be calculated from the root mean square of the estimated d-axis current ^ I _d and the estimated q-axis current strength ^ I _q in equation (8). However, if the amount of current offset at a first current detection time substantially coincides with that at a second current detection time (see (c) in FIG. 3), the calculation may be omitted.

[Gleichung 13] $$I_{d2}=I_p\text{cos}\left({\theta }_{\text{mod}2}\right)+I_{ofst}$$

When the amounts of the current offsets at the first current detection time point and the second current detection time point substantially coincide with each other, a minimum phase current becomes I _d1 and a maximum phase current I _d2 expressed by equations (12) and (13), respectively.
[Equation 12] $$I_{\mathrm{<figure-callout\; id="1"\; label="d"\; filenames="DE112017002259T5\_0025.png,DE112017002259T5\_0026.png"\; state="\{\{state\}\}">d</figure-callout>}1}=I_p\text{cos}\left({\theta }_{\text{mod}1}\right)+I_{Ofst}$$

[Equation 13] $$I_{d2}=I_p\text{cos}\left({\theta }_{\text{mod}2}\right)+I_{Ofst}$$

[Gleichung 15] $${\theta }_{\text{mod2}}=MOD\left(\theta +{\theta }_{Vdq}-60°\mathrm{,120}°\right)-60°-{\theta }_{Idq}+{\theta }_{Vdq}$$

The first current _phase angle θ _mod1 and the second current _phase angle θ _mod2 are expressed by the equations (14) and (15), respectively. MOD (A, B) is here a type of so-called mathematical modular operator and indicates a remainder obtained at parts of A through B.
[Equation 14] $${\mathrm{<figure-callout\; id="1"\; label="\theta \; mod"\; filenames="DE112017002259T5\_0025.png,DE112017002259T5\_0026.png"\; state="\{\{state\}\}">\theta \; </figure-callout>}}_{\text{mod}}=MOD\left(\theta +{\theta }_{Vdq}\mathrm{,\; 120}°\right)-60°-{\theta }_{Idq}+{\theta }_{Vdq}$$

[Equation 15] $${\theta }_{\text{mod2}}=MOD\left(\theta +{\theta }_{Vdq}-60°\mathrm{,\; 120}°\right)-60°-{\theta }_{Idq}+{\theta }_{Vdq}$$

Accordingly, equations (12) and (13) show partially sinusoidal waveforms of currents including offset amounts, which in 6 are shown.

With equations (12) and (13), the amount of current _offset I is _expressed by Equation (16).
[Equation 16] $$I_{Ofst}=\frac{I_{\mathrm{<figure-callout\; id="1"\; label="d"\; filenames="DE112017002259T5\_0025.png,DE112017002259T5\_0026.png"\; state="\{\{state\}\}">d</figure-callout>}1}\text{cos}\left({\theta }_{\text{mod}2}\right)-I_{d2}\text{cos}\left({\theta }_{\text{mod 1}}\right)}{\text{cos}\left({\theta }_{\text{mod}2}\right)-\text{cos}\left({\theta }_{\text{mod}1}\right)}$$

The peak value I _p is not included in equation (16). Therefore, the amount of current offset can be made using a DC current I _d1 which is detected at the first current detection time preset, a DC level I _d2 which is detected at the second current _{detection timing} , and the phases θ _mod1 and θ _{mod2 are} derived. Therefore, the computational load is reduced.

In the above description, the amounts of current offsets represented by the minimum phase current I _d1 which is obtained at the first current detection timing and the maximum phase current I _d2 which are obtained at the second current detection timing, are generated equal. However, if the amounts of the current offsets differ from each other, the amounts of the current offsets may be at the first current detection time and at the second current detection duration using equations (12) and (13) and an estimated current peak I _p be calculated individually.

As described above, according to the present embodiment, since the magnitude of a current offset based on motor constant and a DC bus current detected at a predetermined timing can be calculated, erroneous detection due to a supply shortage or a ground fault of a motor winding can be prevented. Thus, since the accuracy of detecting the amount of current offset is improved, the accuracy of detecting a motor current based on a DC bus current is improved. As a result, a three-phase motor can be controlled properly.

By performing the calculation of the amount of current offset at a predetermined timing as described above, an error of a current detection circuit when detecting a DC bus current can be detected. For example, when a detected amount of current offset is greater than a predetermined value (eg, 10% of a rated current), it is determined that an abnormality has occurred in a current detection circuit or the like, thereby preventing a collapse of an inverter and a power supply. Further, if it is provided to determine that an abnormality has occurred in the current-detecting circuit, if the amount of current offset fluctuates more than a predetermined amount when a three-phase motor to be driven is replaced, it can be determined that the motor constant is not appropriate, if determined is that an anomaly has occurred.

In the present embodiment, since motor constants of a three-phase motor are used to derive the magnitude of a current offset, the derivative is very sensitive to a change in motor constant. Therefore, below, a measuring means for a winding resistance R having the highest sensitivity among the motor constants will be described with reference to FIG 7 described.

[Gleichung 18] $$V_d^{*}=R{I}_d^{*}$$

7 FIG. 15 is a diagram showing a relationship between a d-axis voltage and a d-axis current in the present embodiment. In the present embodiment, the winding resistance R is measured based on the relationship between a d-axis command voltage and a d-axis command current while the d-axis command voltage is being changed. Here, the relationships between the d-axis command voltage and the d-axis command current in a case where there is a current offset and a case where no current offset is present are expressed by equations (17) and (18) (here It is assumed that the q-axis current is zero.
[Equation 17] $$V_d*=R\text{'}{I}_d^{*}+R\text{'}{I}_{Ofst}$$

[Equation 18] $$V_d^{*}=R{I}_d^{*}$$

As in equations (17) and (18) and 7 3, the relationship between the d-axis command voltage and the d-axis command current is represented by a straight line, where the slope of the line indicates the winding resistance and the amount of current offset is expressed as the size of a portion of the line. Therefore, as in 7 shown by applying the d-axis voltage to a three-phase motor while changing the operating point of the d-axis voltage, plotting the d-axis command current at that time and deriving the slope from an approximate straight line, the winding resistance without being affected by the amount a current offset can be obtained. By using the value of the winding resistance, even if the magnitude of the winding resistance varies due to a factor such as a temperature, the amount of current offset can be obtained with high accuracy.

In the case where the amount of current offset is calculated using motor constants as in the present embodiment, when replacing a three-phase motor due to deviations of the motor constants, it may be impossible to calculate an accurate amount of amperage offset, and torque pulsation may be due an excessive amount of amperage offset of the three-phase motor occur.

On the other hand, in the present embodiment, the amount of current offset is determined by using detected values (FIG. I _d1 and I _d2 ) one through the current detector 120 detected DC bus current, while motor constants are used for the current phase calculation. As a result, the influence of the deviation of the motor constant can be reduced. For example, if the speed of a motor is low and the motor can be considered to be in a nearly locked state, it can be assumed that the d-axis current command I _d * is almost zero. At this time, according to the present embodiment, the amount of a current offset generated by a DC current due to the q-axis current can be detected even if there is a current detection error due to the amount of current offset, the current value can be controlled so that it is constant, and thus the torque pulsation can be reduced.

In the present embodiment, using a detected value of a DC bus current detected at a common DC bus current detection timing as described above, as shown in FIG 3 shown is the amount of amperage offset I _ofst of the amperage detector 120 through the in 4 shown offset detector 240 be calculated. In the amperage corrector 250 ( 1 ), a value of a DC bus current that does not include the amount of current offset is obtained from the detected current value (FIG. I _d1 and I _d2 ) and the amount of current offset I _ofst and the three-phase motor currents ( I _u . I _v and I _w ) are calculated based on the value of the DC bus current that does not include the amount of current offset. Consequently, a motor can be controlled with high accuracy.

Next, the minimum pulse width TPS at which a DC bus current can be detected will be explained with reference to FIG 8th described.

8th is a pulse waveform diagram illustrating the operation of the pulse shift calculator 230 ( 1 ) in the present embodiment. A section (a) shows a sawtooth wave-like timer count indicating the carrier period for the PWM pulse generation, (b) shows a PWM section with a current voltage command in a common PWM PWM pulse, and (c) shows a DC bus current waveform I _dc , The pulse width between phases including a U-phase PWM pulse width U _pw , a V-phase PWM pulse width V _pw and a W-phase PWM pulse width W _pw is smaller than a time period Tad required for sampling by an A / D converter, and thus the detection using a microcomputer or the like is difficult. When a small motor current flows, a line voltage is applied to a motor and a motor current flows in accordance with the signal differences between PWM pulses of three phases. However, if the minimum pulse width TPS is not secured, the DC bus current I _dc can not be reliably detected and thus a motor current can not be controlled properly.

As shown in (d), therefore, the DC bus current I _dc can be detected by the minimum pulse width TPS is generated by phase shifting (pulse shift) an intermediate phase waveform of a PWM pulse. The pulse widths ( U _pw . V _pw and W _pw ) of the respective phases shown in (d) are the same as the pulse widths of the respective phases of (b). On the falling edge side of a PWM pulse, a U-phase pulse based on a V-phase pulse as a reference is delayed in phase by a pulse shift amount Tt2, the pulse width being widened such that the inter-phase pulse width between the U-phase pulse and the V Phase pulse to the minimum pulse width TPS, and an A / D sampling is performed.

On the rising edge side of the PWM pulse, the intermediate-phase pulse width between the U-phase pulse and the V-phase pulse decreases, and as shown in (b), a pulse whose polarity relative to the inter-phase pulse between the U-phase pulse and the second phase pulse is generated V-phase pulse is inverted without pulse shift. As a result, while generating a sufficient A / D sampling time, the average value of an applied voltage of a motor within a PWM section can be made equivalent to that in the case without a pulse shift shown in (b), and thus a motor can be tuned by adjusting a voltage applied to the motor and a phase are controlled. At this time, a DC bus current waveform I _dc shown in (e) shows that the area of a current pulse becomes smaller on the rising side of a PWM pulse edge (it becomes smaller in the drawing and becomes a negative-sized area) and the area of the PWM pulse edge PWM pulse edge on the falling side of the current pulse increases. The total area of the current pulses within a PWM section shown in (e) is equivalent to the area of the current pulse within a PWM section shown in (c), however, current values detected by A / D detection are equivalent to detected DC bus current values I ₁ and I ₂ , wherein only one of the pulse shift corresponding current intensity Ips became large.

As described above, in a case where a motor current detects a current value near zero, the amount of current offset is detected even at pulse shift, and the detected current values are offset-corrected, whereby motor current can be calculated with high accuracy and a motor can be properly controlled.

As described above, according to the present embodiment, since the amount of current offset is estimated based on a current phase when the amount of current offset is not included, the amount of current offset included in detected values of DC bus current without erroneous detection with high accuracy be detected. Therefore, an offset correction amount can be detected even in the case of a ground fault or power failure of a motor, and thus the accuracy of detecting a motor current can be improved, and a motor can be controlled with high accuracy. The amount of current offset can be detected even if the duty cycle of a PWM is close to 100%.

Second Embodiment

Next, an electric power steering apparatus as an example of an electric system to which a motor control apparatus according to the present invention is applied will be described.

9 FIG. 14 is a configuration diagram of an electric power steering apparatus according to a second embodiment of the present invention. FIG.

As in 9 As shown, an electrical actuator includes a torque transmitting mechanism 902 and the engine control device 500 that the torque-transmitting mechanism 902 Torque delivers. The engine control device 500 includes, for example, an engine 300 that outputs a torque, and an inverter device 100 that the engine 300 is driven, and the first embodiment described above is applied thereto.

The electric power steering apparatus includes an electric actuator, a steering wheel 900 , a steering detector 901 and an operation amount command unit 903 wherein the operating force of the steering wheel steered by a driver 900 undergoes torque assistance by the electric actuator.

A torque command τ * for the electric actuator is a steering assist torque command of the steering wheel 900 and is from the operation amount command unit 903 according to a through the steering detector 901 detected actuation force of the steering wheel 900 generated. By an engine torque outputted from the electric actuator according to the torque command τ *, the operating force (steering force) transmitted from the driver to the steering wheel becomes 900 exercised, reduced.

The engine control device 500 receives the torque command τ * and corrects the amount of current offset using preset motor constants of the motor 300 as described above, and controls a motor current such that the output torque τ _{m of} the motor 300 coincides with the torque command value τ *.

The output torque τ _{m of} the motor 300 that comes from one directly with a rotor of the engine 300 connected output shaft is output via the torque-transmitting mechanism 902 that uses a deceleration mechanism such as a worm, a wheel or a planetary gear, or a hydraulic mechanism to a rack 910 transmitted to the electric power steering apparatus. As a result, the operating force of the driver is reduced by an electric force (assisted) and the steering angle of the wheels (tires) 920 and 921 is being manipulated.

At this time, an assist amount, ie, the torque command τ *, becomes the operation amount command unit 903 based on an amount of operation, the steering angle or as a steering torque from the steering detector 901 is detected, which is embedded in a steering shaft to detect a steering state, and generates a state amount such as a speed of a vehicle or a state of a road.

According to the present embodiment, by applying the engine control apparatus according to the first embodiment described above, in the case of stopping a vehicle or repeating steering wheel operations at a very low vehicle speed for a long time, for example, to drive the vehicle into a garage, the amount of current offset due to a temperature drift caused when the temperature of an electronic control unit (ECU) rises by a large motor current applied to obtain excessive motor power, are corrected with high accuracy. As a result, deterioration of the engine controllability is prevented, and thus an operation force can be adequately assisted. Even if a variation of a pulse shifting current occurs due to a change in the di / dt characteristic of an inverter driving circuit, a high-accuracy current can be detected, and thus an operating force can be easily assisted even when a steering wheel is turned back.

Third Embodiment

Next, a vehicle brake apparatus will be described as an example of an electric system to which a motor control apparatus according to the present invention is applied.

10 FIG. 10 is a system block diagram showing a configuration of a vehicle brake device according to a third embodiment. FIG.

The motor control apparatus according to the above-described first embodiment is applied to an assist control unit 706 and a motor 731 in 10 applied. A microcomputer for control in the support control unit 706 with an inverter device and a control unit thereof programmed to perform a vehicle braking operation. In addition, the engine 731 integrated on a brake assist device 700 attached and is also designed so that it has a housing 712 with the support control unit 706 forms a one-piece structure.

The vehicle brake device includes a brake pedal 701 , the brake assist device 700 , a power amplification mechanism 800 and wheel mechanisms 850a to 850d , The brake assist device 700 includes a support mechanism 720 , a master cylinder 721 and a storage tank 712 , Inside the master cylinder 721 are a primary piston 726 and a secondary piston 727 arranged so that a primary fluid chamber 721a and a secondary fluid chamber 721b be formed. An operation amount of the brake pedal 701 which the driver steps on is via an entrance bar 722 in the support mechanism 720 entered and by moving the primary piston 726 to the primary fluid chamber 721a transfer, whereby hydraulic pressure is generated. When the secondary piston 727 due to the through the primary fluid chamber 721a In addition, hydraulic pressure generated by the generated hydraulic pressure also becomes hydraulic pressure through the secondary fluid chamber 721b generated. The wheel mechanisms 850a to 850d are each with wheel cylinders 851 Mistake. Every wheel cylinder 851 Brakes a vehicle using one of the master cylinder 721 applied hydraulic pressure from.

Further, a brake operation amount generated by a stroke sensor 702 or one on the brake pedal 701 attached tread force sensor (not shown) is detected in the assist control unit 706 entered the support mechanism 720 controls. The support control unit 706 controls the engine 731 such that it is in a rotational position corresponding to the inputted brake operation amount. Then, the torque of the engine via a deceleration device 723 to a rotation-to-translation conversion device 725 transferred, which converts a rotational power into a translational power, the primary piston 726 pushes and the hydraulic pressure of the primary fluid chamber 721a raised and the secondary piston 727 pushes to the hydraulic pressure of the secondary fluid chamber 721b to increase.

The power amplification mechanism 800 feeds the hydraulic pressures of hydraulic fluids contained in the fluid chambers 721a and 721b are pressurized, via mains 750a and 750b and transmits the hydraulic pressures according to a command of a power amplification control unit 830 to the wheel mechanisms 850a to 850d , As a result, a braking force for the vehicle can be obtained.

The support control unit 706 controls the amount of shift of the primary piston 726 to the amount of pressure of the primary piston 726 adapt. Since the shift amount of the primary piston 726 is not detected directly, the angle of rotation of the drive motor 731 based on a signal from a rotational position sensor (not shown) mounted in the engine 731 is provided, and the shift amount of the primary piston 726 is the propulsive amount of a ball screw 725 , which is a rotation-to-translation conversion device, is calculated.

When the drive motor 731 due to an error stops and the axis of the ball screw 725 can not be controlled so that it returns, the axis of the ball screw 725 by the reaction force of a return spring 728 returned to an initial position, whereby the braking operation of the driver is not interrupted. Therefore, instability of vehicle behavior due to braking of a brake can be avoided.

The power amplification mechanism 800 includes two systems of hydraulic pressure adjustment mechanisms 810a and 810b for adjusting hydraulic fluids of two diagonal wheels among the four wheels. Even if a system fails, the power boosting mechanism can 800 Therefore, a vehicle stably stop and the braking forces of the wheel mechanisms 850a and 850b of two diagonal wheels can be customized. As both systems of hydraulic pressure adjustment mechanisms 810a and 810b operate in the same way, below is the hydraulic pressure adjustment mechanism 810a described.

The fluid pressure adjustment mechanism 810a includes a pump 853 for amplifying a main pressure caused by a hydraulic pressure from a main pipe 750a is generated, a pump motor 852 to drive the pump 853 , a shut-off valve 811 for controlling the supply of hydraulic fluid to a wheel cylinder 851 , a shut-off valve 812 for controlling the supply of hydraulic fluid to the pump 853 , Input valves 814a and 814b for controlling the supply of hydraulic fluid from the main conduit 750a or the pump 853 in every wheel cylinder 851 and outlet valves 813a and 813b for controlling the pressure reduction of the wheel cylinders 851 ,

For example, when the hydraulic pressure is controlled for the anti-lock brake control, when signals from the wheel rotation sensors become 853 in the wheel mechanisms 850a and 850b from the power amplification control unit 830 are processed and a wheel lock is detected during a braking operation, each of the input / output valves (electromagnetic type) and the pump 853 operated to set a hydraulic pressure of a hydraulic fluid to a hydraulic pressure value at which respective wheels do not block. The hydraulic pressure adjusting mechanism may also be used to control the hydraulic pressure to stabilize vehicle behavior.

In the vehicle brake device, the assist control unit includes 706 the motor control apparatus described in the first embodiment described above, and the motor control apparatus is used to control the shift amount of the primary piston 726 used to thereby help a driver stably depress a brake pedal. Therefore, in the vehicle brake device, highly accurate and stable operation can be continued or an abnormality can be detected correctly.

When the charge capacity of a battery 200 Also, if the normal power supply is deteriorated, the amount of assistance also deteriorates. Therefore, the design is such that a brake assist operation using an auxiliary power supply 400 continues as power supply. In this case, it is necessary to suppress a large-current output because the auxiliary power supply 400 an emergency measure is.

When the power supply to the auxiliary power supply 400 can switch the support control unit 706 On the other hand, according to the present embodiment, the amount of current from the power supply is limited to about 1/10 of a normal amount. Even in such a case where the amount of a current is small, the motor control apparatus can accurately control a power supply current by detecting a DC bus current with high accuracy and controlling a motor current. As a result, a stable brake assist control can be continued even if an abnormality occurs in a battery.

Further, according to the present embodiment, when the hydraulic pressure of a hydraulic fluid is substantially constant, the amount of current offset within a short period using a current phase angle determined by a voltage command for estimating the amount of current offset (current detection error) in the detected value is included in a DC bus current strength calculated (see equation (11) above).

It is to be noted that the present invention is not limited to the above-described embodiment but includes various applied examples. For example, the embodiments described above are described in detail to explain the present invention in a readily understandable manner, and are not necessarily limited to those having all the described configurations. In addition, the configurations of other embodiments may be added to and removed from any embodiment and replace part of the configuration thereof.

LIST OF REFERENCE NUMBERS

100:

Inverter device

110:

Current controller

111:

Three-phase / dq computer

120:

Current detector

130:

Inverter circuit

150:

Rotary position detector

200:

battery

220:

PWM generator

230:

Pulse shift calculator

240:

displacement detector

241:

Speed computing unit

242:

dq-axis voltage instruction converter

243:

Voltage phase calculation unit

244:

Current phase calculation unit

245:

Phase angle estimation unit

246:

Offset derivation unit

300:

Three-phase motor

320:

Rotary position sensor

400:

Auxiliary power supply

500:

Motor controller

700:

Brake assist device

701:

brake pedal

702:

stroke sensor

706:

Assist Control Unit

712:

storage tank

720:

support mechanism

721a:

Primary fluid chamber

721b:

Secondary fluid chamber

722:

input rod

723:

delay means

725:

Ball screw

726:

primary piston

727:

secondary piston

728:

Return spring

731:

engine

750a, 750b:

main

800:

strengthening mechanism

810a, 810b:

Hydraulic pressure adjustment mechanism

811:

Absperrausgangsventil

812:

Absperreingangsventil

813a, 813b:

outlet valve

814a, 814b:

inlet valve

830:

Gain control unit

850a, 850b, 850c, 850d:

wheel mechanism

851:

wheel cylinder

852:

pump motor

853:

pump

900:

steering wheel

901:

steering detector

902:

Torque-transmitting mechanism

903:

Operation amount command unit

910:

rack

920, 921:

wheel

QUOTES INCLUDE IN THE DESCRIPTION

This list of the documents listed by the applicant has been generated automatically and is included solely for the better information of the reader. The list is not part of the German patent or utility model application. The DPMA assumes no liability for any errors or omissions.

Cited patent literature

• JP 2014128087 A [0004]

Claims (17)

Motor control device comprising: a power converter; a three-phase motor configured to be driven by the power converter; and a control unit configured to detect a motor current flowing through the three-phase motor based on a detected value of DC bus current, generate a command voltage based on the motor current, and control the power converter using the command voltage the control unit calculates a current phase for a case where the DC bus current does not include a current offset, and estimates the amount of the current offset contained in the detected value of the DC bus current based on the current phase. Motor control device after Claim 1 wherein the current phase is calculated from the command voltage, the motor constant of the three-phase motor and a rotational speed of the three-phase motor. Motor control device after Claim 2 wherein the command voltage is a d-axis command voltage and a q-axis command voltage and the motor constants include a winding resistance, a d-axis inductance, a q-axis inductance, and an induction voltage constant of the three-phase motor. Motor control device after Claim 3 wherein the d-axis command voltage and the q-axis command voltage are denoted by V _d * and V _q * respectively, the winding resistance, the d-axis inductance, the q-axis inductance and the induced voltage constant are denoted by R, L _d , L _q and K _e respectively, the rotational speed is denoted by ω and the current _phase (θ _Idq ) is calculated by equation (9). [Formula 1] $${\theta }_{Idq}={\text{tan}}^{-1}\left\{\frac{R{V}_d^{*}+\omega L_q\left(V_q^{*}-\omega K_e\right)}{-\omega L_d{V}_d^{*}+R\left(V_q^{*}-\omega K_e\right)}\right\}$$

Motor control device after Claim 1 wherein the current phase is calculated from the command voltage when the three-phase motor is stopped. Motor control device after Claim 5 wherein the command voltage comprises a d-axis command voltage and a q-axis command voltage. Motor control device after Claim 6 wherein the d-axis command voltage and the q-axis command _voltage are denoted by V _d * and V _q *, respectively, and the current _phase (θ _Idq ) is calculated according to Equation (11). [Formula 2] $${\theta }_{Idq}={\text{tan}}^{-1}\left(\frac{V_d^{*}}{V_q^{*}}\right)$$

Motor control device after Claim 1 wherein the control unit calculates a voltage phase of the command voltage and estimates the amount of the current offset based on the voltage phase, the current phase, the detected value of the DC bus current, and the rotational position of the three-phase motor. Motor control device after Claim 8 wherein the detected value of the DC bus current strength comprises two detected values detected at different timings. Motor control device after Claim 9 , in which the current _phase is denoted by θ _Idq , the voltage _phase is denoted by θ _Vdq , the two detected values are denoted by I _d1 and I _d2 , the rotational position of the three-phase _{motor is denoted} by θ, and the magnitude of the current _offset I is _determined by equations (14). , (15) and (16). [Formula 3] $${\theta }_{\text{mod 1}}=MOD\left(\theta +{\theta }_{Vdq}\mathrm{,\; 120}°\right)-60°-{\theta }_{Idq}+{\theta }_{Vdq}$$

[Formula 4] $${\theta }_{\text{mod2}}=MOD\left(\theta +{\theta }_{Vdq}-60°\mathrm{,\; 120}°\right)-60°-{\theta }_{Idq}+{\theta }_{Vdq}$$

[Formula 5] $$I_{Ofst}=\frac{I_{d1}\text{cos}\left({\theta }_{\text{mod2}}\right)-I_{d2}\text{cos}\left({\theta }_{\text{mod 1}}\right)}{\text{cos}\left({\theta }_{\text{mod2}}\right)-\text{cos}\left({\theta }_{\text{mod 1}}\right)}$$

Motor control device after Claim 3 wherein the control unit derives the winding resistance based on an instruction current and the command voltage. Motor control device after Claim 1 wherein the controller determines that an anomaly has occurred if the magnitude of the current offset is greater than a predetermined value. Motor control device after Claim 1 in which, when the three-phase motor is replaced by another three-phase motor having different motor constants, an anomaly is determined based on the amount of the current offset. Motor control device comprising: a power converter; a three-phase motor configured to be driven by the power converter; and a control unit configured to detect a motor current flowing through the three-phase motor based on a detected value of DC bus current, generate a command voltage based on the motor current, and control the power converter using the command voltage the control unit includes an offset detector configured to detect the amount of current offset included in the detected value of the DC bus current magnitude; the offset detector estimates a current phase of the motor current from the voltage command and the motor constant, and a current is controlled by correcting the amount of current offset based on the current phase. A power steering apparatus comprising: a steering wheel; and an electric actuator configured to assist an operating force of the steering wheel with torque, the electric actuator following the motor control device Claim 1 includes. A vehicle braking apparatus comprising: a piston configured to be moved by an operation of an electric motor to apply a brake hydraulic pressure to a wheel mechanism of a vehicle; and a control device configured to control the electric motor based on an operation amount of a brake pedal to move the piston, the control device following the engine control device Claim 1 includes. Vehicle brake device according to Claim 16 further comprising: a battery which is a normal power supply of the control device; and an auxiliary power supply serving as a power source of the control device when the battery is abnormal, the control by the motor control device Claim 1 is continued even when a power supply is switched to the auxiliary power supply.

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