JP2017134010A - Electrostatic capacitance sensor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide an electrostatic capacitance sensor that can suppress a temperature drift of an output signal, and has high resolution power.SOLUTION: An electrostatic capacitance sensor includes: a first and second charge amplifiers 5A and 5B that amplify a first reception electrode 4A, a second reception electrode 4B, and a reception signal; a first integration circuit 7A that integrates a signal form an oscillator 2A; an inversion amplifier 12A that inverts a phase of an AC signal; a second integration circuit 7B that integrates the AC signal having the phase inverted; a first synthesis unit 15A that synthesizes the first charger amplification 5A and an output of the first integration circuit 7A; a second synthesis unit 15B that synthesizes the second charger amplification 5B and an output of the second integration circuit 7B; a wave detector 9A that outputs s signal of an exclusive negative sum of a first synthesis signal synthesized by the first synthesis unit 15A and the second synthesis signal synthesized by the second synthesis unit 15B; and a phase detection unit 10 that counts a detection signal by the number of the AC signals from the oscillator 13, and outputs the counted detection signal as a detection signal of a detected area where an object O exists.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a capacitance sensor that detects the position of an object. Specifically, it is possible to suppress temperature drift of an output signal without being affected by impedance of a transmission / reception electrode section, and to achieve high resolution. It is related with the electrostatic capacitance sensor provided with.

  A number of proximity sensors that detect changes in capacitance due to the approach of an object have been proposed as detection means using capacitance used for detecting the position of an object, that is, a capacitance sensor.

  Patent Document 1 proposes a proximity sensor 51 as shown in FIG.

  The proximity sensor 51 includes an AC signal generation source 52, a transmission electrode (antenna) 53 connected to the AC signal generation source 52 via a buffer 56A, a first reception electrode (antenna) 54A, and a first reception. A second receiving electrode (antenna) 54B having the same temperature characteristics as the electrode 54A, a first phase shifter provided between the transmitting electrode 53 and the first receiving electrode 54A, the transmitting electrode 53, A second phase shift portion provided between the second reception electrode 54B, a signal phase shift portion 62 provided between the transmission electrode 53 and the second reception electrode 54B, and a first phase shift portion. A first synthesizing unit that synthesizes the phase-shifted signal and the signal received by the first receiving electrode 54A, and the second phase-shifting unit and the signal phase-shifted by the signal phase-shifting unit and the second A second synthesizing unit that synthesizes the signal received by the receiving electrode 54 </ b> B, and LPF (low And a pass filter) 60.

  The first phase shift unit functions as a low-pass filter including a resistor 57A and a capacitor 55A. The second phase shift unit functions as a low-pass filter including a resistor 57B and a capacitor 55B.

  The signal phase unit 62 is constituted by an inverter, for example.

  A comparator 58A is added to the first synthesis unit, a comparator 58B is added to the second synthesis unit, and the phase detection unit 59A is composed of, for example, an EXOR element.

  In FIG. 10, an object O is a detected object having an appropriate dielectric constant such as a human finger that moves within the detected area.

  Next, the operation of the proximity sensor 51 will be described.

The amplitudes of the reception signal Eb1 received by the first reception electrode 54A and the reception signal Eb2 received by the second reception electrode 54B change depending on the presence or absence of the object O.
In the following description, a state in which the object O is far from the proximity sensor 51 (a state in which the object O is not in proximity) is referred to as a first state. The state in which the object O is close to the proximity sensor 51 is referred to as a second state.

  The reception signal Eb1 of the first reception electrode 54A is a rectangular wave having a small amplitude in the first state.

Further, the reception signal Eb2 of the second reception electrode 54B is a rectangular wave in the second state with an amplitude increased as compared with the first state.
The oscillation signal Ea which is a rectangular wave is composed of a resistor 57A and a capacitor 55A.
It is deformed by the first phase shifter functioning as a low-pass filter, and becomes a triangular wave signal LPF (Ea) having a small amplitude.

  The triangular wave signal LPF (Ea) is converted into a rectangular wave signal DLPF (Ea) that is delayed by 90 ° from the oscillation signal Ea by being binarized again around the intermediate potential indicated by the broken line (not shown).

  In this example, the values of the resistor 57A and the capacitor 55A are set so that the phase delay is 90 °.

  In the first state, the reception signal Eb1 and the triangular wave signal LPF (Ea) are combined at the connection point of the first reception electrode 54A, the resistor 57A, and the capacitor 55A, and the combined signal D1 (Eb1 + LPF (Ea) in the first state is combined. )) Is generated.

  In the second state, the reception signal Eb2 and the triangular wave signal LPF (Ea) are combined at the connection point of the first reception electrode 54A, the resistor 57A and the capacitor 55A, and the combined signal D2 (Eb2 + LPF (Ea) in the second state is combined. )) Is generated.

  In FIG. 11, F1 and F2 are re-binarized signals.

  The re-binarized signal F1 is a signal obtained by binarizing the synthesized signal D1 (Eb1 + LPF (Ea)) in the first state again with the comparator 58A using the intermediate potential indicated by the broken line as a reference.

  The re-binarized signal F2 is a signal obtained by binarizing the composite signal D2 (Eb2 + LPF (Ea)) in the second state again with the comparator 58A using the intermediate potential indicated by the broken line as a reference.

  As shown in FIG. 11, the phase of the re-binarized signal F2 slightly advances by P1 with respect to the re-binarized signal F1.

  The phase difference P1 between the re-binarized signal F1 and the re-binarized signal F2 is a detection amount that correlates with the degree of proximity of the object O.

  FIG. 12 shows the waveforms of the respective parts when the inverted oscillation signal Ea ′ is generated by shifting the oscillation signal Ea by 180 degrees and used in place of the oscillation signal Ea. The re-binarized signal F2 ′ is slightly delayed in phase by P2 from the re-binarized signal F1 ′.

The phase difference P2 between the re-binarized signal F1 ′ and the re-binarized signal F2 ′ is a detection amount that correlates with the degree of proximity of the object O, and is opposite to the phase difference P1 described above.
FIG. 13 shows a mechanism for obtaining a DC voltage from the re-binarized signals F1 and F2, the re-binarized inverted signal F1 ′, and the re-binarized inverted signal F2 ′, each of which has the opposite phase shift direction. Has been.
The uppermost stage of FIG. 13 shows the synthesized signal D1 in the first state (Eb1 + LPF (Ea)) and the re-binarized signal F1 obtained by binarizing it, and the next stage shows the inverted synthesis of the first state. A signal D1 ′ (Eb1 + LPF (Ea ′)) and a re-binarized inverted signal F1 ′ obtained by binarizing the signal D1 ′ are shown.

  In the first state, the pulse signal G1 is based on the re-binarized signal F1 output from the comparator 58A and the re-binarized inverted signal F1 ′ output from the comparator 58B, and a phase detector (detector). This is a signal output from 59A.

  The pulse signal G1 is generated such that a pulse having a pulse width P3 is generated in a portion where the re-binarized signal F1 and the re-binarized inverted signal F1 ′ coincide with each other.

  That is, the pulse signal G1 corresponds to a result of performing an EXOR operation on the re-binarized signal F1 and the re-binarized inverted signal F1 'by the phase detection unit 59A.

  In FIG. 13, the next stage of the pulse signal G1 shows the synthesized signal D2 (Eb2 + LPF (Ea)) in the second state and the binarized signal F2 obtained by binarizing the synthesized signal D2. Inverted composite signal D2 ′ (Eb2 + LPF (Ea ′)) in state 2 and re-binarized inverted signal F2 ′ obtained by binarizing this signal are shown.

  Further, in the second state, the pulse signal G2 shown in the next stage is based on the re-binarized signal F2 output from the comparator 58A and the re-binarized inverted signal F2 ′ output from the comparator 58B. This is a signal output from the detector 59A.

  The pulse signal G2 is generated so that a pulse having a pulse width P4 is generated at a portion where the re-binarized signal F2 and the re-binarized inverted signal F2 'coincide with each other.

  That is, the pulse signal G2 corresponds to a result of performing an EXOR operation on the re-binarized signal F2 and the re-binarized inverted signal F2 'by the phase detection unit 59A.

  Since the amplitude of the reception signal Eb2 is larger than the amplitude of the reception signal Eb1, the amount by which each rectangular wave signal shifts in the opposite direction increases, and the portions where the signals do not match each other increase.

Therefore, the pulse width P4 is a pulse having a wider width than the pulse width P3.
The logical operation results of the pulse signal G1 and the pulse signal G2 are smoothed by the LPF 60, converted into a DC voltage proportional to the pulse widths P3, P4, etc., and output from the output terminal 61.

International Publication No. 2013/058332

  The conventional techniques described above have the following problems.

  That is, the transmission electrode 53, the first reception electrode 54A, and the second reception electrode 54B are generally expected to have various variations depending on the nature of the object O and the shape of its mounting portion.

  For this reason, the transmission electrode 53, the first reception electrode 54A, and the second reception electrode 54B have different capacities depending on their shapes and sizes.

  According to the experiments by the inventors of the present application, the capacitance of each electrode must be different from several pf to several hundred pf, and the values of the resistors 57A and 57B are the design of the electrode section. It must be changed each time depending on.

  This means that a detection circuit must be prepared for each different electrode part. This makes it difficult to obtain a general-purpose detection circuit.

  The capacitance value to be detected is a very small value with a resolution of several pF, and when the pulse detected by this circuit is smoothed and converted to a voltage, a drift that reflects the temperature change of the power supply of the circuit system is reflected. In order to solve this problem, it is necessary to use an ideal power supply circuit that does not change in temperature at a voltage level that does not exist in reality, and the dynamic range of analog circuits is limited. Therefore, there is a limit to the means for improving the SN ratio.

  The present invention has been made in view of the above-described conventional circumstances, is not affected by the impedance of the transmission / reception electrode section, can suppress the temperature drift of the output signal, and has high resolution. An object is to provide a high-performance capacitive sensor.

  A capacitance sensor according to the present invention includes a first AC signal generation source that generates an AC signal, a transmission electrode that is connected to the AC signal generation source and radiates a transmission signal to the object, and a capacitance of the object. First reception electrode for receiving a reception signal corresponding to the first reception amplifier, and a first charge amplifier for amplifying a signal received by the first reception electrode without being affected by parasitic capacitance of the transmission electrode and the first reception electrode A second receiving electrode that has the same shape and characteristics as the first receiving electrode, receives a received signal corresponding to the capacitance of the object, and receives a signal received by the second receiving electrode. A second charge amplifier that amplifies the signal without being affected by parasitic capacitances of the transmission electrode and the second reception electrode; a first integration circuit that integrates a signal from the first AC signal generation source; Between the AC signal source and the second integrating circuit A signal phase shifter for inverting the phase of the AC signal, a second integration circuit for integrating the AC signal whose phase has been inverted, and outputs of the first charge amplifier and the first integration circuit. A first synthesis unit to synthesize, a second synthesis unit to synthesize the second charge amplifier and the output of the second integration circuit, and a first synthesized signal synthesized by the first synthesis unit; And a second synthesized signal synthesized by the second synthesizing unit, a detector for outputting an exclusive negative sum of the first and second synthesized signals, and a first AC signal generation source A second AC signal generation source that generates an AC signal having a frequency different from that of the AC signal and a signal from the detector are input, and a value when the signal from the detector is active is generated as a second AC signal. Counted by the number of AC signals from the source, as a detection signal of the detection area where the object exists And most important, comprising a phase detector for force, the.

  According to the first aspect of the present invention, it is possible to eliminate the influence of the impedance of the transmission / reception electrode unit by the configuration of the first and second charge amplifiers, and to suppress the temperature drift of the output signal by the configuration of the phase detection unit. A high-performance capacitive sensor can be realized and provided.

  According to the second aspect of the present invention, it is possible to eliminate the influence of the impedance of the transmission / reception electrode unit by the configuration of the first charge amplifier, and to suppress the temperature drift of the output signal by the configuration of the phase detection unit. A capacitance sensor can be realized and provided.

  According to the invention described in claim 3, the first receiving electrode and the second receiving electrode are constituted by two electric wires forming a pair having the same temperature characteristic in the form of a flexible flat cord. 1 and the first receiving electrode and the second receiving electrode can be easily formed in a shape corresponding to the shape of the object to be detected. Further, the first receiving electrode, It is possible to realize and provide a high-performance capacitive sensor that can contribute to improvement in resolution of the detection signal by equalizing the detection characteristics of the two receiving electrodes.

  According to the invention of claim 4, in the invention of any one of claims 1 to 3, the circuit scale can be reduced by employing moving average means for performing pseudo moving average, It is possible to realize and provide a high-performance capacitive sensor capable of improving the resolution of the detection signal.

  According to the invention of claim 5, in the invention of claim 2, the first receiving electrode is an electric wire in the form of a flexible flat cord, so that the same effect as that of the invention of claim 2 can be obtained. It is possible to realize and provide a high-performance capacitive sensor that is easy to play and can easily form the first receiving electrode in a shape corresponding to the shape of the detection target object.

FIG. 1 is a schematic circuit diagram showing a configuration of a proximity sensor which is an example of a capacitance sensor according to Embodiment 1 of the present invention. FIG. 2 is a waveform diagram for explaining the operation of the proximity sensor according to the first embodiment. FIG. 2 is a waveform diagram illustrating another operation of the proximity sensor according to the first embodiment. FIG. 2 is a waveform diagram illustrating still another operation of the proximity sensor according to the first embodiment. FIG. 5 is a circuit diagram schematically illustrating the configuration of the charge amplifier in the proximity sensor according to the first embodiment. FIG. 6 is a circuit diagram schematically showing the structure of the charge amplifier converted equivalently to the charge amplifier shown in FIG. FIG. 7 is a circuit diagram schematically illustrating a configuration of an integration circuit in the proximity sensor according to the first embodiment. FIG. 8 is a waveform diagram for explaining a method of counting signals that are logical operation results of the gate signal and the transmission signal in the first embodiment. FIG. 9 is a schematic circuit diagram showing the configuration of the proximity sensor according to Embodiment 2 of the present invention. FIG. 10 is a schematic circuit diagram showing the configuration of a conventional proximity sensor. FIG. 11 is a waveform diagram for explaining the operation of a conventional proximity sensor. FIG. 12 is a waveform diagram for explaining another operation of the conventional proximity sensor. FIG. 13 is a waveform diagram for explaining still another operation of the conventional proximity sensor.

  The object of the present invention is to provide a high-performance capacitive sensor with high resolution, which is capable of suppressing temperature drift of the output signal without being affected by the impedance of the transmission / reception electrode section. A first AC signal generation source that generates an AC signal, a transmission electrode that is connected to the AC signal generation source and radiates a transmission signal to the object, and a first signal that receives a reception signal corresponding to the capacitance of the object Receiving electrode, a first charge amplifier that amplifies a signal received by the first receiving electrode without being affected by parasitic capacitance of the transmitting electrode, the first receiving electrode, and the same as the first receiving electrode A second receiving electrode having a shape and characteristics and receiving a received signal corresponding to the capacitance of the object; and a signal received by the second receiving electrode is parasitic on the transmitting electrode and the second receiving electrode. Being affected by capacity A second charge amplifier for amplifying, a first integration circuit for integrating a signal from the first AC signal generation source, and a first integration circuit provided between the first AC signal generation source and the second integration circuit. A signal phase shifter that inverts the phase of the alternating current signal, a second integrating circuit that integrates the alternating current signal that has been inverted in phase, and outputs of the first charge amplifier and the first integrating circuit. A first synthesis unit to synthesize, a second synthesis unit to synthesize the second charge amplifier and the output of the second integration circuit, and a first synthesized signal synthesized by the first synthesis unit; And a second synthesized signal synthesized by the second synthesizing unit, a detector for outputting an exclusive negative sum of the first and second synthesized signals, and a first AC signal generation source A second AC signal generation source for generating an AC signal having a different period from the AC signal of the detector, and from the detector And the value when the signal from the detector is active is counted by the number of AC signals from the second AC signal generation source, and is output as a detection signal of the detection area where the object exists And a phase detector provided with the moving average means.

  Hereinafter, a capacitance sensor according to an embodiment of the present invention will be described with reference to the drawings.

Example 1
FIG. 1 is a schematic circuit diagram illustrating a configuration of a proximity sensor 1 that is a capacitance sensor according to the first embodiment.

  As shown in FIG. 1, the proximity sensor 1 includes an oscillator 2A that is a first AC signal generation source, a buffer (amplifier) 6A, and a transmission electrode (antenna) 3, and an AC generated by the oscillator 2A. The signal is supplied to the transmission electrode 3 with sufficient energy by the buffer 6A, and the oscillation signal Ea is radiated to the detection region.

  The oscillator 2A is a first AC signal generation source and is configured by using, for example, a crystal resonator. In the first embodiment, an AC signal is supplied from a CPU (not shown) that controls the circuit from the outside. Yes.

  The stability of the frequency and intensity of the transmission signal Ea radiated from the transmission electrode 3 affects the stability of the output of the proximity sensor 1. In particular, the oscillation signal Ea should not have a wavelength change longer than the period of the transmission signal of the oscillator 13 which is the second AC signal generation source used in the phase detector 10 described later.

  The proximity sensor 1 includes a first reception electrode (antenna) 4A that receives a reception signal Eb from a detection area, and a first charge amplifier 5A that is an amplifier that amplifies the reception signal Eb. Yes.

  Similarly, the proximity sensor 1 includes a second reception electrode (antenna) 4B that receives the reception signal Eb from the detection region, and a second charge amplifier 5B that is an amplifier that amplifies the reception signal Eb. Have.

  Further, the proximity sensor 1 generates a first integration circuit 7A that integrates the oscillation signal from the oscillator 2A, and a reference signal that inverts and amplifies the oscillation signal from the oscillator 2A and detects by the detector 9A described later. And a second integrating circuit 7B for integrating the output signal of the inverting amplifier 12A.

  In the proximity sensor 1 shown in FIG. 1, the AC signal from the oscillator 2A is integrated by the first integrating circuit 7A to generate a triangular wave. The amplitude of this triangular wave and the amplitude of the input signal from the first receiving electrode 4A The ratio determines the amplification factor. The same applies to the second integration circuit 7B.

  As a result of the inventor's experiment, it has been found that it is optimal to improve the S / N ratio to the same level.

  In the first embodiment, when the triangular wave is compared at a duty of 50%, the phase of the compared rectangular wave signal is the phase of the rectangular wave signal generated by the oscillator 2A that is the first AC signal generation source. Is assumed to be delayed by 90 °.

  The proximity sensor 1 further includes a first synthesizer 15A that synthesizes each output signal of the first charge amplifier 5A and the first integration circuit 7A, a comparator 8A, and the second charge amplifier 5B. And a second synthesizer 15B for synthesizing the respective output signals of the second integration circuit 7B, a comparator 8B, a detector 9A for detecting the output signals of the comparators 8A and 8B, and an output signal of the detector 9A. A phase detector 10 for detecting (counting) the signal in the time axis direction, an oscillator 13 as a second AC signal generating source for supplying a transmission signal that is an AC signal to the phase detector 10, and an output terminal 11 ing.

  The comparators 8A and 8B binarize the output signals of the first and second combiners 15A and 15B, respectively, and supply them to the detector 9A as reference signals.

  The first receiving electrode 4A and the second receiving electrode 4B are set to have the same temperature characteristics.

  Further, the first receiving electrode 4A and the second receiving electrode 4B are configured using, for example, a VFF (vinyl flexible flat) cord that can be easily changed in shape and processed, paired, and have the same length. Thus, it is configured as a linear electrode arranged in parallel or a ring-shaped electrode arranged concentrically. In addition, when it is desired to increase the capacity in order to enhance the signal, it may be configured by using a product obtained by superposing or cutting the alpet plates in a rectangular shape.

  With the configuration of the first receiving electrode 4A and the second receiving electrode 4B, it is easy to make the first receiving electrode 4A and the second receiving electrode 4B have a shape corresponding to the shape of the object O. Furthermore, it is possible to contribute to improving the resolution of the detection signal by equalizing these detection characteristics.

  The inverting amplifier 12A is an example of a signal phase shift unit. The comparator 8A is configured as a part of the first synthesis unit 15A, and the comparator 8B is configured as a part of the second synthesis unit 15B.

  The object O is a detected object having an appropriate dielectric constant such as a human finger that moves within the detected area.

  Next, the operation of the proximity sensor 1 according to the first embodiment will be described with reference to FIGS. Here, FIG. 2 to FIG. 4 are waveform diagrams for explaining the operation of the proximity sensor 1.

  In the proximity sensor 1 of the first embodiment, the AC signal generated by the oscillator 2A is amplified by the buffer 6A, and is radiated as the oscillation signal Ea to the detection region by the transmission electrode 3.

  The oscillation signal Ea radiated from the transmission electrode 3 forms an electric field in the detection region by the electric charge generated by the transmission electrode 3.

  The first receiving electrode 4A and the second receiving electrode 4B generate charges from an electric field including contributions from polarization due to the atmosphere, dielectric, object O, etc. existing in the detection region.

That is, the first reception electrode 4A and the second reception electrode 4B receive the reception signal Eb corresponding to the electric field formed in the detection region.
At this time, if the object O existing in the detection region does not move at all, the electric field formed by the oscillation signal Ea transmitted from the transmission electrode 3 becomes a steady state, and the first reception electrode 4A and the second reception electrode 4B. Received signal Eb has a stable phase and amplitude.

  On the other hand, for example, when an object O having an appropriate dielectric constant such as a human finger moves within the detection region, the amplitude of the reception signal Eb received by the first reception electrode 4A and the second reception electrode 4B changes. To do.

Even if the object O moves at the distance of the object O that the proximity sensor 1 generally assumes in use (for example, the diameter of the ring antenna or the length of the rod antenna), the first receiving electrode 4A And there is almost no change in the phase of the reception signal Eb received by the second reception electrode 4B.
In the proximity sensor 1 according to the first embodiment, the oscillation signal Ea is amplified by the buffer 6A and is a rectangular wave oscillation signal whose amplitude is the power supply voltage in the circuit, and is emitted to the space by the transmission electrode 3.

In this case, the amplitude of the reception signal Eb received by the first reception electrode 4A and the second reception electrode 4B varies depending on the presence or absence of the object O.
In the following description, a state where the object O is far from the proximity sensor 1 (a state where the object O is not close) is referred to as a first state, and a state where the object O is close to the proximity sensor 1 is referred to as a second state.

  The reception signal Eb1 shown in FIG. 2 is a rectangular wave having a small amplitude in the first state. The reception signal Eb2 is a rectangular wave in the second state with an amplitude increased as compared with the first state.

  An oscillation signal which is a rectangular wave from the oscillator 2A is converted into a triangular wave signal LPF (Ea) having a small amplitude by the first integrating circuit 7A.

  This triangular wave signal LPF (Ea) is binarized again around the intermediate potential indicated by the broken line in FIG. 2 to become a rectangular wave signal DLPF (Ea) delayed by 90 ° from the oscillation signal Ea (not shown). )

This phase delay is an example, and is determined by the constant of the first integrating circuit 7A.
In this example, the amplitude of the triangular wave is set to be about the same as the signal amplitude from the first receiving electrode 4A amplified by the charge amplifier 5A, and the phase delay from the received signal is set to about 90 °. .

  In the first state, the reception signal Eb1 amplified by the first combiner 15A and the triangular wave signal LPF (Ea) are combined at the connection point of the first charge amplifier 5A and the first integration circuit 7A, and the first A composite signal D1 (Eb1 + LPF (Ea)) in the state 1 is generated.

  In the second state, the amplified received signal Eb2 and the triangular wave signal LPF (Ea) are synthesized by the synthesizer 15A at the connection point of the first charge amplifier 5A and the first integrating circuit 7A, and the first The combined signal D2 (Eb2 + LPF (Ea)) in the state 2 is generated by the combiner 15A.

  The re-binarized signal F1 shown in FIG. 2 is a signal obtained by binarizing the synthesized signal D1 (Eb1 + LPF (Ea)) in the first state again with the comparator 8A using the intermediate potential indicated by the broken line as a reference.

  The re-binarized signal F2 is a signal obtained by binarizing the composite signal D2 (Eb2 + LPF (Ea)) in the second state again with the comparator 8A using the intermediate potential indicated by the broken line as a reference.

As shown in FIG. 2, the re-binarized signal F2 slightly advances in phase by P1 with respect to the re-binarized signal F1.
The reason why the phase of the re-binarized signal F2 slightly advances is as follows.

  That is, the phase of the LPF (Ea) is different by 90 ° between the reception signal Eb1 in the first state and the reception signal Eb2 in the second state.

  Further, the received signal Eb2 in the second state has a larger amplitude than the received signal Eb1 in the first state. Furthermore, in the second state, the ratio of adding signals having different phases to LPF (Ea) is larger.

  As will be described later, the phase difference between the synthesized signal D1 (Eb1 + LPF (Ea)) in the first state and the synthesized signal D2 (Eb2 + LPF (Ea)) in the second state correlates with the degree of proximity of the object O. It is a detected amount.

  That is, since the difference in phase determines the SN ratio of the proximity sensor 1, it is desirable to take it as large as possible.

  In FIG. 3, an inverted oscillation signal Ea ′ is an output signal of the inverting amplifier 12A and is a 180 ° phase shift signal of the oscillation signal Ea. Hereinafter, the synthesis of the inverted oscillation signals Ea ′ and Eb1 and Eb2 will be described.

  The square wave inverted oscillation signal Ea 'is transformed by the integrating circuit 7B to become a triangular wave signal LPF (Ea') having a small amplitude.

  The triangular wave signal LPF (Ea ′) becomes a rectangular wave signal DLPF (Ea ′) delayed by 90 ° from the inverted oscillation signal Ea ′ by binarization (not shown).

Here, the rectangular wave signal DLPF (Ea ′) can be regarded as a rectangular wave having a phase advanced by 90 ° from Ea.
In the first state, the reception signal Eb1 and the triangular wave signal LPF (Ea ′) are combined at the connection point of the charge amplifier 5B and the integration circuit 7B, and the inverted combined signal D1 ′ (Eb1 + LPF (Ea ′)) in the first state. Is generated.

  In the second state, the reception signal Eb2 and the triangular wave signal LPF (Ea ′) are combined at the connection point of the charge amplifier 5B and the integration circuit 7B, and the inverted combined signal D2 ′ (Eb2 + LPF (Ea ′)) in the second state. Is generated.

  The re-binarized inverted signal F1 ′ is obtained by binarizing the inverted combined signal D1 ′ (Eb1 + LPF (Ea ′)) in the first state again by the combiner 15B and the comparator 8B with reference to the intermediate potential indicated by the broken line. Signal. The re-binarized inverted signal F2 ′ is obtained by binarizing the inverted composite signal D2 ′ (Eb2 + LPF (Ea ′)) in the second state again with the synthesizer 15B and the comparator 8B using the intermediate potential indicated by the broken line as a reference. Signal.

  As shown in FIG. 3, the re-binarized inverted signal F2 'is slightly delayed in phase by P2 with respect to the re-binarized inverted signal F1'.

The reason why the phase is slightly delayed is as follows.
That is, the amplitude of the amplified received signal Eb2 in the second state is larger than that of the amplified received signal Eb1 in the first state, and the phase of the second state is different from that of the LPF (Ea). This is because the ratio of adding signals is large.
The phase difference between the inverted composite signal D1 ′ (Eb1 + LPF (Ea ′)) in the first state and the inverted composite signal D2 ′ (Eb2 + LPF (Ea ′)) in the second state correlates with the degree of proximity of the object O. It is a detection amount. That is, since the difference in phase determines the SN ratio of the proximity sensor 1, it is desirable to take it as large as possible.

  As shown in FIGS. 2 and 3, when the oscillation signal Ea and the amplified received signal Eb are combined, and when the inverted oscillation signal Ea ′ and the amplified received signal Eb are combined, the same amplified reception is performed. Even if the signal Eb is synthesized, the phase shift direction of each synthesized signal is opposite to the change in the amplitude of the amplified received signal Eb.

Next, a mechanism for obtaining a DC voltage from the re-binarized signals F1 and F2 and the re-binarized inverted signals F1 ′ and F2 ′, which are opposite in phase shift direction, will be described with reference to FIG.
4 shows the synthesized signal D1 (Eb1 + LPF (Ea)) in the first state and the rebinarized signal F1 obtained by binarizing the synthesized signal D1 (Eb1 + LPF (Ea)). A signal D1 ′ (Eb1 + LPF (Ea ′)) and a rebinarized inverted signal F1 ′ obtained by binarizing the signal D1 ′ are shown.

  In the first state, the pulse signal G1 is a rebinarized signal F1 output from the first combiner 15A and the comparator 8A, and a rebinarized inverted signal output from the second combiner 15B and the comparator 8B. This is a signal output from the detector 9A based on F1 ′.

  The pulse signal G1 is generated so that a pulse having a pulse width P3 is generated at a portion where the re-binarized signal F1 and the re-binarized inverted signal F1 'coincide with each other.

  That is, the pulse signal G1 corresponds to the result of performing an EXOR operation on the re-binarized signal F1 and the re-binarized inverted signal F1 '.

  In FIG. 4, the next stage of the pulse signal G1 shows the synthesized signal D2 (Eb2 + LPF (Ea)) in the second state and the re-binarized signal F2 obtained by binarizing the synthesized signal D2. Inverted composite signal D2 ′ (Eb2 + LPF (Ea ′)) in the state of 2 and a re-binarized inverted signal F2 ′ obtained by binarizing this signal are shown.

  In the second state, the pulse signal G2 is based on the re-binarized signal F2 output from the first combiner 15A and the re-binarized inverted signal F2 ′ output from the second combiner 15B. This is a signal output from the detector 9A.

  The pulse signal G2 is generated so that a pulse having a pulse width P4 is generated at a portion where the re-binarized signal F2 and the re-binarized inverted signal F2 'coincide with each other.

  That is, the pulse signal G2 corresponds to the result of performing an EXOR operation on the re-binarized signal F2 and the re-binarized inverted signal F2 '.

  Since the amplitude of the reception signal Eb2 is larger than the amplitude of the reception signal Eb1, the amount by which each rectangular wave signal shifts in the opposite direction increases, and the portions where the signals do not match each other increase.

Therefore, the pulse width P4 is a pulse having a wider width than the pulse width P3.
The logical operation results of the pulse signal G1 and the pulse signal G2 shown in FIG. 4 are obtained by the phase detection unit 10 with the second frequency having a very large frequency compared to the transmission signal of the oscillator 2A that is the first AC signal generation source. Is counted as the wave number of the transmission signal 29 of the oscillator 13 which is an AC signal generation source, and the count result is converted into a numerical value or a DC voltage proportional to the pulse width and output to the output terminal 11.

  As a counting means in this case, for example, as shown in FIG. 8, the second alternating current is used only while the value of the gate signal 28 (pulse signal G1 or G2) is 1 (active) per cycle of the oscillator 2A. It can be realized by a logic circuit that counts the number of transmission signals 29 output from the oscillator 13 that is a signal generation source.

That is, a logic circuit that counts the signal 30 that is a logical operation result of the gate signal 28 and the transmission signal 29 may be configured in the phase detection unit 10.
In this means, the number of the transmission signals 29 is counted, so that the result is not influenced by the power supply voltage as compared with the conventional method of viewing the voltage level obtained by smoothing the gate signal 28 with the LPF. is there.

  In general, the power supply voltage fluctuates depending on the temperature or the like, but the proximity sensor 1 according to the first embodiment, which can obtain only a very small output compared to the power supply voltage, is a great feature.

  In this method, the resolution of the counting result can be increased as much as the frequency of the transmission signal 29 is increased, but it is not possible to obtain a resolution higher than the phase noise of the circuit system.

  However, if a certain count value is obtained as the wave number of the gated oscillator 13 with respect to one cycle of the oscillation signal of the oscillator 2A, the moving average circuit over the cycles of the oscillator 2A for the designated number of times. If the moving average is performed, the average value can be made with high accuracy, that is, with high resolution. However, the time resolution is lost an average number of times.

  The moving average is generally performed by storing the data for the average number of times, subtracting the oldest data instead of adding the latest data, and dividing the result by the average number of times. The amount of data to be stored increases as the number increases.

  In the first embodiment, an integration circuit for integrating the number of transmission signals of the oscillator 2A and an averaging circuit are incorporated in the phase detection unit 10, and new values are continuously integrated by the integration circuit over a certain period. After the average value is obtained by the averaging circuit, each time a new value is added, the value obtained by dividing the integrated value by the specified numerical value is subtracted to obtain the amount of phase shift, thereby obtaining the pulse width of the pulse signal G1 or G2. Is output to the output terminal 11 as a high-resolution numerical value or a DC voltage proportional to.

Here, for the sake of simplicity, the conventional moving average method will be described taking four moving averages as an example. If the data is Xi, the moving average value Si is
Si = Xi + Xi + 1 + Xi + 2 + Xi + 3 (1)
Si + 1 = Xi + 1 + Xi + 2 + Xi + 3 + Xi + 4 (2)
Therefore, according to (2)-(1), Si + 1-Si = Xi + 4-Xi, that is,
Si + 1 = Si + Xi + 4−Xi.
In the moving average method in this case, the value of the moving average value Si + 1 can be obtained by adding new data Xi + 4 to the previous Si and subtracting the first data (i.e., the previous four times) Xi.

In the pseudo moving average method employed by the present invention, the value of the data Xi is not stored, but based on the prediction that the value of the data Xi will not change so much as Si / 4. The value is replaced with Si / 4.
Therefore, in the pseudo moving average method described above,
As Si + 1 = Si + Xi + 4−Si / 4, the moving average value Si + 1 is obtained by this equation, and it was confirmed that the result of the experiment by the pseudo moving average method is good.

Furthermore, it has been found from the calculation mechanism that / 2 can be calculated much faster than division when calculated by 1-bit right shift, / 4 by 2-bit right shift, and / 256 by 8-bit right shift (however, The average number of times is 2 to the nth power).
For example, if 1100 (12) is shifted by 1 bit, it becomes 0110 (6), and if further shifted by 1 bit, it becomes 0011 (3).

  Formula 1 shows a mathematical expression of the pseudo moving average method of the first embodiment in the case where 256 moving averages are performed.

  In this case, there is an advantage that it is not necessary to store 256 pieces of data Xi and the configuration of the phase detector 10 can be simplified.

  Equation 2 shows a mathematical expression of the conventional moving average method when performing the moving average 256 times.

  In this case, a storage capacity for storing 256 pieces of data Xi is required, and the circuit configuration of the phase detection unit 10 becomes large.

  Here, the specified value that divides the integrated value is a value that is a power of 2.

  According to such a configuration, it is not necessary to store a large amount of data in the phase detection unit 10, the circuit configuration of the phase detection unit 10 can be reduced, and as a result, a high sensitivity can be achieved. A detection signal can be obtained.

  FIG. 5 schematically shows the configuration of the first or second charge amplifier 5A or 5B in the first embodiment.

  In FIG. 5, the output terminal 20 indicates the conducting wire portion of the transmitting electrode 3, the input terminal 21 indicates the conducting wire portion of the first receiving electrode 4 </ b> A or the second receiving electrode 4 </ b> B, the conducting wire portion of the transmitting electrode 3, A measured capacitance C1 in the space between the first receiving electrode 4A and the conducting wire portion of the second receiving electrode 4B is indicated by a capacitor 17.

  In addition, a capacitor 18 having a parasitic capacitance (capacitance formed by the transmission electrode 3, the first reception electrode 4A, or the second reception electrode 4B) C2 is inserted between the input terminal 21 and the ground in terms of form. Is equivalent.

  The first or second charge amplifier 5A or 5B that amplifies the capacitance C1 to be measured is composed of an operational amplifier 19 using the charge of the input terminal 21 as an inverting input and a capacitor 16 having a feedback capacitor C3. Is shown by the output section 23 of the first or second charge amplifier 5A or 5B.

  In this configuration, the input portion of the first or second charge amplifier 5A or 5B serves as the output terminal 20.

  In the first or second charge amplifier 5A or 5B schematically shown in FIG. 5, the input signal from the input unit (the output terminal 20) is Vin, and the output signal Vout of the output unit 23 is the amplification factor Vout. / Vin is calculated as follows.

  In the first or second charge amplifier 5A or 5B schematically shown in FIG. 5, when the Thevenin theorem is applied between the input signal Vin and the output signal Vout, the circuit shown in FIG. 5 is shown in FIG. It is equivalently converted to a circuit.

  That is, in FIG. 6, not the capacitor 17 representing the measured capacitance C1, but the measured capacitance C1 between the input section (the output terminal 20) of the first or second charge amplifier 5A or 5B and the operational amplifier 19. The capacitor 17 indicating the parasitic capacitance C2 and the capacitor 18 indicating the parasitic capacitance C2 are arranged in parallel, and the capacitor 18 indicating the parasitic capacitance C2 is no longer grounded.

  However, the input signal Vin at the input unit (the output terminal 20) is replaced with a signal multiplied by C1 / (C1 + C2).

  Here, the amplification factor Vout / Vin of the first or second charge amplifier 5A or 5B is easily calculated as follows.

  That is, if the impedance of the input unit (the output terminal 20) is Zi = 1 / jω (C1 + CC2) and the impedance of the feedback unit Zr = 1 / jωC3, the amplification factor of the inverting amplifier 19 is Zr / Zi = (C1 + C2) / Although it is C3, since the input voltage is multiplied by C1 / (C1 + C2), the amplification factor Vout / Vin = (C1 + C2) / (C3 × C1) / (C1 + C2) = C1 / C3.

  When rewritten, the output signal Vout = (C1 / C3) × Vin, which indicates that it does not depend on the value of the parasitic capacitance C2.

  Therefore, by using the first or second charge amplifier 5A or 5B having the above-described configuration, the proximity sensor 1 according to the first embodiment is not affected by the parasitic capacitance C2, that is, the transmission electrode 3, The reception signal Eb1 or Eb2 obtained by amplifying the reception signal Eb from the object O can be obtained without being affected by the change in capacitance of the first and second reception electrodes 4A and 4B.

  FIG. 7 schematically shows a configuration of an integration circuit 7C corresponding to the integration circuit 7A or the integration circuit 7B.

  An input signal of the oscillator 2A, which is the first AC signal generation source, is input to the input unit 24 of the integration circuit 7C. A resistor 26 is connected to the input unit of the operational amplifier 19; A capacitor 27 is connected between the output unit 25 and an integration result is obtained from the output unit 25.

  In this case, since the transmission signal of the oscillator 2A is a rectangular wave signal, the output signal of the integrating circuit 7C is a triangular wave signal.

  The inclination of the inclined portion of the triangular wave signal can be freely set by the values of the resistor 26 and the capacitor 27.

(Example 2)
Next, with reference to FIG. 9, a proximity sensor 1A according to Embodiment 2 of the present invention will be described.

  In addition, in the proximity sensor 1A of Example 2 shown in FIG. 9, the same code | symbol is attached | subjected to the same element as the case of the proximity sensor 1 of Example 1 mentioned above, and the detailed description is abbreviate | omitted.

  In the proximity sensor 1A according to the second embodiment of the present invention, the proximity sensor 1 according to the first embodiment described above compares the phases of the combined signal (Eb + LPF (Ea)) and the inverted combined signal (Eb + LPF (Ea ′)). As a result, the S / N ratio can be improved and a large output signal can be obtained as compared with the case where only one of the changes is observed, but each circuit element on the inverted signal side is omitted in consideration of an increase in circuit scale. The overall configuration is simplified.

  That is, in the proximity sensor 1A, the second receiving electrode 4B, the second charge amplifier 5B, the second integrating circuit 7B, the comparator 8B, the inverting amplifier 12A, and the second combiner 15B are omitted.

According to the second embodiment having such a configuration, it is possible to realize the proximity sensor 1 </ b> A having a simple configuration suitable for use in applications where the S / N ratio of the output signal is not strictly required.
However, the technical scope of the present invention is not limited to the contents of the above-described embodiments, but extends to the invention described in the claims and equivalents thereof.

  The capacitance sensor of the present invention can be widely applied to uses such as detection of articles on a belt conveyor in a factory that manufactures various articles, cargo detection in the distribution field, and detection of proximity objects in automobiles.

DESCRIPTION OF SYMBOLS 1 Proximity sensor 1A Proximity sensor 2A Oscillator 3 Transmission electrode 4A 1st receiving electrode 4B 2nd receiving electrode 5A 1st charge amplifier 5B 2nd charge amplifier 6A Buffer 7A 1st integration circuit 7B 2nd integration circuit 7C Integration circuit 8A Comparator 8B Comparator 9A Detector 10 Phase detector 11 Output terminal 12A Inverting amplifier 13 Oscillator 15A First combiner 15B Second combiner 16 Capacitor 17 Capacitor 18 Capacitor 19 Operational amplifier 19 Inverting amplifier 20 Output terminal 21 Input terminal 23 output unit 24 input unit 25 output unit 26 resistor 27 capacitor 28 gate signal 29 transmission signal 30 signal C1 measured capacitance C2 parasitic capacitance C3 feedback capacitance

Claims (5)

A first AC signal source for generating an AC signal;
A transmission electrode connected to the AC signal source and emitting a transmission signal to an object;
A first receiving electrode that receives a received signal corresponding to the capacitance of the object;
A first charge amplifier that amplifies a signal received by a first reception electrode without being affected by parasitic capacitance of the transmission electrode and the first reception electrode;
A second receiving electrode having the same shape and characteristics as the first receiving electrode and receiving a received signal corresponding to the capacitance of the object;
A second charge amplifier that amplifies a signal received by the second reception electrode without being affected by parasitic capacitance of the transmission electrode and the second reception electrode;
A first integrating circuit for integrating a signal from the first alternating signal generation source;
A signal phase shifter for inverting the phase of the AC signal provided between the first AC signal generation source and the second integrating circuit;
A second integrating circuit for integrating the AC signal having the inverted phase;
A first synthesis unit for synthesizing the first charge amplifier and the output of the first integration circuit;
A second synthesis unit for synthesizing the second charge amplifier and the output of the second integration circuit;
The first synthesized signal synthesized by the first synthesizing unit and the second synthesized signal synthesized by the second synthesizing unit are inputted, and an exclusive negative sum of the first and second synthesized signals is inputted. A detector for outputting a signal;
A second AC signal generation source for generating an AC signal having a different period from the AC signal of the first AC signal generation source;
The signal from the detector is input, the value when the signal from the detector is active is counted by the number of AC signals from the second AC signal generation source, and the detection area where the object exists is detected. A phase detector that outputs as a signal;
A capacitance sensor comprising: A first AC signal source for generating an AC signal;
A transmission electrode connected to the AC signal source and emitting a transmission signal to an object;
A first receiving electrode that receives a received signal corresponding to the capacitance of the object;
A first charge amplifier that amplifies a signal received by a first reception electrode without being affected by parasitic capacitance of the transmission electrode and the first reception electrode;
A first integrating circuit for integrating an AC signal from the first AC signal source;
A first synthesis unit for synthesizing the first charge amplifier and the output of the first integration circuit;
A detector for outputting an exclusive negative sum signal of the first synthesized signal synthesized by the first synthesis unit and the AC signal from the first AC signal generation source;
A second AC signal generation source for generating an AC signal having a different period from the AC signal of the first AC signal generation source;
The signal from the detector is input, the value when the signal from the detector is active is counted by the number of AC signals from the second AC signal generation source, and the detection area where the object exists is detected. A phase detector that outputs as a signal;
A capacitance sensor comprising:   2. The capacitance sensor according to claim 1, wherein the first receiving electrode and the second receiving electrode are two electric wires that form a pair of flexible flat cords and have equivalent temperature characteristics. .   The amount of phase shift between the first combined signal and the second combined signal detected by the phase detection unit is obtained every time a new value is added over a period to obtain an average value and a new value is added. 4. The capacitance sensor according to claim 1, wherein the capacitance value is obtained by pseudo moving average means for subtracting a value obtained by dividing the average value by a numerical value corresponding to the average number of times. 5.   The capacitance sensor according to claim 2, wherein the first receiving electrode is an electric wire in a flexible flat cord form.

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