JP2017022990A - Control device for synchronous motor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To smoothly shift from a startup operation to a stable operation while ensuring startup torque for a synchronous motor.SOLUTION: A control device for synchronous motor comprises: an arithmetic part 20 that calculates an estimated angle velocity ωe' by which the γ component Iγ of a current vector obtained by converting a fixed coordinate system current vector to a γ-δ rotation coordinate system current vector is made zero; an integration part 60 that obtains an estimated rotation angle θe' by integrating estimated angular velocities; a voltage output part 30 that outputs a voltage that accelerates at startup until reaching a specified velocity, estimates and outputs the γ component of a voltage vector on a γ-δ rotation coordinate system from an equation of Vγ=-ωe'×Lq×Iδ on the basis of the estimated velocity angle until a stable operation from the reach of the specified angle velocity, and also converts and outputs the δ component Vδ of the voltage vector on the rotation coordinate system; and a first conversion part 40 that, using the estimated rotation angle obtained by the integration part, converts a voltage vector on the γ-δ rotation coordinate system output from the voltage output part, into a voltage vector.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a control device for a synchronous motor.

  In Patent Document 1, in a control apparatus for a synchronous motor, the synchronous motor is started by other control operation in which a voltage is applied to a stator coil to drive the synchronous motor regardless of the rotor position, the amplitude of the current vector is reduced, and the position of the rotor is determined. It is described that the control is switched to the self-regulated operation in which the estimation is performed and the vector control reflecting the estimated position of the rotor is performed. Thereby, according to Patent Document 1, since the d-axis current 0A that is the d-axis component of the current vector is fixed, even if there is a large load fluctuation, the switching from the other control operation to the self-control operation can be performed smoothly. It is supposed to be possible.

JP 2008-167553 A

  In the technique described in Patent Document 1, since the d-axis current is fixed at 0 A by reducing the amplitude of the current vector in a state where the position of the rotor is unknown, it is considered that the q-axis current related to the torque is also reduced. If the q-axis current becomes small, the starting torque of the synchronous motor may be insufficient, and the starting property of the synchronous motor may be deteriorated.

  The present invention has been made in view of the above, and provides a control apparatus for a synchronous motor that can smoothly make a transition from a start-up operation to a steady operation while ensuring start-up torque of the synchronous motor. Objective.

  In order to solve the above-described problems and achieve the object, a synchronous motor control device according to one aspect of the present invention is a synchronous motor control device that controls a sensorless synchronous motor, and drives the synchronous motor. A drive unit, and a current vector in a fixed coordinate system when the synchronous motor is driven by the drive unit is converted into a current vector (Iγ, Iδ) in a γ-δ rotating coordinate system, and a γ component of the converted current vector A calculation unit that calculates an estimated angular velocity ωe ′ of a rotating magnetic field that makes Iγ zero, an integration unit that integrates the estimated angular velocity ωe ′ calculated by the calculation unit to obtain an estimated rotation angle θe ′, and activation of the synchronous motor At this time, a voltage that accelerates until the specified angular velocity is reached by synchronous operation is output, and after reaching the specified angular velocity, the q-axis inductance value L And the γ component Vγ of the voltage vector on the γ-δ rotating coordinate system based on the calculated δ component Iδ of the converted current vector and the calculated estimated angular velocity ωe ′, Vγ = −ωe ′ · Lq · Iδ A voltage output unit that estimates and outputs with a mathematical expression and outputs a δ component Vδ of a voltage vector with a predetermined value on the γ-δ rotating coordinate system, and an estimated rotation angle θe obtained by the integrating unit A first conversion unit that converts a voltage vector on the γ-δ rotating coordinate system output from the voltage output unit into a voltage vector in the fixed coordinate system using the ', and the drive unit includes: The synchronous motor is driven so that the synchronous motor operates at a voltage corresponding to the voltage vector converted by the converter.

  In the synchronous motor control device according to the present invention as set forth in the invention described above, the voltage output unit outputs a voltage value capable of securing a synchronous angular velocity as the Vδ during the synchronous operation.

  In the synchronous motor control device according to the present invention, in the above invention, the voltage output unit may be configured such that, during the synchronous operation, when the synchronous angular velocity is ω *, Vγ is Vγ = −ω * · Lq · The output is estimated by the formula of Iδ.

  In the synchronous motor control device according to the present invention as set forth in the invention described above, the voltage output unit outputs the predetermined value by increasing the predetermined value.

  The control device for a synchronous motor according to the present invention has an effect that the transition from the start-up operation to the steady operation can be smoothly performed while securing the start-up torque of the synchronous motor.

FIG. 1 is a diagram illustrating a configuration of a synchronous motor control device according to an embodiment. FIG. 2 is a diagram showing current characteristics at the time of startup. FIG. 3 is a diagram illustrating the relationship among the UVW fixed coordinate system, the γ-δ coordinate system, and the dq coordinate system in the embodiment. FIG. 4 is a diagram for explaining the positioning process before the start of activation in the embodiment. FIG. 5 is a diagram showing a voltage vector and a current vector at the time of overvoltage. FIG. 6 is a diagram illustrating a voltage vector and a current vector in an appropriate state. FIG. 7 is a diagram illustrating a configuration of a speed estimation processing unit in the embodiment. FIG. 8 is a diagram illustrating a processing flow of the start-up operation in the embodiment. FIG. 9 is a diagram illustrating voltage control at startup in the embodiment. FIG. 10 is a diagram illustrating a transition characteristic from the startup operation to the steady operation.

  Hereinafter, embodiments of a control apparatus for a synchronous motor according to the present invention will be described in detail with reference to the drawings. In addition, this invention is not limited by this embodiment.

(Embodiment)
A synchronous motor control device 1 according to an embodiment will be described with reference to FIG. The control device 1 receives a command rotational speed from the outside. The control device 1 performs a steady operation in which a sensorless synchronous motor (hereinafter referred to as a synchronous motor) M is controlled in a vector manner so as to operate at a command rotational speed. The synchronous motor M includes, for example, a brushless permanent magnet motor and a load.

  When starting the synchronous motor M, the control device 1 does not start in this steady operation, but first performs the start-up operation of the synchronous motor M in the start-up operation mode provided for start-up, and then the start-up operation mode. Is shifted to the steady operation mode and the steady operation of the synchronous motor M is performed in the steady operation mode.

  In this start-up operation mode, in order to start the synchronous motor M from a stop when no induced voltage is detected, the voltage and frequency that can be optimally started according to the environment (temperature, load fluctuation, etc.) in which the synchronous motor M is used. A method of determining the target value of the product VF and performing the synchronous operation so that the product VF converges to the target value (hereinafter referred to as VF control) can be considered. According to this VF control, when fluctuations occur in the load assumed in advance (for example, when a viscosity increase due to the temperature of the lubricating oil or a differential pressure of the compressor occurs), vibrations etc. deviate from the assumed starting characteristics. It may occur. Further, when a certain amount of load fluctuation is taken into consideration, the starting torque is set higher, so that a voltage higher than the voltage corresponding to the appropriate starting torque is applied. When the voltage is excessive compared to the voltage corresponding to the appropriate starting torque, surplus power is generated in addition to the power used for the rotation of the actual load, and this power is a large current on the d-axis (flux axis) side as an ineffective component. May occur.

  As described above, there is a possibility that tuning is difficult in the synchronous activation by the VF control, but another problem may occur due to the activation voltage set high with a margin. That is, since the starting voltage is determined in accordance with the assumed maximum load, the load is mostly light during actual use. In other words, it starts up in an almost excessive voltage state. The surplus power due to excessive voltage appears as d-axis current as described above for the dq rotation axis current, and in most cases, a d-axis current larger than the q-axis current used as torque flows. . In addition, after the mode transition (switching from the synchronous activation to the normal operation), the dq axis current is controlled to the optimum value by the controller for high efficiency operation by the position detection operation. That is, the q-axis current has a value corresponding to the load for torque control, and the d-axis current has a value of 0 or less.

  However, the reversal of the dq axis current (Iq <Id) due to excessive voltage at the time of start-up has an adverse effect upon mode transition from the start-up operation mode to the steady operation mode. In other words, the current controller and the speed controller during steady operation return an inverted current to an appropriate value, so that excessive correction is performed. This correction may cause hunting of the current and rotation speed, and may cause a step-out stop or a forced stop due to overcurrent limitation.

  As countermeasures against this, slowing down the responsiveness of the controller causes performance degradation such as lack of disturbance response capability in steady state. Or measures such as mode transition when the VF characteristics and load characteristics almost coincide with each other during synchronous operation, or acceleration by fully synchronizing and mitigating controller switching shocks using stability due to moment of inertia. There is. In any case, it is further difficult to adjust the product VF of voltage and frequency.

  As another starting method, a method of superimposing a high-frequency signal that does not contribute to the rotation of the motor and estimating the position of the rotor from the salient pole ratio of the inductance is also conceivable. However, this method may require complicated calculations and may not be used with the synchronous motor M having a salient pole ratio of 1.

  A problem that may occur when shifting from the startup operation mode to the steady operation mode will be specifically described with reference to FIG. FIG. 2 shows an example of current characteristics at the time of startup by synchronous operation.

  FIG. 2 shows an example of characteristics when the synchronous activation is performed at a constant voltage of 16 V and a constant speed of 5 rps. As shown in FIG. 2, although the synchronous motor M continues to rotate, the d-axis current greatly exceeds the q-axis current. This is considered to be due to an overvoltage state with respect to the load and excessive power being generated on the d-axis side. In this state, if the load increases or the rotational speed is increased, an appropriate synchronous operation state should be obtained.

When the synchronous motor M is shifted to a steady operation, the rotation speed is controlled by the torque current Iq in an appropriate operation, and the d-axis current becomes a value of 0 or less. Therefore, if the mode transition is performed in the state as shown in FIG. 2, the current controller outputs an excessive adjustment voltage so as to bring the d-axis current close to 0, and the overshoot of this adjustment amount is also applied to other controllers. Will have an impact. That is, it is considered difficult to smoothly transition from the startup operation to the steady operation.
Therefore, in the present embodiment, the following analysis was performed for the above problem, particularly for the synchronous activation by VF.

  That is, it is considered that the inefficient start-up characteristics due to excessive voltage and the cause of d-q axis current reversal are mismatches between the voltage with respect to the load and the rotational speed. Assuming that the load is indefinite within a certain range, the factors that can be adjusted at startup are the voltage and the rotational speed of the rotating magnetic field. Since the voltage needs to secure a starting current to some extent, if the voltage is kept constant and the number of rotations of the rotating magnetic field is varied according to the load, surplus power due to excessive voltage can be shifted to the starting torque. It should be. That is, if the rotational speed is increased and the surplus power is consumed by increasing the rotational speed at light loads, current reversal due to excessive reactive power can be prevented and smooth mode transition can be performed. That is, instead of predetermining the target value of the product VF of voltage and frequency, but adjusting the number of revolutions so that the voltage and load are appropriately balanced, securing of starting torque due to excessive voltage and seamless switching at the time of mode transition Can be realized at the same time.

  Further, in this embodiment, in order to reduce the amount of calculation, the actual magnetic flux position need not be estimated without using the dq rotation coordinate system that requires the actual magnetic flux position (d-axis position) to be estimated. Use a -δ rotating coordinate system. FIG. 3 shows the relationship among the UVW fixed coordinate system, the γ-δ coordinate system, and the dq coordinate system.

  The d axis is the direction of the magnetic pole N of the rotor in the synchronous motor M, and the axis orthogonal to the rotation direction is the q axis. The control axis corresponding to the dq axis is the γ-δ axis, and the rotation angle from the α axis (U phase) to the γ axis is θe. Since the rotor position is not detected in the start-up operation, the phase difference Δθ between the γ axis and the d axis is also shown. The rotor rotates counterclockwise at an angular velocity ωe.

  The control device 1 does not estimate the position of the rotor, but it is necessary to determine only the initial phase of the rotor when starting up. Although there is a method of superimposing a high frequency on the determination of the initial phase, the control device 1 may take into account the fact that an SPM electric motor (saliency ratio 1) may be used as the synchronous motor M and the ease of processing. As shown in FIG. 4, positioning control by specific phase energization is performed.

  Positioning is performed by a general method, and the relative position of the rotating coordinate system with respect to the fixed coordinate system is not particularly particular. However, in the control device 1, for example, as shown in FIG. Energize the positioning so that the phases match.

  The control device 1 uses a d-axis (= γ-axis) current to start rotation, but most of the current flows as a d-axis current during positioning. Therefore, in order to distinguish the positioning current from the current at the start-up, the control device 1 inserts a zero voltage period T1 (see FIG. 9) for resetting the current between the positioning and the start-up. Since the time constant due to resistance and inductance in the electric circuit of the synchronous motor M is small and the current immediately disappears, the zero voltage period T1 is provided for a very short time to ensure this time constant.

  Next, the configuration of the synchronous motor control device 1 according to the embodiment will be specifically described with reference to FIG. FIG. 1 is a diagram illustrating a configuration of a synchronous motor control device 1. In FIG. 1, the voltage and current on the γ-axis side are Vγ * and Iγ *, and the δ-axis side is similarly Vδ * and Iδ * (* represents a command value).

  The control device 1 includes a drive unit 10, a detection unit 50, a calculation unit 20, a voltage output unit 30, an integration unit 60, and a first conversion unit 40.

  The drive unit 10 drives the synchronous motor M by supplying three-phase AC signals U, V, and W to the synchronous motor M. The internal configuration of the drive unit 10 will be described later.

The detection unit 50 detects (picks up) the amplitude of at least two-phase currents. Specifically, the detection unit 50 includes a current sensor 51 and a current sensor 52. The current sensor 51 detects the amplitude of the U-phase current i _U and supplies it to the computing unit 20. The current sensor 52 detects the amplitude of the W-phase current i _W and supplies it to the computing unit 20. Each of the current sensor 51 and the current sensor 52 may AD-convert the current value and supply it to the arithmetic unit 20 as a signal that can be controlled by a digital computer.

The arithmetic unit 20 converts the current vector (i _U , i _W ) in the fixed coordinate system (UVW coordinate system) when the driving unit 10 drives the synchronous motor M into the current vector (γ-δ coordinate system) in the rotational coordinate system (γ-δ coordinate system). i _γ , i _δ ). The rotating coordinate system (γ-δ coordinate system) has a δ axis and a γ axis that intersect each other. Then, the calculation unit 20 calculates the estimated angular velocity ωe ′ of the rotating magnetic field that makes the γ component Iγ of the current vector in the converted rotating coordinate system zero.
Specifically, the calculation unit 20 includes a second conversion unit (three-phase to two-phase converter (UVW / γ-δ)) 21 and a speed estimation processing unit 22.

The second converter 21 receives the detected value of the amplitude of the U-phase current i _U from the current sensor 51 and receives the detected value of the amplitude of the W-phase current i _W from the current sensor 52. The second conversion unit 21 receives the estimated rotation angle θe ′ of the rotating coordinate system from the integration unit 60. For example, the second conversion unit 21 converts the current vector (i _U , i _W ) in the fixed coordinate system (UVW coordinate system) to the current in the rotational coordinate system (γ-δ coordinate system) by using the following formulas 1 and 2. Convert to vector (i _γ , i _δ ).
i _γ = (√2) {i _U × cos (θe ′ + π / 6) −i _W × sin (θe ′)}
i _δ = − (√2) {i _U × sin (θe ′ + π / 6) + i _W × cos (θe ′)}

  The second conversion unit 21 outputs the γ component of the current vector in the converted rotational coordinate system (γ-δ coordinate system), that is, the γ-axis current Iγ to the speed estimation processing unit 22, and the δ component of the current vector, that is, δ. The shaft current Iδ is output to the voltage output unit 30.

  The speed estimation processing unit 22 receives the γ-axis current Iγ from the second conversion unit 21. The speed estimation processing unit 22 calculates an estimated angular velocity ωe ′, which is an estimated value of the angular velocity ωe (see FIG. 3) of the rotating magnetic field that makes the γ-axis current Iγ zero, according to the received γ-axis current Iγ. The speed estimation processing unit 22 supplies the calculated estimated angular velocity ωe ′ to the integration unit 60 and the voltage output unit 30.

  The voltage output unit 30 estimates the γ component Vγ of the voltage vector on the rotating coordinate system (γ-δ coordinate system) based on the estimated angular velocity ωe ′ calculated by the calculation unit 20. At the same time, the voltage output unit 30 sets the δ component Vδ of the voltage vector to a predetermined value on the rotating coordinate system (γ-δ coordinate system).

  Specifically, the voltage output unit 30 includes a fixed voltage output unit 31 and an estimated voltage output unit 32.

  Fixed voltage output unit 31 sets Vδ to a predetermined value (for example, a fixed value), and outputs the value to first conversion unit 40 as δ-axis voltage command Vδ *.

  For example, when starting the synchronous motor M, the driving is started at the starting voltage from the positioning state shown in FIG. 4 with Vγ * = 0 and Vδ * = predetermined values (for example, fixed values). The starting voltage Vδ * set in advance in the fixed voltage output unit 31 is set in advance to a value that sets a higher overvoltage state in consideration of the load fluctuation. The control device 1 adjusts the angular velocity of the rotating magnetic field so that the voltage Vδ * that generates the torque and the load are appropriately balanced.

  The estimated voltage output unit 32 receives the estimated angular velocity ωe ′ from the speed estimation processing unit 22 of the calculation unit 20 and receives the δ-axis current Iδ from the second conversion unit 21 of the calculation unit 20. The estimated voltage output unit 32 estimates the γ component Vγ of the voltage vector using the estimated angular velocity ωe ′ and the δ-axis current Iδ, and uses the estimated value as the γ-axis voltage command Vγ * to the first conversion unit 40. Output. The principle of estimating the γ component Vγ of the voltage vector by the estimated voltage output unit 32 will be described later.

  The integration unit 60 integrates the estimated angular velocity ωe ′ to obtain the estimated rotation angle θe ′.

  Specifically, the integration unit 60 includes an integrator 61. The integrator 61 integrates the estimated angular velocity ωe ′ of the rotating magnetic field to calculate an estimated rotational angle θe ′ that is an estimated value of the phase angle θe (see FIG. 3) of the rotating coordinate system that rotates with the rotating magnetic field. The integrator 61 outputs the calculated estimated rotation angle θe ′ to the first conversion unit 40 and the calculation unit 20, respectively.

  The first conversion unit 40 uses the estimated rotation angle θe ′ obtained by the integration unit 60 to convert the voltage vector on the rotation coordinate system (γ-δ coordinate system) estimated by the voltage output unit into a fixed coordinate system ( To a voltage vector in the UVW coordinate system).

Specifically, the first converter 40 includes a two-phase to three-phase converter (γ-δ / UVW) 41. The two-phase to three-phase converter 41 converts the γ-axis voltage command Vγ * and the δ-axis voltage command Vδ *, that is, the voltage vector (v _γ , v _δ ) in the rotating coordinate system (γ-δ coordinate system) to the voltage output unit 30. Receive from. The two-phase to three-phase converter 41 receives the estimated rotation angle θe ′ from the integration unit 60. The two-phase to three-phase converter 41 uses, for example, the following Equation 3 to Equation 5 to convert the voltage vector (v _γ , v _δ ) in the rotating coordinate system (γ-δ coordinate system) to a fixed coordinate system (UVW coordinate system). To voltage vectors (v _U , v _v , v _W ).
_{v U = (√ (2/3)} ) {v γ × cos (θe ') - v δ × sin (θe')} ··· Equation 3
_{v v = (√ (1/2)} × v γ + √ (1/6) × v δ) × sin (θe ') + (√ (1/2) × v δ -√ (1/6) × v _γ ) × cos (θe ′) Equation 4
_{v W = (√ (1/6)} × v δ -√ (1/2) × v γ) × sin (θe ') - (√ (1/6) × v γ + √ (1/2) × v _δ ) × cos (θe ′) Equation 5

The two-phase to three-phase converter 41 outputs the voltage vector (v _U , v _v , v _W ) in the fixed coordinate system (UVW coordinate system) in the converted rotating coordinate system (γ-δ coordinate system) to the drive unit 10. To do. Although Equations 3 to 5 are equations for three-phase demodulation, the two-phase to three-phase converter 11 may perform two-phase modulation to increase the voltage utilization rate.
The drive unit 10 drives the synchronous motor M so that the synchronous motor M operates at a voltage corresponding to the voltage vector converted by the first conversion unit 40.

Specifically, the drive unit 10 includes a PWM converter 11 and a driver 13. The PWM converter 11 outputs voltage vectors (v _U , v _v , v _W ) in a fixed coordinate system (UVW coordinate system), that is, U-phase voltage command Vu *, V-phase voltage command Vv *, and W-phase voltage command Vw *. Received from the first converter 40. The PWM converter 11 converts the U-phase voltage command Vu *, the V-phase voltage command Vv *, and the W-phase voltage command Vw * into PWM signals and supplies them to the driver 13. Thereby, the PWM converter 11 drives the synchronous motor M via the driver 13.

  The driver 13 receives the PWM signal from the PWM converter 11. The driver 13 has a plurality of switching elements (not shown), for example, performs a power conversion operation by switching the plurality of switching elements at a predetermined timing according to the PWM signal, and generates the generated three-phase AC signals U, V, By supplying W to the synchronous motor M, the synchronous motor M is driven.

Next, the estimation principle of the γ component Vγ of the voltage vector by the estimated voltage output unit 32 will be described. For example, the voltage / current equation in the steady state of the synchronous motor M is expressed by the following equations 6 and 7.
Vd = R · Id−ωe · Lq · Iq Equation 6
Vq = R · Iq + ωe · Ld · Id + ωe · φ Equation 7
FIG. 5 shows Expressions 6 and 7 in a state where the applied voltage is excessive.

Further, assuming that the magnet torque is more dominant than the reluctance torque at the time of starting, and assuming that the maximum condition Id = 0 of the magnet torque, Expressions 6 and 7 become Expressions 8 and 9 below.
Vd = −ωe · Lq · Iq Equation 8
Vq = R · Iq + ωe · φ Equation 9
A vector diagram of the equations 8 and 9 is shown in FIG.

  If the state shown in FIG. 5 is brought close to the state shown in FIG. 6, it can be said that the state is suitable for synchronous operation.

  In the state shown in FIG. 5, if the current vector I is in the second quadrant (Id ≦ 0), in order to shift the synchronous motor M to the appropriate state shown in FIG. It is necessary to adjust the q-axis current by increasing the shaft voltage. That is, when adjusting the synchronous motor M, it is necessary to simultaneously adjust both the γ-axis voltage Vγ and the δ-axis voltage Vδ, or both the synchronization speed and the δ-axis voltage Vδ.

  On the other hand, when the q-axis voltage is excessive, as shown in FIG. 5, the current vector I is in the first quadrant (Id> 0), and there is room for load increase / decrease and acceleration, so the q-axis voltage Vq is fixed. If only the synchronization speed or the d-axis voltage Vd is adjusted, it is considered that the synchronization motor M can be adjusted to an appropriate operation state.

  As an idea of the starting method, the d-axis voltage Vd is adjusted so that the synchronization speed becomes appropriate with respect to the q-axis voltage Vq. In other words, the synchronization speed is varied according to the q-axis voltage Vq and the load state, and the load variation and the DC voltage variation are flexibly dealt with.

According to FIGS. 5 and 6, if the voltage on the d-axis side is set to −ωe · Lq · Iq and the induced voltage ωe · Ld · Id is set to 0, the state of FIG. 5 is shifted to the optimum state of FIG. You can see that That is, the state of FIG. 6 becomes the state of FIG. 5 if the q-axis voltage Vq is constant and the d-axis voltage Vd is controlled to an appropriate value and the synchronization speed is increased. An appropriate value (for example, an optimum value) of the d-axis voltage Vd is given by the following Expression 10.
Vd = −ωe · Lq · Iq Equation 10

  The induced voltage ωe · Ld · Id on the q-axis side becomes 0 when the d-axis current Id becomes 0 regardless of the value of the angular velocity, so the synchronization speed is adjusted by the controller so that the d-axis current Id becomes 0. Then, it is considered that the optimum state of FIG. 6 is reached.

  That is, in the above concept, the estimated voltage output unit 32 replaces the d-axis voltage Vd with the γ-axis voltage Vγ, replaces the d-axis current Id with the γ-axis current Iγ, and replaces the q-axis voltage Vq with the δ-axis. Control equivalent to that obtained by replacing the voltage Vδ with the q-axis current Iq by the δ-axis current Iδ is performed.

  Next, the configuration of the speed estimation processing unit 22 will be described with reference to FIG. FIG. 7 is a diagram illustrating a configuration of the speed estimation processing unit.

The speed estimation processing unit 22 (see FIG. 1) is based on the concept of setting Id to 0, and can be realized with the configuration of FIG. That is, when the rotor accelerates immediately after startup, a positive current corresponding to the excess power due to the startup torque and overvoltage appears on the d-axis in proportion to the angular velocity. The speed estimation processing unit 22 in FIG. 7 multiplies the γ-axis current Iγ by coefficients K _P and K _I by the coefficient multiplier 224 and the coefficient multiplier 221, respectively, and integrates the output of the coefficient multiplier 221 by the integrator 222. The adder 225 adds the output of 222 and the output of the coefficient unit 224 to calculate and output the estimated angular velocity ωe ′.

  The estimated angular velocity ωe ′ output from the velocity estimation processing unit 22 becomes the estimated rotation angle θe ′ in the integration unit 60 (see FIG. 1), and the coordinate conversion processing performed by the two-phase / three-phase converter 41 (see FIG. 1). The conversion angle is advanced according to the angular velocity. With the updated conversion angle, a current is newly converted into orthogonal rotation coordinates by coordinate conversion processing by the second conversion unit 21 (see FIG. 1), and a γ-axis current Iγ corresponding to the d-axis current Id is calculated. .

  The speed estimation processing unit 22 performs re-correction according to the γ-axis current Iγ corresponding to the d-axis current Id. By this sequential estimation process, the angular velocity converges at an angular velocity at which the γ-axis current Iγ corresponding to the d-axis current Id becomes zero. Actually, the dq axis current is unknown, but when the estimated angular velocity ωe ′ converges, the phase difference Δθ (see FIG. 3) between the d axis and the γ axis also converges to 0, and the dq value Instead of γ-δ values are substituted.

Further, in order to converge the estimated angular velocity ωe ′, Vd (= Vγ) is simultaneously calculated using the estimated angular velocity ωe ′. When Equation 10 is rewritten using the q-axis inductance value Lq, the substitute value, and the estimated angular velocity, Vγ shown in Equation 11 below is obtained.
Vγ = −ωe ′ · Lq · Iδ Expression 11

Estimated angular velocity ωe ′ and estimated rotational angle θe ′ are obtained with δ-axis voltage Vδ as a constant value and γ-axis current Iγ as input to speed estimation processing unit 22, and estimated voltage output unit 32 gives γ-axis voltage Vγ according to Equation 11. If this is the case, it is considered that the synchronous motor M converges to the state of FIG. 6 at a rotational speed balanced by the δ-axis voltage Vδ and the load.

  Next, the function of the limiter 223 in FIG. 7 will be described.

  When starting from the initial state shown in FIG. 4, the rotor magnetic pole and the q-axis current Iq start from a phase difference of 90 °, and the phase angle gradually decreases as the rotor rotates. It becomes. Further, immediately after startup, torque for moving the rotor from the stopped state is required, and the influence of overvoltage overlaps, resulting in overshooting the rotational speed of the rotor. The above Equations 6 to 9 are convergence values in the steady state, and the transient state term is omitted, and thus cannot cope with such a transient state. In other words, the overshooted rotor becomes vibrational, and the speed estimation processing unit 22 corrects in the deceleration direction in order to suppress the vibration, but the estimated angular speed may oscillate depending on the phase between the rotor and the correction direction. In particular, when the estimated angular velocity ωe ′ fluctuates in the deceleration direction when the rotor is accelerated, there is a high possibility that the rotor is pulled back in the deceleration direction suddenly and steps out.

Therefore, the speed estimation processing unit 22 adjusts the coefficients K _P and K _I so as to suppress oscillation, and inserts a limiter 223 before the coefficient unit 224. In other words, the limiter 223 limits the γ component Iγ of the current vector so as not to become a predetermined value or less, and the calculation unit 226 sets the estimated angular velocity ωe of the rotating magnetic field to zero the γ component Iγ of the current vector limited by the limiter 223. 'Is calculated. Thereby, the vibration accompanying pulsation can be relieved and the starting torque of the synchronous motor M can be improved.

  That is, even if the γ-axis current Iγ (d-axis current Id) fluctuates to a negative value due to instantaneous overshoot, the rotation direction is determined, so the rotation speed of the rotor and the rotating magnetic field will not be negative. . Therefore, if the current value is limited by the limiter 223, a state in which the rotation speed falls to a negative value due to the oscillation phenomenon and the rotation direction is reversed can be canceled. For example, if a lower limiter is inserted so that the negative value of the γ-axis current Iγ (d-axis current Id) is zero, the speed is estimated even if the γ-axis current Iγ (d-axis current Id) becomes negative due to overshoot. Since the deviation (input) of the processing unit 22 becomes 0, the instantaneous correction by the proportional control becomes invalid and the current rotational speed can be maintained.

  Since the synchronous operation can be optimized (for example, optimized) by the method as described above, when shifting to the steady operation, the shift can be performed easily and smoothly without special control.

  For example, the rotation speed and the rotation angle inherit the estimated rotation speed ωe ′ and the estimated rotation angle θe ′, and the dq axis voltages are Vγ and Vδ as Vd and Vq. Taking normal vector control as an example, the initial value (integrator) of the speed estimation processing unit 22 is the δ-axis current immediately before the transition. The integrator of the q-axis current controller is initialized with a value obtained by subtracting the d-axis interference voltage from the δ-axis voltage. The interference voltage is calculated from the γ-axis current Iγ and the estimated angular velocity ωe ′. The integrator of the d-axis current controller is initialized with a value obtained by adding the q-axis interference voltage to the γ-axis voltage. The interference voltage is calculated from the δ-axis current Iδ and the estimated angular velocity ωe ′. With the initialization of these controllers, it is transferred to steady operation.

  FIG. 8 shows a summary of the activation process. That is, the positioning energization process (step S1) is performed, and when a predetermined time elapses (Yes in step S2), the 0 current process for resetting the current is performed (step S3), and when the 0 voltage period T1 elapses (step S4). Yes), the above-described processing (synchronous operation processing) including estimating the angular velocity ωe of the rotating magnetic field that makes the γ-axis current Iγ zero is performed (step S5), and the predetermined period T2 (see FIG. 9) has elapsed. Then (Yes in step S6), the operation mode is shifted from the startup operation mode to the steady operation mode assuming that the synchronous motor M is in an appropriate state (step S7).

  As described above, in the embodiment, the calculation unit 20 converts the current vector in the fixed coordinate system when the synchronous motor M is driven by the drive unit 10 into the current vector in the γ-δ rotation coordinate system, and is converted. The estimated angular velocity of the rotating magnetic field that makes the γ component of the current vector in the rotating coordinate system zero is calculated. The integrating unit 60 integrates the estimated angular velocity calculated by the calculating unit 20 to obtain an estimated rotation angle. Based on the q-axis inductance value Lq, the δ component of the current vector converted by the calculation unit 20, and the estimated angular velocity calculated by the calculation unit 20, the voltage output unit 30 γ component of the voltage vector on the γ-δ rotational coordinate system. Is estimated and output by Equation 11 above, and the δ component of the voltage vector is output with a predetermined value on the γ-δ rotating coordinate system. The first conversion unit 40 converts the voltage vector on the γ-δ rotation coordinate system output from the voltage output unit 30 into the voltage vector in the fixed coordinate system using the estimated rotation angle obtained by the integration unit 60. . The drive unit 10 drives the synchronous motor M so that the synchronous motor M operates at a voltage corresponding to the voltage vector converted by the first conversion unit 40. As a result, the synchronous operation can be optimized (for example, optimized) so that the γ-axis current Iγ (d-axis current Id) becomes zero without detecting the position of the rotor, so that the starting torque of the synchronous motor is secured. Thus, the transition from the starting operation to the steady operation can be performed smoothly.

  In the embodiment, the drive unit 10 performs positioning energization for positioning the rotor in the period T1 (see FIG. 9) before starting the start, resets the current, and then starts the start. In the period T2 (see FIG. 9), the synchronous motor M is driven so that the synchronous motor M operates at a voltage corresponding to the voltage vector converted by the first converter 40. Thereby, the synchronous motor M can be started appropriately.

  In the embodiment, the limiter 223 limits the γ component Iγ of the current vector so that it does not become a predetermined value or less, and the calculation unit 226 rotates the γ component Iγ of the current vector limited by the limiter 223 to zero. The estimated angular velocity ωe ′ of the magnetic field is calculated. Thereby, the vibration accompanying pulsation can be relieved and the starting torque of the synchronous motor M can be improved.

  For example, in the case where the position detection operation using the induced voltage is performed after the mode transition, there may be a case where the rotation speed at which the induced voltage can be sufficiently detected cannot be reached with the constant voltage at the start-up. In such a case, the δ-axis voltage Vδ may be increased in a pattern as shown in FIG. That is, the voltage output unit 30 increases the rotation number of the rotating magnetic field and the rotor synchronized with the rotation value by gradually increasing a predetermined value in the period T2 after the start of startup, while performing γ-δ rotation. On the coordinate system, the δ component Vδ of the voltage vector may be changed to a voltage (voltage for performing mode transition) that can obtain a rotational speed that is equal to or greater than a value at which the induced voltage can be detected. At this time, as shown in FIG. 9, a constant voltage section may be provided after the δ-axis voltage Vδ is increased, and after the rotation speed is converged during this period, the operation may be shifted to a steady operation. In addition, this method is not limited to the voltage control pattern shown in FIG. 9, and after increasing the q-axis voltage in a ramp shape immediately after startup, the rotational speed is converged by fixing at an arbitrary constant voltage. Also good. As the δ-axis voltage Vδ (q-axis voltage Vq) increases, the angular velocity also increases, and the increased amount appears as the γ-axis current Iγ (d-axis current Id), so that the estimated angular velocity can be increased as a result.

  As described above, by changing the synchronization speed via the δ-axis voltage Vδ according to the load state, it is possible to flexibly cope with load fluctuations and to cope with a wide range of starting loads.

  Further, in the control device 1, it is possible to switch to the control of the speed estimation processing unit 22 shown in FIG. That is, after the synchronization speed is ensured in the overvoltage state by the synchronous activation by the VF control, the mode may be shifted to a state where the mode can be shifted by the method of the present embodiment. In this case, the VF control voltage is the δ-axis voltage Vδ, and the γ-axis voltage Vγ is given by Equation 11. As the value of the speed ω, the synchronous angular speed of the VF control is used as an electrical angle. When the designated rotational speed is reached, the rotational speed is kept constant and the method of this embodiment is switched, but it is confirmed that the current vector I is in the first quadrant. That is, the γ-axis current Iγ may be a positive value larger than zero. If the γ-axis current Iγ is a negative value, a process such as maintaining the rotational speed and increasing the δ-axis voltage Vδ is necessary. At this time, the rotational speed can be converged even if the γ-axis current Iγ is a negative value, but there is a case where the torque is insufficient with respect to the load. Therefore, it is safer to switch from the overvoltage state. Since the proportional control does not function by the limiter, it converges to the true value by the integral control, so it takes time to converge. Then, after reaching an arbitrary rotational speed, the angular velocity of the electrical angle corresponding to the rotational speed is set as the initial value of the integrator 222 in FIG. 7 and then switched to the method of the present embodiment. After securing a stable time until convergence, the mode shifts to steady operation.

  FIG. 10 shows the switching characteristics by simulation when the mode is shifted from the startup operation mode to the steady operation mode and control is transferred to normal vector control. The starting voltage and load characteristics are the same as those in FIG. 2, and the control device is replaced with the control device 1 of the present embodiment from that in FIG. 10A is a voltage characteristic, FIG. 10B is a current characteristic, FIG. 10C is an estimated rotational speed (angular velocity), FIG. 10D is an axis error Δθ, and FIG. 10E is a rotational angle ( It is a graph of (turned back at an electrical angle of 360 °). The axis error Δθ corresponds to the phase difference Δθ between the d axis and the γ axis shown in FIG. As can be seen from FIG. 10D, the axis error Δθ converges to zero. That is, according to the present invention, since the γ-δ axis coincides with the dq axis without performing the dq axis estimation, the same effect as the dq axis estimation can be obtained. Thereby, all measured values are smoothly shifted to the steady operation (the vertical one-dot chain line represents the timing at which the mode shifts).

  As described above, the synchronous motor control device according to the present invention is useful for controlling the synchronous motor.

DESCRIPTION OF SYMBOLS 1 Control apparatus 10 Drive part 20 Calculation part 30 Voltage output part 40 1st conversion part 50 Detection part 60 Integration part

Claims (4)

A control device for a synchronous motor for controlling a sensorless synchronous motor,
A drive unit for driving the synchronous motor;
A current vector in a fixed coordinate system when driving the synchronous motor by the drive unit is converted into a current vector (Iγ, Iδ) in a γ-δ rotating coordinate system, and the γ component Iγ of the converted current vector is set to zero. A calculation unit for calculating an estimated angular velocity ωe ′ of the rotating magnetic field to be
An integration unit for integrating the estimated angular velocity ωe ′ calculated by the calculation unit to obtain an estimated rotation angle θe ′;
When starting the synchronous motor, it outputs a voltage that accelerates until the specified angular velocity is reached by synchronous operation, and after reaching the specified angular velocity until the steady operation of the synchronous motor, the q-axis inductance value Lq and the converted current are output. Based on the δ component Iδ of the vector and the calculated estimated angular velocity ωe ′, the γ component Vγ of the voltage vector is expressed as Vγ = −ωe ′ · Lq · Iδ on the γ-δ rotating coordinate system.
A voltage output unit that outputs the δ component Vδ of the voltage vector with a predetermined value on the γ-δ rotating coordinate system;
A first conversion for converting a voltage vector on the γ-δ rotation coordinate system output from the voltage output unit into a voltage vector in the fixed coordinate system using the estimated rotation angle θe ′ obtained by the integration unit. And
With
The drive unit drives the synchronous motor so that the synchronous motor operates at a voltage corresponding to the voltage vector converted by the conversion unit.   2. The synchronous motor control device according to claim 1, wherein the voltage output unit outputs a voltage value capable of ensuring a synchronous angular velocity as the Vδ during the synchronous operation. In the synchronous operation, the voltage output unit sets Vγ to Vγ = −ω * · Lq · Iδ when the synchronous angular velocity is ω *.
The synchronous motor control device according to claim 2, wherein the synchronous motor control device outputs the estimated value by the following formula.   4. The synchronous motor control device according to claim 1, wherein the voltage output unit increases the predetermined value and outputs the increased value. 5.

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