JP2016136823A - Power conversion control device, power conversion unit and power system - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a power conversion control device that can reduce the cost while securing the performance and function of the power conversion control device.SOLUTION: A power conversion control device 3 controls a switching circuit 2s comprising a first phase switching part for electrically connecting a DC power source to a first phase of an AC load 1, and a second phase switching part for electrically connecting the DC power source to a second phase of the AC load 1. The power conversion control device 3 has a first current sensor 5 and a second current sensor 6. The first current sensor 5 detects current flowing in a secondary side current path through which the first phase switching part and the first phase of the AC load 1 are connected to each other. The second current sensor 6 detects current flowing in a primary side current path through which the DC power source and the second phase switching part are connected to each other.SELECTED DRAWING: Figure 4

Description

  The present disclosure relates to a power conversion control device, a power conversion unit, and a power system.

  Conversion of DC power output from a DC power source to AC power to be applied to an AC load is performed using a power converter. The power conversion device is controlled by the power conversion control device. In one example, the AC load is a three-phase motor, and the power conversion device is an inverter. In this case, the power conversion control device can be referred to as a motor control device. In another example, the AC load is a power grid, and the power conversion device is a grid interconnection inverter. In this case, the power conversion control device can be referred to as a grid interconnection inverter control device.

  Conventionally, a power conversion control device having at least two current sensors has been used so that currents in different phases of an AC load can be detected. According to such a power conversion control device, the current vector (or three-phase current) applied to the AC load can be specified. The power conversion control device controls the power conversion device using the specified current vector.

  Various studies have been made on power conversion control devices having current sensors (see Patent Documents 1 to 3).

JP 2004-159391 A JP 2012-50215 A JP 2009-33876 A

  In the prior art, it is difficult to reduce the cost of the power conversion control device while ensuring the performance and function of the power conversion control device. The present disclosure aims to provide a novel technique that can alleviate this difficulty.

That is, this disclosure
The DC power source is converted into the first phase of the AC load so that the DC power output from the DC power source is converted into AC power to be applied to an AC load having different first and second phases. A power conversion control device for controlling a switching circuit having a first phase switching unit electrically connected and a second phase switching unit electrically connecting the DC power source to the second phase of the AC load. And
A first current sensor that detects a current flowing through the secondary current path in a secondary current path that connects the first phase switching unit and the first phase of the AC load;
A second current sensor for detecting a current flowing through the primary-side current path in a primary-side current path connecting the DC power source and the second-phase switching unit;
A power conversion control device is provided.

  The combination of the first current sensor and the second current sensor of the present disclosure is suitable for reducing the cost of the power conversion control device while ensuring the performance and function of the power conversion control device.

Block diagram of three-phase motor, inverter and motor control device The figure for demonstrating the internal structure of an inverter, and arrangement | positioning of the current sensor in 1st Embodiment. Diagram for explaining the dq coordinate system Diagram for explaining αβ coordinate system The block diagram for demonstrating the structure of the motor control apparatus which concerns on 1st Embodiment. Diagram for explaining the situation where the time required to extract the switching element current cannot be secured The figure for demonstrating the case where a 1st driving | operation is performed and the case where a 2nd driving | running is performed The block diagram for demonstrating the internal structure of the magnetic flux and torque estimation part of 1st Embodiment Block diagram for explaining the internal configuration of the magnetic flux command calculation unit Block diagram for explaining the internal configuration of the overcurrent detection unit Diagram for explaining analog signal output from overcurrent detection circuit The graph which shows the result of the simulation performed in order to show the difference of the gain characteristic obtained by the magnetic flux estimation part of 1st Embodiment, and the gain characteristic obtained by the conventional magnetic flux estimation part The graph which shows the result of the simulation performed in order to show the difference of the phase characteristic obtained by the magnetic flux estimation part of 1st Embodiment, and the phase characteristic obtained by the conventional magnetic flux estimation part The graph which shows the result of the simulation done in order to show the estimation accuracy of the magnetic flux by the conventional magnetic flux estimation part The graph which shows the result of the simulation performed in order to show the estimation precision of the magnetic flux by the magnetic flux estimation part of 1st Embodiment The block diagram for demonstrating the internal structure of the magnetic flux and torque estimation part of the modification 1-1. The block diagram for demonstrating the internal structure of the magnetic flux and torque estimation part of the modification 1-2. The block diagram for demonstrating the internal structure of the magnetic flux and torque estimation part of the modification 1-3 The graph which shows the result of the simulation performed in order to show the difference of the gain characteristic obtained by the magnetic flux estimation part of modification 1-3, and the gain characteristic obtained by the conventional magnetic flux estimation part The graph which shows the result of the simulation performed in order to show the difference of the phase characteristic obtained by the magnetic flux estimation part of modification 1-3, and the phase characteristic obtained by the conventional magnetic flux estimation part Graph to explain suitable correction factors The graph which shows the result of the simulation performed in order to show the estimation precision of the magnetic flux by the magnetic flux estimation part of the modification 1-3 The figure for demonstrating arrangement | positioning of the current sensor in the internal structure of an inverter, and the modification 1-4. The figure for demonstrating arrangement | positioning of the current sensor in the internal structure of an inverter, and the modification 1-5 The block diagram for demonstrating the structure of the motor control apparatus which concerns on 2nd Embodiment. The block diagram for demonstrating the structure of the grid connection inverter control apparatus which concerns on 3rd Embodiment.

The first aspect of the present disclosure is:
The DC power source is converted into the first phase of the AC load so that the DC power output from the DC power source is converted into AC power to be applied to an AC load having different first and second phases. A power conversion control device for controlling a switching circuit having a first phase switching unit electrically connected and a second phase switching unit electrically connecting the DC power source to the second phase of the AC load. And
A first current sensor that detects a current flowing through the secondary current path in a secondary current path that connects the first phase switching unit and the first phase of the AC load;
A second current sensor for detecting a current flowing through the primary-side current path in a primary-side current path connecting the DC power source and the second-phase switching unit;
A power conversion control device is provided.

  From the viewpoint of cost reduction, it is advantageous to keep the number of current sensors provided in the secondary current path to a minimum and to use the current sensors in the primary current path together. Since the secondary current path is floating, it is highly necessary to use an insulated current sensor as a current sensor provided in the secondary current path, and the insulated current sensor is likely to be expensive. is there. When a current sensor is provided in the primary current path, a path connected to the ground potential can be used as the current path for providing the current sensor, and therefore an inexpensive non-insulated current sensor can be used. Because. Further, from the viewpoint of ensuring the estimation accuracy of the current vector of the AC load and ensuring the control performance of the power conversion control device, the secondary side current path and the primary side current path are detected so that currents of different phases are detected. There are advantages to providing a current sensor for each. When the frequency of the AC power is high, the period in which the current flows continuously through the primary side current path tends to be short, and the detection value by the current sensor in the primary side current path is difficult to use for control. This is because the current vector can be estimated with sufficient accuracy only from the detected value of the current sensor in the current path. In addition, when the frequency of the AC power is low, it is difficult to accurately estimate the current vector only from the detection value of the current sensor in the secondary current path, but the detection value of the current sensor in the primary current path is It is because estimation accuracy can be ensured by using together. As understood from the above description, the combination of the first current sensor and the second current sensor in the first mode is high even if the frequency of the AC power is low while the power conversion control device is configured at low cost. Also, it is easy to ensure the control performance of the power conversion control device. In the present specification, the primary side is closer to the DC power supply side than each switching unit, and the secondary side is closer to the AC load side than each switching unit.

The second aspect of the present disclosure includes, in addition to the first aspect,
Provided is a power conversion control device including only the first current sensor as a current sensor for detecting a current flowing through a secondary current path existing on the AC load side with respect to each switching unit of the switching circuit.

  According to the second aspect, it is easy to reduce the cost of the power conversion control device.

In the third aspect of the present disclosure, in addition to the first aspect or the second aspect,
The first current sensor is a current transformer;
The second current sensor is a shunt resistor, and provides a power conversion control device.

  The shunt resistor is suitable as the second current sensor because it can be configured at low cost as a current sensor for the primary-side current path. Since the current transformer is highly reliable as an insulating sensor, it is suitable as the first current sensor.

The fourth aspect of the present disclosure includes, in addition to the first to third aspects,
Overcurrent detection that is provided outside the inverter including the switching circuit and determines whether or not an overcurrent has occurred by receiving an analog signal (in the hardware part) indicating the detection value of the second current sensor. A power conversion control device is provided.

  The overcurrent detection unit according to the fourth aspect reliably shows the detection value of the second current sensor even when the frequency of the AC power is high and the period of current flowing through the current path on the primary side is short. An analog signal can be received. Therefore, the configuration of the fourth aspect is suitable for detection of overcurrent generated in the current path connecting the DC power supply and the second-phase switching unit.

The fifth aspect of the present disclosure includes, in addition to the fourth aspect,
The overcurrent detector is
Constructed by hardware, the analog signal indicating the detection value of the second current sensor is input, and the analog signal indicating the detection value of the second current sensor larger than the current threshold is input over a certain period. An overcurrent detection circuit that continuously outputs an analog signal indicating that an overcurrent has been detected;
An overcurrent determination unit that determines whether or not an overcurrent has occurred using the analog signal output from the overcurrent detection circuit is provided.

  The overcurrent detection circuit of the fifth aspect continuously outputs an analog signal indicating that an overcurrent has been detected over a certain period. Since such an analog signal is easier to process than an analog signal output instantaneously, the overcurrent determination unit can easily determine whether or not an overcurrent has occurred.

The sixth aspect of the present disclosure includes, in addition to the first to fifth aspects,
The power conversion control device includes:
When the amplitude of the fundamental component of the AC voltage to be applied to the second phase of the AC load is greater than or equal to a threshold amplitude, the detection value of the first current sensor is used instead of the detection value of the second current sensor. Performing a first operation for specifying a current vector applied to the AC load,
When the amplitude of the fundamental component of the AC voltage to be applied to the second phase of the AC load is less than the threshold amplitude, both the detection value of the first sensor and the detection value of the second sensor are used. Provided is a power conversion control device for performing a second operation for specifying a current vector applied to the AC load.

The seventh aspect of the present disclosure includes, in addition to the first to fifth aspects,
The power conversion control device includes:
By controlling the switching circuit in a PWM manner by comparing a carrier signal and a command signal representing a fundamental wave component of an AC voltage to be applied to the AC load,
For the second phase of the AC load, when a modulation rate that is a ratio of the amplitude of the command signal to the amplitude of the carrier signal is equal to or greater than a threshold modulation rate, the detection value of the second current sensor is not used. Performing a first operation for specifying a current vector applied to the AC load using a detection value of one current sensor;
For the second phase of the AC load, when the modulation rate is less than the threshold modulation rate, both the detection value of the first sensor and the detection value of the second sensor are applied to the AC load. Second operation to identify the current vector
The threshold modulation rate provides a power conversion control device in a range of 0.5 to 0.9.

The eighth aspect of the present disclosure includes, in addition to the first to fifth aspects,
The power conversion control device includes:
When the frequency of the fundamental component of the AC power to be applied to the second phase of the AC load is equal to or higher than a threshold frequency, the detection value of the first current sensor is used instead of the detection value of the second current sensor. Performing a first operation for specifying a current vector applied to the AC load,
When the frequency of the fundamental component of the AC power to be applied to the second phase of the AC load is less than the threshold frequency, both the detection value of the first sensor and the detection value of the second sensor are used. Provided is a power conversion control device for performing a second operation for specifying a current vector applied to the AC load.

The ninth aspect of the present disclosure includes, in addition to the eighth aspect,
The power conversion control device is provided in which the threshold frequency is in a range of 500 to 2500 Hz.

  According to the 6th-9th aspect, a 1st driving | operation and a 2nd driving | operation can be switched appropriately.

The tenth aspect of the present disclosure includes, in addition to the first to ninth aspects,
The power conversion control device controls the switching circuit by a PWM method by comparing a carrier signal and a command signal representing a fundamental wave component of an AC voltage to be applied to the AC load,
Provided is a power conversion control device in which a frequency of the carrier signal is 20 to 50 kHz.

  If the lower limit of the frequency of the carrier signal is set as described above, current ripple can be suppressed. By setting the upper limit of the frequency of the carrier signal as described above, it is possible to avoid a situation in which the current flowing on the primary side is continued for a short period and the detection value by the second current sensor cannot be used for control. easy.

The eleventh aspect of the present disclosure includes, in addition to the first to tenth aspects,
The current vector applied to the AC load is identified by cooperating with the first current sensor without cooperating with the second current sensor, or cooperating with the first sensor and the second sensor. A two-phase current estimator,
A magnetic flux estimator for estimating a rotating magnetic flux vector applied to the AC load using the estimated current vector and a command voltage vector representing an AC voltage to be applied to the AC load; A conversion control device is provided.

  According to the two-phase current estimation unit and the magnetic flux estimation unit of the eleventh aspect, it is possible to control the switching circuit using the estimated values of the current vector and the rotating magnetic flux vector applied to the AC load.

The twelfth aspect of the present disclosure includes, in addition to the eleventh aspect,
The magnetic flux estimator is
An α-axis induced voltage estimator that estimates the α-axis component of the induced voltage generated in the AC load using the estimated α-axis component of the current vector and the α-axis component of the command voltage vector;
A β-axis induced voltage estimator that estimates the β-axis component of the induced voltage generated in the AC load using the estimated β-axis component of the current vector and the β-axis component of the command voltage vector;
An α-axis magnetic flux estimator that temporarily estimates the α-axis component of the rotating magnetic flux vector using the estimated α-axis component of the induced voltage;
A β-axis magnetic flux estimator that temporarily estimates the β-axis component of the rotating magnetic flux vector using the estimated β-axis component of the induced voltage;
An α-axis component correction unit that corrects the estimated α-axis component of the rotating magnetic flux vector using the estimated β-axis component of the induced voltage or the temporarily estimated β-axis component of the rotating magnetic flux vector;
A β-axis component correction unit that corrects the estimated β-axis component of the rotating magnetic flux vector using the estimated α-axis component of the induced voltage or the temporarily estimated α-axis component of the rotating magnetic flux vector; A power conversion control device is provided.

The thirteenth aspect of the present disclosure, in addition to the eleventh aspect,
The magnetic flux estimator is
An α-axis induced voltage estimator that temporarily estimates the α-axis component of the induced voltage generated in the AC load using the estimated α-axis component of the current vector and the α-axis component of the command voltage vector;
A β-axis induced voltage estimator that temporarily estimates the β-axis component of the induced voltage generated in the AC load using the estimated β-axis component of the current vector and the β-axis component of the command voltage vector;
An α-axis component correcting unit that corrects the estimated β-axis component of the induced voltage using the estimated β-axis component of the induced voltage or the estimated β-axis component of the rotating magnetic flux vector;
A β-axis component correction unit that corrects the estimated β-axis component of the induced voltage using the estimated α-axis component of the induced voltage or the estimated α-axis component of the rotating magnetic flux vector;
An α-axis magnetic flux estimation unit that estimates the α-axis component of the rotating magnetic flux vector using the corrected α-axis component of the induced voltage;
And a β-axis magnetic flux estimator that estimates the β-axis component of the rotating magnetic flux vector using the corrected β-axis component of the induced voltage.

  In the conventional magnetic flux estimation unit, it is difficult to ensure the magnetic flux estimation accuracy (typically integration accuracy) when there is an error in the initial value of the magnetic flux estimation value, an offset error, or the like. On the other hand, the correction unit of the twelfth aspect or the thirteenth aspect can alleviate a decrease in estimation accuracy based on these errors. Therefore, according to the magnetic flux estimation unit of the twelfth aspect or the thirteenth aspect, the rotational magnetic flux vector can be estimated with high accuracy.

The fourteenth aspect of the present disclosure includes, in addition to the eleventh to thirteenth aspects,
The power conversion control device operates as a motor control device that controls a three-phase motor as the AC load,
The rotation magnetic flux vector provides a power conversion control device which is a flux linkage vector or a rotor magnetic flux vector of the three-phase motor.

The fifteenth aspect of the present disclosure includes, in addition to the fourteenth aspect,
The rotating magnetic flux vector is the flux linkage vector,
The power conversion control device provides a power conversion control device for controlling the three-phase motor so that the flux linkage vector follows the command magnetic flux vector.

The sixteenth aspect of the present disclosure includes, in addition to the fifteenth aspect,
The power conversion control device includes:
A torque estimating unit for estimating a motor torque applied to the three-phase motor using the estimated current vector and the estimated flux linkage vector;
Provided is a power conversion control device for specifying the command magnetic flux vector so that the motor torque follows the command torque.

The seventeenth aspect of the present disclosure includes, in addition to the fourteenth aspect,
The rotating magnetic flux vector is the rotor magnetic flux vector;
The power conversion control device includes:
A phase estimation unit for estimating the phase of the estimated rotor magnetic flux vector;
The coordinate system of the command current vector and the estimated current vector is unified using the estimated phase, and the current vector is converted into the command using the estimated current vector and the command current vector after the unification. Provided is a power conversion control device for controlling the three-phase motor so as to follow a current vector.

  The power conversion control device according to the fourteenth to seventeenth aspects preferably operates as a motor control device.

According to an eighteenth aspect of the present disclosure,
A power conversion unit having the power conversion control device and the switching circuit according to any one of the first to seventeenth aspects is provided.

  According to the eighteenth aspect, the same effect as that of the first aspect can be obtained.

The nineteenth aspect of the present disclosure includes, in addition to the eighteenth aspect,
Provided is a power conversion unit in which a semiconductor element containing silicon carbide or gallium nitride is used for the switching circuit.

  A switching circuit using a semiconductor element containing silicon carbide or gallium nitride is suitable for high-speed switching.

According to a twentieth aspect of the present disclosure,
An electric power system comprising the DC power source, the power conversion unit, and the AC load according to the eighteenth or nineteenth aspect is provided.

  According to the twentieth aspect, the same effect as that of the first aspect can be obtained.

The twenty-first aspect of the present disclosure includes
The DC power source is converted into the first phase of the AC load so that the DC power output from the DC power source is converted into AC power to be applied to an AC load having different first and second phases. A power conversion control device for controlling a switching circuit having a first phase switching unit electrically connected and a second phase switching unit electrically connecting the DC power source to the second phase of the AC load. And
Magnetic flux estimation for estimating a rotating magnetic flux vector applied to the AC load using an estimated current vector applied to the AC load and a command voltage vector representing an AC voltage to be applied to the AC load A power conversion control device is provided.

  In the twenty-first aspect, the current vector applied to the AC load may be specified using the two-phase current estimation unit of the eleventh aspect, or the current vector may be specified by another method. May be. A secondary-side current path that connects the first-phase switching unit and the first phase of the AC load; a secondary-side current path that connects the second-phase switching unit and the second phase of the AC load; A two-phase current estimation unit may be used in which each current sensor is provided, and a current vector is specified from detection values of these current sensors. Of the current path on the primary side connecting the DC power supply and the switching unit of each phase, a portion after branching from the DC power supply to the switching unit of the first phase of the switching circuit, and the second of the switching circuit from the DC power supply A current sensor may be provided in each of the parts after branching to the phase switching unit, and a two-phase current estimation unit that specifies a current vector from the detection values of these current sensors may be used. As in Modification 1-5 to be described later, a position for detecting the bus current flowing through the portion before branching from the DC power supply to the switching unit of each phase of the switching circuit according to the one shunt current detection method (single shunt current detection method) It is also possible to adopt a mode in which a current sensor is provided in the base and a current vector is specified based on a bus current and a switching pattern generated by each switching unit of the switching circuit.

The twenty-second aspect of the present disclosure includes, in addition to the twenty-first aspect,
The magnetic flux estimator is
An α-axis induced voltage estimator that estimates the α-axis component of the induced voltage generated in the AC load using the estimated α-axis component of the current vector and the α-axis component of the command voltage vector;
A β-axis induced voltage estimator that estimates the β-axis component of the induced voltage generated in the AC load using the estimated β-axis component of the current vector and the β-axis component of the command voltage vector;
An α-axis magnetic flux estimator that temporarily estimates the α-axis component of the rotating magnetic flux vector using the estimated α-axis component of the induced voltage;
A β-axis magnetic flux estimator that temporarily estimates the β-axis component of the rotating magnetic flux vector using the estimated β-axis component of the induced voltage;
An α-axis component correction unit that corrects the estimated α-axis component of the rotating magnetic flux vector using the estimated β-axis component of the induced voltage or the temporarily estimated β-axis component of the rotating magnetic flux vector;
A β-axis component correction unit that corrects the estimated β-axis component of the rotating magnetic flux vector using the estimated α-axis component of the induced voltage or the temporarily estimated α-axis component of the rotating magnetic flux vector; A power conversion control device is provided.

The twenty-third aspect of the present disclosure includes, in addition to the twenty-first aspect,
The magnetic flux estimator is
An α-axis induced voltage estimator that temporarily estimates the α-axis component of the induced voltage generated in the AC load using the estimated α-axis component of the current vector and the α-axis component of the command voltage vector;
A β-axis induced voltage estimator that temporarily estimates the β-axis component of the induced voltage generated in the AC load using the estimated β-axis component of the current vector and the β-axis component of the command voltage vector;
An α-axis component correcting unit that corrects the estimated β-axis component of the induced voltage using the estimated β-axis component of the induced voltage or the estimated β-axis component of the rotating magnetic flux vector;
A β-axis component correction unit that corrects the estimated β-axis component of the induced voltage using the estimated α-axis component of the induced voltage or the estimated α-axis component of the rotating magnetic flux vector;
An α-axis magnetic flux estimation unit that estimates the α-axis component of the rotating magnetic flux vector using the corrected α-axis component of the induced voltage;
A β-axis magnetic flux estimator that estimates the β-axis component of the rotating magnetic flux vector using the corrected β-axis component of the induced voltage;
A power conversion control device is provided.

The twenty-fourth aspect of the present disclosure includes, in addition to the twenty-first to twenty-third aspects,
The power conversion control device operates as a motor control device that controls a three-phase motor as the AC load,
The rotation magnetic flux vector provides a power conversion control device which is a flux linkage vector or a rotor magnetic flux vector of the three-phase motor.

According to a twenty-fifth aspect of the present disclosure, in addition to the twenty-fourth aspect,
The rotating magnetic flux vector is the flux linkage vector,
The power conversion control device provides a power conversion control device for controlling the three-phase motor so that the flux linkage vector follows the command magnetic flux vector.

The twenty-sixth aspect of the present disclosure includes, in addition to the twenty-fifth aspect,
The power conversion control device includes:
A torque estimator for estimating a motor torque applied to the three-phase motor using the estimated current vector and the estimated flux linkage vector;
Provided is a power conversion control device for specifying the command magnetic flux vector so that the motor torque follows the command torque.

The twenty-seventh aspect of the present disclosure includes, in addition to the twenty-fourth aspect,
The rotating magnetic flux vector is the rotor magnetic flux vector;
The power conversion control device includes:
A phase estimation unit for estimating the phase of the estimated rotor magnetic flux vector;
The coordinate system of the command current vector and the estimated current vector is unified using the estimated phase, and the current vector is converted into the command using the estimated current vector and the command current vector after the unification. Provided is a power conversion control device for controlling the three-phase motor so as to follow a current vector.

According to a twenty-eighth aspect of the present disclosure,
A power conversion unit comprising the power conversion control device and the switching circuit according to any one of the twenty-first to twenty-seventh aspects is provided.

The twenty-ninth aspect of the present disclosure includes, in addition to the twenty-eighth aspect,
Provided is a power conversion unit in which a semiconductor element containing silicon carbide or gallium nitride is used for the switching circuit.

The thirtieth aspect of the present disclosure includes
A power system comprising the DC power supply, the power conversion unit and the AC load according to the 28th or 29th aspect is provided.

  Note that the power conversion control device according to the present disclosure can also be used as a generator control device. Further, the present disclosure can be regarded as a control method including one or a plurality of steps.

  Hereinafter, each embodiment will be described with reference to the drawings.

(First embodiment)
In the first embodiment, the power conversion control device is a motor control device, the power conversion device is an inverter, and the AC load is a three-phase motor. The motor control device and the inverter constitute a power conversion unit. The DC power source, the power conversion unit, and the AC load constitute a power system. As shown in FIG. 1, the motor control device 3 can be connected to an inverter 2 and an AC load 1.

(Three-phase motor 1)
The three-phase motor 1 has a rotor and a stator. The rotor includes a permanent magnet 1m (see FIG. 3A). The stator has three-phase armature windings. The armature winding corresponding to the U phase of the three-phase motor 1 may be referred to as a U phase winding. The armature winding corresponding to the V phase of the three-phase motor 1 may be referred to as a V phase winding. The armature winding corresponding to the W phase of the three-phase motor 1 may be referred to as a W phase winding. The three-phase motor 1 may be a motor having a magnetic saliency or a motor having no magnetic saliency. Specifically, the three-phase motor 1 is a permanent magnet synchronous motor, an embedded magnet synchronous motor, or the like.

(Inverter 2)
The inverter 2 is specifically a PWM (Pulse Width Modulation) inverter. The inverter 2 converts the DC voltage into a voltage vector (three-phase AC voltage) by PWM control using the switching circuit 2s. The inverter 2 applies a voltage vector to the three-phase motor 1. In the present embodiment, the inverter 2 is realized by a power module. An example of the power module is an IPM (Intelligent Power Module). The DC voltage is supplied from a DC power source (not shown).

  The internal configuration of the inverter 2 of the present embodiment will be described with reference to FIG. The inverter 2 includes a switching circuit 2s. The switching circuit 2s includes a U-phase half-bridge circuit, a V-phase half-bridge circuit, and a W-phase half-bridge circuit. Each half-bridge circuit has a pair of switching elements. Each half-bridge circuit and the smoothing capacitor 4 are connected to a positive output terminal and a negative output terminal of a DC power supply (not shown). Specifically, in each half bridge circuit, the pair of switching elements are connected in series between the positive output terminal and the negative output terminal of the DC power supply. A DC voltage from a DC power supply is applied to each half bridge circuit. The U-phase half-bridge circuit includes a high-voltage side switching element 8u and a low-voltage side switching element 9u, and the V-phase half-bridge circuit includes a high-voltage side switching element 8v and a low-voltage side switching element 9v. The W-phase half-bridge circuit includes a high-voltage side switching element 8w and a low-voltage side switching element 9w. The switching elements 8u, 8v, 8w on the high voltage side are also called upper arms 8u, 8v, 8w. The switching elements 9u, 9v, 9w on the low voltage side are also called lower arms 9u, 9v, 9w. A diode 10u is connected in parallel to the switching element 8u, a diode 10u is connected in parallel to the switching element 8u, a diode 10v is connected in parallel to the switching element 8v, and a diode 10w is connected in parallel to the switching element 8w. A diode 11u is connected in parallel to 9u, a diode 11v is connected in parallel to the switching element 9v, and a diode 11w is connected in parallel to the switching element 9w. The forward directions of the diodes 10u, 10v, 10w, 11u, 11v, and 11w are directions from the low voltage side to the high voltage side of the DC power supply. The diodes 10u, 10v, 10w, 11u, 11v, and 11w each function as a free wheel diode.

  The U-phase half-bridge circuit (first-phase switching unit) electrically connects a DC power source to the U-phase (first phase) of the three-phase motor 1 (AC load). The V-phase half-bridge circuit (second phase switching unit) electrically connects the DC power source to the V phase (second phase) of the three-phase motor 1. The W-phase half-bridge circuit (third-phase switching unit) electrically connects the DC power source to the W-phase (third phase) of the three-phase motor 1. The switching circuit 2s is controlled by the motor control device 3 (power conversion control device) so that the DC power output from the DC power source is converted into three-phase AC power to be applied to the three-phase motor 1. .

  Hereinafter, the motor control device may be described based on the dq coordinate system. The motor control device may be described based on the αβ coordinate system. The dq coordinate system and the αβ coordinate system are two-dimensional orthogonal coordinate systems. The dq coordinate system and the αβ coordinate system will be described with reference to FIGS. 3A and 3B.

  The dq coordinate system shown in FIG. 3A is a rotating coordinate system. The d-axis and the q-axis rotate at the same speed as the rotation speed (number of rotations) of the magnetic flux generated by the permanent magnet 1m. The counterclockwise direction is the phase advance direction. The permanent magnet 1 m represents a permanent magnet provided on the rotor of the three-phase motor 1. The d-axis is set as an axis extending in the direction of the magnetic flux generated by the permanent magnet 1m. The q-axis is set as an axis obtained by rotating the d-axis by 90 degrees in the advance direction. The U axis corresponds to the U phase winding, the V axis corresponds to the V phase winding, and the W axis corresponds to the W phase winding. The U-axis, V-axis, and W-axis do not rotate even when the rotor rotates. That is, the U axis, the V axis, and the W axis are fixed axes. The angle (phase) θ is a lead angle of the d axis viewed from the U axis. The angle θ is also referred to as a rotor position or a magnetic pole position. The rotational speed ω represents the rotational speed of the rotor. In this specification, unless otherwise specified, an angle means an electrical angle. The angle between the d-axis and the q-axis, the angle θ and the rotational speed ω are values based on the electrical angle.

  The αβ coordinate system shown in FIG. 3B is a fixed coordinate system. The α axis and the β axis are fixed axes. The counterclockwise direction is the phase advance direction. The α axis is set as an axis extending in the same direction as the U axis. The β axis is set as an axis obtained by rotating the α axis by 90 degrees in the advance direction.

Hereinafter, the current value (not the current that actually flows) transmitted as information is denoted by a reference numeral, such as current i _u , current i _v, and current i _w . As shown in FIG. 1, the motor control device 3 detects a current flowing into and out of one phase (U phase) of the three-phase motor 1, and generates an analog signal indicating the detected value (current i _u ). Further, the motor control device 3 detects a current flowing into and out of the phase (V phase) different from the phase in which the current flows in the inverter 2, and generates an analog signal indicating the detected value (current i _v ). The motor control device 3 controls the inverter 2 and the three-phase motor 1 using the current _iu , the current _iv, and the command torque T ^* . Specifically, according to the motor control device 3, the alternating current and the alternating current applied to the three-phase motor 1 can be controlled. The command torque T ^* is generated by a host controller (not shown).

  As shown in FIGS. 2 and 4, the motor control device 3 includes a current sensor 5, a current sensor 6, a two-phase current estimation unit 22, a magnetic flux / torque estimation unit 23, a subtraction unit 24, a command amplitude calculation unit 25, a command magnetic flux. A calculation unit 26, a command voltage calculation unit 29, a two-phase three-phase conversion unit (two-phase three-phase coordinate conversion unit) 30, and an overcurrent detection unit 31 are provided.

  In the present embodiment, the two-phase current estimation unit 22, the magnetic flux / torque estimation unit 23, the subtraction unit 24, the command amplitude calculation unit 25, the command magnetic flux calculation unit 26, the command voltage calculation unit 29, and the two-phase three-phase conversion unit 30 are It is provided by a control application executed in a DSP (Digital Signal Processor) or a microcomputer. However, these elements may include those configured by a logic circuit. The DSP or microcomputer may include peripheral devices such as a core, a memory, an A / D conversion circuit, and a communication port. In the present embodiment, the current sensor 5, the current sensor 6, and the overcurrent detection unit 31 include hardware.

(Control by motor control device 3)
The operation of the motor control device 3 will be briefly described. The current sensor 5 detects a current flowing into and out of the U phase of the three-phase motor 1 (an analog signal indicating the current i _u is generated). By the current sensor 6, current to and from the flow to the V phase of the inverter 2 (the analog signal is generated indicating the current i _v) detected. In the first operation, the two-phase current estimation unit 22, the current i _u, the two-phase currents i _.alpha..beta is generated. In the second operation, the two-phase current estimation unit 22, the current i _u and the current i _v, 2-phase current i _.alpha..beta is generated. From the two-phase current i _αβ and the command two-phase voltage v _αβ ^* , the magnetic flux / torque estimation unit 23 _obtains the estimated magnetic flux ψ _αβ , the phase θ _s of the estimated magnetic flux ψ _αβ , and the estimated torque T. That is, the motor torque and the armature flux linkage are estimated by the magnetic flux / torque estimation unit 23. The subtraction unit 24 obtains the difference between the estimated torque T and the command torque T ^* (torque error ΔT = T ^* −T). The command magnetic flux | ψ _S ^* | is specified by the command amplitude calculator 25 from the command torque T ^* . The command magnetic flux calculation unit 26 specifies the command magnetic flux vector ψ _αβ ^* from the command magnetic flux | ψ _S ^* |, the torque error ΔT, and the phase θ _s . The command voltage calculation unit 29 specifies the command two-phase voltage v _αβ ^* from the command magnetic flux vector ψ _αβ ^* , the estimated magnetic flux ψ _αβ, and the two-phase current i _αβ . The two-phase / three-phase conversion unit 30 converts the command two-phase voltage v _αβ ^* into the command three-phase voltage v _uvw ^* . The inverter 2 operates based on the command three-phase voltage v _uvw ^* . By such control, the three-phase motor 1 is controlled such that the amplitude of the armature flux linkage and the motor torque follow the command magnetic flux | ψ _S ^* | and the command torque T ^* , respectively.

  The motor control device 3 is provided with an overcurrent detection unit 31. The overcurrent detection unit 31 has a configuration suitable for determining whether or not an overcurrent has occurred in the V phase.

In this specification, like the current i _u , the current i _v, and the current i _w , the two-phase current i _αβ means a current value transmitted as information, not a current that actually flows through the three-phase motor 1. Similarly, the command two-phase voltage v _αβ ^* , the estimated magnetic flux ψ _αβ , the phase θ _s , the estimated torque T, the command torque T ^* , the command magnetic flux | ψ _S ^* |, the command magnetic flux vector ψ _αβ ^*, and the command three-phase voltage v _uvw ^* Etc. also mean values transmitted as information.

  Next, components of the motor control device 3 will be described.

(Current sensor 5)
The current sensor 5 (first current sensor) is a secondary side that connects the U-phase half-bridge circuit (first-phase switching unit) and the U-phase (first phase) of the three-phase motor 1 (AC load). Current in the wiring (current path) is detected. In the present embodiment, the motor control device (power conversion control device) 3 is closer to the three-phase motor 1 side (AC load side) than the U-phase, V-phase, and W-phase half-bridge circuits (each switching unit) of the switching circuit 2s. Only the current sensor 5 is provided as a current sensor for detecting the current flowing through the current path existing in (). The current sensor 5 detects a current flowing into and out of the U phase of the three-phase motor 1, generates an analog signal indicating the detected value (U phase current i _u ), and outputs the analog signal. In the present embodiment, the current sensor 5 is an insulating current sensor. Specifically, the current sensor 5 is a current transformer. A shunt resistor can also be used as the current sensor 5. However, when a shunt resistor is used, it is necessary to amplify the voltage while ensuring insulation by using a circuit including an insulation amplifier together.

(Current sensor 6)
The current sensor 6 (second current sensor) is configured to measure the current flowing through the wiring in the primary side wiring (current path) connecting the DC power source and the V-phase half-bridge circuit (second phase switching unit). To detect. As shown in FIG. 2, in the present embodiment, the current sensor 6 is interposed between the bus (specifically, the negative bus) 12 and the V-phase half-bridge circuit in the inverter 2, and the V-phase Measure the current flowing into and out of the half-bridge circuit. Specifically, the current sensor 6 is interposed in series between the bus 12 and the switching element 9v. The phase of the current detected by the current sensor 6 is different from the phase of the current detected by the current sensor 5. In the present embodiment, the current sensor 6 is a non-insulated current sensor. Specifically, in this embodiment, a shunt resistor is used as the current sensor 6 for the purpose of configuring the motor control device 3 at low cost. In the example shown in FIG. 2, not only the V phase but also the U phase and the W phase are provided with shunt resistors. According to this configuration, the resistance value between the three phases is prevented from being unbalanced, and three-phase equilibrium is easily ensured. In the present embodiment, current detection using U-phase and W-phase shunt resistors is not performed. It is not essential to provide shunt resistors for the U phase and the W phase.

Hereinafter, a current flowing between the bus 12 and each half bridge circuit may be referred to as a switching element current. Further, the detected values of the U-phase, V-phase, and W-phase switching element currents may be expressed as U-phase switching element current i _u , V-phase switching element current _iv, and w-phase switching element current i _w . According to this notation, the current sensor 6 of the present embodiment detects the V-phase switching element current and generates an analog signal representing the detected value (V-phase switching element current i _v ).

(Two-phase current estimation unit 22)
The two-phase current estimation unit 22 of the present embodiment estimates the two-phase current i _αβ (α-axis current i _α and β-axis current i _β ). The estimated two-phase current i _αβ is given to the magnetic flux / torque estimator 23 and the command voltage calculator 29. The two-phase current estimation unit 22 performs different operations when the motor control device 3 is performing the first operation and when the second operation is being performed.

In the first operation, the two-phase current estimation unit 22 identifies the current vector applied to the three-phase motor 1 (AC load) in cooperation with the first current sensor 5 without cooperating with the second current sensor 6. To do. That is, the two-phase current estimation unit 22 specifies the two-phase current i _αβ using the detection value of the first current sensor 5 without using the detection value of the second current sensor 6. Specifically, in the first operation, the two-phase current estimation unit 22 specifies the two-phase current i _αβ using only the detection value of the first current sensor 5 as the detection value fed back from the three-phase motor 1. More specifically, in the first operation, the two-phase current estimation unit 22 calculates the two-phase current i _αβ from the U-phase current i _u according to the equation (1). In Equation (1), s is a Laplace operator, and ω _n is a fundamental frequency of AC power. For details of the technique for estimating the two-phase current i _αβ from only the U-phase current i _u , refer to Patent Document 2.

In the second operation, the two-phase current estimation unit 22 specifies a current vector applied to the three-phase motor 1 (AC load) in cooperation with the first sensor 5 and the second sensor 6. That is, the two-phase current estimation unit 22 specifies the two-phase current i _αβ using both the detection value of the first current sensor 5 and the detection value of the second current sensor 6. Specifically, in the second operation, the two-phase current estimation unit 22 uses only the detection value of the first current sensor 5 and the detection value of the second current sensor 6 as detection values fed back from the three-phase motor 1. To identify the two-phase current i _αβ . More specifically, in the second operation, the two-phase current estimation unit 22, according to equation (2), calculates the two-phase currents i _.alpha..beta from the U-phase current i _u and the V-phase switching element current i _v.

The reason why there are two operations of the first operation and the second operation will be described. In the motor control device 3, a current is detected by the current sensor 6, an analog signal indicating the detection value is sampled every sampling period t _s. As long as the current is detected by the current sensor 6 at the sampling timing, it seems that control using the detected value can be performed, but this is not the case. Sampling involves other processes such as quantization, and a series of processes including sampling (process for extracting an analog signal) is not performed instantaneously, but a certain time t _ex (time required for A / D conversion, etc.) This is because it is done over time. That is, although the current sensor 6 to the current sensor 6 is V-phase current flows through the ON period of the switching element 9v (OFF period of switching element 8v) generates a V-phase switching element current i _v, if the period is short a series of steps for extracting an analog signal indicating a V-phase switching element current i _v can not be used V-phase switching element current i _v since not complete control of the inverter 2 and the three-phase motor 1. Specifically, 2-phase current estimation unit 22, when estimating the two-phase currents i _{.alpha..beta,} it can not be used V-phase switching element current i _v. In this specification, such a situation may be referred to as “a situation where the time required for extraction cannot be secured”.

The “situation in which the time required for extraction cannot be secured” when a PWM inverter is used as the inverter 2 will be described with reference to FIGS. 5A and 5B. FIG. 5A shows the relationship between the carrier signal 13 and a command signal (voltage level) V representing the fundamental wave component of the AC voltage to be applied to the V phase of the three-phase motor 1. In the example shown in FIG. 5A, the carrier signal 13 is a periodic triangular wave signal. The period of the carrier signal 13 is referred to as a carrier period. The carrier signal 13 is generated by a duty generation unit (not shown) included in the motor control device 3. The voltage level V is generated by using the V-phase component of the command three-phase voltage v _uvw ^{* in} the duty generation unit. In the duty generation unit, the voltage level V and the carrier signal 13 are compared. When the voltage level V crosses the carrier signal 13 in the process of increasing the level of the carrier signal 13, the duty generator switches the duty signal so that the upper arm of the V phase is turned off and the lower arm is turned on. Supply to circuit 2s. Thereby, the V-phase current flows through the switching element 9v and the current sensor 6 (shunt resistor). After the carrier signal 13 reaches the maximum level, the level of the carrier signal 13 decreases. When the voltage level V crosses the carrier signal 13 in the process of decreasing the level of the carrier signal 13, the duty generator switches the duty signal so that the lower arm of the V phase is turned off and the upper arm is turned on. Supply to circuit 2s. As a result, the V-phase current does not flow through the switching element 9v and the current sensor 6 (shunt resistor). That is, in the period when the carrier signal 13 is above the voltage level V, the current sensor 6 detects the current of the V-phase, and generates an analog signal representative of the V-phase switching element current i _v.

When the carrier signal 13 is a period that exceeds the voltage level V too short, it can not be extracted analog signal indicative of the V-phase switching element current i _v. In this case, the ON period of the switching element 9v is short, and a period during which current continues to flow through the wiring between the switching element 9v and the bus 12 is not sufficiently ensured. Such a situation occurs when the maximum value of the amplitude of the voltage level V is in the range of “not extractable” in FIG. 5A. Such a situation may occur when the amplitude of the voltage level V is large, or when the amplitude (modulation rate) of the voltage level V with respect to the amplitude of the carrier signal 13 is large. Since the AC voltage to be applied to the V phase of the three-phase motor 1 is large when the three-phase motor 1 should be operated at a high frequency, the above situation may occur when the frequency of the voltage level V is high. I can say that. Such a situation is likely to occur even when the carrier frequency is high (the carrier cycle is short). In this case, in order to extract an analog signal indicating the V-phase switching element current _iv , the maximum value of the voltage level V must be set low. There is a restriction that the voltage applied to the three-phase motor 1 must be limited.

  As is clear from the above description, the first operation should be performed when a “situation where the time required for extraction cannot be secured” occurs (or is likely to occur), and the second operation should be performed otherwise. is there. In order to switch the operation based on such a standard, as shown in FIG. 5B, when the voltage level 16 (voltage level V having a large maximum value) is generated by the duty generation unit, the first operation is performed, When level 15 (voltage level V having a small maximum value) is generated by the duty generation unit, the second operation may be performed.

Specifically, in the present embodiment, the first operation is performed when the amplitude of the fundamental wave component of the AC voltage to be applied to the V phase (second phase) of the three-phase motor 1 is equal to or greater than the threshold amplitude V _th. . The second operation is performed when the amplitude of the fundamental component of the AC voltage to be applied to the V phase of the three-phase motor 1 is less than the threshold amplitude V _th . When the motor control device 3 (or power conversion unit or power system) is configured so that the amplitude of the fundamental component of the AC voltage output from the inverter 2 can be increased up to the amplitude V _fmax , the threshold amplitude V _th can be set to an amplitude of 50 to 90% of the amplitude V _fmax , for example. For example, the amplitude V _fmax is the amplitude of the fundamental wave component of the AC voltage output from the inverter 2 when the modulation factor becomes 1.0.

  Regarding the V phase (second phase) of the three-phase motor 1, the first operation is performed when the modulation factor is 0.5 or more, and the second operation is performed when the modulation factor is less than 0.5. You may make it (refer FIG. 5B). Note that the threshold modulation rate for switching between the first operation and the second operation is not limited to 0.5, and can be selected from a range of 0.5 to 0.9.

The first operation is performed when the frequency of the fundamental component of the AC power to be applied to the V phase (second phase) of the three-phase motor 1 is equal to or higher than the threshold frequency ω _th , and the frequency is less than the threshold frequency ω _th . Sometimes the second operation may be performed. The threshold frequency ω _th is in the range of 500 to 2500 Hz, for example.

  Numerical examples relating to switching between the first operation and the second operation are shown below. From the viewpoint of reducing the current ripple, it is better that the number of periods of the carrier signal corresponding to one period of the fundamental wave of the AC power is large. For example, when the frequency of the fundamental wave of AC power is 500 Hz (when the number of pole pairs of the three-phase motor 1 is 1, the rotation speed is 30000 rpm) and the switching frequency (carrier frequency) is 10 kHz, one period of the fundamental wave is This corresponds to 20 periods of the carrier signal. If there are 20 cycles, the current ripple is suppressed to some extent. From the viewpoint of suppressing current ripple, the carrier frequency is preferably 20 kHz or higher when the frequency of the fundamental wave of AC power is 500 Hz or higher. For high-speed switching exceeding 20 kHz, it is preferable to use a semiconductor switching element containing SiC (silicon carbide) or GaN (gallium nitride). As described above, since it is not preferable to use a high carrier frequency for current detection using the current sensor 6 (shunt resistor), it is considered that the carrier frequency should be 10 or less kHz or less in the prior art. It was done. However, according to the study by the present inventors, even when the carrier frequency is 50 kHz (the carrier period is 20 μs), if a 50% off period is secured with respect to the carrier period, the time required for extraction is several μs (10 μs). The signal indicating the current measurement value can be extracted using the microcomputer described below. Therefore, setting the carrier frequency to 50 kHz or less is appropriate from the viewpoint of avoiding “a situation where the time required for extraction cannot be secured”.

(Magnetic flux / torque estimation unit 23)
The magnetic flux / torque estimation unit 23 estimates the armature linkage magnetic flux and torque from the two-phase current i _αβ and the command two-phase voltage v _αβ ^* (specifies the estimated magnetic flux _ψαβ and the estimated torque T). As shown in FIG. 6, in the present embodiment, the magnetic flux / torque estimation unit 23 includes a magnetic flux estimation unit 23a and a torque estimation unit 23b.

(Magnetic flux estimation unit 23a)
The magnetic flux estimator 23a includes a two-phase current i _αβ (α-axis current i _α and β-axis current i _β ) and a command two-phase voltage v _αβ ^* (command α-axis voltage v _α ^* and command β-axis voltage v _β ^* ). From these, the second estimated magnetic flux ψ _2αβ (second estimated magnetic flux ψ _2α , ψ _2β ) and the phase θ _s of the second estimated magnetic flux ψ _2αβ are specified. Specifically, the magnetic flux estimation unit 23a calculates the first estimated magnetic flux ψ _1α from the α-axis current i _α and the command α-axis voltage v _α ^* according to the equation (3). According to the equation (4), the first estimated magnetic flux ψ _1β is calculated from the β-axis current i _β and the command β-axis voltage v _β ^* . According to the equation (5), the second estimated magnetic flux ψ _2α is calculated from the first estimated magnetic flux ψ _1α and the first estimated magnetic flux ψ _1β . According to the equation (6), the second estimated magnetic flux ψ _2β is calculated from the first estimated magnetic flux ψ _1β and the first estimated magnetic flux ψ _1α . According to Expression (7), the phase θ _{s of} the second estimated magnetic flux ψ _2αβ is calculated from the second estimated magnetic flux ψ _2α and the second estimated magnetic flux ψ _2β . R _a in the equations (3) and (4) is _a winding resistance per phase of the three-phase motor 1. s is a Laplace operator. ω _c is a cutoff frequency of incomplete integration (first-order lag system). e _α and e _β are an α component and a β component of the induced voltage e _αβ of the three-phase motor 1. Ω ^* in the equations (5) and (6) is a command speed (command electric angular speed) of the three-phase motor 1. The command speed ω ^* is given from the host controller. For the effects obtained based on the equations (5) and (6), refer to the description below using FIGS. 9A to 10B.

A specific internal configuration of the magnetic flux estimation unit 23a will be described. As shown in FIG. 6, the magnetic flux estimation unit 23a includes an α-axis induced voltage estimation unit 46, a β-axis induced voltage estimation unit 47, an α-axis magnetic flux estimation unit (first integrator) 40, and a β-axis magnetic flux estimation unit. (Second integrator) 41, block 42, block 43, adder (adder) 44, and subtractor (subtractor) 45. The α-axis induced voltage estimation unit 46 calculates and outputs an α-axis induced voltage e _α from the α-axis current i _α and the command α-axis voltage v _α ^* . The β-axis induced voltage estimation unit 47 calculates and outputs a β-axis induced voltage e _β from the β-axis current i _β and the command β-axis voltage v _β ^* . The α-axis induced voltage e _α and the β-axis induced voltage e _β are alternating voltages (signals) having different phases. The α-axis magnetic flux estimation unit 40 integrates the α-axis induced voltage e _α to obtain the first estimated magnetic flux ψ _1α and outputs it. The β-axis magnetic flux estimation unit 41 integrates the β-axis induced voltage e _β to obtain the first estimated magnetic flux ψ _1β and outputs it. Block 42, the first estimated magnetic flux [psi _{l [alpha],} multiplied by ω _c / ω ^*, and outputs the multiplication result ψ 1α ω _c / ω ^_*. Block 43 is the first estimated magnetic flux [psi _{l [beta],} multiplied by ω _c / ω ^*, and outputs the multiplication result ψ 1β ω _c / ω ^_*. The adding unit 44 obtains and outputs the second estimated magnetic flux ψ _2α by adding the multiplication result ψ _1β ω _c / ω ^* to the first estimated magnetic flux ψ _1α . The subtracting unit 45 obtains and outputs the second estimated magnetic flux ψ _2β by subtracting the multiplication result ψ _1α ω _c / ω ^* from the first estimated magnetic flux ψ _1β . The α-axis induced voltage estimator 46 and the α-axis magnetic flux estimator 40 realize the calculation of Expression (3). The calculation of Expression (4) is realized by the β-axis induced voltage estimation unit 47 and the β-axis magnetic flux estimation unit 41. The calculation of Expression (5) is realized by the block 43 and the addition unit 44. The calculation of Expression (6) is realized by the block 42 and the subtraction unit 45. Although omitted in FIG. 6, the magnetic flux estimating unit 23a calculates and outputs the phase θ _s of the second estimated magnetic flux ψ _2αβ from the second estimated magnetic flux ψ _2α and the second estimated magnetic flux ψ _2β. It has an estimation part.

In short, the magnetic flux estimator 23a has an estimated current vector (two-phase current i _αβ ) and a command voltage vector (command two-phase voltage v _αβ ^* ) representing an AC voltage to be applied to the AC load (three-phase motor 1). _{Is used} to estimate the rotating magnetic flux vector (second estimated magnetic flux ψ _2αβ ) applied to the AC load. Specifically, the magnetic flux estimator 23a uses the estimated α-axis component (α-axis current i _α ) of the current vector and the α-axis component (command α-axis voltage v _α ^* ) of the command voltage vector. An α-axis induced voltage estimator 46 for estimating the α-axis component (α-axis induced voltage e _α ) of the induced voltage generated in the (three-phase motor 1), and the β-axis component (β-axis current of the estimated current vector β axis for estimating the β axis component (β axis induced voltage e _β ) of the induced voltage generated in the AC load using i _β ) and the β axis component of the command voltage vector (command β axis voltage v _β ^* ) Α-axis magnetic flux estimation for temporarily estimating the α-axis component (first estimated magnetic flux ψ _1α ) of the rotating magnetic flux vector using the induced voltage estimation unit 47 and the α-axis component (α-axis induced voltage e _α ) of the estimated induced voltage. Unit 40 and the β-axis component of the rotating magnetic flux vector using the estimated β-axis component of the induced voltage (β-axis induced voltage e _β ) (First estimated magnetic flux [psi _{l [beta])} beta -axis magnetic flux estimator 41 provisionally estimating, tentatively estimated beta-axis component of the rotating flux vector of the rotating flux vector which is tentatively estimated using (first estimated magnetic flux [psi _{l [beta])} Using an α-axis component correction unit (block 43 and addition unit 44) that corrects the α-axis component (first estimated magnetic flux ψ _1α ) and a temporarily estimated α-axis component (first estimated magnetic flux ψ _1β ) of the rotating magnetic flux vector. And a β-axis component correction unit (block 42 and subtraction unit 45) that corrects the β-axis component (first estimated magnetic flux ψ _1β ) of the rotational magnetic flux vector that is temporarily estimated.

  With regard to the above description, in this embodiment, the power conversion control device operates as the motor control device 3 that controls the three-phase motor 1 as an AC load, and the rotating magnetic flux vector is linked to the three-phase motor 1. It is a magnetic flux vector. However, it is also possible to configure a motor control device that uses a rotor magnetic flux vector (magnet magnetic flux) as the rotating magnetic flux vector (refer to the second embodiment for details).

The magnetic flux estimation unit 23a adopts and _outputs the second estimated magnetic flux ψ _2αβ (estimated magnetic flux ψ _2α , ψ _2β ) as the estimated magnetic flux ψ _αβ of the armature linkage flux. However, the first estimated magnetic flux ψ _1αβ (estimated magnetic flux ψ _1α , ψ _1β ) may be adopted and output as the estimated magnetic flux ψ _αβ of the armature linkage flux. The phase θ _s can also be specified by changing the estimated magnetic flux ψ _2α and the estimated magnetic flux ψ _2β on the right side of the equation (7) to the estimated magnetic flux ψ _1α and the estimated magnetic flux ψ _1β . In addition, the magnetic flux estimation unit 23a performs estimation using a two-phase voltage obtained by three-phase to two-phase conversion of a detected value of the voltage applied to the three-phase motor 1 instead of the command two-phase voltage v _αβ ^*. The magnetic flux _ψαβ may be obtained.

(Torque estimation unit 23b)
The torque estimation unit 23b estimates the motor torque from the two-phase current i _αβ (α-axis current i _α and β-axis current i _β ) and the estimated magnetic flux ψ _αβ (estimated magnetic flux ψ _α , ψ _β ). In this embodiment, the torque estimation part 23b calculates | requires the estimated torque T using Formula (8). P _n in Equation (8) is the _number of pole pairs of the three-phase motor 1.

Estimated magnetic flux [psi _.alpha..beta obtained in flux-torque estimator 23 is given to the command voltage calculating unit 29. The phase θ _s is given to the command magnetic flux calculator 26. The estimated torque T is given to the subtracting unit 24. In this embodiment, the magnetic flux estimation unit 23a and the torque estimation unit 23b are integrated to form the magnetic flux / torque estimation unit 23. However, the magnetic flux estimation unit 23a and the torque estimation unit 23b are independent from each other. It can also be.

(Subtraction unit 24)
The subtraction unit 24 obtains a difference (torque error ΔT: T ^* −T) between the estimated torque T and the command torque T ^* . A known operator may be used as the subtraction unit 24.

(Command amplitude calculator 25)
The command amplitude calculator 25 specifies the command magnetic flux | ψ _S ^* | from the command torque T ^* . The command magnetic flux | ψ _S ^* | is a scalar. The command magnetic flux | ψ _S ^* | of the present embodiment is for minimizing the motor current. Control that controls a three-phase motor so that the ratio of the motor torque value to the motor current value is maximized while setting the motor torque as the target value is known as Maximum Torque Per Ampere (MTPA) control. It is In the present embodiment, the command amplitude calculation unit 25 is configured to identify the command magnetic flux | ψ _S ^* | from the command torque T ^* using a table. The command amplitude calculator 25 may be configured to specify the command magnetic flux | ψ _S ^* | by calculation.

The method of specifying the command magnetic flux | ψ _S ^* | when executing the maximum torque / current control will be understood by those skilled in the art from the following description. When a motor having no magnetic saliency is used as the three-phase motor 1, the armature flux linkage amplitude | ψ _S | and the motor torque T are approximated by equations (9) and (10). | Ψ _a0 | is a constant given as the amplitude of the magnetic flux generated by the permanent magnet 1 m in the three-phase motor 1. In the following description, | ψ _a0 | may be referred to as a magnetic flux parameter. L is the inductance per phase of the armature winding of the three-phase motor 1. i _q is a q-axis current. P _n is the number of pole pairs of the motor. From equations (9) and (10), equation (11) is derived. By substituting T for the command torque T ^* and | ψ _S | for the command magnetic flux | ψ _S ^* |, respectively, a relational expression between the command torque T ^* and the command magnetic flux | ψ _S ^* | Using this relational expression, the command magnetic flux | ψ _S ^* | can be obtained from the command torque T ^* . Of course, a conversion table can also be created.

When a motor having magnetic saliency is used as the three-phase motor 1, the armature flux linkage amplitude | ψ _S | and the motor torque T are approximated by equations (12) and (13). L _d is the d-axis inductance. L _q is a q-axis inductance. i _d is a d-axis current. The d-axis current i _d and the q-axis current i _q generally satisfy the relationship of Expression (14). Expressions (12), (13), and (14) provide a conversion table that can specify the amplitude | ψ _S | of the armature flux linkage from the motor torque T without using the variables i _d and i _q . By replacing T with the command torque T ^* and | ψ _S | with the command magnetic flux | ψ _S ^* |, the command magnetic flux | ψ _S ^* | can be identified from the command torque T ^* .

(Command magnetic flux calculation unit 26)
The command magnetic flux calculation unit 26 specifies the command magnetic flux vector _ψαβ ^* that the armature linkage magnetic flux should follow from the command magnetic flux | ψ _S ^* |, the phase θ _s, and the torque error ΔT. The command magnetic flux calculation unit 26 in the present embodiment specifies the command magnetic flux vector _ψαβ ^* according to the block diagram shown in FIG. Specifically, the command magnetic flux calculation unit 26 includes a PI control unit 38, an addition unit 36, and a vector generation unit 37. The PI control unit 38 specifies the phase correction amount Δθ _S ^* by proportional-integral control for converging ΔT to zero. The adding unit 36 obtains the sum (θ _S ^* : θ _S + Δθ _S ^* ) of the phase θ _S and the phase correction amount Δθ _S ^* . The vector generation unit 37 specifies the command magnetic flux vector _ψαβ ^* from the command magnetic flux | ψ _S ^* | and the total θ _S ^* . Specifically, the vector generation unit 37 obtains the command magnetic flux vector _ψαβ ^* using the equations (15) and (16). The command magnetic flux vector _ψαβ ^* is given to the command voltage calculation unit 29.

(Command voltage calculation unit 29)
The command voltage calculation unit 29 calculates a command two-phase voltage v _αβ ^* (command α-axis voltage v _α ^* and command β-axis from the difference between the command magnetic flux vector ψ _αβ ^* and the estimated magnetic flux ψ _αβ and the two-phase current i _αβ. The voltage v _β ^* ) is specified. In the present embodiment, the command voltage calculation unit 29 obtains the command two-phase voltage v _αβ ^* using Equation (17). T _s in equation (17) is a control period (sampling period). Incidentally, when the 3-phase motor 1 is rotated at a high speed, the voltage drop based on the winding resistance R _a is very small. For this reason, the command voltage calculation unit 29 ignores the second term on the right side of Equation (17) and specifies the command two-phase voltage v _αβ ^* from the difference between the command magnetic flux vector ψ _αβ ^* and the estimated magnetic flux ψ _αβ. It may be configured as follows. The command two-phase voltage v _αβ ^* is given to the two-phase / three-phase converter 30.

(Two-phase three-phase converter 30)
The two-phase three-phase conversion unit 30 converts the command two-phase voltage v _αβ ^* into the command three-phase voltage v _uvw ^* . Thereafter, a voltage vector corresponding to the command three-phase voltage v _uvw ^* is generated by the inverter 2 and applied to the three-phase motor 1.

(Overcurrent detection unit 31)
Overcurrent detection unit 31 of the present embodiment, by receiving the analog signal indicating a V-phase switching element current i _v in hardware portion determines whether an overcurrent has occurred. Specifically, the overcurrent detection unit 31, using an analog signal indicative of the analog signal and the V-phase switching element current i _v indicates a U-phase current i _u, 3-phase motor 1 the U-phase, V-phase and W-phase Whether or not an overcurrent has occurred is determined. The overcurrent detection unit 31 is provided outside the inverter 2. As illustrated in FIG. 8A, the overcurrent detection unit 31 includes an overcurrent detection circuit 31a and an overcurrent determination unit 31b.

(Overcurrent detection circuit 31a)
The overcurrent detection circuit 31a is configured by hardware. Overcurrent detection circuit 31a includes an analog signal indicative of the V-phase switching element current i _v, by using the current threshold (overcurrent determination value) i _vmax, generates an analog signal S _OC, and outputs. The current threshold i _vmax is a constant. The analog signal S _OC is held at a relatively high level (H level) for a predetermined period t _c from the time when the V-phase switching element current i _v larger than the current threshold value i _vmax is input to the overcurrent detection circuit 31a. . In other periods, the analog signal S _OC is held at a relatively low level (L level). In the present embodiment, the analog signal S _OC is a voltage, the H level means that the analog signal S _OC is a relatively high voltage, and the L level means that the analog signal S _OC is a relatively low voltage. Means. The overcurrent detection circuit 31a typically includes a comparator circuit and a latch circuit. FIG. 8B shows a relationship _among the current threshold value i _vmax , the V-phase switching element current i _v, and the analog signal S _OC . In schematic 8B, V-phase switching element current i _v is the V-phase switching element current i _v at emphasizing aims period is short non-zero is a delta function representation in four positions. In practice, V-phase switching element current i _v forms a pulse waveform (switching pulse waveform) having a width corresponding to the on period of the switching element 9v.

(Overcurrent determination unit 31b)
The overcurrent determination unit 31b performs V-phase overcurrent detection using the analog signal _SOC . Specifically, the overcurrent judging unit 31b, for each sampling period t _s, over the predetermined time t _ex, implementing the step of extracting the analog signal S _OC. The overcurrent determination unit 31b determines that an overcurrent has occurred when the analog signal _SOC is at the H level, and determines that no overcurrent has occurred when the analog signal _SOC is at the L level. .

As described above, when the ON period of the switching element 9v is short, the motor control device 3 extracts an analog signal indicating the detected value of the V-phase current (V-phase switching element current i _v ) by the current sensor 6. I can't. However, according to the overcurrent detection unit 31 of the present embodiment, it is possible to detect the V-phase overcurrent even in such a case. Overcurrent detection circuit 31a, because they are constituted by hardware, the reception of the analog signal indicative of the V-phase switching element current i _v, V-phase switching element current i _v and the comparison and the analog signal between a current threshold i _vmax S _OC This is because the output can be always performed. That is, an analog of H level indicating that an overcurrent from the time when the analog signal indicating the larger V-phase switching element current i _v than the current threshold value i _vmax is input to the overcurrent detection circuit 31a for a period of time t _c is detected This is because the signal S _OC is continuously output from the overcurrent detection circuit 31a and is continuously output, so that the analog signal S _OC can be extracted by the overcurrent determination unit 31b. In the present embodiment, the period t _c has a length equal to or longer than the time _tex . Thereby, the overcurrent detection by the overcurrent determination unit 31b can be reliably performed.

In this embodiment, the U-phase current is detected in the wiring connecting the U-phase half-bridge circuit and the three-phase motor 1 instead of the wiring connecting the DC power supply and the U-phase half-bridge circuit. An analog signal representing the U-phase current i _u is generated. Since this analog signal has not a switching pulse waveform but a waveform close to a sine wave, the analog signal does not change discontinuously during the extraction of the analog signal (during the ON period of the switching element 9u). The instantaneous value of the current flowing through the switching element 9u is almost the same as the instantaneous value of the current flowing through the U phase of the three-phase motor 1. On the other hand, the former current is zero during the OFF period of the switching element 9u. Includes a period in which the latter current is non-zero, even if the switching element 9u is in the OFF period, the current continues to flow through the three-phase motor 1 due to the effect of the inductance component of the three-phase motor 1, and the other current passes through the diode 10u. The switching circuit 2s is configured so that the current can continue to flow through the three-phase motor 1 by circulating the current between the phases. This is because if the “situation where the time required for extraction cannot be secured” does not occur, even if the current flowing through the wiring between the bus 12 and the switching element 9u is measured, the three-phase motor 1 flows in and out. This is why the same current detection value can be obtained even if the measured current is measured, and that the analog signal representing the U-phase current i _u has a waveform close to a sine wave, not a switching pulse waveform). That is, there is no situation where the time required for extracting the analog signal indicating the U-phase current i _u cannot be secured. Therefore, the overcurrent detection unit 31 can detect the U-phase overcurrent by directly applying the U-phase current i _u to the overcurrent determination unit 31b (see FIG. 8A). Although not shown, in this embodiment, a W-phase current i _w (= −i _u −i _v ) is calculated from the U-phase current i _u and the V-phase switching element current i _v using a hardware arithmetic circuit. calculates the using an overcurrent detecting circuit which compares the W-phase current i _w and the current threshold value i _vmax (circuit similar to the overcurrent detection circuit 31a) generating an analog signal (the same signal as the analog signal S _OC) doing. The generated analog signal is supplied to the overcurrent determination unit 31b to detect the W-phase overcurrent.

Without even decline again, if you can secure the time required for extraction of the analog signal indicative of the V-phase switching element current i _v (if the second performing operation), the overcurrent detection unit 31, overcurrent detection circuit 31a without using, by providing a V-phase switching element current i _v directly overcurrent determination unit 31b, it is possible to perform the overcurrent detection of the V phase. In this case, the W-phase overcurrent can also be detected without using the overcurrent detection circuit.

(Problems of conventional technology)
For the purpose of promoting the understanding of the effects of the present embodiment, the problems of the prior art will be described. In Patent Document 1, a current sensor detects one phase (for example, U phase) of the three phase currents of a three-phase motor, and the remaining two phases (for example, V phase and W phase) are detected from the detected current. Techniques for estimating are described. Patent Document 2 describes a technique for detecting a one-phase current in a multiphase motor with a current sensor and estimating a two-phase current (α, β-axis current or d, q-axis current) from the detected current. Yes. According to these techniques, the number of current sensors can be reduced to one. Therefore, according to these techniques, a power conversion control device can be realized at low cost. However, these techniques cannot detect an overcurrent that occurs between two phases other than the phase in which the current is detected. Also, with these techniques, when the rotational speed is low, such as when the motor is started, it is possible to ensure the estimation accuracy of the two-phase current other than the phase in which the current is detected, or to ensure the estimation accuracy of the two-phase current. Difficult to do. For this reason, with these techniques, it is difficult to accurately control the motor when the rotational speed is low, such as when the motor is started. The same applies to controlling an AC load other than the motor.

  A technique is also known in which a switching element current of each phase of a PWM inverter is detected by a shunt resistor, and a motor is controlled using a current detection value. When only this technique is employed, as described with reference to FIG. 5A, a certain period must be secured in which the amplitude of the command signal representing the fundamental component of the AC voltage to be applied continues to exceed the amplitude of the carrier signal. For example, the motor control device cannot extract an analog signal indicating the switching element current. It is conceivable to reduce the carrier signal frequency (switching frequency) to solve this problem. However, if the switching frequency is too low, the current ripple increases.

(Effect of this embodiment: ensuring performance at low cost)
The motor control device 3 of the present embodiment has a current flowing through the current path on the three-phase motor 1 side (secondary side) with respect to the U-phase, V-phase, and W-phase half-bridge circuits (each switching unit) of the switching circuit 2s. Only the current sensor 5 is provided as a current sensor for detecting the current. In order to detect the current at this position, an insulating current sensor such as a current transformer is required. This is because the current path on the secondary side is floating, and it is highly necessary to use an insulating current sensor as the current sensor 5. That is, it is highly necessary to use an expensive current sensor as the current sensor 5. On the other hand, the motor control device 3 detects the current flowing through the current path on the DC power source side (primary side) from the U-phase, V-phase, and W-phase half-bridge circuits (each switching unit) of the switching circuit 2s. A current sensor 6 is provided as a sensor. For the current detection at this position, a non-insulating current sensor such as a shunt resistor can be used. This is because it is easy to use a current path connected to the ground potential as a primary-side current path where the current sensor 6 is provided. That is, an inexpensive current sensor can be used as the current sensor 6. Thus, in this embodiment, it is not necessary to use two expensive current sensors, and the three-phase motor 1 can be controlled by a combination of one expensive current sensor and one inexpensive current sensor. Therefore, the motor control device 3 of the present embodiment can be configured at a low cost. Note that it is difficult to reduce the cost even if shunt resistors are provided in different two-phase current paths on the secondary side. This is because it is necessary to use an insulation amplifier or the like in combination with the shunt resistor in order to ensure insulation, which is costly.

  Furthermore, according to the combination of the current sensor 5 and the current sensor 6, the current vector flowing through the AC load can be estimated with sufficient accuracy both when the speed of the three-phase motor 1 is low (when starting) and when it is high. .

(Effect of this embodiment: Overcurrent detection at low cost)
In addition, the motor control device 3 of the present embodiment includes an overcurrent detection unit 31. According to the overcurrent detection unit 31, even if the on period of the switching element of the phase to which the current sensor 6 is connected is short, the overcurrent of that phase can be detected. Further, the remaining one-phase current can be calculated from the detected value of the detected two-phase current among the three phases, and the overcurrent can be detected from the calculated value. That is, overcurrents in all phases can be detected.

(Effect of this embodiment: Correction of magnetic flux estimation)
As described with reference to FIG. 6, in this embodiment, the output (α-axis induced voltage e _α ) of the α-axis magnetic flux estimator 40 is used and the output (β-axis induced voltage e _β ) of the β-axis magnetic flux estimator 41 is used. Thus, the correction is performed by the α-axis component correction unit (block 43 and addition unit 44). Similarly, the output of the β-axis magnetic flux estimator 41 (β-axis induced voltage e _β ) is used as the output of the α-axis magnetic flux estimator 40 (α-axis induced voltage e _α ). And the subtraction unit 45). That is, the first estimated magnetic flux [psi _{l [alpha]} and the first estimated magnetic flux [psi _{l [beta]} is corrected to the second estimated magnetic flux [psi _2.alpha and second estimated magnetic flux [psi _2.beta, the second estimated magnetic flux [psi _2.alpha and second estimated magnetic flux [psi _2.beta is estimated magnetic flux [psi _alpha and is taken as the estimated magnetic flux [psi _beta. When the α-axis component is expressed as a complex number with the real-axis component and the β-axis component as the imaginary-axis component, the relationship between the induced voltage e _αβ and the second estimated magnetic flux ψ _2αβ can be expressed by Expression (18).

In the case of using a conventional incomplete integration, as the estimated magnetic flux [psi _alpha and estimated flux [psi _beta (estimated magnetic flux [psi _{.alpha..beta),} the same estimated magnetic flux and the first estimated magnetic flux [psi _{l [alpha]} and the first estimated magnetic flux [psi _{l [beta]} (estimated flux _ψ 1αβ) Is used. The relationship between the induced voltage e _αβ and the estimated magnetic flux ψ _1αβ can be expressed by equation (19).

The gain characteristics of the equations (18) and (19) are shown in FIG. 9A. The phase characteristics of Equation (18) and Equation (19) are shown in FIG. 9B. In FIG. 9A, a line 50 indicates the gain characteristic of the equation (19), that is, the gain characteristic when incomplete integration is used. A line 51 indicates the gain characteristic of Expression (18), that is, the gain characteristic in the present embodiment. In FIG. 9B, a line 52 indicates the phase characteristic of the equation (19), that is, the phase characteristic when incomplete integration is used. A line 53 indicates the phase characteristic of the equation (18), that is, the phase characteristic in the present embodiment. Lines 50 to 53 are for ω _c = 40 rad / s. Ideally, the phase characteristic of the integrator is a phase of −90 degrees regardless of the frequency. The phase of the line 52 deviates greatly from −90 degrees in the low frequency region. On the other hand, the phase of the line 53 is substantially −90 degrees even in the low frequency region. From line 52 and line 53, than the case of using the estimated magnetic flux [psi _1Arufabeta based on an incomplete integration in accordance with the prior art as the estimated magnetic flux [psi _{.alpha..beta,} better in the case of using the estimated magnetic flux [psi _2Arufabeta based on the present embodiment as the estimated magnetic flux [psi _.alpha..beta It can be seen that good phase characteristics can be obtained.

The magnetic flux estimation accuracy will be described with reference to FIGS. 10A and 10B. 10A and 10B, line 54 represents the true α-axis magnetic flux. Line 55 represents the true β-axis magnetic flux. “True” means that the magnetic flux actually applied to the three-phase motor 1 is represented without error. Line 56 represents the first estimated magnetic flux ψ _1α . Line 57 represents the first estimated magnetic flux ψ _1β . Line 58 represents the second estimated magnetic flux ψ _2α . A line 59 represents the second estimated magnetic flux ψ _2β . As represented by the lines 56 and 58, the initial values of the estimated magnetic fluxes ψ _1α and ψ _2α are intentionally given an error. Specifically, the initial value of the true α-axis magnetic flux is 0.1 Wb, and the initial values of the true β-axis magnetic flux and the estimated magnetic fluxes ψ _1α , ψ _1β , ψ _2α , ψ _2β are 0 Wb. The scale of the horizontal axis (time axis) in FIGS. 10A and 10B is set so that a transient phenomenon immediately after the start of estimation is grasped. From the lines 54 and 56, it is understood that the phase of the first estimated magnetic flux ψ _1α is delayed from the phase of the true α-axis magnetic flux, and from the lines 55 and 57, the phase of the first estimated magnetic flux ψ _1β is true. It is understood that the phase is delayed from the phase of the β-axis magnetic flux. This is because, when Equation (19), that is, incomplete integration is used, the three-phase motor 1 operates at an operating point deviated from the high-efficiency operating point, a large motor current flows, and the three-phase motor 1 It means that a big loss occurs. This can also adversely affect control when the speed of the three-phase motor 1 is low, such as during startup. On the other hand, it can be understood from the lines 54 and 58 that the phase of the second estimated magnetic flux ψ _2α coincides with the phase of the true α-axis magnetic flux with high accuracy, and from the lines 55 and 59, the second estimated magnetic flux ψ _It can be seen that the phase of _2β coincides with the phase of the true β-axis magnetic flux with high accuracy. This means that according to the equation (18), that is, according to the present embodiment, the three-phase motor 1 can be operated with high efficiency even when the speed of the three-phase motor 1 is low, such as at the time of starting. Means.

  In the present embodiment, the current sensor 5 and the current sensor 6 are provided so as to measure two-phase currents of the U phase and the V phase, and the motor control device 3 is controlled so as to perform control using these detected values. Is configured. However, even if the current sensor 5 and the current sensor 6 are provided so as to measure other two-phase currents, the same control can be performed. Moreover, the matter demonstrated in this embodiment is applicable also to a power conversion unit, a power system, and a control method other than a power conversion control apparatus. Moreover, the matter demonstrated in this embodiment is applicable also when using a power conversion control apparatus as a generator control apparatus. This also applies to the modified examples and embodiments described later.

(Modification 1-1)
Instead of the magnetic flux estimator 23a (FIG. 6) of the first embodiment, a magnetic flux estimator 73a shown in FIG. 11 may be used. Hereinafter, the magnetic flux estimation unit 73a will be described. In the modified example 1-1, the same parts as those in the first embodiment are denoted by the same reference numerals, and the description thereof may be omitted.

The magnetic flux estimation unit 73 a includes a block 70 and a block 71 instead of the block 42 and the block 43. The block 70 performs calculation using the α-axis induced voltage e _α and outputs a calculation result e _α ω _c / ω ^* (s + ω _c ). The calculation result e _α ω _c / ω ^* (s + ω _c ) is the same as the output from the block 42 of the first embodiment. The block 71 performs a calculation using the β-axis induced voltage e _β and outputs a calculation result e _β ω _c / ω ^* (s + ω _c ). The calculation result e _β ω _c / ω ^* (s + ω _c ) is the same as the output from the block 43 of the first embodiment. In addition section 44, by the operation result to the first estimated magnetic flux _{ψ 1α e β ω c / ω} * (s + ω c) is added together, the second estimated magnetic flux [psi _2.alpha is obtained. The subtraction unit 45 subtracts the calculation result e _α ω _c / ω ^* (s + ω _c ) from the first estimated magnetic flux ψ _1β to obtain the second estimated magnetic flux ψ _2β .

In short, the magnetic flux estimator 73a uses the estimated α-axis component (α-axis current i _α ) of the current vector and the α-axis component (command α-axis voltage v _α ^* ) of the command voltage vector to generate an AC load (three-phase). An α-axis induced voltage estimation unit 46 for estimating an α-axis component (α-axis induced voltage e _α ) of the induced voltage generated in the motor 1), and a β-axis component (β-axis current i _β ) of the estimated current vector And β-axis induced voltage estimation for estimating the β-axis component (β-axis induced voltage e _β ) of the induced voltage generated in the AC load using the β-axis component of the command voltage vector (command β-axis voltage v _β ^* ) And an α-axis magnetic flux estimator 40 for temporarily estimating the α-axis component (first estimated magnetic flux ψ _1α ) of the rotating magnetic flux vector using the estimated α-axis component of the induced voltage (α-axis induced voltage e _α ). , Using the estimated β-axis component of the induced voltage (β-axis induced voltage e _β ), the β-axis component of the rotating magnetic flux vector ( Β-axis magnetic flux estimator 41 that temporarily estimates the first estimated magnetic flux ψ _1β ) and the α-axis component of the rotating magnetic flux vector that is temporarily estimated using the estimated β-axis component of the induced voltage (β-axis induced voltage e _β ) Temporarily estimated using an α-axis component correction unit (block 71 and addition unit 44) that corrects (first estimated magnetic flux ψ _1α ) and the α-axis component (α-axis induced voltage e _α ) of the estimated induced voltage. A β-axis component correction unit (block 70 and subtraction unit 45) that corrects the β-axis component (first estimated magnetic flux ψ _1β ) of the rotating magnetic flux vector.

  Similar to the magnetic flux estimator 23a of the first embodiment, the magnetic flux estimator 73a realizes Expression (18). Therefore, according to the modified example 1-1, the same effect as that of the first embodiment can be obtained.

(Modification 1-2)
Instead of the magnetic flux estimator 23a (FIG. 6) of the first embodiment, a magnetic flux estimator 83a shown in FIG. 12 may be used. Hereinafter, the magnetic flux estimation unit 83a will be described. In the modified example 1-2, the same parts as those in the first embodiment are denoted by the same reference numerals, and description thereof may be omitted.

  Unlike the magnetic flux estimator 23a, the magnetic flux estimator 83a does not include the α-axis magnetic flux estimator 40, the β-axis magnetic flux estimator 41, the block 42, the block 43, the adder 44, and the subtractor 45. Instead, the magnetic flux estimating unit 83a includes a block 82, a block 83, an adding unit 84, a subtracting unit 85, an α-axis magnetic flux estimating unit 80, and a β-axis magnetic flux estimating unit 81.

Block 82 multiplies the omega _c / omega ^* the alpha-axis induced voltage e _alpha, and outputs the multiplication result e _α ω _c / ω ^*. Block 83 multiplies the omega _c / omega ^* the beta axis induced voltage e _beta, and outputs the multiplication result e _β ω _c / ω ^*. The adding unit 84 _{obtains an} addition result e _cα = e _α + e _β ω _c / ω ^* by adding the multiplication result e _β ω _c / ω ^* to the α-axis induced voltage e _α . Subtracting unit 85 obtains the beta axis induced voltage subtraction result by subtracting the multiplication result _e α ω _c / ω ^* from _{e β e cβ = e β -e} α ω c / ω *. The α-axis magnetic flux estimating unit 80 integrates the addition result _ecα to obtain an estimated magnetic flux ψ _2α and outputs it. Estimated magnetic flux [psi _2.alpha is the same as the second estimated magnetic flux [psi _2.alpha the first embodiment. The β-axis magnetic flux estimation unit 81 integrates the subtraction result _ecβ to obtain an estimated magnetic flux ψ _2β and outputs it. Estimated magnetic flux [psi _2.beta is the same as the second estimated magnetic flux [psi _2.beta the first embodiment.

In short, the magnetic flux estimator 83a uses the estimated α-axis component of the current vector (α-axis current i _α ) and the α-axis component of the command voltage vector (command α-axis voltage v _α ^* ) to generate an AC load (three-phase). An α-axis induced voltage estimation unit 46 that temporarily estimates the α-axis component (α-axis induced voltage e _α ) of the induced voltage generated in the motor 1), and the β-axis component of the estimated current vector (β-axis current i _β ) And the β-axis component of the command voltage vector (command β-axis voltage v _β ^* ) and the β-axis induction for temporarily estimating the β-axis component (β-axis induced voltage e _β ) of the induced voltage generated in the AC load. Α-axis that corrects the α-axis component (α-axis induced voltage e _α ) of the induced voltage temporarily estimated using the voltage estimation unit 47 and the β-axis component (β-axis induced voltage e _β ) of the temporarily estimated induced voltage A component correction unit (block 83 and addition unit 84), and an α-axis component (α-axis alpha axis e _alpha) and beta axis component correcting section that provisionally correct beta-axis component of the estimated induced voltage (beta-axis induced voltage e _beta) with (block 82 and the subtraction unit 85), the corrected induced voltage An α-axis magnetic flux estimator 80 that estimates the α-axis component (estimated magnetic flux ψ _2α ) of the rotating magnetic flux vector using the component (addition result e _cα ), and the β-axis component (subtraction result e _cβ ) of the corrected induced voltage And a β-axis magnetic flux estimator 81 that estimates the β-axis component (estimated magnetic flux ψ _2β ) of the rotating magnetic flux vector.

  Similar to the magnetic flux estimator 23a of the first embodiment, the magnetic flux estimator 83a realizes Expression (18). Therefore, according to Modification 1-2, the same effect as that of the first embodiment can be obtained.

(Modification 1-3)
Instead of the magnetic flux estimator 23a of the first embodiment, a magnetic flux estimator 93a shown in FIG. 13 can be used. Hereinafter, the magnetic flux estimation unit 93a will be described. In the modified example 1-3, the same parts as those in the first embodiment are denoted by the same reference numerals, and description thereof may be omitted.

  Unlike the magnetic flux estimation unit 23a, the magnetic flux estimation unit 93a does not include the α-axis magnetic flux estimation unit 40, the β-axis magnetic flux estimation unit 41, the block 42, the block 43, the addition unit 44, and the subtraction unit 45. Instead, the magnetic flux estimating unit 93a includes a block 98, a block 99, an adding unit 94, a subtracting unit 95, an α-axis magnetic flux estimating unit 90, a β-axis magnetic flux estimating unit 91, a block 92, and a block. 93.

A block 98 multiplies the α-axis induced voltage e _α by the correction coefficient b to obtain a multiplication result be _α . A block 99 multiplies the β-axis induced voltage e _β by the correction coefficient b to obtain a multiplication result be _β . Addition unit 94, the multiplication result to the BE _alpha, obtain one control cycle before the multiplication result Aw _c [psi _2.beta adding to sum _{e cα = be α + aω c} ψ 2β. Subtracting unit 95 obtains the subtraction result _{e cβ = be β -aω c ψ} 2α by subtracting the multiplication result Aw _c [psi _2.alpha of one control period before the multiplication result BE _beta. The α-axis magnetic flux estimating unit 90 integrates the addition result _ecα to obtain an estimated magnetic flux ψ _2α and outputs it. The β-axis magnetic flux estimation unit 91 _obtains an estimated magnetic flux ψ _2β by integrating the subtraction result _ecβ and outputs it.

In short, the magnetic flux estimator 93a uses the estimated α-axis component (α-axis current i _α ) of the current vector and the α-axis component (command α-axis voltage v _α ^* ) of the command voltage vector to generate an AC load (three-phase). An α-axis induced voltage estimation unit 46 that temporarily estimates the α-axis component (α-axis induced voltage e _α ) of the induced voltage generated in the motor 1), and the β-axis component of the estimated current vector (β-axis current i _β ) And the β-axis component of the command voltage vector (command β-axis voltage v _β ^* ) and the β-axis induction for temporarily estimating the β-axis component (β-axis induced voltage e _β ) of the induced voltage generated in the AC load. Α-axis component correction that corrects the α-axis component (α-axis induced voltage e _α ) of the induced voltage temporarily estimated using the voltage estimating unit 47 and the β-axis component (estimated magnetic flux ψ _2β ) of the estimated rotating magnetic flux vector Section (block 98, block 93 and adder section 94) and the estimated rotating magnetic flux vector Axis component (estimated magnetic flux [psi _2.alpha) beta axis component correcting section that provisionally correct beta-axis component of the estimated induced voltage (beta-axis induced voltage e _beta) with (block 99, block 92 and subtraction unit 95), An α-axis magnetic flux estimator 90 that estimates the α-axis component (estimated magnetic flux ψ _2α ) of the rotating magnetic flux vector using the α-axis component of the corrected induced voltage (addition result e _cα ), and the β-axis of the corrected induced voltage A β-axis magnetic flux estimator 91 that estimates the β-axis component (estimated magnetic flux ψ _2β ) of the rotating magnetic flux vector using the component (subtraction result e _cβ ). Note that the β-axis component of the estimated rotating magnetic flux vector used by the α-axis component correcting unit is specifically the β-axis component of the rotating magnetic flux vector estimated one control cycle before. In addition, the α-axis component of the estimated rotating magnetic flux vector used by the β-axis component correcting unit is specifically the α-axis component of the rotating magnetic flux vector estimated before one control cycle.

When the α-axis component is expressed as a complex number with the real-axis component and the β-axis component as the imaginary-axis component, the relationship between the induced voltage e _αβ and the estimated magnetic flux ψ _2αβ can be expressed by equation (20).

FIG. 14A shows the gain characteristic of the equation (20). The phase characteristic of Expression (20) is shown in FIG. 14B. Lines 50 and 52 in FIGS. 14A and 14B are the same as the lines 50 and 52 in FIG. 9A and FIG. 9B (gain characteristics and phase characteristics when incomplete integration is used). A line 60 in FIG. 14A shows the gain characteristic of Expression (20), that is, the gain characteristic in the case of Modification 1-3. A line 61 in FIG. 14B indicates the phase characteristic of Expression (20), that is, the phase characteristic in the case of Modification 1-3. Line 50, line 52, line 60 and line 61 are for ω _c = 40 rad / s. Lines 60 and 61 are for a = 5 and b = 1. As described above, the phase characteristic of the integrator is ideally a phase of −90 degrees regardless of the frequency. The phase of the line 61 is approximately −90 degrees even in the low frequency region. On the other hand, the gain characteristic decreases as shown by the line 60. In order to improve the gain characteristic, the correction coefficient b may be changed according to the frequency as shown by a line 62 in FIG.

The magnetic flux estimation accuracy will be described with reference to FIG. Lines 54 and 55 in FIG. 16 are the same as lines 54 (corresponding to true α-axis magnetic flux) and line 55 (corresponding to true β-axis magnetic flux) in FIGS. 10A and 10B. A line 63 represents the estimated magnetic flux ψ _2α . Line 64 represents the estimated magnetic flux ψ _2β . As represented by the line 63, an error is intentionally given to the initial value of the estimated magnetic flux ψ _2α . Specifically, the initial values of the estimated magnetic fluxes ψ _2α and ψ _2β are 0 Wb. The scale of the horizontal axis (time axis) in FIG. 16 is set so that a transient phenomenon immediately after the start of estimation is grasped. From the line 54, the line 55, the line 63, and the line 64, it can be seen that the phase delays of the estimated magnetic fluxes ψ _2α and ψ _2β are almost eliminated after one cycle has elapsed from the start of control. This means that according to the modified example 1-3, the three-phase motor 1 can be operated with high efficiency.

(Modification 1-4)
Hereinafter, the motor control device of Modification 1-4 will be described. In Modification 1-4, the same parts as those in the first embodiment may be denoted by the same reference numerals and description thereof may be omitted.

  As shown in FIG. 17, the motor control device of Modification 1-4 includes a current sensor 7 (third current sensor) in addition to the current sensor 5 and the current sensor 6. The current sensor 7 detects a current flowing through the wiring in the wiring (current path) connecting the DC power supply and the W-phase half-bridge circuit (third-phase switching unit). Specifically, the current sensor 7 is interposed in series between the bus 12 and the W-phase half-bridge circuit in the inverter 2 and measures the current flowing into and out of the W-phase half-bridge circuit. . Similar to the current sensor 6, the current sensor 7 is a non-insulated current sensor, for example, a shunt resistor.

  Further, the motor control device of Modification 1-4 includes a two-phase current estimation unit that may perform an operation different from the two-phase current estimation unit 22 in place of the two-phase current estimation unit 22 of the first embodiment. Have. Hereinafter, for the purpose of distinguishing from the two-phase current estimation unit 22, the two-phase current estimation unit of Modification 1-4 may be referred to as a two-phase current estimation unit X.

As can be understood from the description using FIG. 5A, when the command signal (voltage level) V representing the fundamental component of the AC voltage to be applied to the V phase of the three-phase motor 1 is large, the V phase switching element liable a situation that can not secure the time required for extraction of the analog signal indicative of the current i _v. Similarly, when the command signal (voltage level) W representing the fundamental component of the AC voltage to be applied to the W phase of the three-phase motor 1 is large, an analog signal indicating the W phase switching element current i _w is extracted. It is easy to invite a situation where the required time cannot be secured.

In consideration of the above, in Modification 1-4, when the maximum value of voltage level W is greater than or equal to the maximum value of voltage level V, two-phase current estimation unit X performs the same operation as two-phase current estimation unit 22. (Operation using the V-phase switching element current i _v is performed according to the situation together with the U-phase current i _u ).

If the maximum value of the voltage level W less than the maximum value of the voltage level V is 2-phase current estimation unit X is different from the 2-phase current estimation unit 22, W-phase switching in accordance with the situation with U-phase current i _u The operation using the operation using the element current i _w is performed. Specifically, when the voltage level W is high, the two-phase current estimation unit X performs the _conversion from the U-phase current i _u to the two-phase current i _αβ according to the equation (1), as in the first operation of the first embodiment. The 1st driving | operation which specifies is performed. When the voltage level W is small, the two-phase current estimation unit X specifies the two-phase current i _αβ from the U-phase current i _u and the W-phase switching element current i _w . Unlike the equation (2), the latter operation is an operation according to the equation (21), but the technical significance is common to the second operation of the first embodiment. Therefore, in this specification, the latter operation is also referred to as the second operation. Also in the modified example 1-4, whether the first operation or the second operation is performed can be determined based on the threshold amplitude V _th , the modulation rate, or the threshold frequency ω _th as in the first embodiment. it can.

  In Modification 1-4, the lower one of the voltage level V and the voltage level W is used as a criterion for determining whether to perform the first operation or the second operation. The first operation is performed when the maximum value of the voltage level used as the determination criterion is large, and the second operation is performed when the maximum value is small. Therefore, according to the modified example 1-4, the second operation can be positively performed as compared with the first embodiment.

Incidentally, in the modified example 1-4, V-phase switching element current i _v and W-phase switching element current i _w of the second operation from being determined to perform using, V-phase switching element current i _v and W-phase switching elements An analog signal indicating the current i _w is extracted, and then the V-phase switching element current i _v or the W-phase switching element current i _w is used by the two-phase current estimation unit X. However, after the analog signal indicating the V-phase switching element current i _v or the W-phase switching element current i _w that may be used in the second operation is extracted, the V-phase switching element current _iv or the W-phase switching element is extracted. It is determined whether or not the second operation using the current i _w is performed, and when it is determined that the second operation is performed, the V-phase switching element current i _v or the W-phase switching element current i _w is estimated as the two-phase current. It may be used by part X.

  In short, the power conversion control device (motor control device) of Modification 1-4 is two current sensors (current sensor 6 and current sensor 7) that can be selected as the second current sensor, and is different from each other in the switching circuit 2s. Detection of the current sensor that has two current sensors that detect current flowing into and out of the switching unit (half-bridge circuit) and that satisfies the conditions for performing the second operation among the detection values of the two current sensors The value is used as a detection value of the second current sensor.

(Modification 1-5)
Hereinafter, the motor control device of Modification 1-5 will be described. In Modification 1-5, the same portions as those in the first embodiment are denoted by the same reference numerals, and description thereof may be omitted.

In Modification 1-5, as shown in FIG. 18, a current sensor 6 b is used instead of the current sensor 6. Typically, the current sensor 6b is a shunt resistor. The current sensor 6b is provided on the bus 12 and is configured to detect the bus current i _dc . A method for estimating the three-phase current from the bus current i _dc is called a one-shunt current detection method (single shunt current detection method). For details of the single shunt current detection method, refer to Patent Document 3 and the like. By single shunt current detecting method, it is possible to generate a V-phase switching element current i _v. Even using the V-phase switching element current i _v, it is possible to perform the same control as the first embodiment.

In short, in the modified example 1-5, the second current sensor includes each of the switching circuit 2s from the DC power source among the current paths connecting the DC power source and the second-phase switching unit (V-phase half-bridge circuit). The power conversion control device (motor control device) detects the current flowing through the bus 12 that is the portion before branching to the phase switching unit, and the power conversion control device (motor control device) detects the detected value (U-phase current i _u ) of the first sensor The second operation is performed using the detected value of the current flowing through the bus 12 obtained by the second sensor and the switching pattern generated by each switching unit of the switching circuit 2s. The switching pattern indicates the ON / OFF timing of the switching element of each half bridge circuit (switching unit), and is generated based on the duty signal described above.

(Second Embodiment)
Hereinafter, a motor control device (power conversion control device) 103 according to the second embodiment will be described with reference to FIG. Note that in the second embodiment, the same parts as those in the first embodiment are denoted by the same reference numerals, and description thereof may be omitted.

  The motor control device 103 includes a current sensor 5, a current sensor 6, a two-phase current estimation unit 22, a magnetic flux estimation unit 123, a coordinate conversion unit 110, a command current calculation unit 111, a command voltage calculation unit 112, a coordinate conversion unit 113, and a phase estimation. Unit 114 and a two-phase / three-phase conversion unit 30.

(Outline of control by the motor control device 103)
Hereinafter, an outline of the operation of the motor control device 103 will be described. The magnetic flux estimation unit 123 estimates the magnetic flux from the two-phase current i _αβ and the command two-phase voltage v _αβ ^* (the estimated magnet magnetic flux ψ _mαβ is specified). Hereinafter, the α-axis component of the estimated magnet flux [psi _Emuarufabeta described as α-axis estimated magnet flux [psi _m.alpha, it may be described a β-axis component and β-axis estimated magnet flux _ψ mβ. The phase estimation unit 114 _identifies the phase θ _{d of} the estimated magnet flux ψ _mαβ from the estimated magnet flux ψ _mαβ . The coordinate transformation unit 110 identifies the two-phase current i _dq from the two-phase current i _αβ and the phase θ _d . Hereinafter, the d-axis component of the two-phase current i _dq may be referred to as a d-axis current i _d, and the q-axis component may be referred to as a q-axis current i _q . The command current calculation unit 111 identifies the command two-phase current i _dq ^* from the command torque T ^* . Hereinafter, the d-axis component of the command two-phase current i _dq ^* may be referred to as a d-axis command current i _d ^*, and the q-axis component may be referred to as a q-axis command current i _q ^* . The command voltage calculation unit 112 identifies the command two-phase voltage v _dq ^* from the command two-phase current i _dq ^* and the two-phase current i _dq . The coordinate conversion unit 113 specifies the command two-phase voltage v _αβ ^* from the command two-phase voltage v _dq ^* and the phase θ _d .

In the second embodiment, the command voltage calculation unit 112 matches the command two-phase current i _dq ^* obtained from the command torque T ^* with the current vector i _dq obtained by coordinate conversion of the two-phase current i _αβ. Thus, the command two-phase voltage v _αβ ^* is obtained. In this respect, the command voltage calculation unit 112 is different from the command voltage calculation unit 29 of the first embodiment.

  Next, components of the motor control device 103 will be described.

(Magnetic flux estimation unit 123)
The magnetic flux estimation unit 123 _identifies the estimated magnet magnetic flux ψ _mαβ (estimated magnet magnetic flux ψ _mα , ψ _mβ ) from the two-phase current i _αβ and the command two-phase voltage v _αβ ^* . In the present embodiment, the magnetic flux estimating unit 123 obtains the estimated magnet magnetic flux ψ _m by using the equations (3) and (4) described above and the following equations (22) and (23). L in the equations (22) and (23) is an armature reaction magnetic flux estimation inductance.

(Phase estimation unit 114)
The phase estimation unit 114 specifies the phase θ _d of the estimated magnet magnetic flux ψ _mαβ from the estimated magnet magnetic flux ψ _mαβ (estimated magnet magnetic flux ψ _mα , ψ _mβ ). Specifically, the phase estimation unit 114 calculates and outputs the phase θ _d according to the equation (24).

It should be noted that after obtaining the phase θ _d by the equation (24), the phase θ _d ^ is obtained by configuring a PLL (phase locked loop) that realizes the equation (25), and the phase θ _d is substituted for the phase θ _d. ^ Can also be adopted.

(Coordinate converter 110)
The coordinate conversion unit 110 converts the two-phase current i _αβ into the two-phase current i _dq using the phase θ _d . Specifically, the coordinate conversion unit 110 converts the two-phase current i _αβ into the two-phase current i _dq according to the equation (26).

(Command current calculation unit 111)
The command current calculation unit 111 generates a command two-phase current i _dq ^* from the command torque T ^* . Specifically, the command current calculation unit 111 obtains the command two-phase current i _dq ^* using Expression (27) and Expression (28).

(Command voltage calculation unit 112)
The command voltage calculation unit 112 generates a command two-phase voltage v _dq ^* from the command two-phase current i _dq ^* and the two-phase current i _dq . Specifically, the command voltage calculation unit 112 obtains the command two-phase voltage v _dq ^* using Equation (29) and Equation (30). In Equations (29) and (30), K _pi is a proportional gain of the current control PI control, and K _ii is an integral gain of the current control PI control.

As can be understood from the description of the phase estimation unit 114, the coordinate conversion unit 110, the command current calculation unit 111, the command voltage calculation unit 112, and the like, in the second embodiment, the rotation magnetic flux vector is a rotor magnetic flux vector (magnet magnetic flux). ), and the power conversion controlling unit 103, a phase estimator 114 for estimating the phase theta _d of the estimated rotor flux vector (estimated magnet flux [psi _{Emuarufabeta),} the command current vector (command using the phase theta _d The coordinate system of the two-phase current i _dq ^* ) and the estimated current vector (two-phase current i _αβ ) is unified, and the estimated current vector (two-phase current i _dq ) and the command current vector (command 2 The three-phase motor 1 is controlled using the phase current i _dq ^* ) so that the current vector follows the command current vector (command two-phase current i _dq ^* ).

(Coordinate converter 113)
The coordinate conversion unit 113 converts the command two-phase voltage v _dq ^* into the command two-phase voltage v _αβ ^* using the phase θ _d . Specifically, the coordinate conversion unit 113 converts the command two-phase voltage v _dq ^* into the command two-phase voltage v _αβ ^* according to the equation (31).

(Third embodiment)
Hereinafter, the power conversion control apparatus of 3rd Embodiment is demonstrated, referring FIG. Note that in the third embodiment, portions similar to those in the first embodiment or the second embodiment are denoted by the same reference numerals, and description thereof may be omitted.

  In the third embodiment, the power conversion control device is a grid interconnection inverter control device 203, the power conversion device is an inverter (grid interconnection inverter) 2, and the AC load is a three-phase power grid 201. The grid interconnection inverter control device 203 and the inverter 2 constitute a power conversion unit. The DC power source, the power conversion unit, and the AC load constitute a power system. As shown in FIG. 20, the grid interconnection inverter control device 203 can be connected to the inverter 2 and the three-phase power system 201.

  The grid interconnection inverter control device 203 includes a current sensor 5, a current sensor 6, a voltage sensor 206, a two-phase current estimation unit 22, a three-phase two-phase conversion unit (three-phase two-phase coordinate conversion unit) 223, a coordinate conversion unit 210, A command voltage calculation unit 212, a coordinate conversion unit 213, a phase estimation unit 214, an addition unit 224, and a two-phase / three-phase conversion unit 30 are provided.

(Outline of control by the power conversion control device 203)
Hereinafter, an outline of the operation of the power conversion control device 203 will be described. The voltage sensor 206 detects a line voltage between the U phase and the V phase of the three-phase power system 201 and a line voltage between the V phase and the W phase, and detects the detected values (line voltage v _uv and A line voltage v _wv ) is generated. The three-phase two-phase converter 223 identifies the two-phase voltage v _αβ from the line voltage v _uv and the line voltage v _wv . The phase estimation unit 214 identifies the phase θ _{d of} the two-phase voltage v _αβ from the two-phase voltage v _αβ . The coordinate converter 210 identifies the two-phase current i _dq from the two-phase current i _αβ and the phase θ _d . The command voltage calculation unit 212 identifies the command voltage deviation v _dq ^* from the command two-phase current i _dq ^* and the two-phase current i _dq . The command two-phase current i _dq ^* is given from the _host controller. The coordinate conversion unit 213 identifies the command voltage deviation v _αβ ^* from the command voltage deviation v _dq ^* and the phase θ _d . The adder 224 identifies the command two-phase voltage v _αβ ^** from the command voltage deviation v _αβ ^* and the two-phase voltage v _αβ . The two-phase / three-phase converter 30 identifies the command three-phase voltage v _uvw ^* from the command two-phase voltage v _αβ ^** . By such control, the current vector output from the inverter 2 to the three-phase power system 201 is controlled to follow the command current vector (command two-phase current i _dq ^* ).

In the third embodiment, the phase θ _d is obtained using the two-phase voltage V _αβ . Command two-phase current i _dq ^* is given from the _host controller. The command two-phase voltage v _αβ ^** is specified by adding the two-phase voltage v _αβ to the command voltage deviation v _αβ ^* . In these points, the third embodiment is different from the second embodiment.

  Next, components of the grid interconnection inverter control device 203 will be described.

(Voltage sensor 206)
The voltage sensor 206 detects the line voltage between the U phase and the V phase of the three-phase power system 201 and the line voltage between the V phase and the W phase, and detects the detected values (line voltage v _uv and A line voltage v _wv ) is generated.

(Three-phase to two-phase converter 223)
The three-phase two-phase converter 223 converts the line voltage v _uv and the line voltage v _wv into a two-phase voltage v _αβ . Specifically, the three-phase to two-phase conversion unit 223 _obtains a two-phase voltage v _αβ (v _α , v _β ) by performing three-phase to two-phase conversion on the line voltages v _uv and v _wv using the equation (32). ,Output.

(Phase estimation unit 214)
The phase estimation unit 214 identifies the phase θ _d of the two-phase voltage v _αβ from the two-phase voltage v _αβ . Specifically, the phase estimation unit 214 obtains the phase θ _d according to the equation (33).

Incidentally, after obtaining a phase theta _d in equation (33), we obtain the phase theta _d ^ by configuring the PLL for realizing equation (34), adopting a phase theta _d ^ instead of phase theta _d You can also.

(Coordinate converter 210)
The coordinate conversion unit 210 converts the two-phase current i _αβ into the two-phase current i _dq using the phase θ _d . For this conversion, the same conversion formula as Expression (26) is used.

(Command voltage calculation unit 212)
The command voltage calculation unit 212 identifies the command voltage deviation v _dq ^* from the command two-phase current i _dq ^* and the two-phase current i _dq . Specifically, the command voltage calculation unit 212 specifies the command voltage deviation v _dq ^* according to the equations (35) and (36).

(Coordinate conversion unit 213)
The coordinate conversion unit 213 converts the command voltage deviation v _dq ^* into the command voltage deviation v _αβ ^* using the phase θ _d . For this conversion, the same conversion equation as equation (31) is used.

(Adder 224)
The adder 224 identifies the command two-phase voltage v _αβ ^** from the command voltage deviation v _αβ ^* and the two-phase voltage v _αβ . In the third embodiment, unlike the second embodiment, the command two-phase voltage v _αβ ^** is input to the two-phase three-phase converter 30 and converted into the command three-phase voltage v _uvw ^* .

DESCRIPTION OF SYMBOLS 1 3 phase motor 1m Permanent magnet 2 Inverter 2s Switching circuit 3,103,203 Power conversion control device 4 Smoothing capacitor 5,6,6b, 7 Current sensor 8u, 8v, 8w, 9u, 9v, 9w Switching element 10u, 10v, 10w, 11u, 11v, 11w Diode 12 Bus 13 Carrier signal V, 15, 16 Voltage level 22 Two-phase current estimator 23 Magnetic flux / torque estimator 23a, 40, 41, 73a, 80, 81, 83a, 90, 91, 93a, 123 Magnetic flux estimation unit 23b Torque estimation unit 24, 45, 85, 95 Subtraction unit (subtractor)
25 Command amplitude calculation unit 26 Command magnetic flux calculation unit 29, 112, 212 Command voltage calculation unit 30 2 phase 3 phase conversion unit (2 phase 3 phase coordinate conversion unit)
31 Overcurrent detection unit 31a Overcurrent detection circuit 31b Overcurrent determination unit 36, 44, 84, 94, 224 Adder (adder)
37 vector generation unit 38 PI control unit 42, 43, 70, 71, 82, 83, 92, 93, 98, 99 block 46, 47 induced voltage estimation unit 50 to 64 line 110, 113, 210, 213 coordinate conversion unit 111 Command current calculation unit 114, 214 Phase estimation unit 201 Three-phase power system 206 Voltage sensor 223 Three-phase two-phase conversion unit (three-phase two-phase coordinate conversion unit)

Claims (20)

The DC power source is converted into the first phase of the AC load so that the DC power output from the DC power source is converted into AC power to be applied to an AC load having different first and second phases. A power conversion control device for controlling a switching circuit having a first phase switching unit electrically connected and a second phase switching unit electrically connecting the DC power source to the second phase of the AC load. And
A first current sensor that detects a current flowing through the secondary current path in a secondary current path that connects the first phase switching unit and the first phase of the AC load;
A second current sensor for detecting a current flowing through the primary-side current path in a primary-side current path connecting the DC power source and the second-phase switching unit;
A power conversion control device.   2. The power conversion control device according to claim 1, comprising only the first current sensor as a current sensor that detects a current flowing through a secondary current path existing on the AC load side of each switching unit of the switching circuit. . The first current sensor is a current transformer;
The power conversion control device according to claim 1, wherein the second current sensor is a shunt resistor.   2. An overcurrent detection unit that is provided outside an inverter including the switching circuit and that determines whether or not an overcurrent has occurred by receiving an analog signal indicating a detection value of the second current sensor. The power conversion control device according to any one of? The overcurrent detector is
Constructed by hardware, the analog signal indicating the detection value of the second current sensor is input, and the analog signal indicating the detection value of the second current sensor larger than the current threshold is input over a certain period. An overcurrent detection circuit that continuously outputs an analog signal indicating that an overcurrent has been detected;
The power conversion control device according to claim 4, further comprising: an overcurrent determination unit that determines whether an overcurrent has occurred using the analog signal output from the overcurrent detection circuit. The power conversion control device includes:
When the amplitude of the fundamental component of the AC voltage to be applied to the second phase of the AC load is greater than or equal to a threshold amplitude, the detection value of the first current sensor is used instead of the detection value of the second current sensor. Performing a first operation for specifying a current vector applied to the AC load,
When the amplitude of the fundamental component of the AC voltage to be applied to the second phase of the AC load is less than the threshold amplitude, both the detection value of the first sensor and the detection value of the second sensor are used. The power conversion control device according to claim 1, wherein a second operation for specifying a current vector applied to the AC load is performed. The power conversion control device includes:
By controlling the switching circuit in a PWM manner by comparing a carrier signal and a command signal representing a fundamental wave component of an AC voltage to be applied to the AC load,
For the second phase of the AC load, when a modulation rate that is a ratio of the amplitude of the command signal to the amplitude of the carrier signal is equal to or greater than a threshold modulation rate, the detection value of the second current sensor is not used. Performing a first operation for specifying a current vector applied to the AC load using a detection value of one current sensor;
For the second phase of the AC load, when the modulation rate is less than the threshold modulation rate, both the detection value of the first sensor and the detection value of the second sensor are applied to the AC load. Second operation to identify the current vector
The power conversion control device according to claim 1, wherein the threshold modulation rate is in a range of 0.5 to 0.9. The power conversion control device includes:
When the frequency of the fundamental component of the AC power to be applied to the second phase of the AC load is equal to or higher than a threshold frequency, the detection value of the first current sensor is used instead of the detection value of the second current sensor. Performing a first operation for specifying a current vector applied to the AC load,
When the frequency of the fundamental component of the AC power to be applied to the second phase of the AC load is less than the threshold frequency, both the detection value of the first sensor and the detection value of the second sensor are used. The power conversion control device according to claim 1, wherein a second operation for specifying a current vector applied to the AC load is performed.   The power conversion control device according to claim 8, wherein the threshold frequency is in a range of 500 to 2500 Hz. The power conversion control device controls the switching circuit by a PWM method by comparing a carrier signal and a command signal representing a fundamental wave component of an AC voltage to be applied to the AC load,
The power conversion control device according to claim 1, wherein the frequency of the carrier signal is 20 to 50 kHz. The current vector applied to the AC load is identified by cooperating with the first current sensor without cooperating with the second current sensor, or cooperating with the first sensor and the second sensor. A two-phase current estimator,
A magnetic flux estimator for estimating a rotating magnetic flux vector applied to the AC load using the estimated current vector and a command voltage vector representing an AC voltage to be applied to the AC load. Item 11. The power conversion control device according to any one of Items 1 to 10. The magnetic flux estimator is
An α-axis induced voltage estimator that estimates the α-axis component of the induced voltage generated in the AC load using the estimated α-axis component of the current vector and the α-axis component of the command voltage vector;
A β-axis induced voltage estimator that estimates the β-axis component of the induced voltage generated in the AC load using the estimated β-axis component of the current vector and the β-axis component of the command voltage vector;
An α-axis magnetic flux estimator that temporarily estimates the α-axis component of the rotating magnetic flux vector using the estimated α-axis component of the induced voltage;
A β-axis magnetic flux estimator that temporarily estimates the β-axis component of the rotating magnetic flux vector using the estimated β-axis component of the induced voltage;
An α-axis component correction unit that corrects the estimated α-axis component of the rotating magnetic flux vector using the estimated β-axis component of the induced voltage or the temporarily estimated β-axis component of the rotating magnetic flux vector;
A β-axis component correction unit that corrects the estimated β-axis component of the rotating magnetic flux vector using the estimated α-axis component of the induced voltage or the temporarily estimated α-axis component of the rotating magnetic flux vector; The power conversion control device according to claim 11. The magnetic flux estimator is
An α-axis induced voltage estimator that temporarily estimates the α-axis component of the induced voltage generated in the AC load using the estimated α-axis component of the current vector and the α-axis component of the command voltage vector;
A β-axis induced voltage estimator that temporarily estimates the β-axis component of the induced voltage generated in the AC load using the estimated β-axis component of the current vector and the β-axis component of the command voltage vector;
An α-axis component correcting unit that corrects the estimated β-axis component of the induced voltage using the estimated β-axis component of the induced voltage or the estimated β-axis component of the rotating magnetic flux vector;
A β-axis component correction unit that corrects the estimated β-axis component of the induced voltage using the estimated α-axis component of the induced voltage or the estimated α-axis component of the rotating magnetic flux vector;
An α-axis magnetic flux estimation unit that estimates the α-axis component of the rotating magnetic flux vector using the corrected α-axis component of the induced voltage;
The power conversion control device according to claim 11, further comprising a β-axis magnetic flux estimation unit that estimates a β-axis component of the rotating magnetic flux vector using the corrected β-axis component of the induced voltage. The power conversion control device operates as a motor control device that controls a three-phase motor as the AC load,
The power conversion control device according to any one of claims 11 to 13, wherein the rotating magnetic flux vector is an interlinkage magnetic flux vector or a rotor magnetic flux vector of the three-phase motor. The rotating magnetic flux vector is the flux linkage vector,
The power conversion control device according to claim 14, wherein the power conversion control device controls the three-phase motor so that the flux linkage vector follows a command magnetic flux vector. The power conversion control device includes:
A torque estimating unit for estimating a motor torque applied to the three-phase motor using the estimated current vector and the estimated flux linkage vector;
The power conversion control device according to claim 15, wherein the command magnetic flux vector is specified so that the motor torque follows the command torque. The rotating magnetic flux vector is the rotor magnetic flux vector;
The power conversion control device includes:
A phase estimation unit for estimating the phase of the estimated rotor magnetic flux vector;
The coordinate system of the command current vector and the estimated current vector is unified using the estimated phase, and the current vector is converted into the command using the estimated current vector and the command current vector after the unification. The power conversion control device according to claim 14, wherein the three-phase motor is controlled so as to follow a current vector.   A power conversion unit comprising the power conversion control device and the switching circuit according to claim 1.   The power conversion unit according to claim 18, wherein a semiconductor element containing silicon carbide or gallium nitride is used for the switching circuit.   An electric power system comprising the DC power supply according to claim 18, a power conversion unit, and an AC load.

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