JP2016092848A - Power converter and power conditioner using the same - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a power converter in which occurrence of discontinuity of an output can be suppressed, and an air conditioner using the same.SOLUTION: A controller 6 switches plural modes which are different in connection state of a DC power source 100 and a capacitor to a first output point 103 and a second output point 104 by controlling plural switches. The plural modes are classified into a maintenance mode in which no current flows in the capacitor, a charging mode in which current flows in the capacitor, and a discharging mode in which current flows in the capacitor in the opposite direction to the charging mode. The controller 6 switches the maintenance mode, the charging mode and the discharging mode so that the voltage of the capacitor is kept to a target voltage obtained by adding a prescribed value to the half magnitude of the voltage of the DC power source 100.SELECTED DRAWING: Figure 1

Description

  The present invention generally relates to a power converter and a power conditioner using the same, and more particularly to a power converter that converts power from a DC power source and a power conditioner using the same.

  In recent years, with the spread of residential solar power generation devices, fuel cells, power storage devices, and the like, various circuits have been proposed and provided as power conversion devices that convert the output of these DC power sources into AC. For example, Patent Documents 1 and 2 disclose a power converter that generates an AC output converted from a DC voltage source into a plurality of voltage levels (“Multi-level power converter” in Patent Document 1 and “Converter Circuit” in Patent Document 2). Is disclosed.

  According to the description of Patent Document 1, the power conversion device is a 5-level inverter that outputs a 5-level voltage, and includes two DC capacitors, two flying capacitors, and 10 switching elements. ing. In this power converter, in the state where the DC voltage E is applied to the series circuit of two DC capacitors, the voltage of each DC capacitor becomes E / 2 and the voltage of each flying capacitor becomes E / 4. By controlling the switching element, a five-level voltage is output.

JP 2014-64431 A (paragraphs [0002] to [0006], FIGS. 16 and 17) Japanese Patent No. 4369425

  By the way, the power conversion device having the above-described configuration can realize a sinusoidal AC output by switching a plurality of modes having different output voltages with a PWM (Pulse Width Modulation) signal. However, since there is generally an upper limit value and a lower limit value of the duty ratio in PWM control, even in a power converter, it is difficult to make the duty ratio of a specific mode as close as possible to 100% or 0%. is there. Therefore, for example, the power conversion device switches the mode in which the output voltage is E / 4 and the mode E / 2 from the state in which the mode in which the output voltage is zero (0) and the mode in E / 4 are switched by PWM control. When moving to, output discontinuity may occur.

  This invention is made | formed in view of the said reason, and it aims at providing the power converter device which can suppress generation | occurrence | production of the discontinuity of an output, and a power conditioner using the same.

  The power conversion device of the present invention includes a plurality of switches and capacitors, and is electrically connected between a first input point on the high potential side of the DC power supply and a second input point on the low potential side of the DC power supply. By controlling the plurality of switches and the plurality of switches, the plurality of modes in which the DC power supply and the capacitor are connected to the first output point and the second output point are switched, and the first output point and A control unit that changes the magnitude of the output voltage generated between the second output point and the second output point in a plurality of stages, wherein the plurality of modes have no current flowing through the capacitor and the absolute value of the output voltage is zero or The sustain mode is the same as the voltage of the DC power supply, and the absolute value of the output voltage is reduced by subtracting the voltage of the capacitor from the voltage of the DC power supply by passing a current through the capacitor. The charging mode and the discharging mode in which the absolute value of the output voltage is the same as the voltage of the capacitor by flowing a current in the opposite direction to the charging mode into the capacitor, and the control unit In order to maintain the voltage of the capacitor at a target voltage obtained by adding a specified value having a smaller absolute value than the voltage to a half of the voltage of the DC power supply, the sustain mode, the charge mode or The discharge mode is switched.

  A power conditioner according to the present invention includes the power conversion device described above and a disconnector electrically connected between the first output point, the second output point, and a system power supply. .

  The present invention has an advantage that occurrence of output discontinuity can be suppressed.

It is a circuit diagram which shows the structure of the power converter device which concerns on Embodiment 1. FIG. 2A is an explanatory diagram of a first mode of the power conversion device according to the first embodiment, and FIG. 2B is an explanatory diagram of a second mode of the power conversion device according to the first embodiment. 3A is an explanatory diagram of a third mode of the power conversion device according to the first embodiment, and FIG. 3B is an explanatory diagram of a fourth mode of the power conversion device according to the first embodiment. 4A is an explanatory diagram of a fifth mode of the power conversion device according to the first embodiment, and FIG. 4B is an explanatory diagram of a sixth mode of the power conversion device according to the first embodiment. 5A is an explanatory diagram of a seventh mode of the power conversion device according to the first embodiment, and FIG. 5B is an explanatory diagram of an eighth mode of the power conversion device according to the first embodiment. It is a wave form diagram of the output voltage of the power converter device which concerns on Embodiment 1. FIG. It is the schematic which shows the structure of the power conditioner which concerns on Embodiment 1. FIG. 3 is a flowchart illustrating an operation of the power conversion apparatus according to the first embodiment. FIG. 4 is a waveform diagram showing a relationship between a signal wave and a reference wave in a reference example of the first embodiment. FIG. 3 is a waveform diagram illustrating a relationship between a signal wave and a reference wave in the first embodiment. 3 is a timing chart illustrating an operation of the power conversion apparatus according to the first embodiment. 6 is a timing chart illustrating an operation of the power conversion device according to the comparative example of the first embodiment. FIG. 6 is a waveform diagram showing a relationship between a signal wave and a reference wave in the second embodiment. 6 is a timing chart illustrating an operation of the power conversion apparatus according to the second embodiment.

(Embodiment 1)
As shown in FIG. 1, the power conversion device 1 according to the present embodiment includes a conversion circuit 10 and a control unit 6.

  The conversion circuit 10 includes a plurality of switches and capacitors. The conversion circuit 10 is electrically connected between a first input point 101 on the high potential side of the DC power supply 100 and a second input point 102 on the low potential side of the DC power supply 100. In the example of FIG. 1, the plurality of switches include first to eighth switching elements Q1 to Q8, a first bidirectional switch 13, and a second bidirectional switch 14. In the example of FIG. 1, the capacitor is composed of a first capacitor C1 and a second capacitor C2.

  The control unit 6 switches the plurality of modes in which the DC power supply 100 and the capacitor are connected to the first output point 103 and the second output point 104 by controlling the plurality of switches. Thereby, the control unit 6 changes the magnitude of the output voltage V20 generated between the first output point 103 and the second output point 104 in a plurality of stages.

  Here, the plurality of modes are classified into a sustain mode in which no current flows through the capacitor, a charge mode in which current flows through the capacitor, and a discharge mode in which current opposite to the charge mode flows through the capacitor. The In the sustain mode, the absolute value of the output voltage V20 is zero or the same as the voltage of the DC power supply 100. In the charging mode, the absolute value of the output voltage V20 has a magnitude obtained by subtracting the capacitor voltage from the voltage of the DC power supply 100. In the discharge mode, the absolute value of the output voltage V20 is the same as the voltage of the capacitor.

  The capacitor voltage here is the sum of the voltage V1 of the first capacitor C1 and the voltage V2 of the second capacitor C2. The charging mode is a mode in which the capacitor is charged by flowing a current through the capacitor, and the discharging mode is to discharge the capacitor by flowing a current in the opposite direction to the charging mode into the capacitor. Mode.

  The control unit 6 maintains the voltage of the capacitor at a target voltage obtained by adding a specified value having a smaller absolute value than the voltage of the capacitor to a half of the voltage of the DC power supply 100. The maintenance mode is switched between the charging mode and the discharging mode.

  According to this configuration, the power conversion device 1 maintains the voltage of the capacitor at a target voltage obtained by adding a specified value whose absolute value is smaller than the voltage of the capacitor to half the voltage of the DC power supply 100. To do. That is, in this power conversion device 1, the voltage of the capacitor is not maintained at a half of the voltage of the DC power supply 100, but is slightly larger than the half of the voltage of the DC power supply 100 (specified). The target voltage is maintained at a high (or low) target voltage. Therefore, when the power conversion device 1 realizes a sinusoidal AC output by switching a plurality of modes with a PWM signal, even if there is an upper limit value and a lower limit value of the duty ratio, output discontinuity occurs. There is an advantage that it can be suppressed. The detailed reason for suppressing the occurrence of output discontinuity will be described later.

  Specifically, in the present embodiment, as shown in FIG. 1, the conversion circuit 10 includes a first conversion circuit 11 and a second conversion circuit 12, a first bidirectional switch 13, and a second bidirectional switch. 14. The first conversion circuit 11 and the second conversion circuit 12 are electrically connected in parallel between the first input point 101 and the second input point 102. The first bidirectional switch 13 and the second bidirectional switch 14 are electrically connected between the first conversion circuit 11 and the second conversion circuit 12.

  The first conversion circuit 11 includes first to fourth switching elements Q1 to Q4 and a first capacitor C1. Here, the first conversion circuit 11 sets a connection point between the second switching element Q <b> 2 and the third switching element Q <b> 3 as the first output point 103.

  The first to fourth switching elements Q1 to Q4 are electrically connected in series between the first input point 101 and the second input point 102. The first to fourth switching elements Q1 to Q4 are in the order of the first switching element Q1, the second switching element Q2, the third switching element Q3, and the fourth switching element Q4 from the first input point 101 side. Connected in series. The first capacitor C1 is electrically connected in parallel with the series circuit of the second switching element Q2 and the third switching element Q3.

  The second conversion circuit 12 includes fifth to eighth switching elements Q5 to Q8 and a second capacitor C2. Here, the second conversion circuit 12 sets a connection point between the sixth switching element Q6 and the seventh switching element Q7 as the second output point 104.

  The fifth to eighth switching elements Q5 to Q8 are electrically connected in series between the first input point 101 and the second input point 102. The fifth to eighth switching elements Q5 to Q8 are in the order of the fifth switching element Q5, the sixth switching element Q6, the seventh switching element Q7, and the eighth switching element Q8 from the first input point 101 side. Connected in series. The second capacitor C2 is electrically connected in parallel with the series circuit of the sixth switching element Q6 and the seventh switching element Q7.

  The first bidirectional switch 13 is a connection point between the first connection point 201, which is a connection point between the first switching element Q1 and the second switching element Q2, and a connection point between the seventh switching element Q7 and the eighth switching element Q8. It is electrically connected to a certain second connection point 202.

  The second bidirectional switch 14 is a connection point between the third connection element 203, which is a connection point between the third switching element Q3 and the fourth switching element Q4, and a connection point between the fifth switching element Q5 and the sixth switching element Q6. It is electrically connected to a certain fourth connection point 204.

  As described above, the first to eighth switching elements Q1 to Q8, the first bidirectional switch 13 and the second bidirectional switch 14 constitute the plurality of switches, and the first capacitor C1 and the second switch The capacitor C2 constitutes the capacitor. Further, the controller 6 maintains the sum of the voltage V1 of the first capacitor C1 and the voltage V2 of the second capacitor C2 (capacitor voltage) at the target voltage.

  Hereinafter, the power converter 1 according to the present embodiment and the power conditioner 20 (see FIG. 7) using the power converter 1 will be described in detail. However, the configuration described below is only an example of the present invention, and the present invention is not limited to the following embodiment, and the technical idea according to the present invention is not deviated from this embodiment. Various changes can be made in accordance with the design or the like as long as they are not.

  In the present embodiment, the case where the power conditioner 20 is a residential power conditioner that is used by being electrically connected to a photovoltaic power generation device serving as the DC power source 100 is exemplified. However, the use of the power conditioner 20 is illustrated. It is not intended to limit. The power conditioner 20 may be used by being electrically connected to a DC power source 100 other than a solar power generation device, such as a household fuel cell or a power storage device, or a non-residential such as a store, factory, office, etc. May be used. Furthermore, the power converter 1 is not intended to limit the application to the power conditioner 20, and the power converter 1 may be used other than the power conditioner 20.

<Configuration of power converter>
The power converter 1 of this embodiment is electrically connected to the DC power supply 100 as shown in FIG. Here, since the DC power source 100 is composed of a solar power generation device, the power conversion device 1 is connected to the DC power source 100 via a connection box.

  The power conversion device 1 according to the present embodiment includes a conversion circuit (first conversion circuit 11, second conversion circuit 12, first bidirectional switch 13, and second bidirectional switch 14) 10 and a control unit 6. The generator 3, the filter circuit 5, the first detector 21, and the second detector 22 are further provided.

  The first input point 101 and the second input point 102 serve as a pair of input terminals in the power converter 1, and the DC power source 100 is electrically connected between the pair of input terminals (the first input point 101 and the second input point 102). Connected.

  The first output point 103 of the first conversion circuit 11 and the second output point 104 of the second conversion circuit 12 are electrically connected to the third output point 105 and the fourth output point 106 through the filter circuit 5, respectively. Has been. In the present embodiment, the third output point 105 and the fourth output point 106 serve as a pair of output terminals in the power conversion device 1. Hereinafter, the first output voltage V10 generated between the third output point 105 and the fourth output point 106 is also simply referred to as “output voltage V10”.

  In the present embodiment, the output voltage V <b> 10 of the power conversion device 1 is an AC voltage, and the third output point 105 and the fourth output point 106 are electrically connected to the system power supply (commercial power system) 7. Further, the third output point 105 and the fourth output point 106 are electrically connected to a load 8 that operates by receiving supply of AC power.

  Specifically, the pair of output terminals of the power conversion device 1 is connected to the load 8 and the system power supply 7 by being electrically connected to an interconnection breaker provided in the distribution board. That is, the power conversion device 1 converts the DC power input from the DC power source 100 into AC power, and the AC power is transferred from the pair of output terminals (the third output point 105 and the fourth output point 106) to the load 8 and the system. Output to power supply 7. In FIG. 1, the system power supply 7 is a single-phase three-wire system having a U-phase and a W-phase. However, the system power supply 7 is not limited to this example, and may be a single-phase two-wire system.

  Next, the structure of each part of the power converter device 1 will be described in detail.

  The power conversion apparatus 1 uses the input terminal on the high potential (positive electrode) side of the DC power supply 100 among the pair of input terminals connected to the DC power supply 100 as the first input point 101, and the low potential (negative electrode) of the DC power supply 100. The input terminal on the side is the second input point 102. Therefore, a DC voltage output from the DC power supply 100 is applied as an input voltage between the first input point 101 and the second input point 102.

  Here, it is assumed that the input terminal (second input point 102) on the low potential side of the DC power supply 100 is the circuit ground of the power converter 1, and the potential thereof is 0 [V]. Then, the potential of the first input point 101 is expressed by E [V] using the DC voltage E [V] output from the DC power supply 100.

  As described above, the first conversion circuit 11 includes the first to fourth switching elements Q1 to Q4 connected in series between the first input point 101 and the second input point 102, and the first capacitor C1. doing. For each of the first to fourth switching elements Q1 to Q4, a depletion type n-channel MOSFET (Metal-Oxide-Semiconductor Field Effect Transistor) is used here as an example.

  The drain of the first switching element Q1 is electrically connected to the first input point 101. The drain of the second switching element Q2 is electrically connected to the source of the first switching element Q1. The drain of the third switching element Q3 is electrically connected to the source of the second switching element Q2. The drain of the fourth switching element Q4 is electrically connected to the source of the third switching element Q3. Further, the source of the fourth switching element Q 4 is electrically connected to the second input point 102.

  Here, the source of the second switching element Q 2 (the drain of the third switching element Q 3) is the first output point 103. Furthermore, the source of the first switching element Q1 (the drain of the second switching element Q2) is the first connection point 201, and the source of the third switching element Q3 (the drain of the fourth switching element Q4) is the third connection. A point 203 is obtained.

  One end of the first capacitor C1 is electrically connected to the drain (first connection point 201) of the second switching element Q2, and the other end is electrically connected to the source (third connection point 203) of the third switching element Q3. Connected. In other words, the first capacitor C1 has one end electrically connected to the first input point 101 via the first switching element Q1 and the other end connected to the second input point 102 via the fourth switching element Q4. Electrically connected.

  As described above, the second conversion circuit 12 includes the fifth to eighth switching elements Q5 to Q8 connected in series between the first input point 101 and the second input point 102, and the second capacitor C2. doing. Here, the second conversion circuit 12 has basically the same configuration as the first conversion circuit 11, and the fifth to eighth switching elements Q5 to Q8 correspond to the first to fourth switching elements Q1 to Q4. The second capacitor C2 corresponds to the first capacitor C1. Here, depletion type n-channel MOSFETs are used for the fifth to eighth switching elements Q5 to Q8, similarly to the first to fourth switching elements Q1 to Q4.

  That is, the drain of the fifth switching element Q5 is electrically connected to the first input point 101. The drain of the sixth switching element Q6 is electrically connected to the source of the fifth switching element Q5. The drain of the seventh switching element Q7 is electrically connected to the source of the sixth switching element Q6. The drain of the eighth switching element Q8 is electrically connected to the source of the seventh switching element Q7. Further, the source of the eighth switching element Q8 is electrically connected to the second input point 102.

  Here, the source of the sixth switching element Q 6 (the drain of the seventh switching element Q 7) is the second output point 104. Further, the source of the fifth switching element Q5 (the drain of the sixth switching element Q6) is the fourth connection point 204, and the source of the seventh switching element Q7 (the drain of the eighth switching element Q8) is the second connection point. A point 202 is obtained.

  The second capacitor C2 has one end electrically connected to the drain (fourth connection point 204) of the sixth switching element Q6 and the other end electrically connected to the source (second connection point 202) of the seventh switching element Q7. Connected. In other words, the second capacitor C2 has one end electrically connected to the first input point 101 via the fifth switching element Q5 and the other end connected to the second input point 102 via the eighth switching element Q8. Electrically connected.

  The circuit constant (capacitance) of the second capacitor C2 is equal to the circuit constant (capacitance) of the first capacitor C1.

  In FIG. 1, first to eighth diodes D1 to D8 are connected in antiparallel to the first to eighth switching elements Q1 to Q8, respectively. The first to eighth diodes D1 to D8 are parasitic diodes of the first to eighth switching elements Q1 to Q8, respectively. That is, the parasitic diode of the first switching element Q1 constitutes the first diode D1, and similarly, the parasitic diodes of the second, third,... Switching elements Q2, Q3,. , D3. For example, the first diode D1 is connected in a direction in which the drain side of the first switching element Q1 is a cathode and the source side is an anode.

  The first conversion circuit 11 and the second conversion circuit 12 configured as described above are electrically connected in parallel between the first input point 101 and the second input point 102. That is, the first conversion circuit 11 and the second conversion circuit 12 are connected in parallel between both ends of the DC power supply 100.

  The first bidirectional switch 13 is electrically connected between the first connection point 201 and the second connection point 202. That is, the first connection point 201 of the first conversion circuit 11 is electrically connected to the second connection point 202 of the second conversion circuit 12 via the first bidirectional switch 13. Here, the first bidirectional switch 13 includes a ninth switching element Q9 and a tenth switching element Q10 electrically connected in series between the first connection point 201 and the second connection point 202. have. The first bidirectional switch 13 is connected in the order of the ninth switching element Q9 and the tenth switching element Q10 from the first connection point 201 side.

  More specifically, a depletion type n-channel MOSFET is used for each of the ninth and tenth switching elements Q9 and Q10, similarly to each of the first to eighth switching elements Q1 to Q8. The source of the ninth switching element Q9 is connected to the first connection point 201, and the drain of the ninth switching element Q9 is connected to the drain of the tenth switching element Q10. The source of the tenth switching element Q <b> 10 is connected to the second connection point 202. In short, the ninth switching element Q9 and the tenth switching element Q10 are connected in anti-series between the first connection point 201 and the second connection point 202 so that the drains are connected to each other. .

  The second bidirectional switch 14 is electrically connected between the third connection point 203 and the fourth connection point 204. That is, the third connection point 203 of the first conversion circuit 11 is electrically connected to the fourth connection point 204 of the second conversion circuit 12 via the second bidirectional switch 14. Here, the second bidirectional switch 14 includes the twelfth switching element Q12 and the eleventh switching element Q11 electrically connected in series between the third connection point 203 and the fourth connection point 204. have. The second bidirectional switch 14 is connected in the order of the twelfth switching element Q12 and the eleventh switching element Q11 from the third connection point 203 side.

  More specifically, a depletion type n-channel MOSFET is used for each of the eleventh and twelfth switching elements Q11 and Q12, similarly to each of the first to eighth switching elements Q1 to Q8. The source of the eleventh switching element Q11 is connected to the fourth connection point 204, and the drain of the eleventh switching element Q11 is connected to the drain of the twelfth switching element Q12. The source of the twelfth switching element Q12 is connected to the third connection point 203. In short, the eleventh switching element Q11 and the twelfth switching element Q12 are connected in anti-series between the third connection point 203 and the fourth connection point 204 so that the drains are connected to each other. .

  Also, ninth to twelfth diodes D9 to D12 are connected in antiparallel to the ninth to twelfth switching elements Q9 to Q12, respectively. The ninth to twelfth diodes D9 to D12 are parasitic diodes of the ninth to twelfth switching elements Q9 to Q12, respectively. That is, the parasitic diode of the ninth switching element Q9 constitutes the ninth diode D9, and similarly, the parasitic diodes of the tenth, eleventh and twelfth switching elements Q10, Q11 and Q12 are the tenth, eleventh and twelfth elements, respectively. Diodes D10, D11, D12. For example, the ninth diode D9 is connected in such a direction that the drain side of the ninth switching element Q9 is a cathode and the source side is an anode.

  In the present embodiment, the first bidirectional switch 13 is configured to be able to switch between operation states including an all-off state and an all-on state. The all-off state of the first bidirectional switch 13 is a state in which a bidirectional current is interrupted between the first connection point 201 and the second connection point 202. The all-on state of the first bidirectional switch 13 is a state in which bidirectional current passes between the first connection point 201 and the second connection point 202. Similarly, the second bidirectional switch 14 is configured to be able to switch the operation state including the all-off state and the all-on state. The all-off state of the second bidirectional switch 14 is a state in which bidirectional current is interrupted between the third connection point 203 and the fourth connection point 204. The all-on state of the second bidirectional switch 14 is a state in which bidirectional current passes between the third connection point 203 and the fourth connection point 204.

  Furthermore, in the present embodiment, the operation state of the first bidirectional switch 13 is such that the current flowing from the second connection point 202 to the first connection point 201 is cut off, and the first connection point 201 to the second connection point 202. It further includes a half-on state for passing a flowing current. Further, the operating state of the second bidirectional switch 14 blocks the current flowing from the third connection point 203 to the fourth connection point 204 and allows the current flowing from the fourth connection point 204 to the third connection point 203 to pass. It further includes a semi-on state.

  Therefore, in the power conversion device 1 of the present embodiment, bidirectional current can pass between the first connection point 201 and the second connection point 202 by setting the first bidirectional switch 13 to the all-on state. Can create a unique state. Moreover, the power converter device 1 of this embodiment can pass a bidirectional | two-way electric current between the 3rd connection point 203 and the 4th connection point 204 by making the 2nd bidirectional switch 14 all the ON states. Can create a unique state.

  That is, the first bidirectional switch 13 is turned off when the ninth and tenth switching elements Q9 and Q10 are both off, and the ninth and tenth switching elements Q9 and Q10 are both on. All are turned on. Further, in the first bidirectional switch 13, when the tenth switching element Q10 is on and the ninth switching element Q9 is off, the direction of current is restricted to one direction by the ninth diode D9. Semi-on state.

  Further, the second bidirectional switch 14 is fully turned off when the eleventh and twelfth switching elements Q11 and Q12 are both off, and the eleventh and twelfth switching elements Q11 and Q12 are both on. All are turned on. Further, in the second bidirectional switch 14, when the twelfth switching element Q12 is on and the eleventh switching element Q11 is off, the direction of the current is limited to one direction by the eleventh diode D11. Semi-on state.

  As described above, the bidirectional switch (each of the first bidirectional switch 13 and the second bidirectional switch 14) in the present embodiment has three operation states consisting of a full-off state, a full-on state, and a half-on state. Can be switched.

  In other words, the first bidirectional switch 13 is electrically connected between the positive terminal of the first capacitor C1 and the negative terminal of the second capacitor C2. The second bidirectional switch 14 is electrically connected between the negative terminal of the first capacitor C1 and the positive terminal of the second capacitor C2. That is, the first capacitor C1 of the first conversion circuit 11 and the second capacitor C2 of the second conversion circuit 12 are connected in a crossed manner via the first bidirectional switch 13 and the second bidirectional switch 14. Has been.

  Furthermore, the gates of the first to eighth switching elements Q1 to Q8 and the ninth to twelfth switching elements Q9 to Q12 are electrically connected to the control unit 6, respectively. The control unit 6 can individually switch on / off the first to fourth switching elements Q <b> 1 to Q <b> 4 and thereby controls the first conversion circuit 11. The control unit 6 can individually switch on / off the fifth to eighth switching elements Q5 to Q8, and thereby controls the second conversion circuit 12. The controller 6 can individually switch on / off the ninth and tenth switching elements Q9 and Q10, and thereby controls the first bidirectional switch 13. The controller 6 can individually switch on / off the eleventh and twelfth switching elements Q11 and Q12, and thereby controls the second bidirectional switch 14.

  Note that the control unit 6 may be provided individually for each of the first conversion circuit 11, the second conversion circuit 12, the first bidirectional switch 13, and the second bidirectional switch 14.

  In the present embodiment, the control unit 6 includes a drive circuit 61 that supplies a drive signal to the first to twelfth switching elements Q1 to Q12, and a microcomputer (microcomputer) 62 that supplies a signal to the drive circuit 61.

  The drive circuit 61 is configured to drive (control) each element individually by giving a drive signal to the control terminals (gates) of the first to twelfth switching elements Q1 to Q12. The microcomputer 62 is configured to control the drive circuit 61 by giving a PWM signal to the drive circuit 61. That is, the control unit 6 individually controls the first to twelfth switching elements Q1 to Q12 by a drive signal generated by the drive circuit 61 in response to an instruction from the microcomputer 62.

  Here, it is preferable that the drive circuit 61 also has a function as a short-circuit prevention circuit that prevents two or more switching elements from being simultaneously turned on to prevent a short-circuit current from flowing. That is, when switching elements of a specific combination are simultaneously turned on, for example, the first input point 101 and the second input point 102 are short-circuited, and the current from the DC power supply 100 may flow as a short-circuit current to the switching element. There is. Therefore, it is preferable that the drive circuit 61 is configured such that such specific combination of switching elements does not turn on at the same time. For example, when the drive signals input to the gates of a specific combination of switching elements simultaneously become H level, the drive circuit 61 forcibly lowers the drive signal to L level to simultaneously turn on the specific combination of switching elements. It is configured not to let you.

  The generation unit 3 generates a reference wave composed of at least one carrier wave. In the present embodiment, a sawtooth wave is an example of a carrier wave. Therefore, the generation unit 3 is configured with a sawtooth wave oscillator. The reference wave (one or a plurality of carrier waves) generated by the generating unit 3 is output to the microcomputer 62 of the control unit 6 and used to generate a PWM signal in the microcomputer 62. That is, the microcomputer 62 generates a PWM signal by a sawtooth wave comparison method that compares a reference wave that is a sawtooth wave and a signal wave based on the target value of the output voltage V10. Since the power converter 1 of this embodiment is used for the power conditioner 20 (see FIG. 7) as described above, the signal wave based on the target value of the output voltage V10 has the same waveform based on the same sine wave as that of the system power supply 7. Become. Specific examples of the reference wave and details of the PWM signal generation method will be described later.

  The generation of the PWM signal can be realized if there is a function of generating a reference wave and a function of comparing the reference wave and the signal wave. Therefore, in the present embodiment, the generation unit 3 that generates the reference wave is provided separately from the microcomputer 62 that generates the PWM signal, but the generation unit 3 may be incorporated in the control unit (microcomputer 62) 6. The control unit 6 may be configured to use a comparator (comparator) instead of the microcomputer 62 to compare the signal wave with the reference wave and generate a PWM signal. Further, the sawtooth wave is merely an example of the carrier wave, and the carrier wave may be, for example, a triangular wave. In this case, the generation unit 3 is configured by a triangular wave oscillator.

  As shown in FIG. 1, the filter circuit 5 has a pair of inductors L1 and L2 and a third capacitor C3. One inductor L <b> 1 is electrically connected between the first output point 103 and the third output point 105. The other inductor L 2 is electrically connected between the second output point 104 and the fourth output point 106. However, if the inductors L1 and L2 are electrically connected between at least one of the first output point 103 and the second output point 104 and the output terminal (the third output point 105 and the fourth output point 106). Any one of the inductors L1 and L2 may be omitted.

  The third capacitor C <b> 3 is electrically connected between the third output point 105 and the fourth output point 106. In other words, the filter circuit 5 is a series circuit of an inductor L1, a third capacitor C3, and an inductor L2 that are electrically connected between the first output point 103 and the second output point 104.

  The first detection unit 21 is configured to detect the voltage of the first capacitor C1. Here, the 1st detection part 21 detects the magnitude | size of the voltage V1 which generate | occur | produces at the both ends of the 1st capacitor C1, using the 1st connection point 201 side as a positive electrode. The first detection unit 21 includes a pair of voltage dividing resistors connected in series between the first connection point 201 and the third connection point 203, for example. However, the structure of the 1st detection part 21 is not restricted to this, What is necessary is just a structure which can detect the value (magnitude | size) of the voltage (both ends voltage) V1 which generate | occur | produces at the both ends of the 1st capacitor C1. The first detection unit 21 outputs the value of the voltage V <b> 1 as a detection result to the microcomputer 62 of the control unit 6.

  The second detection unit 22 is configured to detect the voltage of the second capacitor C2. Here, the 2nd detection part 22 detects the magnitude | size of the voltage V2 which generate | occur | produces at the both ends of the 2nd capacitor C2 by making the 4th connection point 204 side into a positive electrode. The second detection unit 22 includes, for example, a pair of voltage dividing resistors connected in series between the fourth connection point 204 and the second connection point 202. However, the configuration of the second detection unit 22 is not limited to this, and may be any configuration that can detect the value (magnitude) of the voltage (voltage across both ends) V2 generated at both ends of the second capacitor C2. The second detection unit 22 outputs the value of the voltage V <b> 2 that is the detection result to the microcomputer 62 of the control unit 6.

  In the present embodiment, the detection circuit that detects the voltage of the capacitors (the first capacitor C1 and the second capacitor C2) is configured by the first detection unit 21 and the second detection unit 22. Further, as will be described in detail later, the voltage of the capacitor detected by the detection circuit includes the detection result of the first detection unit 21 (voltage V1 of the first capacitor C1) and the detection result of the second detection unit 22 (second capacitor C2). Voltage V2). The operation of the control unit 6 using the detection results of the first detection unit 21 and the second detection unit 22 will be described later.

<Basic operation of power converter>
The basic operation of the power conversion device 1 configured as described above will be briefly described with reference to FIGS. 2A, 2B, 3A, 3B, 4A, 4B, 5A, and 5B. In the drawing, a thick arrow represents a current path, and a switching element with a dotted circle represents an on-state element.

  The basic operation of the power conversion device 1 here is a period from when the supply of power from the DC power supply 100 is started until the first capacitor C1 and the second capacitor C2 are charged to the reference voltage (hereinafter referred to as “starting period”). The operation of the power conversion apparatus 1 after elapse of “)”. That is, the operation of the power conversion device 1 from the state where the first capacitor C1 and the second capacitor C2 are charged to the reference voltage is defined as the basic operation of the power conversion device 1.

  In the present embodiment, the control unit 6 maintains the voltage of the capacitor at a target voltage obtained by adding a specified value whose absolute value is smaller than the voltage of the capacitor to ½ the voltage of the DC power supply 100. It is configured. The voltage of the capacitor here is the sum of the voltage V1 of the first capacitor C1 and the voltage V2 of the second capacitor C2, and the circuit constant (capacitance) of the first capacitor C1 and the circuit constant (capacitance) of the second capacitor C2. Are equivalent. Accordingly, the target reference voltage of each of the first capacitor C1 and the second capacitor C2 is ½ of the target voltage of the capacitor voltage.

  That is, if 1/2 of the specified value is a predetermined value, the reference voltage for the first capacitor C1 is the applied voltage applied between the first input point 101 and the second input point 102 from the DC power supply 100. This is a voltage obtained by adding a predetermined value to the size of 1/4. Similarly, the reference voltage for the second capacitor C2 is also obtained by adding a predetermined value to 1/4 of the applied voltage applied between the first input point 101 and the second input point 102 from the DC power supply 100. Voltage.

  In the following, it is assumed that the output voltage of the DC power supply 100 is E [V], the potential of the first input point 101 is E [V], and the potential of the second input point 102 is 0 [V]. Here, assuming that the predetermined value is ΔE [V], the voltage across each of the first capacitor C1 and the second capacitor C2 charged to the reference voltage is E / 4 + ΔE [V]. However, the predetermined value ΔE [V] is sufficiently smaller than the reference voltage. Therefore, in the “basic operation of power conversion device” column and the “configuration of power conditioner” column, for convenience of explanation, the predetermined value ΔE is ignored and the reference voltage is assumed to be E / 4 [V]. . In other words, the voltage across each of the first capacitor C1 and the second capacitor C2 charged to the reference voltage is E / 4 [V].

  Hereinafter, a potential difference between the first output point 103 and the second output point 104, that is, a voltage generated between the first output point 103 and the second output point 104 will be described as the second output voltage V20. The second output voltage V20 is also simply referred to as “output voltage V20”.

  Since the third output point 105 and the fourth output point 106 are electrically connected to the system power supply 7, the potential difference between the third output point 105 and the fourth output point 106, that is, the third output point 105-the second output point. The first output voltage V10 generated between the four output points 106 is equal to the output voltage of the system power supply 7. The potential difference between the first output point 103 and the third output point 105 and the potential difference between the second output point 104 and the fourth output point 106 are absorbed by the filter circuit 5.

  The power conversion device 1 switches the first conversion circuit 11, the second conversion circuit 12, the first bidirectional switch 13, and the second bidirectional switch 14 to a total of eight modes 1 to 8. As a result, the power conversion device 1 converts the DC voltage (E [V]) applied between the first input point 101 and the second input point 102 into an AC voltage, and the first output point 103 and the first input point 103 An output voltage V20 is generated between the two output points 104. In the following description, the first to twelfth switching elements Q1 to Q12 are assumed to be in the “off” state when the on / off state is not mentioned. Further, it is assumed that the voltage drop in the first to twelfth switching elements Q1 to Q12 and the voltage drop in the first to twelfth diodes D1 to D12 are negligible.

  Here, the control part 6 controls the 1st-12th switching elements Q1-Q12 according to the following two conditions.

  As a first condition, a one-to-one pair is set by the first to fourth switching elements Q1 to Q4 of the first conversion circuit 11 and the fifth to eighth switching elements Q5 to Q8 of the second conversion circuit 12. On / off is switched for each pair. Here, the first and eighth switching elements Q1, Q8 are paired, the second, seventh switching elements Q2, Q7 are paired, the third, sixth switching elements Q3, Q6 are paired, and the fourth, fifth Switching elements Q4 and Q5 form a pair.

  The second condition is that the second switching element Q2 and the third switching element Q3 are not simultaneously turned on or off. Further, in the first to fourth modes, the first switching element Q1 and the eleventh switching element Q11, and in the fifth to eighth modes, the fourth switching element Q4 and the ninth switching element Q9, Are not simultaneously turned on or off.

  First, in the first mode shown in FIG. 2A, the first and second switching elements Q1 and Q2 of the first conversion circuit 11, the seventh and eighth switching elements Q7 and Q8 of the second conversion circuit 12, and the second Each of the twelfth switching elements Q12 of the bidirectional switch 14 is in an ON state. That is, the second bidirectional switch 14 is in a half-on state. In this state, as shown in FIG. 2A, the first input point 101 is electrically connected to the first output point 103 via the first switching element Q1 and the second switching element Q2. The second input point 102 is electrically connected to the second output point 104 via the eighth switching element Q8 and the seventh switching element Q7. At this time, among the semiconductor elements (switching elements, diodes), there are a total of four elements, i.e., first, second, seventh, and eighth switching elements Q1, Q2, Q7, and Q8, and the twelfth switching element Q12. There is no current flowing through.

  Accordingly, the first output point 103 has the same potential (E [V]) as the first input point 101, and the second output point 104 has the same potential (0 [V]) as the second input point 102. Therefore, the output voltage V20 of the power conversion device 1 generated between the first output point 103 and the second output point 104 is E (= E-0) [V]. Further, at this time, the potential of the third output point 105 becomes a potential obtained by subtracting the voltage across the inductor L1 from the potential of the first output point 103, and the potential of the fourth output point 106 becomes the potential of the second output point 104 to the inductor. The potential is the sum of the voltages at both ends of L2.

  Next, in the second mode shown in FIG. 2B, the first and third switching elements Q1 and Q3 of the first conversion circuit 11, the sixth and eighth switching elements Q6 and Q8 of the second conversion circuit 12, Each of the two bidirectional switches 14 and the twelfth switching element Q12 is in an ON state. That is, the second bidirectional switch 14 is in a half-on state. In this state, as shown in FIG. 2B, the first input point 101 is electrically connected to the first output point 103 via the first switching element Q1, the first capacitor C1, and the third switching element Q3. The The second input point 102 is electrically connected to the second output point 104 via the eighth switching element Q8, the second capacitor C2, and the sixth switching element Q6. At this time, among the semiconductor elements (switching elements, diodes), there are a total of four elements, that is, the first, third, sixth, and eighth switching elements Q1, Q3, Q6, and Q8, and the twelfth switching element Q12. There is no current flowing through.

  Therefore, the potential of the first output point 103 is lower than the potential (E [V]) of the first input point 101 by the voltage across the first capacitor C1 (E / 4 [V]), that is, 3E / 4 ( = EE-4) [V]. The potential of the second output point 104 is higher than the potential of the second input point 102 (0 [V]) by the voltage across the second capacitor C2 (E / 4 [V]), that is, E / 4 ( = 0 + E / 4) [V]. Therefore, the output voltage V20 of the power converter 1 generated between the first output point 103 and the second output point 104 is E / 2 (= 3E / 4-E / 4) [V]. Further, at this time, the potential of the third output point 105 becomes a potential obtained by subtracting the voltage across the inductor L1 from the potential of the first output point 103, and the potential of the fourth output point 106 becomes the potential of the second output point 104 to the inductor. The potential is the sum of the voltages at both ends of L2.

  Next, in the third mode shown in FIG. 3A, the second switching element Q2 of the first conversion circuit 11, the seventh switching element Q7 of the second conversion circuit 12, and the second bidirectional switch 14 11 and 12 switching elements Q11 and Q12 are in an ON state, respectively. That is, the second bidirectional switch 14 is in an all-on state. In this state, the second output point 104 passes through the seventh switching element Q7, the second capacitor C2, the eleventh switching element Q11, the twelfth switching element Q12, the first capacitor C1, and the second switching element Q2. And electrically connected to the first output point 103. At this time, among the semiconductor elements (switching elements, diodes), there are a total of four elements, ie, the second, seventh, eleventh, and twelfth switching elements Q2, Q7, Q11, and Q12.

  Accordingly, the potential at the first output point 103 is determined by the voltage across the first capacitor C1 (E / 4 [V]) and the voltage across the second capacitor C2 (E / 4 [V]) from the potential at the second output point 104. ) And a higher potential. Therefore, the output voltage V20 of the power converter 1 generated between the first output point 103 and the second output point 104 becomes E / 2 (= E / 4 + E / 4) [V]. Further, at this time, the potential of the third output point 105 becomes a potential obtained by subtracting the voltage across the inductor L1 from the potential of the first output point 103, and the potential of the fourth output point 106 becomes the potential of the second output point 104 to the inductor. The potential is the sum of the voltages at both ends of L2. In this state, since the second bidirectional switch 14 is fully on, the conversion circuit 10 can cause a bidirectional current to flow between the first output point 103 and the second output point 104. it can.

  Next, in the fourth mode shown in FIG. 3B, the third switching element Q3 of the first conversion circuit 11, the sixth switching element Q6 of the second conversion circuit 12, and the second bidirectional switch 14 are switched. 11 and 12 switching elements Q11 and Q12 are in an ON state, respectively. That is, the second bidirectional switch 14 is in an all-on state. In this state, the second output point 104 is electrically connected to the first output point 103 via the sixth switching element Q6, the eleventh switching element Q11, the twelfth switching element Q12, and the third switching element Q3. Connected. At this time, among the semiconductor elements (switching elements, diodes), there are a total of four elements, ie, third, sixth, eleventh, and twelfth switching elements Q3, Q6, Q11, and Q12.

  Therefore, the potential at the first output point 103 is the same as that at the second output point 104. Therefore, the output voltage V20 of the power conversion device 1 generated between the first output point 103 and the second output point 104 becomes 0 [V]. Further, at this time, the potential of the third output point 105 becomes a potential obtained by subtracting the voltage across the inductor L1 from the potential of the first output point 103, and the potential of the fourth output point 106 becomes the potential of the second output point 104 to the inductor. The potential is the sum of the voltages at both ends of L2. In this state, since the second bidirectional switch 14 is fully on, the conversion circuit 10 can cause a bidirectional current to flow between the first output point 103 and the second output point 104. it can.

  On the other hand, in the fifth to eighth modes, the power conversion device 1 switches the operation between the first conversion circuit 11 and the second conversion circuit 12 based on the first to fourth modes. The bi-directional switch 13 and the second bi-directional switch 14 perform operations that are interchanged. That is, in the fifth to eighth modes and the first to fourth modes, the operation of the conversion circuit 10 is the first conversion circuit 11 and the first bidirectional switch 13, and the second conversion circuit 12 and the second mode. The direction switch 14 is replaced with a symmetrical operation.

  That is, in the fifth mode shown in FIG. 4A, the operation of the conversion circuit 10 is symmetric to the fourth mode. Therefore, in the fifth mode, the second switching element Q2 of the first conversion circuit 11, the seventh switching element Q7 of the second conversion circuit 12, and the ninth and tenth switching of the first bidirectional switch 13 are switched. Elements Q9 and Q10 are each in an on state. That is, the first bidirectional switch 13 is fully turned on. In this state, as shown in FIG. 4A, the first output point 103 is connected to the second switching element Q2, the ninth switching element Q9, the tenth switching element Q10, and the seventh switching element Q7 through the second switching element Q2. It is electrically connected to the output point 104. At this time, among the semiconductor elements (switching elements, diodes), there are a total of four elements, ie, the second, seventh, ninth, and tenth switching elements Q2, Q7, Q9, and Q10.

  Therefore, the potential at the first output point 103 is the same as that at the second output point 104. Therefore, the output voltage V20 of the power conversion device 1 generated between the first output point 103 and the second output point 104 becomes 0 [V]. Further, at this time, the potential at the third output point 105 is a potential obtained by adding the voltage across the inductor L1 to the potential at the first output point 103, and the potential at the fourth output point 106 is changed from the potential at the second output point 104 to the inductor. The potential is obtained by subtracting the voltage across L2. In this state, since the first bidirectional switch 13 is fully on, the conversion circuit 10 can cause a bidirectional current to flow between the first output point 103 and the second output point 104. it can.

  Next, in the sixth mode shown in FIG. 4B, the operation of the conversion circuit 10 is symmetric to the third mode. Therefore, in the sixth mode, the third switching element Q3 of the first conversion circuit 11, the sixth switching element Q6 of the second conversion circuit 12, and the ninth and tenth switching of the first bidirectional switch 13 are switched. Elements Q9 and Q10 are each in an on state. That is, the first bidirectional switch 13 is fully turned on. In this state, the first output point 103 passes through the third switching element Q3, the first capacitor C1, the ninth switching element Q9, the tenth switching element Q10, the second capacitor C2, and the sixth switching element Q6. And electrically connected to the second output point 104. At this time, among the semiconductor elements (switching elements, diodes), there are a total of four elements, that is, third, sixth, ninth, and tenth switching elements Q3, Q6, Q9, and Q10.

  Accordingly, the potential at the first output point 103 is determined by the voltage across the first capacitor C1 (E / 4 [V]) and the voltage across the second capacitor C2 (E / 4 [V]) from the potential at the second output point 104. ) And a lower potential. Therefore, the output voltage V20 of the power converter 1 generated between the first output point 103 and the second output point 104 is −E / 2 (= −E / 4−E / 4) [V]. Further, at this time, the potential at the third output point 105 is a potential obtained by adding the voltage across the inductor L1 to the potential at the first output point 103, and the potential at the fourth output point 106 is changed from the potential at the second output point 104 to the inductor. The potential is obtained by subtracting the voltage across L2. In this state, since the first bidirectional switch 13 is fully on, the conversion circuit 10 can cause a bidirectional current to flow between the first output point 103 and the second output point 104. it can.

  Next, in the seventh mode shown in FIG. 5A, the operation of the conversion circuit 10 is symmetric to the second mode. Therefore, in the seventh mode, the second and fourth switching elements Q2 and Q4 of the first conversion circuit 11, the fifth and seventh switching elements Q5 and Q7 of the second conversion circuit 12, and the first bidirectional switch The thirteenth switching elements Q10 are in the on state. That is, the first bidirectional switch 13 is in a half-on state. In this state, as shown in FIG. 5A, the first input point 101 is electrically connected to the second output point 104 via the fifth switching element Q5, the second capacitor C2, and the seventh switching element Q7. The The second input point 102 is electrically connected to the first output point 103 via the fourth switching element Q4, the first capacitor C1, and the second switching element Q2. At this time, among the semiconductor elements (switching elements, diodes), there are a total of four elements, ie, the second, fourth, fifth, and seventh switching elements Q2, Q4, Q5, and Q7, and the tenth switching element Q10. There is no current flowing through.

  Therefore, the potential of the first output point 103 is higher than the potential of the second input point 102 (0 [V]) by the voltage across the first capacitor C1 (E / 4 [V]), that is, E / 4 ( = 0 + E / 4) [V]. The potential of the second output point 104 is lower than the potential of the first input point 101 (E [V]) by the voltage across the second capacitor C2 (E / 4 [V]), that is, 3E / 4 ( = EE-4) [V]. Therefore, the output voltage V20 of the power conversion device 1 generated between the first output point 103 and the second output point 104 becomes −E / 2 (= E / 4-3E / 4) [V]. Further, at this time, the potential at the third output point 105 is a potential obtained by adding the voltage across the inductor L1 to the potential at the first output point 103, and the potential at the fourth output point 106 is changed from the potential at the second output point 104 to the inductor. The potential is obtained by subtracting the voltage across L2.

  Next, in the eighth mode shown in FIG. 5B, the operation of the conversion circuit 10 is symmetric with respect to the first mode. Therefore, in the eighth mode, the third and fourth switching elements Q3 and Q4 of the first conversion circuit 11, the fifth and sixth switching elements Q5 and Q6 of the second conversion circuit 12, and the first bidirectional switch. The thirteenth switching elements Q10 are in the on state. That is, the first bidirectional switch 13 is in a half-on state. In this state, as shown in FIG. 5B, the first input point 101 is electrically connected to the second output point 104 via the fifth switching element Q5 and the sixth switching element Q6. The second input point 102 is electrically connected to the first output point 103 via the fourth switching element Q4 and the third switching element Q3. At this time, among the semiconductor elements (switching elements, diodes), there are a total of four elements, that is, the third, fourth, fifth, and sixth switching elements Q3, Q4, Q5, and Q6, and the tenth switching element Q10. There is no current flowing through.

  Therefore, the first output point 103 has the same potential (0 [V]) as the second input point 102, and the second output point 104 has the same potential (E [V]) as the first input point 101. Therefore, the output voltage V20 of the power conversion device 1 generated between the first output point 103 and the second output point 104 is −E (= 0−E) [V]. Further, at this time, the potential at the third output point 105 is a potential obtained by adding the voltage across the inductor L1 to the potential at the first output point 103, and the potential at the fourth output point 106 is changed from the potential at the second output point 104 to the inductor. The potential is obtained by subtracting the voltage across L2.

  In short, the power conversion apparatus 1 changes the magnitude of the output voltage V20 generated between the first output point 103 and the second output point 104 in a plurality of stages by switching the first to eighth modes.

  More specifically, the first conversion circuit 11 uses the first capacitor C1 as a flying capacitor, and switches the first to fourth, ninth to twelfth switching elements Q1 to Q4 and Q9 to Q12 on / off, thereby The potential at one output point 103 is switched. In principle, the first capacitor C1 is charged in the second and seventh modes and discharged in the third and sixth modes. However, if the first to eighth modes are switched at a relatively high frequency, the first capacitor C1 is operated during basic operation. The voltage across the first capacitor C1 can be regarded as substantially constant (E / 4 [V]).

  The second conversion circuit 12 uses the second capacitor C2 as a flying capacitor, and switches the potential of the second output point 104 by switching on / off the fifth to twelfth switching elements Q5 to Q12. In principle, the second capacitor C2 is charged in the second and seventh modes and discharged in the third and sixth modes. However, if the first to eighth modes are switched at a relatively high frequency, the second capacitor C2 is operated during basic operation. The voltage across the second capacitor C2 can be regarded as substantially constant (E / 4 [V]).

  In short, the control unit 6 switches the pair of modes in which the magnitude of the output voltage V20 is the same and the directions of the currents flowing through the capacitors (the first capacitor C1 and the second capacitor C2) are reversed. Switching between charging and discharging.

  Specifically, when the output voltage V20 is set to E / 2 [V], the control unit 6 switches the second mode and the third mode as a pair of modes so that the capacitor (first capacitor C1 and second capacitor C2) are switched between charging and discharging. In addition, when the output voltage V20 is set to −E / 2 [V], the control unit 6 switches the seventh mode and the sixth mode as a pair of modes, so that the capacitor (the first capacitor C1 and the first mode) Switching between charging and discharging of the second capacitor C2).

  Here, since the first capacitor C1 and the second capacitor C2 are charged in the second and seventh modes in principle, the second mode and the seventh mode are also referred to as “charging modes” below. Since the first capacitor C1 and the second capacitor C2 are discharged in the third and sixth modes in principle, the third mode and the sixth mode are also referred to as “discharge modes” below. Further, among the first to eighth modes, the modes other than the charge mode and the discharge mode (first, fourth, fifth, and eighth modes) are modes that maintain the capacitor voltage without contributing to charging or discharging of the capacitor. Therefore, hereinafter, the first, fourth, fifth, and eighth modes are also referred to as “maintenance modes”. In the following description, the control unit 6 outputs a “charge command” when selecting the charge mode and selects the “discharge command” when selecting the discharge mode when selecting either the charge mode or the discharge mode. Command "is output.

  That is, when the output voltage V20 is set to E / 2 [V], the control unit 6 outputs a charge command when charging the capacitors (the first capacitor C1 and the second capacitor C2), and is in the charging mode. 2 mode is selected. When the output voltage V20 is set to E / 2 [V], the control unit 6 outputs a discharge command when discharging the capacitors (the first capacitor C1 and the second capacitor C2), and the third mode is the discharge mode. Select a mode.

  Similarly, when charging the capacitors (first capacitor C1 and second capacitor C2) when the output voltage V20 is set to -E / 2 [V], the control unit 6 outputs a charging command and performs charging in the charging mode. A certain seventh mode is selected. When the output voltage V20 is set to -E / 2 [V], the control unit 6 outputs a discharge command when discharging the capacitors (the first capacitor C1 and the second capacitor C2), and is in the discharge mode. Select the mode.

  As described above, the control unit 6 charges the capacitor by switching between the charging mode and the discharging mode in which the magnitude of the output voltage V20 is the same and the direction of the current flowing through the capacitor is reversed as a pair of modes. And switching between discharge. However, the capacitor is charged in the charge mode and the capacitor is discharged in the discharge mode only when a forward current described later flows in the conversion circuit 10. In a state where a reverse current, which will be described later, flows in conversion circuit 10, the capacitor is discharged in the charge mode, and the capacitor is charged in the discharge mode. This point will be described later. Hereinafter, the description will be made assuming that the current flowing through the conversion circuit 10 is a forward current unless otherwise specified.

  As described above, in the first to eighth modes, the power conversion device 1 outputs the voltage having the first output point 103 as the high potential side and the second output point 104 as the low potential side as the output voltage V20. Will do. In the first to fourth modes, the power conversion device 1 converts the output voltage V20 generated between the first output point 103 and the second output point 104 to E [V] (first mode), E / Switching is performed in three stages of 2 [V] (second and third modes) and 0 [V] (fourth mode). In the fifth to eighth modes, the power conversion device 1 sets the output voltage V20 generated between the first output point 103 and the second output point 104 to 0 [V] (fifth mode), −E / Switching is performed in three stages of 2 [V] (6th and 7th mode) and -E [V] (8th mode).

  Therefore, the power conversion device 1 switches the output mode V20 to E [V], E / 2 [V], 0 [V], -E / 2 [V] by switching the first to eighth modes. ] And -E [V]. The power conversion apparatus 1 generates the first output voltage V10 that is an AC voltage between the third output point 105 and the fourth output point 106 by appropriately switching the output voltage V20 in five stages.

  Here, the output voltage V10 is equal to the output voltage of the system power supply 7, and has a sinusoidal waveform as shown in FIG. In FIG. 6, the horizontal axis represents the time axis and the vertical axis represents the voltage value. Here, in a state where the output voltage V10 varies in a range of 0 [V] to E [V] (that is, a period corresponding to a positive half-wave in a sine wave) T1 to T3, the power converter 1 It operates by switching the first to fourth modes. In a state where the output voltage V10 fluctuates in the range of 0 [V] to -E [V] (that is, a period corresponding to the negative half-wave in the sine wave) T4 to T6, the power conversion device 1 It operates by switching the mode of ~ 8.

  Table 1 summarizes the first to eighth modes described above.

  Here, the control part 6 switches on / off of the 1st-12th switching elements Q1-Q12 with a PWM signal, and implement | achieves the said 1st-8 mode.

  More specifically, in the states T1 and T3 in which the output voltage V10 varies in the range of 0 [V] to E / 2 [V] in FIG. Repeat the mode switching operation. Here, the controller 6 balances the discharge and charge of the first capacitor C1 and the second capacitor C2 by adjusting the time length in the second mode and the third mode.

  Further, in the state T2 in which the output voltage V10 fluctuates in the range of E / 2 [V] to E [V] in FIG. 6, the control unit 6 performs the operation of switching the first to third modes as shown in Table 1. repeat. Here, the controller 6 balances the discharge and charge of the first capacitor C1 and the second capacitor C2 by adjusting the time length in the second mode and the third mode.

  In FIG. 6, in the states T4 and T6 where the output voltage V10 varies in the range of 0 [V] to −E / 2 [V], the control unit 6 performs the fifth to seventh modes as shown in Table 1. Repeat the switching action. Here, the control unit 6 balances the discharge and charge of the first capacitor C1 and the second capacitor C2 by adjusting the time length in the sixth mode and the seventh mode.

  Furthermore, in the state T5 in which the output voltage V10 varies in the range of −E / 2 [V] to −E [V] in FIG. 6, the control unit 6 switches the sixth to eighth modes as shown in Table 1. Repeat the operation. Here, the control unit 6 balances the discharge and charge of the first capacitor C1 and the second capacitor C2 by adjusting the time length in the sixth mode and the seventh mode.

  It should be noted that states T1, T3, T4, T6 in which the absolute value of the average value of the output voltage V20 is changed within a range from zero to ½ of the voltage of the DC power supply 100 are referred to as “first state” below. Also called. Further, the states T2 and T5 in which the absolute value of the average value of the output voltage V20 is changed within the range of 1/2 the voltage of the DC power supply 100 to the voltage of the DC power supply 100 are hereinafter referred to as “second state”. Also called.

  In the present embodiment, the control unit 6 switches the first to eighth modes described above while changing the duty ratio of the PWM signal, so that the waveform of the output voltage V10 approximates a sine wave. The output voltage V20 generated between the output point 103 and the second output point 104 is controlled. In short, the power conversion device 1 changes the magnitude of the output voltage V20 generated between the first output point 103 and the second output point 104 in five steps by the control unit 6 to thereby generate a sinusoidal AC voltage ( An output voltage V10) is generated between the third output point 105 and the fourth output point 106.

  Note that the fourth mode and the fifth mode are both modes in which the output voltage V20 is 0 [V] and do not contribute to the discharging and charging of the first capacitor C1 and the second capacitor C2. For this reason, it is conceivable to omit either the fourth mode or the fifth mode, but the power conversion device 1 is configured as the fourth mode in consideration of the positive / negative balance of the output voltage V10. By dividing the fifth mode, the switching loss can be reduced and the efficiency is improved.

  According to the power conversion device 1 of the present embodiment, the number of elements through which a current flows (hereinafter referred to as “the number of passing elements”) among the semiconductor elements (switching elements, diodes) is any one of the first to eighth elements as described above. In this mode, it is “4” or less.

  In particular, in the third and fourth modes in which the second bidirectional switch 14 is fully turned on, even if the eleventh switching element Q11 and the twelfth switching element Q12 are counted as separate elements, the number of passing elements is “ 4 ". Similarly, in the fifth and sixth modes in which the first bidirectional switch 13 is fully turned on, even if the ninth switching element Q9 and the tenth switching element Q10 are counted as separate elements, the number of passing elements is “4”. Therefore, when the first and second bidirectional switches 13 and 14 are each composed of one element, the number of passing elements in the third to sixth modes is “3”.

  By the way, it is preferable that the control unit 6 switches between the maintenance mode and the charge mode or the discharge mode so that the first capacitor C1 and the second capacitor C2 repeat charging and discharging around the reference voltage, respectively. The reference voltage here is a voltage (E / 4 [V]) that is 1/4 of the voltage applied from the DC power supply 100 between the first input point 101 and the second input point 102 as described above. ). Thereby, the both-ends voltage of the 1st capacitor C1 and the both-ends voltage of the 2nd capacitor C2 at the time of basic operation are maintained at a reference voltage (E / 4 [V]), respectively.

  Further, in this case, the control unit 6 switches between the maintenance mode and the charge mode or the discharge mode so that the average value of the detection result of the first detection unit 21 and the detection result of the second detection unit 22 becomes the reference voltage. It is preferable.

  By the way, as described above, in the power conversion device 1 of the present embodiment, the second bidirectional switch 14 is fully turned on in the third and fourth modes, and the first bidirectional switch in the fifth and sixth modes. Reference numeral 13 denotes an all-on state. That is, in any of the first to eighth modes, the element through which a current flows among the semiconductor elements (switching elements, diodes) is any one of the first to twelfth switching elements Q1 to Q12, and the diode (first to first modes). No current flows through the twelve diodes D1 to D12). Therefore, the power conversion device 1 can flow a bidirectional current between the first output point 103 and the second output point 104 in any of the first to eighth modes.

  Hereinafter, the current flowing through the conversion circuit 10 in the direction indicated by the thick arrow in FIGS. 2A, 2B, 3A, 3B, 4A, 4B, 5A, and 5B is referred to as “forward current” and is converted in the opposite direction to the forward current. The current flowing through the circuit 10 is referred to as “reverse current”. That is, in the first to fourth modes in which the output voltage V10 varies in the range of 0 [V] to E [V], the current from the first output point 103 to the third output point 105 is the forward current. In the fifth to eighth modes in which the output voltage V10 varies in the range of 0 [V] to -E [V], the current from the second output point 104 to the fourth output point 106 is the forward current.

  In this way, the power conversion device 1 allows the output current to flow between the third output point 105 and the fourth output point 106, the third output point 105, and the second output point because the conversion circuit 10 supports bidirectional current. A phase difference can be set between the output voltage V10 generated between the four output points 106. In short, if there is a phase difference between the output current and the output voltage V10, a period in which the output current is different from the output voltage V10 (for example, the output voltage V10 is positive and the output current is negative) occurs.

  Here, during the period in which the output current and the output voltage V10 have the same sign, the current flowing through the conversion circuit 10 is a forward current, but in the period in which the output current and the output voltage V10 have different signs, the conversion circuit 10 The current flowing through becomes a reverse current. Therefore, when the power converter 1 sets a phase difference between the output current and the output voltage V10, the conversion circuit 10 needs to support a bidirectional current. In the power conversion device 1 of the present embodiment, since the conversion circuit 10 supports bidirectional current, it is possible to set a phase difference between the output current and the output voltage V10.

  In particular, when the power conversion device 1 is used for the power conditioner 20 (see FIG. 7) for the solar power generation device, the power conversion device is used for the purpose of detecting the isolated operation and suppressing the voltage rise of the system power supply 7. 1 may set a phase difference between the output current and the output voltage V10. Further, when the power conversion device 1 is used as a power conditioner for a power storage device, the power conversion device 1 is supplied with power by setting a phase difference between the output current and the output voltage V10. The direction is controlled, and charging and discharging of the power storage device are switched. The power conversion device 1 of the present embodiment can cope with such a use by creating a state in which bidirectional current can pass through the conversion circuit 10.

<Configuration of the inverter>
As shown in FIG. 7, the power conditioner 20 according to the present embodiment includes the power conversion device 1 and the disconnector 9. The disconnector 9 is electrically connected between the first output point 103 (see FIG. 1) and the second output point 104 (see FIG. 1) and the system power supply 7. In the example of FIG. 7, the disconnector 9 is electrically connected between the third output point 105 and the fourth output point 106 and the system power supply 7. In other words, the resolver 9 is connected to the first output point 103 and the second output point 104 via the filter circuit 5 (see FIG. 1). In other words, the circuit breaker 9 has only to be between the first output point 103 and the second output point 104 and the system power supply 7 and is directly connected to the first output point 103 and the second output point 104. Is not essential, and may be connected to the subsequent stage of the filter circuit 5 as in this embodiment.

  Here, the circuit breaker 9 is electrically connected between the first contact point 91 electrically connected between the third output point 105 and the system power supply 7, and between the fourth output point 106 and the system power supply 7. And a second contact portion 92 connected to the. However, the circuit breaker 9 only needs to be electrically connected between at least one of the third output point 105 and the fourth output point 106 and the system power supply 7, and the first contact part 91 and the second contact part 9 Any of 92 may be omitted.

  The power conditioner 20 performs grid connection operation in a steady state, converts DC power input from the DC power supply 100 into AC power by the power conversion device 1, and outputs the AC power to the system power supply 7 and the load 8. Although detailed description is omitted, the power conditioner 20 performs a self-sustained operation in which the disconnector 9 is opened and AC power is output in a state disconnected from the system power supply 7 in the event of an abnormality such as a power failure of the system power supply 7. It is configured as follows.

  According to the power conditioner 20, the first converter circuit 11 and the second converter circuit 12 and the system power supply 7 can be electrically disconnected by opening (disconnecting) the disconnector 9. Therefore, the power conditioner 20 opens the disconnector 9 during the start-up period after the power is turned on and before the power converter 1 starts the basic operation described above, so that the first output point 103 and the second output point are opened. A current path including the filter circuit 5 can be formed between the terminal 104 and the terminal 104.

  The current path here is a current path including the inductor L1, the third capacitor C3, and the inductor L2 constituting the filter circuit 5. By using this current path as a charging path, the power conversion device 1 uses the first capacitor C1 and the second capacitor even if the third output point 105 and the fourth output point 106 are electrically insulated. C2 can be charged.

  Therefore, the power conversion device 1 can charge the first capacitor C1 and the second capacitor C2 even if the third output point 105 and the fourth output point 106 are not connected to the system power supply 7. In other words, the power conversion device 1 operates in a steady state even when no load is connected between the pair of output terminals (the third output point 105 and the fourth output point 106) (no load state). Necessary capacitors (first capacitor C1 and second capacitor C2) can be charged. The steady operation here refers to the operation of the power conversion apparatus 1 after the start-up period has elapsed, that is, after the first capacitor C1 and the second capacitor C2 are charged to the reference voltage (E / 4 [V]). Therefore, it is synonymous with the basic operation described above.

  That is, the control unit 6 always selects the charging mode during the start-up period from the start of power supply from the DC power supply 100 to the conversion circuit 10 until the capacitor is charged to a predetermined voltage (reference voltage). Is preferred. Note that the charging mode here is different from the charging mode (second and seventh modes) in the basic operation, and the circuit between the first output point 103 and the second output point 104 is opened by opening the disconnector 9. In this mode, a current path including the filter circuit 5 is configured. When the first capacitor C1 and the second capacitor C2 are charged to the reference voltage (E / 4 [V]), the power conversion device 1 shifts to the basic operation described above, and enters the charging mode (second, seventh). ) And a discharge mode (third and sixth modes).

  According to this configuration, the power conversion device 1 does not switch between the charging mode and the discharging mode during the start-up period, and always charges the capacitor by selecting the charge mode. Can move to operation.

<Features>
In the following description, the control unit 6 adds a specified value (2ΔE [V]), which is smaller in absolute value than the voltage of the capacitor, to a target voltage (E / The voltage of the capacitor is maintained at (2 + 2ΔE [V]). The specified value (2ΔE [V]) may be a positive value (2ΔE> 0) or a negative value (2ΔE <0) as long as the absolute value is smaller than the capacitor voltage. However, first, the description will be made assuming that the specified value is positive (plus). In this case, the voltage across each of the first capacitor C1 and the second capacitor C2 charged to the reference voltage is slightly higher than E / 4 [V], and becomes E / 4 + ΔE [V]. The predetermined value ΔE [V] may be a value sufficiently smaller than the reference voltage, but as an example, when the voltage E of the DC power supply 100 is 330 [V], the predetermined value ΔE [V] is It is about 2-3 [V].

  Below, how the power converter device 1 of this embodiment switches the 1st-8th mode in basic operation | movement is demonstrated in detail.

  As described above, the control unit 6 determines whether the reference mode (at least one carrier wave) generated by the generation unit 3 and the signal wave based on the target value of the output voltage V10 are in the maintenance mode, A PWM signal for switching between the mode and the discharge mode is generated. For example, in the states T1 and T3 (see FIG. 6) in which the output voltage V10 fluctuates in the range of 0 [V] to E / 2 [V], the control unit 6 uses the PWM signal to change to the fourth mode that is the maintenance mode. And a second mode that is a charging mode or a third mode that is a discharging mode.

  In the present embodiment, the control unit 6 selects one of the charge mode and the discharge mode for each cycle of the reference wave so that only one of the charge mode and the discharge mode is included in one cycle of the reference wave. Configured to select. The power conversion device 1 according to the present embodiment uses one or a plurality of carrier waves having no phase difference as a reference wave, and according to a comparison result of the magnitudes of the reference wave and the signal wave, Switch between discharge modes. The signal wave here is a signal wave based on the target value of the output voltage V10, that is, the same sine wave as that of the system power supply 7.

  Here, as to whether to select the charge mode or the discharge mode, the control unit 6 basically determines according to the detection result of the detection circuit. The detection circuit includes a first detection unit 21 and a second detection unit 22, and the detection result of the detection circuit is an average value of the voltage V1 of the first capacitor C1 and the voltage V2 of the second capacitor C2. More specifically, the detection result of the first detector 21 (the voltage V1 of the first capacitor C1) and the detection result of the second detector 22 (the voltage V2 of the second capacitor C2) are output to the microcomputer 62. The microcomputer 62 obtains the average value Vc of the voltage V1 of the first capacitor C1 and the voltage V2 of the second capacitor C2 obtained individually as described above. The average value Vc is expressed by Vc = (V1 + V2) / 2.

  Here, the control unit 6 operates according to a flowchart shown in FIG. 8, for example.

  That is, the control unit 6 first compares the average value Vc of the voltage V1 of the first capacitor C1 and the voltage V2 of the second capacitor C2 with the reference voltage (E / 4 + ΔE [V]) (S1). At this time, if the average value Vc is equal to or higher than the reference voltage (S1: Yes), the control unit 6 outputs a discharge command (S2). On the other hand, if the average value Vc is less than the reference voltage (S1: No), the control unit 6 outputs a charge command (S3).

  The control unit 6 repeatedly performs the processes of S1 to S3 in the same cycle as that of the reference wave, thereby outputting one of the charge command and the discharge command for each cycle of the reference wave. The voltage V1 of the capacitor C1 and the voltage V2 of the second capacitor C2 are maintained at the reference voltage. As described above, the charge command and the discharge command are basically switched at any time according to the comparison result between the detection result of the detection circuit and the reference voltage.

  However, in the present embodiment, in a predetermined period, the control unit 6 outputs the charge command and the discharge command in a predetermined pattern regardless of the comparison result between the detection result of the detection circuit and the reference voltage. The predetermined period here is a period in which the combination of modes switched by the PWM signal among the first to eighth modes changes. For example, the period before and after the transition from the state T1 to the state T2 or from the state T2 It is a period before and after the time of transition to the state T3. In addition to this, a period before and after the time point when the state transitions from the state T4 to the state T5 and a time period before and after the point when the state transitions from the state T5 to the state T6 are included in the predetermined period. Note that the period including the zero cross point of the output voltage V10 is included in the predetermined period in this embodiment, such as the period before and after the transition from the state T3 to the state T4 and the period before and after the transition from the state T6 to the state T1. I can't.

  For example, in a predetermined period before and after the transition from the state T1 to the state T2, the control unit 6 outputs a discharge command in the state T1 and a charge command in the state T2. In a predetermined period before and after the transition from the state T2 to the state T3, the control unit 6 outputs a charge command to the state T2 and a discharge command to the state T3. In a predetermined period before and after the transition from the state T4 to the state T5, the control unit 6 outputs a discharge command in the state T4 and a charge command in the state T5. In a predetermined period before and after the transition from the state T5 to the state T6, the control unit 6 outputs a charge command in the state T5 and a discharge command in the state T6.

  Hereinafter, as a reference example, a case where four carrier waves CW1 to CW4 having no phase difference are used will be briefly described with reference to FIG. FIG. 9 shows the relationship between the signal wave OS1 (indicated by a thick line) and the carrier waves CW1 to CW4 with the horizontal axis as the time axis and the vertical axis as the voltage. In FIG. 9, since the relationship between the signal wave OS1 and the carrier waves CW1 to CW4 is schematically shown, each of the carrier waves CW1 to CW4 includes only 40 periods in one period of the signal wave OS1. However, it is not limited to this configuration. Actually, each of the carrier waves CW1 to CW4 may be included in several tens to several hundreds of cycles in one cycle of the signal wave OS1.

  In this reference example, the first carrier wave CW1 periodically oscillates between 0 [V] and E / 2 [V], with 0 [V] as the minimum value and E / 2 [V] as the maximum value. It is a sawtooth wave. The second carrier wave CW2 is a sawtooth wave that periodically oscillates between E / 2 [V] and E [V], with E / 2 [V] as the minimum value and E [V] as the maximum value. The third carrier wave CW3 is a sawtooth wave that oscillates periodically between 0 [V] and -E / 2 [V], with 0 [V] being the maximum value and -E / 2 [V] being the minimum value. is there. The fourth carrier wave CW4 is a saw that oscillates periodically between -E / 2 [V] and -E [V], with -E / 2 [V] as the maximum value and -E [V] as the minimum value. It is a wave.

  In the reference example, the signal wave is a sinusoidal waveform that is equal to the waveform of the output voltage V10 shown in FIG. 6 and fluctuates within the range of −E [V] to E [V].

  If the first to fourth carrier waves CW1 to CW4 are used as in the reference example, the control unit 6 generates a PWM signal according to the comparison result between the reference wave and the signal wave OS1 in each of the states T1 to T6. The sustain mode and the charge mode or the discharge mode can be switched. That is, for example, in the states T1 and T3 in which the signal wave OS1 fluctuates in the range of 0 [V] to E / 2 [V], the control unit 6 controls the signal wave OS1 and the reference wave (first to fourth carrier waves). A PWM signal is generated according to a comparison result with the first carrier wave CW1 among CW1 to CW4). Further, for example, in the state T2 in which the signal wave OS1 fluctuates in the range of E / 2 [V] to E [V], the control unit 6 includes the signal wave OS1 and the reference wave (first to fourth carrier waves CW1 to CW1). A PWM signal is generated in accordance with a comparison result with the second carrier wave CW2 in CW4). However, when the four carrier waves CW1 to CW4 are used as in the reference example, the configuration of the generation unit 3 and the processing of the control unit 6 are complicated.

  Therefore, the power conversion device 1 of the present embodiment uses only one continuous carrier wave CW1 as a reference wave, as shown in FIG. 10, and simplifies the configuration of the generation unit 3 and the processing of the control unit 6. FIG. 10 shows the relationship between the signal wave OS1 (indicated by a thick line) and the carrier wave CW1, with the horizontal axis representing the time axis and the vertical axis representing the voltage. In FIG. 10, as in FIG. 9, only one cycle of the carrier wave CW1 is included in one cycle of the signal wave OS1, but the present invention is not limited to this configuration. Actually, the carrier wave CW1 may be included in several tens to several hundreds of periods in one period of the signal wave OS1. The frequency of the carrier wave CW1 is, for example, about 16 kHz.

  In the example of FIG. 10, the carrier wave CW1 is a sawtooth wave that oscillates periodically between 0 [V] and E / 2 [V], with 0 [V] as the minimum value and E / 2 [V] as the maximum value. It is.

  In the present embodiment, the signal wave OS1 is based on the same sine wave as that of the system power supply 7 as shown in FIG. 10 (indicated by a broken line in the figure), but not the sine wave itself, depending on the states T1 to T6. It is the waveform of the shape which shifted the said sine wave suitably. That is, the original sine wave is E / 2 [so that the signal wave OS1 of FIG. 10 falls between 0 [V] which is the minimum value of the carrier wave CW1 and E / 2 [V] which is the maximum value. In a state T2 exceeding V], the waveform is obtained by shifting the original sine wave in the negative (minus) direction by E / 2 [V]. Further, the signal wave OS1 is positive in the original sine wave by E / 2 [V] in the states T4 and T6 where the original sine wave fluctuates in the range of 0 [V] to -E / 2 [V]. The waveform is shifted in the plus direction. Further, the signal wave OS1 is positive (plus) by E [V] in the state T5 where the original sine wave fluctuates in the range of -E2 [V] to -E [V]. It becomes a shifted waveform.

  The power conversion device 1 of the present embodiment generates a PWM signal according to the comparison result between one carrier wave CW1 and the signal wave OS1 as shown in FIG. 10, and switches between the maintenance mode and the charge mode or the discharge mode. . Furthermore, the power conversion device 1 according to the present embodiment is configured to select either the charge mode or the discharge mode in accordance with the charge command and the discharge command.

  Hereinafter, a comparison result between the carrier wave CW1 and the signal wave OS1 will be described as a comparison signal Sj1. The comparison signal Sj1 is a signal that becomes “H” during a period in which the signal wave OS1 is equal to or lower than the carrier wave CW1 (OS1 ≦ CW1). Further, it is assumed that the control unit 6 outputs a command signal Scd1 that becomes “H” during the period for outputting the charge command and becomes “L” during the period for outputting the discharge command.

  By using the signal wave OS1 shown in FIG. 10, the control unit 6 according to the present embodiment performs the sustain mode, the charge mode, and the discharge mode according to the comparison signal Sj1 and the command signal Scd1 according to the determination conditions shown in Table 2. Select one of the modes.

  The operation of the control unit 6 according to the determination conditions shown in Table 2 will be described in more detail with reference to FIG. Hereinafter, the control unit 6 outputs a charge signal Sc1 that is “H” during the period in which the conversion circuit 10 operates in the charge mode, and a discharge signal Sd1 that is “H” during the period in which the conversion circuit 10 operates in the discharge mode. Assume that.

  FIG. 11 shows the operation of the control unit 6 during a predetermined period before and after the transition from the state T1 to the state T2. Here, the predetermined period is a period corresponding to six periods (first to sixth periods P1 to P6) of the carrier wave CW1, centering on a time point when the state T1 is shifted to the state T2.

  In the state T1, the control unit 6 selects the charge mode or the discharge mode during a period in which the signal wave OS1 exceeds the carrier wave CW1 (OS1> CW1) (a period in which the comparison signal Sj1 is “L”). During a period in which the signal wave OS1 is equal to or lower than the carrier wave CW1 (a period in which the comparison signal Sj1 is “H”), the control unit 6 selects the maintenance mode. Then, the control unit 6 selects the charging mode during the period in which the charge command is output (the period in which the command signal Scd1 is “H”) among the periods in which the signal wave OS1 exceeds the carrier wave CW1, and the charge signal Sc1. Set to “H”. The control unit 6 selects the discharge mode during the period in which the discharge command is output (the period in which the command signal Scd1 is “L”) among the periods in which the signal wave OS1 exceeds the carrier wave CW1, and sets the discharge signal Sd1 to “ H ”.

  Here, in a predetermined period (first to sixth cycles P1 to P6) before and after the transition from the state T1 to the state T2, the control unit 6 issues a discharge command to the state T1 and a charge command to the state T2. Output each. Therefore, the discharge command is output (command signal Scd1 is “L”) over the first to third cycles P1 to P3 of the carrier wave CW1 corresponding to the state T1. As a result, in the state T1, the power conversion device 1 is configured so that the sustain mode and the discharge mode are PWM in one cycle (each of the first to third cycles P1 to P3) of the carrier wave CW1 as shown in FIG. It is switched by a signal. Here, the charging mode in the state T1 is the second mode, the discharging mode is the third mode, and the sustaining mode is the fourth mode. Therefore, in each of the first to third periods P1 to P3, the control unit 6 switches the second to fourth modes in the order of the fourth mode and the third mode.

  As a result, in each of the first to third periods P1 to P3 of the state T1, the output voltage V20 generated between the first output point 103 and the second output point 104 is 0 [V as shown in FIG. ] (Fourth mode) and E / 2 + 2ΔE [V] (third mode). Here, the reason why the output voltage V20 in the third mode is not E / 2 [V] but E / 2 + 2ΔE [V] is that the voltage V1 of the first capacitor C1 and the voltage V2 of the second capacitor C2 are respectively E / 4 + ΔE. It is because it was set as [V]. That is, the potential of the first output point 103 in the third mode is higher than the potential of the second output point 104 by the sum of the voltage V1 of the first capacitor C1 and the voltage V2 of the second capacitor C2. Therefore, the output voltage V20 becomes E / 2 + 2ΔE [V].

  On the other hand, in the state T2, the control unit 6 selects the charge mode or the discharge mode during a period in which the signal wave OS1 is equal to or lower than the carrier wave CW1 (a period in which the comparison signal Sj1 is “H”). During a period in which the signal wave OS1 exceeds the carrier wave CW1 (a period in which the comparison signal Sj1 is “L”), the control unit 6 selects the maintenance mode. Then, the control unit 6 selects the charging mode during the period in which the charge command is output (the period in which the command signal Scd1 is “H”) among the periods in which the signal wave OS1 is equal to or lower than the carrier wave CW1, and the charge signal Sc1 Set to “H”. The control unit 6 selects the discharge mode and sets the discharge signal Sd1 to “During the period when the discharge command is output (the period in which the command signal Scd1 is“ L ”) during the period in which the signal wave OS1 is equal to or lower than the carrier wave CW1. H ”.

  Here, in a predetermined period (first to sixth cycles P1 to P6) before and after the transition from the state T1 to the state T2, the control unit 6 issues a discharge command to the state T1 and a charge command to the state T2. Output each. Therefore, the charge command is output (command signal Scd1 is “H”) over the fourth to sixth cycles P4 to P6 of the carrier wave CW1 corresponding to the state T2. As a result, in the state T2, the power conversion device 1 has a PWM mode in which the charge mode and the sustain mode are PWM in one cycle (each of the fourth to sixth cycles P4 to P6) of the carrier wave CW1, as shown in FIG. It is switched by a signal. Here, the charging mode in the state T2 is the second mode, the discharging mode is the third mode, and the sustaining mode is the first mode. Therefore, in each of the fourth to sixth periods P4 to P6, the control unit 6 switches the first to third modes in the order of the second mode and the first mode.

  As a result, the output voltage V20 generated between the first output point 103 and the second output point 104 in each of the fourth to sixth periods P4 to P6 of the state T2 is E / 2 as shown in FIG. -2ΔE [V] (second mode) and E [V] (first mode) are switched in this order. Here, the reason why the output voltage V20 in the second mode is not E / 2 [V] but E / 2−2ΔE [V] is that the voltage V1 of the first capacitor C1 and the voltage V2 of the second capacitor C2 are E / 4 + ΔE [V]. That is, the potential of the first output point 103 in the second mode is lower than the potential (E [V]) of the first input point 101 by the voltage V1 of the first capacitor C1, that is, 3E / 4-ΔE. (= EE−4−ΔE) [V]. The potential of the second output point 104 in the second mode is higher than the potential (0 [V]) of the second input point 102 by the voltage V2 of the second capacitor C2, that is, E / 4-ΔE ( = 0 + E / 4−ΔE) [V]. Therefore, the output voltage V20 generated between the first output point 103 and the second output point 104 becomes E / 2−2ΔE [V].

  By the way, in the state T1, the fourth mode and the third mode are switched by the PWM signal. Therefore, as the signal wave OS1 rises, the duty ratio of the fourth mode becomes close to 0%. The mode duty ratio is close to 100%. In other words, the duty ratio of 0 [V] in the output voltage V20 is close to 0%, and the duty ratio of E / 2 + 2ΔE [V] is close to 100%. However, in general, since there is an upper limit value and a lower limit value of the duty ratio in PWM control, the duty ratio of 0 [V] in the output voltage V20 is completely in the third period P3 in which the signal wave OS1 is maximum. It will not be 0%. Here, the average value of the output voltage V20 is E / 2 [V] in the third period P3.

  In the state T2, since the second mode and the first mode are switched by the PWM signal, the duty ratio of the second mode approaches 100% as the signal wave OS1 decreases, and the first mode The mode duty ratio is close to 0%. In other words, the duty ratio of E / 2-2ΔE [V] is close to 100%, and the duty ratio of E [V] at the output voltage V20 is close to 0%. However, in general, since there is an upper limit value and a lower limit value of the duty ratio in the PWM control, the duty ratio of E [V] in the output voltage V20 is perfect even in the fourth period P4 where the signal wave OS1 is minimum. It will not be 0%. Here, the average value of the output voltage V20 is E / 2 [V] in the fourth period P4.

  Therefore, in the transition from the state T1 to the state T2 (the third period P3 and the fourth period P4), the average value of the output voltage V20 is E / 2 [V]. Here, the output voltage V10 generated between the third output point 105 and the fourth output point 106 corresponds to the average value of the output voltage V20. Therefore, when transitioning from the state T1 to the state T2 (the third period P3 and the fourth period P4), the output voltage V10 continuously changes as shown in FIG.

  Similarly, in the remaining states T3 to T6, the power conversion device 1 according to the present embodiment follows the determination condition in Table 2 according to the comparison result between the carrier wave CW1 and the signal wave OS1, and the charge command or the discharge command. Switching between the maintenance mode, the charging mode, and the discharging mode. Thus, for example, when the state T2 is shifted to the state T3, the output voltage V10 continuously changes. Similarly, the output voltage V10 continuously changes when transitioning from the state T4 to the state T5 and when transitioning from the state T5 to the state T6.

  Note that, as described above, the discharge mode or the charge mode is continuously selected for a predetermined period (several cycles of the carrier wave CW1), and strictly speaking, the voltages of the first capacitor C1 and the second capacitor C2 Variations can also occur. However, voltage fluctuations of the first capacitor C1 and the second capacitor C2 can be suppressed to a negligible level by setting the capacitances of the first capacitor C1 and the second capacitor C2 and the period of the carrier wave CW1.

  Further, the predetermined period in which the control unit 6 outputs the charge command and the discharge command in a predetermined pattern is equivalent to six periods (first to sixth periods P1 to P6) of the carrier wave CW1 as in the example of FIG. It is not limited to the period. The predetermined period may be a period (third and fourth periods P3 and P4) corresponding to two periods of the carrier wave CW1, which is set so as to straddle at least the time when the state shifts.

  By the way, the specified value (2ΔE [V]) is not limited to a positive value (2ΔE> 0) but a negative value (2ΔE <0) as long as the absolute value is smaller than the voltage of the capacitor. There may be. When the specified value is negative (minus), unlike the case where the specified value is positive (plus), the voltage across each of the first capacitor C1 and the second capacitor C2 charged to the reference voltage is E It becomes slightly lower than / 4 [V] (E / 4 + ΔE [V]).

  That is, when the specified value is negative, the output voltage V20 is slightly lower than ½ of the voltage of the DC power supply 100 in the discharge mode and slightly higher than ½ of the voltage of the DC power supply 100 in the charge mode. . Therefore, when the specified value is negative, the control unit 6 reverses the relationship between the discharge mode and the charging mode from the case where the specified value is positive, so that the output voltage is the same as when the specified value is positive. V20 (see FIG. 11) is obtained. That is, when the specified value is negative, for example, in a predetermined period before and after the transition from the state T1 to the state T2, the control unit 6 outputs a charge command in the state T1 and a discharge command in the state T2.

  Therefore, in the state T1, in one cycle of the carrier wave CW1 (each of the first to third cycles P1 to P3), the charging mode (second mode) and the maintenance mode (fourth mode) are PWM signals. It will be switched by. Thus, in each of the first to third periods P1 to P3 of the state T1, the output voltage V20 generated between the first output point 103 and the second output point 104 is 0 [V] (fourth mode). ) And E / 2-2ΔE [V] (second mode). On the other hand, in the state T2, in one cycle of the carrier wave CW1 (each of the fourth to sixth cycles P4 to P6), the discharge mode (third mode) and the sustain mode (first mode) are PWM signals. It will be switched by. Thus, in each of the fourth to sixth periods P4 to P6 of the state T2, the output voltage V20 generated between the first output point 103 and the second output point 104 is E2 + 2 + ΔE [V] (third Mode) and E [V] (first mode).

  As a result, even when the specified value is negative, the average value of the output voltage V20 is E / 2 [V] when shifting from the state T1 to the state T2. Here, the output voltage V10 generated between the third output point 105 and the fourth output point 106 corresponds to the average value of the output voltage V20. Therefore, the output voltage V10 continuously changes when the state T1 is shifted to the state T2.

<Comparative example>
Hereinafter, in order to explain the effect of the present embodiment, the voltage of the capacitor is maintained at a target voltage (E / 2 [V]) that is ½ the voltage of the DC power supply 100. A power conversion device of a comparative example will be described with reference to FIG. Since the power conversion device of the comparative example is common to the power conversion device 1 of the present embodiment with respect to the configuration and operation other than the capacitor voltage, the same reference numerals are given to the same configurations as those of the present embodiment below. The description will be omitted as appropriate.

  In the comparative example, the voltage across each of the first capacitor C1 and the second capacitor C2 charged to the reference voltage is E / 4 [V].

  FIG. 12 shows the operation of the control unit 6 during a predetermined period before and after the transition from the state T1 to the state T2. Here, the predetermined period is a period corresponding to six periods (first to sixth periods P1 to P6) of the carrier wave CW1, centering on a time point when the state T1 is shifted to the state T2.

  First, in each of the first to third periods P1 to P3 of the state T1, the output voltage V20 generated between the first output point 103 and the second output point 104 is 0 [V as shown in FIG. ] (Fourth mode) and E / 2 [V] (third mode).

  On the other hand, in each of the fourth to sixth periods P4 to P6 of the state T2, the output voltage V20 generated between the first output point 103 and the second output point 104 is E / 2 [ V] (second mode) and E [V] (first mode) are switched in this order.

  By the way, in the state T1, the fourth mode and the third mode are switched by the PWM signal. Therefore, as the signal wave OS1 rises, the duty ratio of the fourth mode becomes close to 0%. The mode duty ratio is close to 100%. In other words, the duty ratio of 0 [V] in the output voltage V20 is close to 0%, and the duty ratio of E / 2 [V] is close to 100%. However, in general, since there is an upper limit value and a lower limit value of the duty ratio in PWM control, the duty ratio of 0 [V] in the output voltage V20 is completely in the third period P3 in which the signal wave OS1 is maximum. It will not be 0%. Therefore, in the third period P3, the average value of the output voltage V20 does not become E / 2 [V], but is a value lower than E / 2 [V] (hereinafter referred to as “E / 2−α [V]”). )become.

  In the state T2, since the second mode and the first mode are switched by the PWM signal, the duty ratio of the second mode approaches 100% as the signal wave OS1 decreases, and the first mode The mode duty ratio is close to 0%. In other words, the duty ratio of E / 2 [V] is close to 100%, and the duty ratio of E [V] at the output voltage V20 is close to 0%. However, in general, since there is an upper limit value and a lower limit value of the duty ratio in the PWM control, the duty ratio of E [V] in the output voltage V20 is perfect even in the fourth period P4 where the signal wave OS1 is minimum. It will not be 0%. Therefore, in the fourth period P4, the average value of the output voltage V20 does not become E / 2 [V], but becomes a value exceeding E / 2 [V] (hereinafter referred to as “E / 2 + α [V]”). .

  Therefore, when the state T1 is shifted to the state T2 (the third period P3 and the fourth period P4), the average value of the output voltage V20 is E / 2−α [V] and E / 2 + α [V]. It does not match. Here, the output voltage V10 generated between the third output point 105 and the fourth output point 106 corresponds to the average value of the output voltage V20. Therefore, when the state T1 is shifted to the state T2 (the third period P3 and the fourth period P4), the output voltage V10 changes discontinuously as shown in FIG. 12, resulting in a step.

<Effect>
According to the power conversion device 1 of the present embodiment described above, a specified value having a smaller absolute value than the voltage of the capacitor is added to a target voltage that is ½ of the voltage of the DC power supply 100, and the capacitor voltage is reduced. Maintain voltage. In other words, in this power conversion device 1, the voltage of the capacitor is not maintained at a half of the voltage of the DC power supply 100, but is slightly larger than the half of the voltage of the DC power supply 100 (specified). The target voltage is maintained at a high (or low) target voltage. As a result, if the specified value is positive, as described above, the output voltage V20 generated between the first output point 103 and the second output point 104 is greater than ½ of the voltage of the DC power supply 100 in the discharge mode. It is slightly higher and slightly lower than ½ of the voltage of the DC power supply 100 in the charging mode. If the specified value is negative, the output voltage V20 is slightly lower than ½ of the voltage of the DC power supply 100 in the discharge mode and slightly higher than ½ of the voltage of the DC power supply 100 in the charge mode. Therefore, when the power conversion device 1 realizes a sinusoidal AC output by switching a plurality of modes with a PWM signal, even if there is an upper limit value and a lower limit value of the duty ratio, output discontinuity occurs. There is an advantage that it can be suppressed.

  That is, as described in the “Comparative Example” column, the power conversion apparatus 1 maintains the capacitor voltage at a target voltage that is half the voltage of the DC power supply 100, for example, from the state T1 to the state T2. When shifting to, the output voltage V10 changes discontinuously and causes a step. On the other hand, the power conversion device 1 of the present embodiment maintains the voltage of the capacitor at a target voltage that is slightly higher (or lower) than ½ of the voltage of the DC power supply 100. For example, from the state T1 The average value of the output voltage V20 when shifting to the state T2 can be made the same value. Therefore, in the power conversion device 1 of the present embodiment, for example, when the state is changed from the state T1 to the state T2, the output voltage V10 changes continuously, and a step is hardly generated in the output voltage V10.

  In other words, in the power conversion device 1 of the present embodiment, the output voltage V20 is slightly higher than 1/2 of the voltage of the DC power supply 100 in the discharge mode, and slightly higher than 1/2 of the voltage of the DC power supply 100 in the charge mode. By lowering the value, the lower limit value of the duty ratio in the maintenance mode can be increased. That is, the power converter 1 can adjust the average value of the output voltage V20 to ½ of the voltage of the DC power supply 100 without bringing the duty ratio of the sustain mode as close as possible to 0%. The accuracy of PWM control can be improved by increasing the lower limit value.

  It should be noted that since the output current is zero at the zero cross point of the output voltage V10 at the time of transition from the state T3 to the state T4 and the time of transition from the state T6 to the state T1, there is a problem that occurs in the output voltage V10. Continuity does not matter. For this reason, the power conversion device 1 of the present embodiment suppresses discontinuity of the output voltage V10 at a time other than the zero-cross point of the output voltage V10, such as when the state T1 shifts to the state T2. To do.

  Further, in the power conversion device 1, as in the present embodiment, the conversion circuit 10 includes a first conversion circuit 11 and a second conversion circuit 12, a first bidirectional switch 13 and a second bidirectional switch 14. It is preferable to provide. The power conversion device 1 includes a first conversion circuit 11 and a second conversion circuit 12 that are connected in parallel between both ends of a DC power supply 100, and the first conversion circuit 11 is connected between the first conversion circuit 11 and the second conversion circuit 12. The bidirectional switch 13 and the second bidirectional switch 14 are connected. Here, the first conversion circuit 11 includes four switching elements (first to fourth switching elements Q1 to Q4) and one capacitor (first capacitor C1). Similarly, the second conversion circuit 12 includes four switching elements (fifth to eighth switching elements Q5 to Q8) and one capacitor (second capacitor C2).

  In this configuration, the current input from the DC power supply 100 to the power conversion device 1 is out of ten switching elements (first to eighth switching elements Q1 to Q8 and first and second bidirectional switches 13 and 14). It passes through at most four elements. Therefore, this power conversion device 1 has the advantage that the sum of the conduction loss (loss) of the switching elements is relatively small, and the power conversion efficiency can be further improved.

  Further, the power conversion device 1 generally requires a large heat radiating device (an air cooling device such as a heat sink or a fan) because the calorific value increases as the conduction loss increases. The power conversion device 1 of the present embodiment can be expected to reduce the size of the heat dissipation device by suppressing conduction loss.

  Compared with the configuration described in Patent Document 1, the power conversion device 1 of the present embodiment also has an advantage that the entire device can be reduced in size by the amount that does not require a voltage dividing capacitor. In other words, the power conversion device described in Patent Document 1 divides the DC voltage E by E / 2 by applying the DC voltage E to a series circuit of two DC capacitors. The capacitor is an essential configuration. On the other hand, since the power conversion device 1 of the present embodiment does not require a voltage dividing capacitor, the entire device can be reduced in size accordingly.

  In addition, as in the present embodiment, the control unit 6 uses the PWM signal in the first state (states T1, T3, T4, and T6) to specify the maintenance mode in which the output voltage V20 is zero and the charge mode or discharge mode. It is preferable to switch at. In this case, in the second state (states T2 and T5), the control unit 6 uses the PWM signal as the sustain mode in which the absolute value of the output voltage V20 is the same as the voltage of the DC power supply 100, and the charge mode or the discharge mode. It is preferable to switch. Here, the first state (states T1, T3, T4, T6) is a state in which the absolute value of the average value of the output voltage V20 is changed within a range from zero to half the voltage of the DC power supply 100. It is. The second state (states T2 and T5) is a state in which the absolute value of the average value of the output voltage V20 is changed within a range of 1/2 the voltage of the DC power supply 100 to the voltage of the DC power supply 100.

  According to this configuration, when performing the PWM control using the PWM signal, the control unit 6 has an advantage that the lower limit value of the duty ratio in the maintenance mode can be increased as described above to improve the accuracy of the PWM control. is there.

  Further, according to the power conditioner 20 according to the present embodiment, the disconnector 9 is opened (disconnected), thereby electrically connecting the first conversion circuit 11 and the second conversion circuit 12 to the system power supply 7. Can be separated. Therefore, the power conditioner 20 performs grid connection operation in a steady state, and when the system power supply 7 is abnormal, such as a power failure, opens the disconnector 9 and outputs AC power in a state disconnected from the system power supply 7. You can perform autonomous operation.

  Further, as in the present embodiment, the operating state of the first bidirectional switch 13 is such that the current flowing from the second connection point 202 to the first connection point 201 is cut off, and the first connection point 201 to the second connection point. It is preferable to further include a half-on state in which the current flowing to 202 is passed. In this case, the operating state of the second bidirectional switch 14 blocks the current flowing from the third connection point 203 to the fourth connection point 204 and passes the current flowing from the fourth connection point 204 to the third connection point 203. It is preferable to further include a half-on state.

  According to this configuration, the first bidirectional switch 13 is half-on in a mode in which the current flowing from the first connection point 201 to the second connection point 202 does not need to be interrupted as in the seventh and eighth modes. The state is fine. Therefore, the control unit 6 can continue to turn on the tenth switching element Q10 in the period (states T4 to T6) in which the operation for switching the fifth to seventh modes or the sixth to eighth modes is repeated. In other words, in the fifth and sixth modes, the first bidirectional switch 13 is all on, so that the tenth switching element Q10 is turned off whenever the tenth switching element Q10 is turned off each time the mode is switched to the seventh and eighth modes. Switching loss may occur. The power converter 1 of the present embodiment allows the first bidirectional switch 13 to keep the tenth switching element Q10 on when the fifth to seventh modes or the sixth to eighth modes are switched. The generated switching loss can be reduced.

  Similarly, in the mode in which the current flowing from the fourth connection point 204 to the third connection point 203 does not need to be interrupted as in the first and second modes, the second bidirectional switch 14 may be in a half-on state. . Therefore, the power conversion device 1 of the present embodiment is configured so that the twelfth switching element Q12 is kept on when the first to third modes or the second to fourth modes are switched, so that the second bidirectional switch 14 can reduce the switching loss.

  Further, if the control unit 6 shifts the first bidirectional switch 13 from the half-on state to the fully-on state in a state where the current flows through the first bidirectional switch 13, the first bidirectional switch 13. Switching loss caused by the switch 13 can be further reduced. That is, for example, when switching from the seventh mode to the sixth mode, the control unit 6 turns on the ninth switching element Q9 while the ninth diode D9 is turned on, so that the ninth switching is performed. Zero volt switching of the element Q9 can be realized. Similarly, if the control unit 6 shifts the second bidirectional switch 14 from the half-on state to the fully-on state in a state where the current is flowing through the second bidirectional switch 14, Switching loss caused by the directional switch 14 can be further reduced.

  In the power conversion device 1, the conversion circuit 10 includes the first conversion circuit 11 and the second conversion circuit 12, the first bidirectional switch 13 and the second bidirectional switch 14 as described above. It is not limited and can be changed as appropriate. The number of switching elements is not limited to 12 of the first to twelfth switching elements Q1 to Q12, and can be changed as appropriate.

  The first to eighth switching elements Q1 to Q8 and the ninth to twelfth switching elements Q9 to Q12 are not limited to depletion type n-channel MOSFETs, and other semiconductor switches may be used. For example, a power semiconductor device using a wide band gap semiconductor material such as IGBT (Insulated Gate Bipolar Transistor) or GaN (gallium nitride) is used.

  Also, the specific configuration of the bidirectional switch (each of the first bidirectional switch 13 and the second bidirectional switch 14) is not limited to the configuration described above. The bidirectional switch may be a double gate (dual gate) structure bidirectional switch using a wide band gap semiconductor material such as GaN (gallium nitride).

(Embodiment 2)
The power conversion device 1 according to the present embodiment generates a PWM signal using a signal wave OS1 as shown in FIG. 13 and switches between the maintenance mode and the charge mode or the discharge mode. 1 and different. Since the circuit configuration of the power conversion device 1 of the present embodiment is the same as that of the power conversion device 1 of the first embodiment, the same configurations as those of the first embodiment will be denoted by the same reference numerals and description thereof will be omitted as appropriate. To do.

  That is, in the first embodiment, the control unit 6 uses the signal wave OS1 as shown in FIG. 10, and as shown in Table 2, depending on the states T1 to T6, the control unit 6 is in the sustain mode, the charge mode, and the discharge mode. The determination conditions for selecting the mode to be applied may differ. For example, the condition for selecting the charging mode is that the comparison signal Sj1 is “L” and the command signal Scd1 is “H” in the state T1, whereas the comparison signal Sj1 is “H” and the command signal Scd1 is “H” in the state T2. H ”, and the determination condition is different between the state T1 and the state T2.

  Therefore, in the present embodiment, as shown in FIG. 13, by devising the shape of the signal wave OS1, the determination conditions for selecting the maintenance mode, the charging mode, and the discharging mode are set regardless of the states T1 to T6. It is uniform. FIG. 13 shows the relationship between the signal wave OS1 (indicated by a thick line) and the carrier wave CW1, with the horizontal axis representing the time axis and the vertical axis representing the voltage. In FIG. 13, only 20 periods of the carrier wave CW1 are included in one period of the signal wave OS1, but the present invention is not limited to this configuration. Actually, the carrier wave CW1 may be included in several tens to several hundreds of periods in one period of the signal wave OS1. The frequency of the carrier wave CW1 is, for example, about 16 kHz.

  In the example of FIG. 13, the carrier wave CW1 is a triangular wave that oscillates periodically between 0 [V] and E / 2 [V], with 0 [V] as the minimum value and E / 2 [V] as the maximum value. is there. This carrier wave CW1 is a triangular wave whose voltage change rate per unit time is the same when rising and falling. However, the carrier wave CW1 shown in FIG. 13 is merely an example, and the carrier wave CW1 may be different when the voltage change rate per unit time is rising or falling, and may be, for example, a sawtooth wave.

  In the present embodiment, as shown in FIG. 13, the signal wave OS1 is based on the same sine wave as that of the system power supply 7 (indicated by a broken line in the figure), but the sine wave is appropriately shifted according to the states T1 to T6. In addition, the waveform is inverted. That is, the original sine wave is E / 2 [, so that the signal wave OS1 in FIG. 13 falls between 0 [V] which is the minimum value of the carrier wave CW1 and E / 2 [V] which is the maximum value. In the state T2 exceeding V], the waveform is obtained by inverting the original sine wave around E / 2 [V]. In other words, the signal wave OS1 in the state T2 has a waveform obtained by inverting the positive / negative of the original sine wave around 0 [V] and shifting it in the positive (plus) direction by E [V]. The signal wave OS1 is a positive signal of the original sine wave centered on 0 [V] in the states T4 and T6 where the original sine wave fluctuates in the range of 0 [V] to -E / 2 [V]. • The waveform is inverted from negative. Furthermore, this signal wave OS1 is positive or negative with respect to the original sine wave centered on 0 [V] in the state T5 where the original sine wave fluctuates in the range of -E2 [V] to -E [V]. The waveform is inverted and inverted about E / 2 [V]. In other words, the signal wave OS1 in the state T5 has a waveform obtained by shifting the original sine wave in the positive (plus) direction by E [V].

  Hereinafter, a comparison result between the carrier wave CW1 and the signal wave OS1 will be described as a comparison signal Sj1. The comparison signal Sj1 is a signal that becomes “H” during a period in which the signal wave OS1 is equal to or lower than the carrier wave CW1 (OS1 ≦ CW1). Further, it is assumed that the control unit 6 outputs a command signal Scd1 that becomes “H” during the period for outputting the charge command and becomes “L” during the period for outputting the discharge command.

  By using the signal wave OS1 shown in FIG. 13, the control unit 6 of the present embodiment follows the determination conditions shown in Table 3 according to the comparison signal Sj1 and the command signal Scd1 regardless of the states T1 to T6. And any one of the charge mode and the discharge mode is selected.

  The operation of the control unit 6 according to the determination conditions shown in Table 3 will be described in more detail with reference to FIG. Hereinafter, the control unit 6 outputs a charge signal Sc1 that is “H” during the period in which the conversion circuit 10 operates in the charge mode, and a discharge signal Sd1 that is “H” during the period in which the conversion circuit 10 operates in the discharge mode. Assume that. In the example of FIG. 14, the period until the carrier wave CW1 changes from the minimum value (0 [V]) to the maximum value (E / 2 [V]) and again changes to the minimum value (0 [V]). , One period of the carrier wave CW1 (each of the first to sixth periods P1 to P6). The control unit 6 outputs one of a charge command and a discharge command in each of the first to sixth periods P1 to P6 of the carrier wave CW1.

  FIG. 14 shows the operation of the control unit 6 during a predetermined period before and after the time point when the state T1 is shifted to the state T2. Here, the predetermined period is a period corresponding to six periods (first to sixth periods P1 to P6) of the carrier wave CW1, centering on a time point when the state T1 is shifted to the state T2.

  As shown in FIG. 14, the control unit 6 performs the charge mode during the period (OS1> CW1) in which the signal wave OS1 exceeds the carrier wave CW1 (period in which the comparison signal Sj1 is “L”) regardless of the states T1 and T2. Or, select the discharge mode. During a period in which the signal wave OS1 is equal to or lower than the carrier wave CW1 (a period in which the comparison signal Sj1 is “H”), the control unit 6 selects the maintenance mode. Then, the control unit 6 selects the charging mode during the period in which the charge command is output (the period in which the command signal Scd1 is “H”) among the periods in which the signal wave OS1 exceeds the carrier wave CW1, and the charge signal Sc1. Set to “H”. The control unit 6 selects the discharge mode during the period in which the discharge command is output (the period in which the command signal Scd1 is “L”) among the periods in which the signal wave OS1 exceeds the carrier wave CW1, and sets the discharge signal Sd1 to “ H ”.

  Here, in a predetermined period (first to sixth cycles P1 to P6) before and after the transition from the state T1 to the state T2, the control unit 6 issues a discharge command to the state T1 and a charge command to the state T2. Output each.

  Therefore, the discharge command is output (command signal Scd1 is “L”) over the first to third cycles P1 to P3 of the carrier wave CW1 corresponding to the state T1. As a result, in the state T1, the power conversion device 1 is configured so that the discharge mode and the sustain mode are PWM in one cycle (each of the first to third cycles P1 to P3) of the carrier wave CW1, as shown in FIG. It is switched by a signal. Here, the charging mode in the state T1 is the second mode, the discharging mode is the third mode, and the sustaining mode is the fourth mode. Therefore, in each of the first to third periods P1 to P3, the control unit 6 switches the second to fourth modes in the order of the third mode, the fourth mode, and the third mode.

  On the other hand, the charge command is output (command signal Scd1 is “H”) over the fourth to sixth periods P4 to P6 of the carrier wave CW1 corresponding to the state T2. As a result, in the state T2, the power conversion device 1 has a PWM mode in which the charge mode and the sustain mode are PWM in one cycle (each of the fourth to sixth cycles P4 to P6) of the carrier wave CW1, as shown in FIG. It is switched by a signal. Here, the charging mode in the state T2 is the second mode, the discharging mode is the third mode, and the sustaining mode is the first mode. Therefore, in each of the fourth to sixth periods P4 to P6, the control unit 6 switches the first to third modes in the order of the second mode, the first mode, and the second mode.

  As a result, in each of the first to third periods P1 to P3 of the state T1, the output voltage V20 is E / 2 + 2ΔE [V] (third mode), 0 [V] (fourth) as shown in FIG. Mode) and E / 2 + 2ΔE [V] (third mode). On the other hand, in each of the fourth to sixth periods P4 to P6 of the state T2, the output voltage V20 is E / 2−2ΔE [V] (second mode), E [V] ( The first mode is switched in the order of E / 2-2ΔE [V] (second mode).

  By the way, in the state T1, the fourth mode and the third mode are switched by the PWM signal. Therefore, as the signal wave OS1 rises, the duty ratio of the fourth mode becomes close to 0%. The mode duty ratio is close to 100%. In other words, the duty ratio of 0 [V] in the output voltage V20 is close to 0%, and the duty ratio of E / 2 + 2ΔE [V] is close to 100%. However, in general, since there is an upper limit value and a lower limit value of the duty ratio in PWM control, the duty ratio of 0 [V] in the output voltage V20 is completely in the third period P3 in which the signal wave OS1 is maximum. It will not be 0%. Here, the average value of the output voltage V20 is E / 2 [V] in the third period P3.

  In the state T2, since the second mode and the first mode are switched by the PWM signal, the duty ratio of the second mode becomes close to 100% as the signal wave OS1 rises. The mode duty ratio is close to 0%. In other words, the duty ratio of E / 2-2ΔE [V] is close to 100%, and the duty ratio of E [V] at the output voltage V20 is close to 0%. However, in general, since there is an upper limit value and a lower limit value of the duty ratio in PWM control, the duty ratio of E [V] in the output voltage V20 is perfect even in the fourth period P4 where the signal wave OS1 is maximum. It will not be 0%. Here, the average value of the output voltage V20 is E / 2 [V] in the fourth period P4.

  Therefore, even in the power conversion device 1 of the present embodiment, when the state transitions from the state T1 to the state T2 (the third period P3 and the fourth period P4), as in the first embodiment, the output voltage V20 All of the average values are E / 2 [V]. Here, the output voltage V10 generated between the third output point 105 and the fourth output point 106 corresponds to the average value of the output voltage V20. Therefore, when the state T1 is shifted to the state T2 (the third period P3 and the fourth period P4), the output voltage V10 continuously changes as shown in FIG.

  Similarly, in the remaining states T3 to T6, the power conversion device 1 according to the present embodiment follows the determination condition in Table 3 according to the comparison result between the carrier wave CW1 and the signal wave OS1, and the charge command or the discharge command. Switching between the maintenance mode, the charging mode, and the discharging mode. Thus, for example, when the state T2 is shifted to the state T3, the output voltage V10 continuously changes. Similarly, the output voltage V10 continuously changes when transitioning from the state T4 to the state T5 and when transitioning from the state T5 to the state T6.

  According to the configuration of the present embodiment described above, the determination conditions for selecting the sustain mode, the charge mode, and the discharge mode are uniformly set regardless of the states T1 to T6 by devising the shape of the signal wave OS1. Can be aligned. That is, as shown in FIG. 13, while based on the same sine wave as that of the system power supply 7, not the sine wave itself but a signal wave OS1 having a shape obtained by appropriately shifting and inverting the sine wave according to the states T1 to T6. By using, the power converter device 1 can simplify determination conditions. As a result, the power conversion apparatus 1 has an advantage that the process related to the determination for selecting the maintenance mode, the charging mode, and the discharging mode can be simplified.

  In addition, the power converter device 1 uses signal wave OS1 as shown in FIG. 13, so that the timing for switching between the charge command and the discharge command is uniform regardless of the states T1 to T6 (E). / 2 [V]).

  Other configurations and functions are the same as those of the first embodiment.

DESCRIPTION OF SYMBOLS 1 Power converter 11 1st conversion circuit 12 2nd conversion circuit 13 1st bidirectional switch 14 2nd bidirectional switch 21 1st detection part 22 2nd detection part 6 Control part 7 System power supply 9 Separator 20 Power Conditioner 100 DC power supply 101 First input point 102 Second input point 103 First output point 104 Second output point 201 First connection point 202 Second connection point 203 Third connection point 204 Fourth connection point C1 First capacitor C2 Second capacitor Q1 to Q8 First to eighth switching elements V1 First capacitor voltage V2 Second capacitor voltage

Claims (4)

A conversion circuit including a plurality of switches and capacitors and electrically connected between a first input point on the high potential side of the DC power supply and a second input point on the low potential side of the DC power supply;
By controlling the plurality of switches, a plurality of modes having different connection states of the DC power supply and the capacitor with respect to the first output point and the second output point are switched, and the first output point and the second output point are A control unit that changes the magnitude of the output voltage generated between
The plurality of modes are a sustain mode in which no current flows through the capacitor and the absolute value of the output voltage is zero or the same as the voltage of the DC power supply, and the absolute value of the output voltage is reduced by flowing current through the capacitor. A charging mode in which the voltage of the capacitor is subtracted from the voltage of the DC power supply, and an absolute value of the output voltage is the same as the voltage of the capacitor by flowing a current in the opposite direction to the charging mode to the capacitor. Are classified as discharge modes,
The control unit maintains the voltage of the capacitor at a target voltage obtained by adding a specified value having a small absolute value compared to the voltage of the capacitor to a half of the voltage of the DC power supply. A power conversion device that switches between a maintenance mode and the charge mode or the discharge mode. The conversion circuit includes a first conversion circuit and a second conversion circuit electrically connected in parallel between the first input point and the second input point, and the first conversion circuit and the second conversion circuit. A first bidirectional switch and a second bidirectional switch electrically connected to each other,
The first conversion circuit includes a first switching element, a second switching element, a third switching element, a fourth switching element from the first input point side between the first input point and the second input point. First to fourth switching elements electrically connected in series in the order of the switching elements, and a second circuit electrically connected in parallel to the series circuit of the second switching element and the third switching element. 1 capacitor, and a connection point between the second switching element and the third switching element is the first output point,
The second conversion circuit includes a fifth switching element, a sixth switching element, a seventh switching element, an eighth switching element from the first input point side between the first input point and the second input point. In the order of the switching elements, the fifth to eighth switching elements electrically connected in series, and the sixth switching element and the seventh switching element connected in parallel with the series circuit. 2 capacitor, and the connection point between the sixth switching element and the seventh switching element is the second output point,
The first bidirectional switch is a connection point between the first switching element and the second switching element, and a connection point between the seventh switching element and the eighth switching element. Connected to the second connection point,
The second bidirectional switch is a connection point between the third switching element and the fourth switching element, and a connection point between the fifth switching element and the sixth switching element. Connected to the fourth connection point,
The plurality of switches includes the first to eighth switching elements, the first bidirectional switch, and the second bidirectional switch,
The capacitor is composed of the first capacitor and the second capacitor,
The power control device according to claim 1, wherein the control unit maintains a sum of a voltage of the first capacitor and a voltage of the second capacitor at the target voltage. The controller is
In the first state in which the absolute value of the average value of the output voltage is changed within a range from zero to ½ of the voltage of the DC power supply, the sustain mode in which the output voltage becomes zero, Switch between charge mode or discharge mode with PWM signal,
In the second state in which the absolute value of the average value of the output voltage is changed within a range of 1/2 the voltage of the DC power supply to the voltage of the DC power supply, the absolute value of the output voltage is the DC 3. The power conversion device according to claim 1, wherein the sustain mode, which is the same as a voltage of a power supply, and the charge mode or the discharge mode are configured to be switched by a PWM signal. The power conversion device according to any one of claims 1 to 3,
A power conditioner comprising: a disconnector electrically connected between the first output point and the second output point and a system power supply.

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