JP2015139244A - High frequency power supply device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To suppress heating generated in a switch of a DC/RF conversion part by preventing an increase in a loss generated in the switch, and to miniaturize a heat sink for radiating heat from the switch.SOLUTION: The high frequency power supply device includes: a DC/RF conversion part 5 for converting the DC output of a variable DC power supply part 4 into a high frequency output; and a control part 8 having an output control part 9 for controlling the variable DC power supply part 4 and a drive signal supply circuit 10 for supplying a drive signal to a switch configuring a switching circuit 5A of the DC/RF conversion part 5. The drive signal supply circuit 10 includes: an inter-control terminal voltage detection part 10C for detecting a voltage between the control terminals of each switch of the switching circuit 5A and an inter-control terminal voltage control part 10E for performing control to maintain a voltage to be detected by the inter-control terminal voltage detection part 10C at a target value. Thus, it is possible to prevent an increase in a conduction loss generated in each switch by performing control to maintain the inter-control terminal voltage of each switch of the switching circuit 5A at the target value.

Description

  The present invention relates to a high frequency power supply device that supplies high frequency power to a load such as a plasma load.

  As a high-frequency power supply device that supplies high-frequency power to a load such as a plasma load, as shown in Patent Document 1, a variable DC power supply unit having a function of adjusting a direct current output and a direct current output from the variable DC power supply unit A DC / RF converter (direct current / high frequency alternating current converter) that converts the output into a high frequency alternating current output by an on / off operation of the switch element, and controls the variable DC power source to set the value from the DC / RF converter. What is used is to output high-frequency power that is maintained at the same level.

  FIG. 19 shows the configuration of a conventional high-frequency power supply device of this type. In the figure, 1 is a high frequency power supply device, 2 is a load to which high frequency power is supplied from the high frequency power supply device 1, and 3 is an impedance matching device provided between the high frequency power supply device 1 and the load 2.

  The high frequency power supply device 1 shown in FIG. 19 includes a variable DC power supply unit (variable DC power supply unit) 4 having a function of adjusting the magnitude of DC power to be output according to the output control signal Sdc, and a variable DC power supply unit. DC / RF converter (DC / RF converter) 5 that converts DC power output from 4 to RF power, a low-pass filter 6 that removes harmonic components from the output of DC / RF converter 5, and a low-pass filter 6 includes a power detection unit 7 inserted between the output terminal 6 and the output terminal 1a of the high-frequency power supply device, and a control unit 8 'for controlling the variable DC power supply unit 4 and the DC / RF conversion unit 5. The power detection unit 7 includes a traveling wave component of the high frequency power output from the high frequency power supply device 1 and a reflected wave component of the high frequency power reflected and returned from the input unit of the impedance matching unit 3 when impedance mismatching occurs. Detection is performed and a traveling wave power detection signal Pf and a reflected wave power detection signal Pr are output.

  For example, as shown in FIG. 20, the DC / RF converter 5 has at least one leg 5a composed of a high-side switch Q1 and a low-side switch Q2 connected in series with each other, and the at least one leg Is composed of a series circuit of an inductor Lr and a capacitor Cr, and the output voltage Vdc of the variable DC power supply section passes through the switching circuit 5A. It is comprised by the class D amplifier provided with the applied series resonance circuit 5B. In the example shown in FIG. 20, the output of the variable DC power supply unit 4 is applied to the primary coil Wp of the transformer 5C through the switching circuit 5A and the series resonance circuit 5B, and the output of the secondary coil Ws of this transformer is applied to the low-pass filter 6. Have been entered. Note that the transformer 5C may be omitted.

  In the DC / RF converter shown in FIG. 20, the switches Q1 and Q2 constituting the switching circuit 5A are MOSFET FET1 and FET2, respectively, and the source of FET1 is connected to the drain of FET2, thereby Q2 is connected in series. A capacitor Cs is connected to both ends of the series circuit of the switches Q1 and Q2 (between the output terminals of the variable DC power supply unit). Between the control terminals of the switches Q1 and Q2 (between the gate and source of the FET1 and FET2 in the illustrated example), the voltage waveform from the drive signal supply circuit 10 'provided in the control unit 8' is sine wave or rectangular wave, respectively. The drive signals S1 and S2 exhibiting are given alternately.

  In the DC / RF converter 5 shown in FIG. 20, the series resonant circuit is selected by appropriately selecting the resonant frequency and Q (quality factor) of the series resonant circuit 5B with respect to the switching frequency of the switches Q1 and Q2. A sinusoidal AC current is passed through 5B and the primary coil Wp of the transformer 5C to convert the output of the variable DC power supply unit 4 into high-frequency AC power.

  The control unit 8 ′ shown in FIG. 19 gives a predetermined output control signal Sdc to the variable DC power supply unit 4 in accordance with the traveling wave component detection signal Pf output from the power detection unit 7, whereby the power detection unit 7 Output control to keep the traveling wave component (power supplied to the load) of the high frequency power detected by the set value, and when the reflected wave component detected by the power detection unit 7 exceeds the set specified value, Reflection protection control is performed to suppress the output of the variable DC power supply unit 4 so as to suppress the reflected wave component below a specified value. The control unit 8 ′ also performs a DC / RF conversion operation for converting the DC output of the variable DC power supply unit 4 into high-frequency power having a predetermined frequency by turning on and off the switch of the switching circuit of the DC / RF conversion unit 5. Let Part 5 do it.

  In the example shown in FIG. 20, a class D amplifier including a half-bridge type switching circuit 5A including one leg 5a including a series circuit of a high-side switch Q1 and a low-side switch Q2 and a series resonance circuit 5B is used to generate DC. / RF conversion unit 5 is configured, but the high-side switch of the leg of the class D amplifier is replaced with a choke coil, and a switching circuit is configured only by the low-side switch, and the output of the DC power supply unit is series-resonated through the choke coil. In some cases, the DC / RF converter is configured by a class E amplifier applied to the circuit.

  In general, the impedance matching unit 3 provided between the high-frequency power supply device 1 and the load 2 controls a variable capacitor or variable inductor, which is an impedance variable element, with a motor so that the load side from the input end of the impedance matching unit is controlled. Since the matching operation is performed by adjusting the observed impedance, impedance matching cannot be achieved instantaneously, and impedance matching usually requires a time of 100 msec to several seconds. Since reflection occurs at the input unit of the impedance matching unit 3 until impedance matching is achieved, the reflected wave power returning from the impedance matching unit 3 to the DC / RF conversion unit 5 increases, and the DC / RF conversion unit Current flowing in the switch constituting the switching circuit 5 increases, and conduction loss caused by the switch increases.

  When the load 2 for supplying high-frequency power is a plasma load, the impedance of the load is unstable, and depends on conditions such as applied power, pressure of gas in the chamber, flow rate of gas supplied into the chamber, and processing time. Since the load impedance changes finely, impedance mismatching occurs frequently, and a lot of loss occurs in the switches constituting the switching circuit of the DC / RF converter 5. In particular, when the impedance seen from the DC / RF converter is close to a state where the impedance is close to a short circuit and close to a state where total reflection occurs, a large current flows through the switch of the switching circuit. A large conduction loss occurs due to the on-resistance of the switch.

JP 2003-143861 A

  As described above, in the high-frequency power supply apparatus in which the DC power obtained from the variable DC power supply unit 4 is converted into high-frequency power by the DC / RF conversion unit 5A including a switching amplifier such as a class D amplifier or a class E amplifier. When the reflected power increases or when the load current increases, a lot of loss occurs in each switch of the switching circuit constituting the switching amplifier.

  Therefore, in the conventional high frequency power supply device, it is necessary to provide a large heat sink for each switch in order to prevent each switch of the DC / RF converter from being damaged due to an excessive temperature rise. There was a problem of becoming. Further, when a large loss occurs in the switch of the DC / RF conversion unit, the efficiency of the apparatus is lowered. Therefore, it is desirable to reduce the loss generated in each switch of the DC / RF conversion unit as much as possible.

  An object of the present invention is to prevent an excessive conduction loss from occurring in each switch of a switching circuit provided in a DC / RF conversion unit that converts an output of a variable DC power supply unit into high-frequency power, and to provide a heat sink provided for the switching circuit. An object of the present invention is to provide a high-frequency power supply device that can be miniaturized.

  The present invention has a variable DC power supply unit capable of controlling a DC output and a switching circuit having a switch composed of a semiconductor amplifying element, and the DC output of the variable DC power supply unit is output at high frequency by the switching operation of the switching circuit. A DC / RF converter that converts the signal into a DC / RF and a signal generator that generates a sine wave or rectangular wave voltage signal having a frequency equal to the output frequency of the DC / RF converter. A high-frequency power supply device including a drive signal supply circuit that provides a drive signal between control terminals of a switch of a switching circuit in synchronization with an output signal of a signal generator so as to cause the switching circuit to perform a switching operation for conversion To do. In the present application, in order to achieve the above object, at least the following first to eleventh inventions are disclosed.

  In the first invention, the drive signal supply circuit detects the voltage between the control terminals of each switch of the switching circuit directly or indirectly, and the detection signal includes information on the voltage between the control terminals of each switch. Between the control terminals that output the voltage, and between the control terminals that control the voltage between the control terminals of each switch detected from the detection signal output from the control terminal voltage detector to maintain the set target value And a voltage controller.

  If the drive signal supply circuit is configured as described above, the voltage between the control terminals of each switch constituting the switching circuit is maintained at a set target value regardless of the output voltage of the variable DC power supply unit. Can do. Even if the voltage between the control terminals is kept constant, the on-resistance of the switch will change if the current flowing through the switch (or the drain current if the switch is a MOSFET) changes, but the fluctuation range of the current flowing through the switch is taken into account. If the voltage between the control terminals is kept at an appropriate value, the on-resistance of each switch can be kept within a suitable range to prevent excessive conduction loss at each switch. When the value of the voltage applied from the variable DC power supply unit to the DC / RF conversion unit is changed, it is possible to prevent the on-resistance of each switch from being out of the proper range and causing excessive conduction loss in each switch. it can. Therefore, according to the present invention, the heat generated in each switch can be suppressed, and the heat sink provided for cooling each switch can be reduced in size.

  The second invention disclosed in the present application is applied to the first invention. In the present invention, the switching circuit is composed of a high-side switch and a low-side switch connected in series to each other and is variable. At least one leg connected between the output terminals of the DC power supply unit is provided. In this case, the drive signal supply circuit includes a transformer having a high-side switch secondary coil and a low-side switch secondary coil connected between the control terminals of the high-side switch and between the control terminals of the low-side switch, and the switches And a reference voltage generator for generating a reference voltage that gives a target value of the voltage between the control terminals. In this case, the voltage control unit between the control terminals includes an error amplification unit that calculates a deviation between the detection signal voltage output from the control terminal voltage detection unit and the reference voltage, an output voltage of the signal generator, and an output voltage of the error amplification unit. And a multiplier for inputting a multiplication output voltage including information of a multiplication value to the primary side of the transformer.

  The multiplication of the output voltage of the signal generator and the output voltage of the error amplifier can be performed by a multiplication element such as a multiplication IC or a double balanced mixer. When the output voltage of these multiplication elements has a level sufficient to generate a drive signal of a level necessary for driving the switch from the transformer, a multiplier is constituted only by the multiplication element. The output voltage of the multiplication element can be input to the transformer as the multiplication output voltage. On the other hand, when the multiplication element cannot generate a voltage of a level necessary for driving the switch, the output of the multiplication element needs to be amplified by a driver amplifier and input to the transformer. In this case, the multiplier includes a multiplication element and a driver amplifier that amplifies the output voltage of the multiplication element and outputs a multiplication output voltage. That is, the multiplier may be composed of only a multiplication element, or may be composed of a multiplication element and a driver amplifier that amplifies its output. The same applies to the multipliers used in other inventions described below.

  The third invention disclosed in the present application is also applied to the first invention. In the present invention as well, the switching circuit is composed of a high-side switch and a low-side switch connected in series to each other, and a variable DC power source is provided. At least one leg connected between the output terminals, and the drive signal supply circuit includes a secondary coil for the high side switch connected between the control terminals of the high side switch and between the control terminals of the low side switch. And a transformer having a secondary coil for the low-side switch, and a reference voltage generator for generating a reference voltage that gives a target value of the voltage between the control terminals of each switch. These configurations are the same as in the second invention, but in the present invention, the voltage control unit between the control terminals performs a calculation to divide the reference voltage by the detection signal voltage output from the voltage detection unit between the control terminals. And a multiplier for inputting a multiplication output voltage including information on a multiplication value of the output voltage of the signal generator and the output voltage of the divider to the primary side of the transformer.

  The fourth invention disclosed in the present application is applied to the second invention or the third invention. In the present invention, the voltage detection unit between the control terminals is configured to detect each voltage from the voltage across the primary coil of the transformer. The voltage information between the control terminals of the switch is obtained.

  The fifth invention disclosed in the present application is applied to the second invention or the third invention. In the present invention, the voltage detection part between the control terminals is based on the voltage across the secondary coil of the transformer. The voltage information between the control terminals of each switch is obtained.

  The sixth invention disclosed in the present application is applied to any one of the first to fifth inventions. In the present invention, the switching circuit includes a high-side switch and a low-side switch connected in series to each other. It consists of a half-bridge type switching circuit having only one leg configured and connected between the output terminals of the variable DC power supply unit.

  The seventh invention disclosed in the present application is applied to any one of the first to fifth inventions. In the present invention, the switching circuit includes a high-side switch and a low-side switch connected in series to each other. It is composed of a full-bridge type switching circuit that includes two legs that are configured and connected between output terminals of the variable DC power supply unit, and that has a configuration in which the two legs are connected in parallel.

  An eighth invention disclosed in the present application is applied to the first invention. In the present invention, a single switch in which a switching circuit is connected between output terminals of a variable DC power supply unit through a choke coil. The drive signal supply circuit includes a reference voltage generation unit that generates a reference voltage that provides a target value of the voltage between the control terminals of the switch. The control terminal voltage control unit includes an error amplifying unit that calculates a deviation between the detection signal voltage output from the control terminal voltage detecting unit and the reference voltage, an output voltage of the signal generator, and an output voltage of the error amplifying unit. And a multiplier that outputs a multiplication output voltage including information on the multiplication value of the multiplication circuit, and the multiplication output voltage is input as a drive signal between the control terminals of the switch of the switching circuit directly or through a transformer.

  A ninth invention disclosed in the present application is applied to the first invention. In the present invention, a single switch in which a switching circuit is connected between output terminals of a variable DC power supply unit through a choke coil. The drive signal supply circuit includes a reference voltage generation unit that generates a reference voltage that provides a target value of the voltage between the control terminals of the switch. In addition, the control terminal voltage control unit is a divider that performs an operation to divide the reference voltage by the detection signal voltage output from the control terminal voltage detection unit, and a multiplication value of the output voltage of the signal generator and the output voltage of the divider. And a multiplier that outputs a multiplication output voltage including the above information, and the multiplication output voltage is input as a drive signal directly or through a transformer between the control terminals of the switches of the switching circuit.

  The tenth invention disclosed in the present application is applied to any of the first to ninth inventions. In the present invention, the target value of the voltage between the control terminals of the switch is the switching circuit. It is set to a value suitable for suppressing the conduction loss caused by the switch to be configured below the set limit value.

  The eleventh invention disclosed in the present application is applied to the tenth invention. In the present invention, the target value of the voltage between the control terminals with respect to the parameter including information on the current flowing through the switch of the switching circuit. A target value calculation unit is provided for calculating. In this case, the reference voltage generator is configured to generate a reference voltage that gives the target value calculated by the target value calculator.

  As described above, if a target value calculation unit for calculating a target value of the voltage between the control terminals of the switch is provided for the parameter including information on the current flowing through the switch of the switching circuit, the current flows through each switch of the switching circuit. Since the target value of the voltage between the control terminals of each switch can be determined according to the current, the conduction loss generated in each switch can be effectively reduced regardless of the state of the load.

  As a parameter including information on the current flowing through the switch of the switching circuit, a detected value of the current output from the switching circuit may be used, or an impedance of a circuit viewed from the switching circuit may be used.

  According to the present invention, the voltage between the control terminals of each switch of the switching circuit is detected directly or indirectly in the drive signal supply circuit that supplies the drive signal to each switch constituting the switching circuit provided in the DC / RF converter. Between the control terminals of each switch detected from the detection signal output from the control signal output by the voltage detection unit between the control terminals and the voltage detection unit between the control terminals that outputs the detection signal voltage including information on the voltage between the control terminals of each switch Between the control terminals of each switch constituting the switching circuit, regardless of the output voltage of the variable DC power supply unit. By keeping the voltage at a set target value, the on-resistance of each switch can be kept within a suitable range to prevent excessive conduction loss in each switch. In, it is possible to prevent with a change of the voltage applied from the variable DC power supply unit to the DC / RF conversion section excessive conduction loss in the switch occurs. Therefore, according to the present invention, heat generated by each switch can be suppressed, and the heat sink provided for cooling each switch can be reduced in size. Further, since loss generated in the switching circuit can be reduced, the efficiency of the device can be improved.

  Further, in the present invention, a target value calculation unit for calculating a target value of the voltage between the control terminals is provided for a parameter including information on the current flowing through the switch of the switching circuit, and according to the current flowing through each switch of the switching circuit. When the target value of the voltage between the control terminals of each switch is determined, the voltage between the control terminals of each switch of the switching circuit of the DC / RF converter is set to an appropriate value regardless of the state of the load. Thus, it is possible to effectively reduce the conduction loss caused by each switch.

It is the block diagram which showed the structure of one Embodiment of this invention. It is the circuit diagram which showed the Example of this invention which actualized each part of embodiment shown in FIG. It is the block diagram which showed the structure of the feedback control system comprised in the voltage control part between control terminals used in the Example shown in FIG. (A) thru | or (E) are the waveform diagrams which showed the voltage waveform of each part observed by the simulation performed using the circuit of FIG. FIG. 3 is a circuit diagram showing a modification of the embodiment shown in FIG. 2. It is the circuit diagram which showed the other Example of this invention which actualized each part of embodiment of FIG. (A) thru | or (E) are the waveform diagrams which showed the voltage waveform of each part observed by the simulation which performed the waveform of the drive signal as a sine wave about the Example of FIG. (A) thru | or (E) is the wave form diagram which showed the voltage waveform of each part observed by the simulation which performed the waveform of the drive signal as a rectangular wave in the Example of FIG. FIG. 7 is a circuit diagram showing a modification of the embodiment shown in FIG. 6. It is the circuit diagram which showed the example which comprised the DC / RF conversion part by D class amplifier using a full bridge type switching circuit. FIG. 2 is a circuit diagram illustrating an example in which the DC / RF conversion unit is configured by a class E amplifier in the embodiment illustrated in FIG. 1. It is the block diagram which showed the structure of other embodiment of this invention. It is the block diagram which showed the structure of other embodiment of this invention. It is the circuit diagram which showed the structural example of the conventional high frequency power supply device. It is the graph which showed an example of the relationship between the input capacity | capacitance of MOSFET, feedback capacity | capacitance, output capacity | capacitance, and the drain-source voltage. (A) thru | or (E) are the waveform diagrams which showed the voltage waveform of each part observed by the simulation performed using the circuit of FIG. FIG. 15 is a waveform diagram for explaining that the gate voltage waveform changes even when the load resistance RL is changed while the DC voltage input to the DC / RF conversion unit is constant in the circuit shown in FIG. 14. It is the graph which showed an example of the relationship between the drain current of MOSFET, and drain-source resistance (ON resistance) by using the gate-source voltage as a parameter. It is the block diagram which showed the structure of the conventional high frequency power supply device. It is the circuit diagram which showed the structural example of the DC / RF conversion part used with the conventional high frequency power supply device.

Hereinafter, preferred embodiments of the present invention will be described in detail with reference to the drawings.
FIG. 1 shows the configuration of an embodiment of a high frequency power supply device according to the present invention. In FIG. 1, 1 is a high frequency power supply device, 2 is a load to which high frequency power is supplied from the high frequency power supply device 1, and 3 is a high frequency power supply. An impedance matching unit provided between the device 1 and the load 2.

  In the present embodiment, it is assumed that the load 2 is a plasma load. The plasma load is a load that generates plasma by high-frequency power supplied from the high-frequency power supply device 1 and is a plasma processing apparatus that performs processing such as etching by irradiating plasma on an object to be processed such as a semiconductor. The plasma load normally includes an electrode for generating plasma in the chamber, and generates plasma when high-frequency power is applied to the electrode. The impedance of the plasma load varies finely according to various conditions such as the magnitude of electric power applied between the electrodes, the pressure of the gas in the chamber, the flow rate of the gas supplied into the chamber, and the time for generating the plasma.

  The impedance matching unit 3 includes a variable capacitor or variable inductor that is an impedance variable element, and an operation mechanism that operates the variable impedance element using a motor as a drive source, so as to maximize the high-frequency power consumed by the load 2. In addition, the output impedance of the high-frequency power supply device 1 and the impedance when the load side is viewed from the high-frequency power supply device are adjusted to have a conjugate relationship. Since the output impedance of the high frequency power supply device is generally designed to be 50Ω, the impedance matching unit 3 operates so that the impedance viewed from the high frequency power supply device 1 when viewed from the load side is equal to 50Ω.

  The illustrated high frequency power supply device 1 includes a variable DC power supply unit 4, a DC / RF conversion unit 5 that converts a direct current output of the variable DC power supply unit 4 into a high frequency output, and a harmonic component from the output of the DC / RF conversion unit 5. The low-pass filter 6 to be removed, the traveling wave component of the high-frequency power supplied to the load 2 and the power detection unit 7 for detecting the reflected wave component of the high-frequency power reflected and returned by the load, the variable DC power supply unit 4 and the DC The control part 8 which controls the / RF conversion part 5 is provided.

  The variable DC power supply unit 4 may be of any type as long as it has a function of adjusting the output DC power in accordance with the output control signal Sdc given from the output control unit described later. The variable DC power supply unit 4 includes, for example, a rectification power supply unit that converts commercial frequency AC power into DC power, and a DC-DC converter that converts a DC output of the rectification power supply unit into DC power having an arbitrary magnitude. Is done. The DC-DC converter is, for example, an inverter that temporarily converts input DC power into AC power, a transformer that transforms the output of the inverter, a rectifier circuit that rectifies the output of the transformer, and a smoother output of the rectifier circuit. And a smoothing circuit for controlling the output of the inverter in accordance with an output control signal (PWM control signal) Sdc to output DC power at an arbitrary level.

  The DC / RF conversion unit 5 used in this embodiment includes a switching circuit 5A, a series resonance circuit 5B in which the output of the variable DC power supply unit 4 is supplied through the switching circuit 5A, and a primary coil connected in series to the series resonance circuit 5B. And a class D amplifier including the transformer 5C.

  In the present embodiment, like the DC / RF converter shown in FIG. 20, the switching circuit 5A is a half-bridge circuit having only one leg composed of a series circuit of a high-side switch Q1 and a low-side switch Q2. It is made up of. The high side switch Q1 and the low side switch Q2 constituting the switching circuit 5A are configured by semiconductor amplifying elements such as MOSFETs (field effect transistors) and bipolar transistors. In the following description, the high side switch Q1 and the low side switch Q2 are Assume that each of them is composed of MOSFETs FET1 and FET2.

  A switch composed of a semiconductor amplifying element such as a MOSFET is turned on in an active region when a drive signal having a peak value higher than a threshold level is applied between its control terminals (in the case of a MOSFET, between a gate and a source). Then, the ON operation is performed in the saturation region while a drive signal having a peak value equal to or higher than a certain value exceeding the threshold level is applied. In the illustrated DC / RF converter 5, the high-side switch and the low-side switch of the switching circuit 5 </ b> A are alternately turned on to resonate the series resonance circuit 5 </ b> B, and the DC output of the variable DC power supply unit 4 is changed to high-frequency AC. Convert to output. The output of the DC / RF conversion unit 5 can be appropriately adjusted by adjusting the output of the variable DC power supply unit 4.

  The low-pass filter 6 is a well-known L-type filter composed of an inductor and a capacitor, removes harmonic components from the output of the DC / RF converter 5, and outputs the high-frequency output from the DC / RF converter 5. Only the desired frequency component is passed.

  The power detector 7 is composed of a directional coupler, and the traveling wave component of the high frequency power supplied from the DC / RF converter 5 to the load 2 through the low pass filter 6 and the high frequency reflected by the load 2 and returned. A reflected wave component of power is detected, and a traveling wave component detection signal Pf and a reflected wave component detection signal Pr are output.

  The control unit 8 that controls the variable DC power supply unit 4 and the DC / RF conversion unit 5 includes an output control unit 9 and a drive signal supply circuit 10. The output control unit 9 supplies the variable DC power supply unit 4 with a predetermined output control signal Sdc, thereby causing the variable DC power supply unit 4 to keep the traveling wave component of the high-frequency power detected by the power detection unit 7 at a set value. A variable DC power supply unit for controlling the high-frequency output control to be controlled and the reflected wave component to be below a specified value when the reflected wave component of the high-frequency power detected by the power detection unit 7 exceeds a specified value (allowable value). The reflection protection control for suppressing the output of 4 is performed. The output control signal Sdc can take an appropriate form according to the configuration of the variable DC power supply unit 4. When the variable DC power supply unit 4 is composed of a rectifying power supply unit and a DC-DC converter, the output control signal is a signal given to the control terminal of the switch of the converter for PWM control of the output of the converter, It is a pulse waveform signal that repeats a high level state and a low level state with a duty ratio.

  The drive signal supply circuit 10 used in this embodiment includes a signal generator 10A, a transformer 10B, a control terminal voltage detection unit 10C, a reference voltage generation unit 10D, and a control terminal voltage control unit 10E. .

  The signal generator 10 </ b> A is a drive signal generation source applied to each switch of the switching circuit 5 </ b> A of the DC / RF converter 5, and is a sine wave or rectangular wave voltage equal to the frequency of the high frequency output output from the DC / RF converter 5. It consists of an oscillator that generates a signal.

  When the switching circuit 5A includes a high-side switch Q1 and a low-side switch Q2 connected in series as shown in FIG. 20, the transformer 10B shown in FIG. 1 is connected to the control terminal from the signal generator 10A. One primary coil to which an AC signal is input through the inter-voltage controller 10E, and a high-side switch secondary coil and a low-side connected in parallel between the control terminals of the high-side switch Q1 and the control terminal of the low-side switch Q2, respectively. A secondary coil for the switch. The secondary coil for the high side switch and the secondary coil for the low side switch are wound so as to induce a voltage that is 180 degrees different in phase, and between the control terminals of the high side switch Q1 and the control terminal of the low side switch Q2. An AC voltage having a phase difference of 180 ° is applied.

  The high-side switch Q1 and the low-side switch Q2 are alternately turned on using the positive half-wave voltage of the AC voltage applied between the respective control terminals as drive signals S1 and S2. When the high side switches Q1 and Q2 are alternately turned on, a resonance current flows from the variable DC power supply unit 4 to the series resonance circuit 5B through the switching circuit 5A, and the DC power output from the variable DC power supply unit 4 is high frequency. Converted to electric power.

  The inter-control-terminal voltage detection unit 10C directly or indirectly detects the voltage between the control terminals of each switch constituting the leg of the switching circuit, and generates a detection signal voltage including information on the voltage between the control terminals of each switch. This is the output part. The detection unit 10C is configured to obtain information on the voltage between the control terminals of each switch from the voltage across the primary coil of the transformer 10B, or to control each switch from the voltage across the secondary coil of the transformer 10B. It can be configured to obtain information on the voltage between terminals. In this embodiment, since the voltage applied between the control terminals of the high-side switch Q1 and the low-side switch Q2 is an AC voltage, the inter-control-terminal voltage detection unit 10C is configured such that the voltage on the primary side or the secondary side of the transformer 10B Or the average value of the rectified output of the voltage on the primary side or the voltage on the secondary side of the transformer 10B.

  The reference voltage generator 10D generates a reference voltage that gives a target value of the voltage between the control terminals of each switch of the switching circuit 5A, and is a constant voltage value corresponding to the target value of the voltage between the control terminals of each switch. To generate a regulated DC voltage.

  The control terminal voltage control unit 10E is a control unit that controls the voltage between the control terminals of each switch detected from the detection signal output from the control terminal voltage detection unit 10C to maintain a set target value. The voltage control unit between control terminals includes, for example, an error amplifying unit that calculates a deviation between a detection signal voltage output from the control terminal voltage detection unit 10C and a reference voltage generated by the reference voltage generation unit 10D, and an output voltage of the signal generator And a multiplier that multiplies the output voltage of the error amplifying unit and inputs the obtained multiplied output voltage to the primary side of the transformer 10B. Multiplication of the output voltage of the signal generator and the output voltage of the error amplifier can be performed by a multiplication element such as a multiplication IC or a double balanced mixer. Since these multiplication elements generally cannot generate an output necessary for driving a switch such as a MOSFET, a normal multiplier includes a multiplication element and a driver amplifier that amplifies the output of the multiplication element. Composed.

  When supplying high-frequency power from the high-frequency power supply device 1 shown in FIG. 1 to the load 2, the operation of matching the output impedance of the high-frequency power supply device 1 with the impedance viewed from the output end of the high-frequency power supply device 1. Is always performed by the impedance matching unit 3. As described above, the impedance matching unit 3 is configured to adjust the impedance when the load side is viewed from the input end of the impedance matching unit by controlling the variable capacitor or the variable inductor, which is an impedance variable element, with a motor. Therefore, impedance matching cannot be achieved instantaneously. Normally, impedance matching requires 100 msec to several sec. Since reflection occurs at the load until impedance matching is achieved, the reflected wave power returning from the load 2 to the DC / RF conversion unit 5 increases, and the switch constituting the switching circuit of the DC / RF conversion unit 5 is used. The flowing current increases and the conduction loss caused by the switch increases.

  When the load 2 for supplying high-frequency power is a plasma load, the impedance of the load is unstable, and depends on conditions such as applied power, pressure of gas in the chamber, flow rate of gas supplied into the chamber, and processing time. The load impedance changes finely. For this reason, an impedance mismatch state frequently occurs at the input unit of the impedance matching unit 3, and the power reflected at the input unit of the impedance matching unit returns to the high frequency power supply device. Therefore, the switching circuit of the DC / RF conversion unit 5 is A lot of conduction loss occurs in the switch to be configured. In particular, when the impedance seen from the DC / RF converter is close to a state where the impedance is close to a short circuit and close to a state where total reflection occurs, a large current flows through the switch of the switching circuit. A large conduction loss occurs due to the on-resistance of the switch.

  The conduction loss that occurs in the switch of the switching circuit 5A is determined by the product of the ON resistance of the switch (ON resistance) and the square of the current flowing through the switch. When the switch is composed of a semiconductor amplifying element such as a MOSFET, the on-resistance varies depending on the voltage between the control terminals of the switch. Therefore, the conduction loss generated in the switch varies depending on the magnitude of the voltage applied between the control terminals of the switch.

  When the current flowing through the switch is relatively small, the conduction loss can be reduced by reducing the on-resistance of the switch. However, when a large current can flow through the switch due to an increase in reflected wave power, etc. If the value is reduced, the current flowing through the switch increases, and the conduction loss increases (because it is proportional to the square of the energization current). Therefore, in order to reduce the conduction loss generated in the switch, it is necessary to keep the on-resistance of the switch at an appropriate value in consideration of the amount of current flowing through the switch.

  However, in this type of high-frequency power supply device, when the DC voltage input from the variable DC power supply unit 4 to the DC / RF conversion unit 5 is changed in order to adjust the high-frequency power applied to the load, the DC / RF conversion unit Since the DC voltage applied to each switch of the switching circuit 5 changes and the input capacitance and feedback capacitance of the semiconductor amplifying element (eg, MOSFET) constituting each switch change, the control terminal of each switch A problem arises in that the voltage applied to the switch changes, and the voltage between the control terminals of each switch deviates from an appropriate value as the current changes. When the voltage between the control terminals of each switch deviates from an appropriate value, the conduction loss generated in each switch increases and the temperature of each switch rises.

  Therefore, in the present invention, the drive signal supply circuit 10 detects the voltage between the control terminals of each switch of the switching circuit 5A directly or indirectly, and generates a detection signal voltage including information on the voltage between the control terminals of each switch. Detected from the detection signal output from the control terminal voltage detection unit 10C, the reference voltage generation unit 10D that generates a reference voltage that gives a target value of the voltage between the control terminals of each switch, and the control terminal voltage detection unit 10C And a control terminal voltage control unit 10E for controlling the voltage between the control terminals of each switch to be maintained at a set target value, so that the switching circuit is independent of the output voltage of the variable DC power supply unit 4. Keeping the voltage between the control terminals of each switch constituting 5A at a set target value, the ON resistance of each switch is suitable for preventing excessive conduction loss from occurring in each switch. And which make it possible to keep to a value in the range.

  In order to accurately control the voltage between the control terminals of the high side switch Q1 and the voltage between the control terminals of the low side switch Q2, the voltage between the control terminals of the high side switch Q1 and the low side switch Q2 is accurately controlled. However, it is preferable to individually control the voltages between the control terminals of both switches, but with this configuration, it is inevitable that the configuration of the control circuit becomes complicated. For this reason, in the present embodiment, the high-side switch and the low-side switch are detected by detecting the voltage across one of the two secondary coils of the transformer 10B or by detecting the voltage across the primary coil of the transformer 10B. The voltage between the control terminals of the high-side switch and the low-side switch is controlled collectively by obtaining information on the voltage between the control terminals of the high-side switch and controlling the voltage input to the transformer 10B based on this information. ing. Normally, there is no great variation in the characteristics of the high-side switch and the low-side switch, so that even if configured as in the present embodiment, no problem occurs.

  The Example which shows the more concrete structural example of each part in the case of comprising the DC / RF conversion part 5 of the high frequency power supply device 1 shown by FIG. 1 by the class D amplifier using the half-bridge type switching circuit 5A. It was shown in 2. In the embodiment shown in FIG. 2, the switching circuit 5A of the DC / RF converter 5 has one leg composed of a series circuit of a high side switch Q1 and a low side switch Q2 in parallel between the output terminals of the variable DC power source 4. It has a connected half-bridge configuration.

  In the switching circuit 5A shown in FIG. 2, the high-side switch Q1 and the low-side switch Q2 are MOSFETs FET1 and FET2, respectively, and the drain of the FET1 is connected to the positive output terminal of the variable DC power supply unit 4 and the source of the FET2 Are connected to the negative output terminal of the variable DC power supply unit 4, and the source of FET1 and the drain of FET2 are connected in common, so that FET1 and FET2 are connected in series. The output voltage of the variable DC power supply unit 4 is applied to both ends of the circuit.

  In this example, the common connection point between the source of FET1 and the drain of FET2 and the source of FET2 are the output terminals of the switching circuit 5A, and the common connection point between the source of FET1 and the drain of FET2 is the inductor Lr. It is connected to one end of a series resonance circuit 5B composed of a series circuit with a capacitor Cr. The other end of the series resonant circuit 5B is connected to one end of the primary coil Wp of the transformer 5C, and the other end of the primary coil Wp of the transformer is connected to the source of the FET2.

  In the example shown in FIG. 2, one end and the other end of the secondary coil Ws of the transformer 5C are output terminals of the DC / RF converter 5, and a high frequency output obtained between these output terminals is the inductor La and A low-pass filter 6 including a capacitor Ca is input. In FIG. 2, RL is connected between the output terminals of the low-pass filter 6 in order to simulate the operation of the circuit shown in FIG. 2 assuming that the impedance matching unit 3 shown in FIG. The load resistance is 50Ω.

  The drive signal supply circuit 10 shown in FIG. 2 includes a transformer 10B including one primary coil W1, a high-side switch secondary coil W21, and a low-side switch secondary coil W22. A secondary coil for switching W21 and a secondary coil for low side switching W22 are connected in parallel between the gate and source of FET1 and between the gate and source of FET2. The secondary coil W21 for the high side switch and the secondary coil W22 for the low side switch are wound so as to induce voltages having the same peak value and different in phase by 180 degrees, and control the FET 1 (high side switch Q1). An alternating voltage having a phase difference of 180 ° is applied between the terminals and between the control terminals of the FET 2 (low-side switch Q2).

  The FET1 and FET2 constituting the high-side switch Q1 and the low-side switch Q2, respectively, use the positive half-wave voltage of the AC voltage applied between the gate and source (between the control terminals) from the transformer 10B as the drive signals S1 and S2. The ON state is maintained while the drive signals S1 and S2 are equal to or greater than the threshold value, and the OFF state is established when the drive signals S1 and S2 drop below the threshold value. Since the drive signals S1 and S2 are 180 degrees out of phase, the FET1 and FET2 are alternately turned on at a timing different by 180 degrees. When FET1 and FET2 are alternately turned on, a resonance current flows from the variable DC power supply unit 4 to the series resonance circuit 5B through the switching circuit 5A, and DC power output from the variable DC power supply unit 4 is converted into high-frequency power. The

  In order to prevent the output of the variable DC power supply unit from being short-circuited due to a period in which FET1 and FET2 are turned on at the same time, the timing at which the drive signal S1 becomes less than the threshold (the timing at which the drive signal S1 disappears) The threshold between the timing when the signal S2 reaches the threshold (the timing when the driving signal S2 is generated) and the timing when the driving signal S2 becomes less than the threshold (the timing when the driving signal S2 disappears) and the driving signal S1 are the threshold. A fixed dead time (a period during which no drive signal is applied to either the high-side switch or the low-side switch) is set between the timing when the value is reached (the timing when the drive signal S1 is generated).

  2 includes a Schottky barrier diode Ds having a cathode connected to one end of the primary coil W1 of the transformer 10B and one end to the anode of the diode Ds. The other end of the signal generator 10A is grounded together with the other end of the signal generator 10A. The capacitor Cb is charged to the negative half-wave peak value of the voltage Vin at both ends of the transformer primary coil W1, and is connected to both ends of the capacitor Cb. The detection signal voltage Vf proportional to the peak value of the voltage Vin across the primary coil of the transformer 10B is generated at both ends of the capacitor Cb. Since the voltage Vin across the primary coil of the transformer 10B reflects the voltage between the gate and source of the FET1 and FET2, the voltage between the gate and source of the FET1 and FET2 from the detection signal voltage Vf appearing across the capacitor C1. Information on (voltage between control terminals) can be obtained.

  Further, the control terminal voltage control unit 10E includes an error amplification unit 10E1 for calculating a deviation between the detection signal voltage Vf output from the control terminal voltage detection unit 10C and the reference voltage Vr obtained from the reference voltage generation unit 10D, and a signal generation And an analog multiplier MUL that multiplies the output voltage E of the multiplier 10A and the output voltage of the error amplifier 10E1.

  The error amplifying unit 10E1 is composed of resistors R1 and R2 and is composed of an adder Ad that adds the reference voltage Vr and the detection signal voltage -Vf, an operational amplifier OP1, and resistors R3 and R4. The first inversion amplifier circuit Am1, in which the output voltage Vr-Vf of Ad is input to the inverting input terminal of the operational amplifier OP1, an operational amplifier OP2, resistors R5 and R6, and a capacitor C1 are included in the first inversion. The output of the amplifier Am1 is composed of a second inverting amplifier circuit Am2 input to the inverting input terminal of the operational amplifier OP2, and the voltage of the difference between the reference voltage Vr and the detection signal voltage Vf from the second operational amplifier circuit Am2. A voltage proportional to ΔV = Vr−Vf is output.

  The multiplier MUL includes a multiplication element that multiplies the output voltage E of the signal generator 10A and the output voltage of the error amplifying unit 10E1, and a driver amplifier that amplifies the output of the multiplication element. A multiplication output voltage of the output voltage E and the output voltage of the error amplifying unit 10E1 is input to the primary side of the transformer 10B as a voltage of a level necessary for driving the MOSFET.

  Note that r shown between the multiplier MUL and the transformer 10B in FIG. 2 is the resistance of the internal impedance of the signal generator 10A, the resistance of the internal impedance of the amplifier provided in the multiplier, and the wiring L is a combined inductance of the inductance component of the internal impedance of the signal generator 10A, the inductance component of the internal impedance of the amplifier provided in the multiplier, and the inductance component of the wiring.

  In the drive signal supply circuit shown in FIG. 2, the voltage Vin across the primary coil W1, the detection signal voltage Vf, the reference voltage Vr, and the input voltage ΔV (= Vr−Vf) of the error amplifying unit are functions of time. Suppose that these Laplace transforms are Vin (s), Vf (s), Vr (s) and ΔV (s), respectively. Note that s = jω (ω is an angular frequency).

  FIG. 3 is a block diagram illustrating a configuration of a feedback control system configured in the inter-control-terminal voltage controller 10E illustrated in FIG. In FIG. 3, H (s) is a transfer function that gives the relationship between the input voltage Vin (s) and the output voltage Vf (s) of the control terminal voltage detector 10C, and G1 (s) is the error amplifier. 10E1 is a transfer function that gives the relationship between the input voltage ΔV (s) = Vr (s) −Vf (s) of the amplifier circuit and the output voltage. G2 (s) is a transfer function that gives the relationship between the input of the multiplier MUL (the output of the error amplifier 10E1) and the output Vin (s) of the multiplier MUL (the voltage across the primary coil of the transformer 10B). is there.

From FIG. 3, the following equations (1) to (3) are established.
Vf (s) = H (s) * Vin (s) (1)
ΔV (s) = Vr (s) −Vf (s) (2)
Vin (s) = G1 (s) * G2 (s) * ΔV (s)
= G1 (s) * G2 (s) * {Vr (s) -Vf (s)}
= G1 (s) * G2 (s) * {Vr (s) -H (s) * Vin (s)} (3)
From equation (3)
{1 + G1 (s) * G2 (s) * H (s)} * Vin (s) = G1 (s) * G2 (s) * Vr (s) (4)
When Vin (s) is obtained from equation (4),
Vin (s) = [G1 (s) * G2 (s) / {1 + G1 (s) * G2 (s) * H (s)}] * Vr (s) (5)

The feedback control system shown in FIG. 3 controls the peak value (DC component) of the AC voltage at both ends of the primary coil of the transformer to be constant. Therefore, when s = 0 in the equation (5), the following equation: Is established.
Vin (0) = [G1 (0) * G2 (0) / {1 + G1 (0) * G2 (0) * H (0)}] * Vr (0) (6)
Here, when s = 0, G1 (0) >> G2 (0), G1 (0) * G2 (0) * H (0) >> 1 holds, and G1 reaches as high a frequency as possible. When (s) >> G2 (s) and G1 (s) * G2 (s) * H (s) >> 1 are established, the following expression is established.
Vin (s) ≒ Vr (s) / H (s) (5) '
Vin (0) ≒ Vr (0) / H (0) (6) '
Here, since H (0) can be regarded as being constant and is not affected by the output voltage of the variable DC power supply unit, the input voltage Vin of the transformer 10B is determined from the variable DC power supply unit to DC by the equation (6) ′. Regardless of the DC voltage input to the / RF converter, it is understood that the voltage is controlled to be equal to a constant voltage value determined by the product of the reference voltage Vr (0) and 1 / H (0) t. . Also, at a frequency where G1 (s) * G2 (s) * H (s) >> 1 holds, even if the DC voltage Vdc input to the DC / RF converter 5 and the load resistance RL change, the response is good. The input voltage Vin can be controlled to be constant.

  In order to confirm the effect of the present invention, the inventor determines whether the DC / RF conversion unit does not perform the control for keeping the voltage between the control terminals of the switch of the DC / RF conversion unit constant or not. A simulation for observing changes in the gate-source voltages of the FET1 and FET2 when the DC voltage Vdc input to 5 was changed was performed using a computer.

  First, the DC voltage Vdc input to the DC / RF conversion unit 5 is changed using the high frequency power supply device shown in FIG. 14 corresponding to the high frequency power supply device shown in FIG. A simulation for observing the change in the voltage between the gate and the source of FET1 and FET2 in the case of being performed was performed. In performing this simulation, in the high frequency power supply device shown in FIG. 14, the amplitude of the output signal of the signal generator 10A was 20 V, the frequency was 13.56 MHz, and the signal waveform was a sine wave. As FET1 and FET2 constituting the high side switch and the low side switch, the input capacitance Ciss, the feedback capacitance Crss, and the output capacitance Coss change as shown in FIG. 15 with respect to the drain-source voltage VDS. The MOSFET shown is used. In FIG. 15, the unit of capacitance (capacitance) is farad, and the unit of drain-source voltage VDS is volt. FIG. 15 is quoted from the MOSFET data sheet published by the manufacturer of the FET. Further, the combined resistance r of the resistance component of the internal impedance of the signal generator 10A and the resistance component of the wiring is 0.5Ω, and the combined inductance L of the inductance component of the internal impedance of the signal generator 10A and the inductance component of the wiring is 20 nH. . The load resistance RL was 50Ω.

  In the high-frequency power supply device shown in FIG. 14, when the DC voltage Vdc input from the variable DC power supply unit 4 to the DC / RF conversion unit 5 is changed between 50 V and 300 V, FIGS. A voltage waveform as shown in Fig. 1 was observed. FIG. 16A shows the change of the direct-current voltage Vdc input from the variable DC power supply unit 4 to the DC / RF conversion unit 5 with respect to time T, and FIG. 16B shows the output of the high-frequency power supply device. The voltage Vout is shown with respect to time T. 16C shows the waveform of the output voltage E of the signal generator 10A, and FIGS. 16D and 16E show the gate-source voltages Va-Vb and Vc-Vd of the FET1 and FET2, respectively. It shows a change. Va−Vb is the potential difference between the potential Va of the terminal a connected to the gate of the FET 1 and the potential Vb of the terminal b connected to the source, and Vc−Vd is the potential Vc of the terminal c connected to the gate of the FET 1 and the terminal d connected to the source. This is a potential difference from the potential Vd. In FIGS. 16A to 16E, the unit of time T on the horizontal axis is μsec, and the unit of voltage on the vertical axis is volts. The same applies to the units of the horizontal and vertical axes in other simulation waveforms described below.

  From FIG. 16, the peak value of the gate-source voltage of the FET 1 and FET 2 is input to the DC / RF conversion unit even though the peak value of the signal generator 10 A, which is the signal source of the drive signals of the FET 1 and FET 2, is constant. It turns out that it fluctuates with the change of the input DC voltage (FIG. 16A).

  FIGS. 17A to 17C show the case where the DC voltage Vdc input to the DC / RF converter 5 is 300 “V” (constant) and the load resistance RL is 10Ω and 50Ω in the circuit of FIG. The result of having simulated the change of the gate-source voltage Va-Vb of FET1 at this time is shown. FIG. 17A shows the change of the DC voltage Vdc input from the variable DC power supply unit 4 to the DC / RF conversion unit 5 with respect to time T. FIG. 17B shows the signal generator 10A. The waveform of the output voltage E is shown. The waveform (a) in FIG. 6C shows the waveform of the gate-source voltage Va-Vb of the FET 1 when the load resistance RL is 10Ω, and the waveform (b) shows the gate-source voltage of the FET 1 when the load resistance RL is 50Ω. The waveform of the source-to-source voltage Va-Vb is shown. From these results, it can be seen that when the load resistance RL varies, the voltage between the gate and the source of the MOSFET varies. From this simulation result, when the load resistance RL is changed from 50Ω to 10Ω, the gate-source voltage decreases. As a result, the on-resistance of each FET increases and the gate-source voltage does not decrease. In comparison, it can be seen that the conduction loss of each FET increases.

  The on-resistance (drain-source resistance when the MOSFET is on) of the MOSFET depends on the drain current and the magnitude of the gate-source voltage. FIG. 18 shows an example of a characteristic curve showing the relationship among the on-resistance, drain current, and gate-source voltage of the MOSFET. FIG. 18 is quoted from the power MOSFET application note issued by Fuji Electric Co., Ltd. In this figure, the horizontal axis ID indicates the drain current, and the vertical axis RDS (on) indicates the drain when on. • Indicates source resistance (ON resistance). The parameter VGS indicates the gate-source voltage (drive signal) of the MOSFET. As can be seen from FIG. 18, the on-resistance RDS (on) of the MOSFET increases as the drain current ID increases and decreases as the gate-source voltage VGS increases.

  As described above, when the DC voltage input to the DC / RF converter 5 changes or the load impedance changes, the magnitude of the gate voltage of the MOSFET of the switching circuit of the DC / RF converter changes. As a result, the on-resistance of the MOSFET increases and the conduction loss of the MOSFET may increase, which may cause an excessive temperature rise in each switch. Therefore, in the conventional high frequency power supply device, it is necessary to use a large heat sink provided for the switch constituting the switching circuit, and the size of the device cannot be avoided.

  Next, regarding the embodiment of the present invention shown in FIG. 2, the change in the gate-source voltage of FET1 and FET2 when the DC voltage input from the variable DC power supply unit 4 to the DC / RF conversion unit 5 is changed. The results of the simulation performed for viewing are shown in FIGS. 4A shows a change in the DC voltage Vdc input from the variable DC power supply unit 4 to the DC / RF conversion unit 5 with respect to time T. FIG. 4B shows a high-frequency power supply device. The output voltage Vout is shown with respect to time T. 4C shows the waveform (sine wave) of the output voltage E of the signal generator 10A. FIGS. 4D and 4E show the gate-source voltages Va-Vb and Vc of the FET1 and FET2, respectively. -Vd changes.

  As is apparent from the results shown in FIGS. 4A to 4E, feedback control is performed to maintain the gate-source voltages of FET1 (high-side switch Q1) and FET2 (low-side switch Q2) at a target value. Thus, it was confirmed that the gate-source voltage of both FETs can be kept constant.

  As described above, if the voltage between the control terminals of each switch of the switching circuit of the DC / RF converter can be maintained at the target value, by setting the target value to an appropriate value, Since the ON resistance of each switch can be maintained at a value suitable for preventing excessive conduction loss from occurring in each switch, it is possible to prevent a situation in which the temperature of each switch is excessively increased. Therefore, it is not necessary to use a large heat sink as a heat sink for radiating heat from the switch constituting the switching circuit, and the apparatus can be miniaturized. Further, if the above control is performed, the loss generated in the DC / RF converter can be reduced, so that not only the efficiency of the apparatus can be improved, but also the temperature rise of each switch is increased. Since it can suppress, the effect that the lifetime of each switch can be extended is also acquired.

  In the embodiment shown in FIG. 2, the inter-control-terminal voltage detection unit 10C is configured to detect the peak value of the voltage across the primary coil of the transformer 10B. However, the peak value of the secondary voltage of the transformer 10B is detected. It is also possible to configure the control terminal voltage detector 10C as described above. For example, as shown in FIG. 5, the inter-control-terminal voltage detector 10C can be configured to detect the peak value of the voltage across the secondary coil W22 of the transformer.

  In the embodiment shown in FIG. 2, the control terminal voltage control unit 4E includes an error amplification unit 10E1 that calculates a deviation between the detection signal voltage output from the control terminal voltage detection unit 10C and the reference voltage, and a signal generator 10A. And the multiplier MUL that multiplies the output voltage of the error amplifying unit by driving the output voltage of the multiplier MUL between the control terminals of the switch (MOSFET in the above example) of the switching circuit 5A through the transformer. Although it is configured to input as a signal, the inter-control-terminal voltage control unit controls the voltage between the control terminals of each switch detected from the detection signal output by the inter-control-terminal voltage detection unit at a set target value. However, the present invention is not limited to the configuration shown in FIG.

  6 shows another embodiment of the present invention in which the configuration of the control terminal voltage controller 10E is different from the example shown in FIG. 2 in the embodiment shown in FIG. In the embodiment shown in FIG. 6, a divider DIV that performs an operation of dividing the reference voltage Vr generated by the reference voltage generator 10D by the detection signal voltage output by the inter-control terminal voltage detector 10C, and the signal generator 10A The inter-control-terminal voltage control unit 10E is configured by a multiplier MUL that multiplies the output voltage E by the output voltage of the divider DIV. Other configurations are the same as those of the embodiment shown in FIG.

In the embodiment shown in FIG. 6, the magnitude of the signal voltage generated by the signal generator 10A is E, the reference voltage generated by the reference voltage generator 10D is Vr, and both ends of the capacitor Cb of the control terminal voltage detector 10C. Vf is the magnitude of the detected voltage obtained by Vdiv, Vdiv is the magnitude of the output voltage of the divider DIV, Vmul is the magnitude of the output voltage of the multiplier MUL, and Vin is the peak value of the voltage across the primary coil of the transformer 10B. Then, the following formula is established.
Vf ≒ Vin (7)
Vdiv = Vr / Vf (8)
Vmul = E x Vdiv
= E x (Vr / Vf)
≒ E x (Vr / Vin) (9)

  In the embodiment shown in FIG. 6, since E and Vr are constant, the output Vmul of the multiplier is the voltage across the primary coil of the transformer 10B reflecting the voltage between the gate and source of FET1 and FET2. It is inversely proportional to the peak value Vin. That is, when the voltage peak value Vin across the primary coil of the transformer 10B is equal to the reference voltage Vr, since (Vr / Vin) = 1, Vmul≈E, and the voltage peak across the primary coil of the transformer 10B. When the value Vin is lower than the reference voltage Vr, since (Vr / Vin)> 1, Vmul> E. When the peak value Vin of the voltage across the primary coil of the transformer 10B is higher than the reference voltage Vr, Vmul <E because (Vr / Vin) <1. Accordingly, when the peak value Vin of the voltage across the primary coil of the transformer 10B rises above the target value, the voltage input to the transformer decreases according to the rate of increase, and the peak value of the voltage across the primary coil of the transformer 10B. When Vin falls below the target value, the voltage input to the transformer rises according to the rate of reduction, and the peak value Vin of the voltage across the transformer 10B approaches the target value.

  With respect to the embodiment of the present invention shown in FIG. 6, the waveform of the signal generated by the signal generator 10A is set as a sine wave (the waveform of the drive signal applied to the FET1 and FET2 is set as a sine wave), FIGS. 7A to 7E show the results of simulations performed to observe changes in the gate-source voltage of FET1 and FET2 when the DC voltage input to the RF converter 5 is changed. 7A shows a change in the DC voltage Vdc input from the variable DC power supply unit 4 to the DC / RF conversion unit 5 with respect to time T. FIG. 7B shows the high-frequency power supply device. The output voltage Vout is shown with respect to time T. FIG. 7C shows the waveform of the output voltage E of the signal generator 10A. FIGS. 7D and 7E show the changes in the gate-source voltages Va-Vb and Vc-Vd of the FET1 and FET2, respectively. Is shown.

  In the embodiment of the present invention shown in FIG. 6, the waveform of the signal generated by the signal generator 10A is set as a rectangular wave (the waveform of the drive signal applied to the FET1 and FET2 is set as a rectangular wave). 8A to 8E show the results of simulations performed to observe changes in the gate-source voltage of FET1 and FET2 when the DC voltage input to the / RF converter 5 is changed. . FIG. 8A shows the change of the DC voltage Vdc input from the variable DC power supply unit 4 to the DC / RF conversion unit 5 with respect to time T, and FIG. 8B shows the output voltage Vout of the high-frequency power supply device over time. Shown against T. FIG. 8C shows the waveform of a rectangular signal voltage E output from the signal generator 10A. FIGS. 8D and 8E show the gate-source voltages Va-Vb of FET1 and FET2, respectively. The change in Vc-Vd is shown.

  From the results shown in FIGS. 7A to 7E and FIGS. 8A to 8E, the gates of FET1 and FET2 can be obtained without performing error amplification as in the embodiment shown in FIG. When the primary voltage of the transformer 10B reflecting the source-to-source voltage rises from the target value, the level of the signal voltage input to the transformer is lowered according to the rise rate, and the primary voltage of the transformer 10B is lowered from the target value. It has been confirmed that the gate-source voltage of both FETs can be maintained at the target value also by performing control to increase the level of the signal voltage input to the transformer depending on the rate of decrease.

  In each of the above embodiments, the inter-control-terminal voltage detection unit 10C is configured to obtain information on the voltage between the control terminals of the FET1 and FET2 from the peak value of the voltage across the primary coil W1 of the transformer 10B. As shown in FIG. 9, the inter-control-terminal voltage detection unit 10C may be configured to obtain information on the voltage between the control terminals of each switch from the voltage at both ends of the secondary coil of the transformer 10B. In the example shown in FIG. 9, the control terminal voltage detector 10 </ b> C detects the peak value of the voltage across the secondary coil W <b> 22 of the transformer 10 </ b> B, and includes a detection signal including information on the voltage between the gate and source of the FET 2. It is configured to output Vc. The other configuration of the embodiment shown in FIG. 9 is the same as that of the embodiment shown in FIG.

  In each of the above-described embodiments, the switching circuit 5A of the DC / RF converting unit 5 is formed of a half bridge circuit including only one leg composed of a series circuit of the high-side switch Q1 and the low-side switch Q2. The high-side switch and the low-side switch are alternately turned on to convert the DC output of the variable DC power supply unit 4 into a high-frequency output. However, the present invention uses such a switching circuit. The present invention is not limited to this, and has a configuration in which at least one leg including a series circuit of a high-side switch and a low-side switch is connected, and the at least one leg is connected in parallel between the output terminals of the variable DC power supply unit 4. High frequency power supply using a DC / RF conversion unit comprising a class D amplifier having a switching circuit having a series resonance circuit It can be widely applied to location.

  For example, as shown in FIG. 10, there are two legs: a leg composed of a series circuit of a high-side switch Q1 and a low-side switch Q2, and a leg composed of a series circuit of a high-side switch Q3 and a low-side switch Q4. The present invention is also applied to a high-frequency power supply device using a DC / RF converter 5 that includes a full-bridge circuit type switching circuit 5A and that applies a voltage between output terminals of the switching circuit to the series resonance circuit 5B. can do. In the DC / RF converter 5 shown in FIG. 10, the high-side switches Q1 and Q4 in one set at the diagonal position of the full bridge circuit are turned on, and the other set in the diagonal position at the high-side switch Q1 and Q4. By alternately generating a state in which the side switches Q2 and Q3 are turned on, a resonance current is passed through the series resonance circuit 5B to convert the DC output of the variable DC converter 4 into a high frequency output.

  Further, according to the present invention, as shown in FIG. 11, the switching circuit 5A includes a single switch Q2 (FET 2) connected between the output terminals of the variable DC power supply unit 4 through the choke coil Lch, and both ends of the switch Q2. The DC / RF converter 5 is constituted by a class E amplifier which is constituted by a capacitor Cp connected in parallel to each other and configured such that the output voltage of the variable DC power supply 4 is applied to the series resonant circuit 5B through the choke coil Lch. It can also be applied when configured. In this case, as the transformer 10B for supplying a drive signal to the switch Q2 constituting the switching circuit, a transformer including one primary coil W1 and one secondary coil W2 is used. Other configurations of the embodiment shown in FIG. 11 are the same as those of the embodiment shown in FIG. In the configuration shown in FIG. 11, the drive signal is supplied to the switch Q2 through the transformer 10B. However, the configuration is such that the output of the multiplier MUL is directly applied between the gate and source of the switch Q2 without using the transformer. You can also In FIG. 11, the capacitor Cp is provided to moderate the rise of the drain-source voltage waveform of the switch Q2.

  As shown in FIG. 11, when the DC / RF conversion unit 5 is configured by a class E amplifier, error amplification is performed using the divider DIV and the multiplier MUL, as in the embodiment shown in FIG. Without reducing the level of the signal voltage input to the transformer according to the rate of increase when the primary voltage of the transformer 10B reflecting the gate-source voltage of the switch Q2 (FET2) rises above the target value, When the primary voltage of 10B drops below the target value, control is performed to keep the voltage between the control terminals of the FET 2 constant by performing control to increase the level of the signal voltage input to the transformer according to the rate of decrease. The voltage control unit 10E1 between the control terminals can be configured.

  In the above embodiment, the reference voltage generator 10D is configured to generate a constant reference voltage. However, the voltage between the control terminals of the switch with respect to the parameter including information on the current flowing through the switch of the switching circuit 5A. It is also possible to provide a target value calculation unit that calculates the target value of the reference voltage generation unit 10D so as to generate a reference voltage that gives the target value calculated by the calculation unit. As a parameter including information on the current flowing through the switch of the switching circuit 5A, the current itself flowing through the switching circuit can be used, or a load-side impedance that is an impedance when the load side is viewed from the switching circuit 5A can be used.

  FIG. 12 shows an embodiment in which a target value of the voltage between the control terminals of the switch is calculated for a parameter including information on the current flowing through the switch of the switching circuit 5A, and a reference voltage that gives the calculated target value is generated. The structure of a form is shown. In this embodiment, from the traveling wave power detection signal Pf and the reflected wave power detection signal Pr output by the power detection unit 7 and the impedance between the switching circuit 5A and the output terminal of the high frequency power supply device, the switching circuit 5A to the load A load-side impedance detection unit 10 that detects the impedance viewed from the load side as a load-side impedance, and calculates a target value of a voltage between the control terminals of each switch of the switching circuit with respect to the load-side impedance detected by the detection unit 10 The target value calculation unit 10G is provided, and the target value calculated by the target value calculation unit 10G is given to the reference voltage generation unit 10D.

  In the embodiment shown in FIG. 12, the load-side impedance detection unit 10F calculates the impedance viewed from the output terminal of the high-frequency power supply device 1 from the traveling wave power detection signal Pf and the reflected wave power detection signal Pr, By combining this impedance and the impedance of the circuit between the switching circuit 5A and the output terminal of the high-frequency power supply device, the load-side impedance that is the impedance of the switching circuit 5A viewed from the load side is detected.

  The target value calculation unit 10G calculates the target value of the voltage between the control terminals of each switch of the switching circuit 5A with respect to the load side impedance detected by the load side impedance detection unit 10F. The calculation of the target value is performed by creating a map that gives a relationship between the load side impedance and the target value based on experiments, and the map is calculated with respect to the load side impedance detected by the load side impedance detection unit 10F. This can be done by searching.

  The reference voltage generator 10D generates a reference voltage having a voltage value that gives the target value calculated by the target value calculator 10G. The other configuration of the high frequency power supply device shown in FIG. 12 is the same as that of the embodiment shown in FIG.

  FIG. 13 shows an embodiment in which a target value of the voltage between the control terminals of the switch is calculated for a parameter including information on the current flowing through the switch of the switching circuit 5A, and a reference voltage that gives the calculated target value is generated. The other structure of a form is shown. In this embodiment, a current detection unit 10H that detects a current flowing through the switching circuit 5A, and a target value of a voltage between control terminals of each switch of the switching circuit with respect to the current detected by the detection unit, using a map. A target value calculation unit 10G for calculation is provided, and the target value calculated by the target value calculation unit 10G is given to the reference voltage generation unit 10D. The reference voltage generation unit 10D generates a reference voltage having a voltage value that gives the target value calculated by the target value calculation unit 10G, as in the embodiment shown in FIG. Also in this case, the reference voltage generator 10D is configured to generate a reference voltage having a voltage value that gives the target value calculated by the target value calculator 10G.

  As in the embodiment shown in FIG. 12 or FIG. 13, a target value calculation unit 10G is provided that calculates a target value of the voltage between the control terminals of the switch for a parameter including information on the current flowing through the switch of the switching circuit 5A. When the reference voltage that gives the target value calculated by the target value calculation unit is generated from the reference voltage generation unit 10D, each switch is controlled according to the current flowing through each switch of the switching circuit 5A. Since the target value of the voltage between the terminals can be determined, it is possible to effectively reduce the conduction loss that occurs in each switch regardless of the state of the load.

DESCRIPTION OF SYMBOLS 1 High frequency power supply device 2 Load 3 Impedance matching device 4 Variable DC power supply part 5 DC / RF conversion part 5A Switching circuit 5B Series resonance circuit 5C Transformer 6 Low-pass filter 7 Power detection part 8 Control part 9 Output control part 10 Drive signal supply circuit 10A Signal generator 10B Transformer 10C Control terminal voltage detection unit 10D Reference voltage generation unit 10E Control terminal voltage control unit 10F Load side impedance detection unit 10G Target value calculation unit 10H Current detection unit Q1 High side switch Q2 Low side switch FET1 High side switch MOSFET comprising
FET2 MOSFET constituting the low-side switch

Claims (11)

A switching circuit having a variable DC power supply unit capable of controlling a DC output and a switch composed of a semiconductor amplifying element is provided, and the DC output of the variable DC power supply unit is converted into a high frequency output by a switching operation of the switching circuit. A DC / RF converter and a signal generator for generating a sine wave or rectangular wave voltage signal having a frequency equal to the output frequency of the DC / RF converter, and converting the output of the variable DC power source to a high frequency output A high-frequency power supply device including a drive signal supply circuit for supplying a drive signal between control terminals of the switch of the switching circuit in synchronization with the output signal of the signal generator so as to cause the switching circuit to perform a switching operation for ,
The drive signal supply circuit includes:
A voltage between control terminals that directly or indirectly detects a voltage between control terminals of each switch of the switching circuit and outputs a detection signal voltage including information on a voltage between the control terminals of each switch; and
A voltage control unit between control terminals that controls the voltage between the control terminals of each switch detected from the detection signal output by the voltage detection unit between the control terminals to maintain a set target value; and
A high frequency power supply device comprising: The switching circuit includes at least one leg configured by a high-side switch and a low-side switch connected in series with each other and connected between output terminals of the variable DC power supply unit,
The drive signal supply circuit includes a transformer having a secondary coil for a high side switch and a secondary coil for a low side switch connected between the control terminals of the high side switch and between the control terminals of the low side switch, A reference voltage generator for generating a reference voltage that gives a target value of the voltage between the control terminals,
The voltage control unit between the control terminals includes an error amplification unit that calculates a deviation between the detection signal voltage output from the control terminal voltage detection unit and the reference voltage, an output voltage of the signal generator, and an error amplification unit. The high frequency power supply device according to claim 1, further comprising a multiplier that inputs a multiplication output voltage including information of a multiplication value with the output voltage to a primary side of the transformer. The switching circuit includes at least one leg configured by a high-side switch and a low-side switch connected in series with each other and connected between output terminals of the variable DC power supply unit,
The drive signal supply circuit includes a transformer having a high-side switch secondary coil and a low-side switch secondary coil connected between the control terminals of the high-side switch and between the control terminals of the low-side switch, and the control of the switch A reference voltage generator for generating a reference voltage that gives a target value of the voltage between the terminals,
The voltage control unit between control terminals includes a divider that performs an operation of dividing the reference voltage by a detection signal voltage output by the voltage detection unit between control terminals, an output voltage of the signal generator, and an output voltage of the divider The high frequency power supply device according to claim 1, further comprising: a multiplier that inputs a multiplication output voltage including information on a multiplication value of the first to the primary side of the transformer.   4. The high frequency power supply device according to claim 2, wherein the inter-control-terminal voltage detection unit is configured to obtain information on a voltage between control terminals of each switch from a voltage between both ends of the primary coil of the transformer.   4. The high frequency power supply device according to claim 2, wherein the inter-control-terminal voltage detection unit is configured to obtain information on a voltage between control terminals of each switch from a voltage between both ends of a secondary coil of the transformer. .   The switching circuit is composed of a high-side switch and a low-side switch connected in series with each other, and includes a half-bridge type switching circuit having only one leg connected between output terminals of the variable DC power supply unit. The high frequency power supply device according to any one of claims 1 to 5.   The switching circuit includes two legs connected to each other between the output terminals of the variable DC power supply unit, the high-side switch and the low-side switch connected in series with each other, and the two legs are connected in parallel. The high frequency power supply device according to claim 1, comprising a full bridge type switching circuit having a configuration. The switching circuit is configured by a single switch connected between output terminals of the variable DC power supply unit through a choke coil,
The drive signal supply circuit includes a reference voltage generation unit that generates a reference voltage that gives a target value of a voltage between control terminals of the switch,
The voltage control unit between the control terminals includes an error amplification unit that calculates a deviation between the detection signal voltage output from the control terminal voltage detection unit and the reference voltage, an output voltage of the signal generator, and an error amplification unit. A multiplier for outputting a multiplication output voltage including information on a multiplication value with the output voltage,
The high-frequency power supply device according to claim 1, wherein the multiplication output voltage is input as a drive signal directly or through a transformer between control terminals of the switch. The switching circuit is configured by a single switch connected between output terminals of the variable DC power supply unit through a choke coil,
The drive signal supply circuit includes a reference voltage generation unit that generates a reference voltage that gives a target value of a voltage between control terminals of the switch,
The voltage control unit between control terminals includes a divider that performs an operation of dividing the reference voltage by a detection signal voltage output by the voltage detection unit between control terminals, an output voltage of the signal generator, and an output voltage of the divider A multiplier for outputting a multiplication output voltage including information on a multiplication value of
The high-frequency power supply device according to claim 1, wherein the multiplication output voltage is input as a drive signal directly or through a transformer between control terminals of the switch.   The target value of the voltage between the control terminals of the switch is set to a value suitable for suppressing a conduction loss generated in the switch constituting the switching circuit below a set limit value. The high frequency power supply device described in 1. A target value calculation unit for calculating a target value of the voltage between the control terminals for a parameter including information on the current flowing through the switch of the switching circuit;
The high-frequency power supply device according to claim 10, wherein the reference voltage generation unit is configured to generate a reference voltage that gives a target value calculated by the target value calculation unit.

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