JP2015053746A - Resonant dc/dc converter and multiphase resonant dc/dc converter - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a resonant DC/DC converter that implements improved output controllability while suppressing an increase in the number of components.SOLUTION: A resonant DC/DC converter 20 includes a transformer including a primary coil L1 and a secondary coil L3, and rectification diodes connected to the secondary coil L3. The rectification diodes are provided with an assist circuit 22 for assisting a resonant operation, and an LC filter connected subsequently to the assist circuit 22. The assist circuit 22 comprises a capacitor C2 and an inductor L5. In a multiphase resonant DC/DC converter, outputs are coupled in the assist circuit 22 or the LC filter and a phase difference between multiple phases is changed to control an output voltage.

Description

  The present invention relates to a resonant DC / DC converter and a multiphase resonant DC / DC converter, and more particularly to output control.

  A DC / DC converter that converts a DC voltage is used in an electric vehicle such as a hybrid vehicle or an electric vehicle, an industrial robot, a machine tool, and a power machine using an electric motor such as an elevator. The DC / DC converter boosts or steps down a DC voltage from a power supply source to a desired voltage, and drives an electric vehicle or a power machine.

  As the DC / DC converter, there is a resonance type converter using electromagnetic induction and resonance. Patent Document 1 describes a one-stone current resonance type DC / DC converter. The current flowing from the DC input power source to the resonance inductance is switched by the semiconductor switch element, and the resonance circuit composed of the resonance inductance and the resonance capacitor is resonated. A voltage based on the resonance and electromagnetic induction appears on the resonance inductance and the output voltage of the DC input power supply is applied to the primary side of the high-frequency transformer, and a load voltage is output from the secondary side of the high-frequency transformer. .

  Patent Document 2 describes a resonance type LLC converter.

Japanese Patent Laid-Open No. 5-260745 US Pat. No. 6,344,795

  In general, a resonance type DC / DC converter tends to have a complicated and expensive circuit configuration, and thus it is required to improve output controllability while suppressing an increase in the number of components as much as possible.

  An object of the present invention is to provide a resonant DC / DC converter capable of improving output controllability while suppressing an increase in the number of parts.

  The present invention provides a transformer including a primary side coil connected to a DC power source and a secondary side coil connected to an output load, a switch element connected to the primary side coil, and a parallel connection to the switch element. A resonant DC / DC converter including a capacitor to be connected to the secondary coil, an auxiliary circuit connected to the rectifier circuit for assisting a resonance operation, and connected to the auxiliary circuit The auxiliary circuit includes a capacitor and an inductor.

  The present invention also provides a transformer including a primary side coil connected to a DC power source and a secondary side coil connected to an output load, a switch element connected to the primary side coil, and the switch element. A multi-phase resonant DC / DC converter comprising a plurality of resonant DC / DC converters including a capacitor connected in parallel and a rectifier circuit connected to the secondary coil, and constituting a multi-phase; The plurality of resonant DC / DC converters are connected to the rectifier circuit and include an auxiliary circuit for assisting a resonance operation and an LC filter connected to the auxiliary circuit, and the auxiliary circuit includes a capacitor and an inductor. The plurality of resonant DC / DC converters are characterized in that outputs are coupled to each other at a subsequent stage of the rectifier circuit.

  In one embodiment of the present invention, the auxiliary circuit and the LC filter are provided in common to the plurality of resonant DC / DC converters, and the plurality of resonant DC / DC converters constitute the auxiliary circuit. In the capacitor, outputs are coupled to each other.

  In another embodiment of the present invention, the auxiliary circuit is provided for each of the plurality of resonant DC / DC converters, and the LC filter is provided in common for the plurality of resonant DC / DC converters. The resonant DC / DC converter is characterized in that outputs are coupled to each other in a capacitor constituting the LC filter, and the inductors constituting the auxiliary circuit are magnetically coupled to each other.

  Yet another embodiment of the invention is characterized in that the phase difference between the polyphases varies between 0 ° and 180 °.

  According to the resonance type DC / DC converter of the present invention, output controllability can be improved by providing an auxiliary circuit in front of the LC filter in addition to the LC filter.

  Further, according to the multiphase resonant DC / DC converter of the present invention, an auxiliary circuit is provided in front of the transformer in addition to the LC filter on the secondary side of the transformer, thereby improving the output controllability. By combining the outputs on the secondary side with each other, an increase in the number of parts can be suppressed.

  Furthermore, according to the multiphase resonant DC / DC converter of the present invention, the output voltage can be easily changed by changing the phase difference between the polyphases, and the output is adaptively controlled according to the size of the load. it can.

It is a circuit block diagram of the resonance type DC / DC converter used as the premise of embodiment. It is a circuit block diagram of the resonance type DC / DC converter of embodiment. It is a graph which shows the time change of the capacitor | condenser C2 voltage of embodiment. It is a graph which shows the time change of the capacitor | condenser C2 voltage of a comparative example. It is a graph which shows the time change of the inductor L4 voltage of embodiment. It is a graph which shows the time change of the inductor L4 voltage of a comparative example. It is a circuit block diagram of other embodiment. It is a circuit block diagram of other embodiment. It is a circuit block diagram which is not output-coupled. It is a circuit block diagram of other embodiment. It is a circuit block diagram of other embodiment. It is a graph which shows the relationship between a phase and an output voltage. It is a graph which shows the relationship between an output current and a frequency. It is a graph which shows the relationship between output current, maximum voltage Vp, and transformer current Ip (1), Ip (2). It is a circuit block diagram of other embodiment. It is a circuit block diagram of other embodiment.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings.

<Structure of pre-requisite single-stone resonant DC / DC converter>
First, a one-stone resonance type DC / DC converter which is a premise of the present embodiment will be described. The resonance type DC / DC converter of this embodiment can be positioned as an extension or improvement of such a one-stone resonance type DC / DC converter.

  FIG. 1 shows a circuit configuration diagram of a one-stone resonance type DC / DC converter 10. The transformer includes a primary coil L1 and a secondary coil L3.

  Regarding the primary side of the transformer, one end of the primary side coil L1 is connected to the positive side of the DC power source Vin via the auxiliary resonance coil L2. The other end of the primary coil L1 is connected to one end of the switch element T1. A diode D1 is connected in reverse parallel to the switch element T1, and a resonance capacitor C1 is connected in parallel to the switch element T1. The other end of the primary coil L1 is connected to one end of the switch element T1 and is also connected to one end of the capacitor C1. The other end of the switch element T1 and the other end of the capacitor C1 are both connected to the negative electrode side of the DC power supply Vin. The switch element T1 is a switching transistor, for example, and is controlled by applying a control signal to the gate terminal.

  Regarding the secondary side of the transformer, the secondary coil L3 is connected to a smoothing capacitor C3 via a rectifier diode, and the capacitor C3 is connected to a load RL via a filter coil L4. The capacitor C3 and the coil (inductor) L4 remove noise with very high frequency components.

  When the switch element T1 switches from on to off, the potential of the capacitor C1 is zero, and the current I flowing through the primary coil at this time is Ion. When the switch element T1 is turned off in this state, the resonance operation is started by the primary coil and the capacitor C1, and the potential of the capacitor C1, that is, the potential of the switch element T1 rises. A negative voltage starts to be applied to the secondary coil L1, and the current I of the first coil L1 starts to decrease. When the negative current is reached, the potential of the capacitor C1 starts to decrease and finally becomes zero. Thereafter, since the negative current of the primary coil L1 flows through the diode D1, the potential of the capacitor C1 remains zero. When the switch element T1 is turned on, the current of the primary coil L1 increases again. When the switch element T1 is turned off when the switch element T1 is turned on again, the above operation is repeated.

  Accordingly, in the circuit of FIG. 1, resonance occurs when the gate terminal of the switch element T1 is driven at the same frequency and duty ratio as the above-described resonance occurs, and functions as a DC / DC converter.

  However, in the configuration in which the smoothing capacitor C3 is connected immediately after the rectifier diode as in the configuration of FIG. 1, the capacitor voltage is sufficiently large with respect to the resonance system constant. The DC voltage is equivalent to the above, and it simply has a smoothing function. In addition, since a high-capacity capacitor becomes a reactor component in high-frequency driving, it is necessary to separately connect a ceramic capacitor or the like in parallel with high-frequency characteristics.

<DC / DC converter of this embodiment>
FIG. 2 shows a circuit configuration of the DC / DC converter 20 of the present embodiment. The configuration on the primary side is the same as in FIG.

  On the other hand, regarding the circuit configuration on the secondary side, the inductor L4 is connected to the secondary coil L3, and further rectifier diodes D2 and D3 as rectifier circuits are connected. In the configuration of FIG. 1, a capacitor C3 and an inductor L4 are connected as an LC filter for removing high frequency noise after the rectifier diode D2. In the configuration of FIG. 2, the capacitor C3 is similarly removed to remove high frequency noise. In addition, an LC filter including the inductor L6 is connected, and an auxiliary circuit 22 for auxiliary amplification of resonance operation is connected between the preceding stage, that is, between the rectifier diodes D2 and D3 and the LC filter. The auxiliary circuit 22 includes a capacitor C2 and an inductor L5 as shown in the figure.

  In the circuit configuration of this embodiment, when an AC voltage is applied to the primary coil L1 of the transformer, an in-phase AC voltage is generated in the secondary coil L3 of the transformer. When the voltage becomes larger than the voltage of the capacitor C2 of the auxiliary circuit 22, the rectifier diode is biased in the forward direction, and energization is started. When energization is started, the capacitor C2 is charged, and the voltage of the capacitor C2 increases.

  Thereafter, when the voltage on the transformer side becomes lower than the voltage of the capacitor C2, the current is turned off because the rectifier diode is reverse-biased. On the other hand, the inductor L5 of the auxiliary circuit 22 always has a current that is substantially the same as the direct current flowing through the load RL or a sum of alternating currents smaller than the direct current. C2 is discharged and the voltage drops. This operation occurs in the same cycle as the resonance process, and the same voltage component as the resonance frequency is generated in the capacitor C2. This AC voltage generated in the capacitor C2 can be made a sufficiently large AC voltage near the maximum output of the converter by optimizing the constants of the capacitor C2 and the inductor L5.

  Further, since the phase of the AC voltage generated in the capacitor C2 includes a phase component opposite to the AC voltage applied to the primary side of the transformer, the coupling of the coils L1 and L3 is compared with the configuration shown in FIG. When the rate is 1, the voltage applied to the inductor L4 corresponding to the parasitic component increases, and the secondary current increases. This increases the output. That is, the resonance phenomenon is amplified by the auxiliary circuit 22 including the capacitor C2 and the inductor L5, and the capacitor C2 and the inductor L5 have an effect of amplifying the resonance phenomenon.

  It should be noted that the auxiliary circuit 22 in the present embodiment has not only a filter function similar to the LC filter connected to the subsequent stage, but also a function of auxiliary amplification of the resonance operation.

  Since the capacitor C2 has a small capacity, it can be handled by a ceramic capacitor or a film capacitor having excellent high-frequency characteristics, and is suitable for high-frequency driving.

  Further, since the inductor L5 also has a filter effect, the capacities of the capacitor C3 and the inductor L6 which are added purely as an LC filter can be reduced, and the entire secondary side circuit can be downsized.

  Next, the circuit configuration shown in FIG. 2 will be described more specifically.

The constants of the circuit elements in FIG. 2 are set as follows, for example.
L1 = L3 = 1.5 μH
L2 = 0.8μH
L4 = 0.1μH
L5 = L6 = 1 μH
C1 = 2.6nF
C2 = 30nF
C3 = 2μF
RL = 20Ω

  FIG. 3 shows a voltage waveform of the capacitor C2 of the auxiliary circuit 22 when the input voltage is 200 V and driving is performed at a frequency of 1.5 MHz. FIG. 4 shows a voltage waveform of the circuit configuration of FIG. 1 in which the capacitor C2 is a smoothing capacitor (2 μF) for comparison. In both figures, the horizontal axis represents time, and the vertical axis represents voltage.

  As shown in FIG. 3, in the circuit configuration of this embodiment, an alternating voltage of about 250 Vpp and 1.5 MHz is generated, but in FIG.

  Further, regarding the voltage of the inductor L4 obtained by analysis, FIG. 5 shows the case of the circuit configuration of the present embodiment, and FIG. 6 shows the case of the circuit configuration of FIG. In both figures, the horizontal axis represents time, and the vertical axis represents voltage. In the circuit configuration of the present embodiment, a voltage of about the maximum voltage Vp = 75V is applied, whereas in the circuit configuration of FIG. 1, the maximum voltage Vp = 54V to 56V. The side current is increased by 30 to 40%. As a result, while the load is the same 20Ω, an output voltage of only 195 V can be obtained with the circuit configuration of FIG. 1, whereas in the circuit configuration of this embodiment, the output is increased to 237 V and about 47%. . The resonance amplification effect of the auxiliary circuit 22 in this embodiment is clear.

  Although the output voltage fluctuates in the above description, it is of course possible to perform control to keep the output voltage constant.

<Resonant DC / DC Converter of Other Embodiment>
In the above embodiment, the single-phase resonance type DC / DC converter has been described. However, in the single-phase drive, the efficiency may be reduced at a low load. For this reason, it is conceivable that multi-phase driving is performed to suppress a decrease in efficiency at low load.

  Therefore, in the present embodiment, a multi-phase resonance type DC / DC converter capable of reducing the cost by using the multi-phase drive and sharing the secondary side components will be described. Even in multi-phase driving, resonance amplification can be performed by providing the auxiliary circuit 22 in the previous stage of the LC filter as in the above-described embodiment.

  7 and 8 show the circuit configuration of the multiphase resonant DC / DC converter of this embodiment. The circuit configuration shown in FIG. 2 is a circuit configuration in which two circuit configurations are connected in parallel for two-phase driving, and two-phase outputs are combined at a predetermined location on the secondary side output unit.

  The two-phase resonance type DC / DC converter 30 shown in FIG. 7 has a configuration in which the secondary sides of the two converters are coupled by a capacitor C <b> 2 that constitutes the auxiliary circuit 22. The secondary side of the first phase of the two-phase configuration includes a secondary coil L2, an inductor L4, and a rectifier diode, and the auxiliary circuit 22 is connected to the rectifier diode, and a capacitor for the LC filter is provided at the subsequent stage of the auxiliary circuit 22. C3 and the inductor L6 are connected. The secondary side of the second phase in the two-phase configuration also includes a secondary coil L2, an inductor L4, and a rectifier diode, and the first phase auxiliary circuit 22 is connected to the rectifier diode. That is, the auxiliary circuit 22 and the LC filter are shared by the first phase and the second phase, and the secondary side outputs of the first phase and the second phase are coupled to each other by the capacitor C2 of the auxiliary circuit 22.

  Further, in the two-phase resonant DC / DC converter 40 shown in FIG. 8, the secondary side of the two converters is coupled by the capacitor C3 of the LC filter, and the inductor L5 of the auxiliary circuit 22 in the preceding stage is magnetically coupled between the two phases. This is the configuration. Of the two-phase configuration, the secondary side of the first phase includes a secondary coil L2, a coil L4, and a rectifier diode, and the auxiliary circuit 22 is connected to the rectifier diode, and a filter capacitor C3 is provided downstream of the auxiliary circuit 22. And the coil L6 are connected. The secondary side of the second phase of the two-phase configuration also includes a secondary coil L2, a coil L4, and a rectifier diode, and an auxiliary circuit 22 'is connected to the rectifier diode. The auxiliary circuit 22 ′ includes a capacitor C 2 ′ and an inductor L 5 ′, and the inductor L 5 ′ is magnetically coupled to the inductor L 5 of the auxiliary circuit 22. The auxiliary circuit 22 'is connected to the capacitor C3. That is, the auxiliary circuits 22 and 22 'are provided for each phase, and the LC filter is shared by the first phase and the second phase. The secondary side outputs of the first phase and the second phase are coupled to each other by the capacitor C3 of the LC filter and the inductors L5 and L5 'of the auxiliary circuits 22 and 22'.

  First, the operation of the two-phase resonant DC / DC converter 30 shown in FIG. 7 will be described.

  FIG. 9 shows a circuit configuration when two converters are connected in parallel. FIG. 10 shows the same circuit configuration as that of FIG. In FIG. 9, the impedance Z after the capacitor C2 as seen from the rectifier diode side of the two converters is assumed to be Z = 2 × Zo. When the two converters have the same specifications and the same operation, the points of each part are the same potential, so no current flows even if they are connected, and can be regarded as the same circuit. Therefore, FIG. 9 and FIG. 10 are substantially the same circuit.

  Next, consider the case where only one side operates even if the two converters have the same specifications. In this case, the impedance Z after the capacitor C2 is Z = 2 × Zo in the case of FIG. 9, but the impedance Z is Z = Zo in the case of FIG. That is, paying attention to the configuration of FIG. 10 (that is, the configuration of FIG. 7), the impedance Z after the rectifier diode is Z = 2 × Zo when two converters operate simultaneously, whereas only one-side operation is performed. Z = Zo, and the impedance Z changes according to the operating conditions.

  Here, in the configuration of FIG. 10 (configuration of FIG. 7), there are an ON state in which the secondary rectifier diode is energized and an OFF state in which it is not energized. Become. For this reason, even if the same operation is performed between the two phases, the situation differs when the phase difference is 0 ° and the phase difference is 180 °.

  When the phase difference is 0 °, the two phases have the same operation, so the impedance Z after the rectifier diode is Z = 2 × Zo as described above. When the phase difference is 180 °, the other rectifier diode is energized when the other side is energized. Is equivalent to one-sided operation due to non-energization, and the impedance Z after the rectifier diode is Z = Zo as described above. However, since the influence of the other operation remains as a change in the initial state at the start of energization, it is slightly different from the pure one-side operation. Eventually, in the configuration of FIG. 10 (configuration of FIG. 7), as the phase difference changes from 180 ° to 0 °, the secondary-side impedance Z increases from Zo to 2 × Zo at most twice. .

  Since the configuration of FIG. 10 (configuration of FIG. 7) is a two-phase drive, high efficiency can be achieved by operating only a single phase at low load. Further, since the secondary side parts are shared after the rectifier diode, the number of parts can be reduced. Furthermore, since the impedance changes due to the phase difference between the two phases as described above, the output voltage is lowered and the phase difference between the two phases is 0 when the impedance is small when the phase difference between the two phases is 180 ° even under the same conditions. When the impedance increases at °, the output voltage increases. As a result, the output voltage can be controlled by the phase difference between the two phases, and the controllability is improved.

  The control of the output voltage by the phase difference is effective in the high output side region because the control range is wider when the secondary side changes in the low impedance region. Since EMI (electromagnetic interference) is dominant in the high output region, the ability to control the output at the same frequency on the high output side is effective in advance in the high frequency region compared to the control method by changing the frequency. It can be said that providing a simple filter is more preferable because EMI can be reduced.

  Next, the circuit configuration of FIG. 8 will be described.

  In the circuit configuration of FIG. 8, the effect of magnetically coupling the two phases by the inductors L5 and L5 'is added. When the inductors L5 and L5 ′ have a positive coupling coefficient (K), if two converter operations at a phase difference of 0 ° are caused to simultaneously flow the same current to the inductors L5 and L5 ′, the effect of magnetic coupling The inductors L5 and L5 ′ have an inductance component that is twice as much as the maximum.

  On the other hand, in the case of the phase difference operation of 180 ° close to the one-side operation, the inductor L5 'and the capacitor C2' are added to the inductor L5. However, since the capacity of the capacitor C3 is sufficiently larger than the capacity of the capacitor C2 (since the capacitor C3 is for filtering and the capacitor C2 is for impedance conversion, C2 << C3), the inductor L5 ′ And the influence of the capacitor C2 ′ can be ignored and can be regarded as a one-sided operation.

  Next, the circuit configuration shown in FIG. 8 will be described more specifically.

FIG. 11 shows the configuration of the DC / DC converter 40. The configuration of the secondary side first phase A in the figure also exists as the same configuration as the configuration of the secondary side second phase. The constants of each element are as follows, for example.
C1 = 2.6nF
C2 = C2 ′ = 30 nF
C3 = 2μF
L1 = L2 = 1.5 μH
L3 = 0.8μH
L4 = 0.1μH
L5 = L5 ′ = L6 = 1 μH
K (L5 / L5 ′) = 1

  FIG. 12 shows the relationship between the phase difference and the output voltage when driving at 1.7 MHz and changing the phase difference between the two phases. In the figure, the horizontal axis represents the phase difference and the vertical axis represents the output voltage. When the load resistance is as large as 500Ω, the influence on the primary side constant is small, so the change in the output voltage is small even if the phase difference between the two phases is changed. On the other hand, when the load resistance is as small as 10Ω, the output voltage changes by changing the phase difference between the two phases, and the impedance becomes larger and the output voltage becomes larger when the phase difference is 0 °. The reason why the voltage change is small under the condition of 500Ω is that the impedance change on the load side does not appear as the change in output voltage because the impedance on the primary side is smaller than the load impedance. By setting the impedance of the primary side and the transformer in accordance with the necessary load region, it becomes possible to always control the output voltage by the phase difference in the necessary load region.

  FIG. 13 shows the relationship between the drive frequency and the output current. In the figure, the horizontal axis represents output current, and the vertical axis represents drive frequency and duty. It can be seen that the control of the output current of 0 to 20 A can be achieved at the drive frequency of 1.46 MHz to 1.9 MHz under the conditions of the input voltage of 170 V to 230 V and the output voltage of 200 V. This control frequency range is a control frequency range that is significantly smaller than the control frequency range in the conventional LLC converter (single-phase drive). Further, all the soft switching operations are held in the range of FIG.

The phase difference between the two phases is 0 ° in the output range of 10A to 20A, and 180 ° below that. The phase difference is 180 ° in the medium load range of 10A or less. When the phase difference is 180 °, the input current from the power source is 180 ° between the two phases, and the fundamental wave cancels out. Therefore, 180 ° phase difference operation is desirable in the middle load range. In FIG. 12, the two-phase operation is shown in order to show the controllability at the time of the two-phase operation. However, the efficiency can be improved by the single-phase operation in the low load region. In summary,
Low load range: Single phase operation Medium load range: 2-phase operation with phase difference of 180 °
High load range: 2-phase operation with 0 ° phase difference
Is desirable.

  The frequency range at a phase difference of 180 ° is 1.3 MHz to 1.9 MHz, and the frequency range at a phase difference of 0 ° is 1.46 MHz to 1.9 MHz. The current ripple on the input side is small and the voltage ripple on the output side is small. From this, it can be said that the steady output range (medium load range) is controlled with a phase difference of 180 °, and the negative maximum output (during high load) is preferably controlled with a phase difference of 0 °.

  FIG. 14 shows the relationship between the maximum value Vp of the element voltage, the maximum value of the transformer current (primary side Ip (1), secondary side Ip (2)) and the output current. In the figure, the horizontal axis represents output current, and the vertical axis represents Vp, Ip (1), Ip (2). This is a result of using a SiC MOSFET as a switching element. However, since the element voltage is 1 kV or less and the maximum value of the element current that is the same as Ip (1) is 35 A or less, a Si MOSFET can also be used. . Alternatively, a GaN MOSFET may be used.

  As mentioned above, although embodiment of this invention was described, this invention is not limited to these, A various deformation | transformation is possible.

  For example, in the present embodiment, a two-phase DC / DC converter has been described. However, a configuration of three or more phases may be used, and another phase that is in phase with the first phase and about 180 ° with respect to the first phase. The control range can be expanded by switching the number of other phases that are phase differences. Further, as a three-phase configuration, for example, a two-phase operation can be performed in an intermediate load region, and an output voltage can be controlled by phase difference control between two operating phases.

  Further, when the circuit configurations of FIGS. 7 and 8 are compared, the circuit configuration of FIG. 7 requires fewer parts, but hard switching tends to occur when the phase difference control is executed. On the other hand, in the circuit configuration of FIG. 8, the number of parts is larger than that of FIG. 7, but soft switching can be realized when the phase difference control is executed as described above. Therefore, considering these characteristics, either FIG. 7 or FIG. 8 may be selected according to the application.

  Furthermore, although the circuit configuration of FIG. 2 is half-wave rectification, it can be extended to double-wave rectification.

  15 and 16 show circuit configurations in the case where the circuit configuration in FIG. 2 is extended to double-wave rectification. FIG. 15 shows the case of C coupling, and FIG. 16 shows the case of L coupling.

  In FIG. 15, the secondary coil L3 is composed of two coils L3-1 and L3-2, and the coil L4 is also composed of coils L4-1 and L4-2. The output ends of the coils L4-1 and L4-2 are both connected to the capacitor C2 via a rectifier diode. The point that the capacitor C2 and the inductor L5 form the auxiliary circuit 22 is the same as in FIG.

  In FIG. 16, the secondary coil L3 is composed of two coils L3-1 and L3-2, and the coil L4 is also composed of coils L4-1 and L4-2. The output end of the coil L4-1 is connected to the auxiliary circuit 22 via a rectifier diode, and the output end of the coil L4-2 is connected to the auxiliary circuit 22 'via a rectifier diode. The auxiliary circuit 22 includes a capacitor C2 and an inductor L5, the auxiliary circuit 22 'includes a capacitor C2' and an inductor L5 ', and the inductors L5 and L5' are magnetically coupled.

  10, 20, 30, 40 DC / DC converter, 22, 22 'auxiliary circuit.

Claims (5)

A transformer comprising a primary coil connected to a DC power source and a secondary coil connected to an output load;
A switch element connected to the primary coil;
A capacitor connected in parallel to the switch element;
A rectifier circuit connected to the secondary coil;
A resonant DC / DC converter comprising:
An auxiliary circuit connected to the rectifier circuit and assisting a resonance operation;
An LC filter connected to the auxiliary circuit;
And the auxiliary circuit is composed of a capacitor and an inductor. A transformer comprising a primary coil connected to a DC power source and a secondary coil connected to an output load;
A switch element connected to the primary coil;
A capacitor connected in parallel to the switch element;
A rectifier circuit connected to the secondary coil;
A multiphase resonant DC / DC converter having a plurality of resonant DC / DC converters comprising
The plurality of resonant DC / DC converters are:
An auxiliary circuit connected to the rectifier circuit and assisting a resonance operation;
An LC filter connected to the auxiliary circuit;
The auxiliary circuit comprises a capacitor and an inductor, and
The plurality of resonant DC / DC converters have outputs coupled to each other at a subsequent stage of the rectifier circuit. The multiphase resonant DC / DC converter according to claim 2,
The auxiliary circuit and the LC filter are provided in common for the plurality of resonant DC / DC converters,
A plurality of the resonance type DC / DC converters have outputs coupled to each other in the capacitor constituting the auxiliary circuit. The multiphase resonant DC / DC converter according to claim 2,
The auxiliary circuit is provided for each of the plurality of resonant DC / DC converters,
The LC filter is provided in common for the plurality of resonant DC / DC converters,
The plurality of resonance type DC / DC converters are characterized in that outputs are coupled to each other in a capacitor constituting the LC filter, and the inductors constituting the auxiliary circuit are magnetically coupled to each other. DC / DC converter. The multiphase resonant DC / DC converter according to any one of claims 2 and 3,
A multiphase resonant DC / DC converter characterized in that a phase difference between polyphases varies between 0 ° and 180 °.

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