JP2014108025A - Ipm type electric rotary machine - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide an IPM type electric rotary machine of low cost and high energy density by implementing high-efficiency rotational drive while reducing the amount of permanent magnets to be used.SOLUTION: The IPM type electric rotary machine comprises a rotor 12 in which permanent magnets 16 are embedded in a V shape, and a stator for housing the rotor, in which the number of slots for each pole and each phase is 2. In the case where the permanent magnets are present to a d-axis side, each of the permanent magnets within a range where a magnet magnetic flux is generated in a direction to cancel an armature magnetic flux at the d-axis side, is replaced with a flux barrier 17c of a small-permeability void. An outer end side inner surface 17b1 of a flux barrier 17b of a side bridge 30 of the rotor sets a midpoint 17b1m within a range where an included angle θ8 (electrical angle) with a (d) axis is 64.7° to 74.2°, and settles an included angle θ9 (mechanical angle) between an extension surface of a d-axis side inner surface 17b1d and a q-axis side inner surface 17b1q within a range of 0° to 37°.

Description

  The present invention relates to an IPM type electric rotating machine, and more particularly to an apparatus that realizes highly efficient rotational driving.

Electric rotating machines mounted on various devices are required to have characteristics corresponding to the mounted devices.
For example, in the case of a drive motor mounted on a hybrid vehicle (HEV: Hybrid Electric Vehicle) together with an internal combustion engine as a drive source, or mounted on an electric vehicle (EV: Electric Vehicle) as a single drive source, the motor rotates at a low speed. It is required to have a wide variable speed characteristic at the same time as generating a large torque in the region.

This type of vehicle is required to improve the energy conversion efficiency of each component, including the electric rotating machine, in order to improve the fuel efficiency. In particular, the in-vehicle electric rotating machine is expected to improve the efficiency in the normal range. It is rare. Furthermore, in-vehicle electric rotating machines are required to have a smaller and higher energy density structure from the viewpoints of installation space restrictions and weight reduction.
By the way, in HEV and EV, generally, a low-speed rotation / low load region of an electric rotating machine is a regular region. For this reason, the ratio of contribution to the torque of the in-vehicle electric rotating machine is larger for the magnet torque than for the reluctance torque according to the magnitude of the armature current, and many high-magnetism permanent magnets are used for higher efficiency. Tend to use.
Because of this tendency, for electric rotating machines, neodymium magnets with a high residual magnetic flux density are placed inside the rotor core in order to improve the energy conversion efficiency, especially in the normal range of low-speed rotation and low load range. An IPM (Interior Permanent Magnet) type which is an embedded permanent magnet type synchronous motor is frequently used. In this IPM type electric rotating machine, a permanent magnet is embedded in the rotor so as to have a V-shape that opens toward the outer peripheral surface, thereby forming a magnetic circuit that can actively use reluctance torque in addition to magnet torque. Has been proposed (for example, Patent Document 1). Further, in the IPM type electric rotating machine, a structure including a flux barrier projecting from both outer end sides of the permanent magnet toward the outer peripheral surface of the rotor is proposed, and the flux barrier is made continuous with the outer peripheral surface side of the rotor. A structure (Patent Document 2) for releasing the structure and a structure for expanding the space of the flux barrier toward the outer peripheral surface of the rotor (Patent Document 3) have also been proposed.

JP 2006-254629 A Japanese Unexamined Patent Publication No. 2011-4480 JP 2012-29524 A

By the way, in recent electric rotating machines, permanent magnets containing rare earths such as Nd, Dy, Tb are often used to increase the magnetic force and heat resistance. Due to the instability, there is a growing need for higher efficiency while reducing the amount of rare earth used.
However, in the HEV and EV, the normal area of the electric rotating machine is the low-speed rotation / low load area. Therefore, in order to increase the magnet torque that contributes to the area, in the IPM type motor described in Patent Document 1, However, there is a tendency to increase the amount of use of high magnetic permanent magnets. This is a direction that hinders the solution of the problem of reducing the amount of rare earth used.
Moreover, in the IPM type electric rotating machine described in Patent Documents 2 and 3, since the flux barriers on both outer end sides of the permanent magnet are darkly expanded on the outer peripheral surface side of the rotor, In the meantime, the magnetic resistance increases, and the cogging torque or the like increases, preventing high-quality rotation.
Accordingly, an object of the present invention is to provide a low-cost and high-energy density electric rotating machine by realizing high-efficiency and high-quality rotating drive while reducing the amount of permanent magnets used.

  A first aspect of the invention relating to an IPM electric rotating machine that solves the above-described problems is a rotor that is embedded with a permanent magnet and rotates integrally with a drive shaft, and that the rotor is housed so as to be relatively rotatable. A stator that accommodates a coil in a slot between a plurality of teeth facing each other and functions as an armature, and is configured to have an electric rotating machine that has 2 slots per phase per pole, The permanent magnet is arranged in a V-shape that opens toward the outer peripheral surface of the rotor, and the permanent magnet extends to the d-axis side in the magnetic flux direction that coincides with the central axis of the permanent magnet for each magnetic pole formed by the permanent magnet. When the magnet is present, the permanent magnet in a range in which the magnet magnetic flux is generated in the direction to cancel the armature magnetic flux generated by the armature on the d-axis side is replaced with a gap having a small magnetic permeability, and the rotor An axial center on the d-axis of the outer peripheral surface of A parallel center adjustment groove is formed, and a pair of side adjustment grooves parallel to the shaft center are formed on both outer end sides of the permanent magnet on the outer peripheral surface, and the rotor is arranged from both outer end sides of the permanent magnet. A flux barrier projecting toward the outer peripheral surface, and between the outer end side inner surface of the flux barrier and the outer peripheral surface of the rotor, the q-axis side in the magnetic flux direction between the magnetic poles of the rotor and the d-axis The inner surface of the flux barrier is formed on the d-axis side inner surface on both sides of the middle point of the both end side corners located on the back side of the outer peripheral surface of the rotor. And the q-axis side inner surface, and when the included angle between the d axis and the straight line passing through the midpoint of the rotor shaft and the inner surface of the flux barrier outer side is 64, .7 ° ≦ θ8 (electrical angle) ≦ 74.2 ° satisfying the relationship, d-axis side The inner surface is extended in a direction parallel to the outer peripheral surface of the rotor from an intermediate point of the outer surface on the outer end side of the flux barrier, and the q-axis side inner surface is an extended surface of the d-axis side inner surface toward the q-axis side. When the included angle between them is θ9, the relationship of 0 ° <θ9 (mechanical angle) ≦ 37 ° is satisfied.

  According to a second aspect of the invention related to the IPM type electric rotating machine that solves the above problem, in addition to the specific matter of the first aspect, the included angle θ8 is 64.9 ° ≦ θ8 (electrical angle) ≦ 74. It is characterized by satisfying the 2 ° relationship.

As described above, according to the first aspect of the present invention, the permanent magnet in the range that generates the magnet magnetic flux in the direction of canceling the armature magnetic flux on the d-axis side is replaced with a gap having a small magnetic permeability. The magnet magnetic flux and the armature magnetic flux do not interfere (cancel) on the d-axis side, and the armature magnetic flux can be restricted from passing through the range. Therefore, the magnet magnetic flux that wastes the armature magnetic flux on the d-axis side can be eliminated, the reluctance torque can be effectively used together with the magnet torque, and the permanent magnet itself can be used while obtaining the torque more than before the replacement of the d-axis side permanent magnet. The amount can be reduced.
Furthermore, by replacing the permanent magnet with a gap, the magnet magnetic flux can be reduced, the induced voltage constant on the high speed rotation side can be reduced, and the output on the high speed rotation side can be improved. Further, the weight can be reduced and the inertia can be reduced.
Further, by reducing the magnetic flux of the magnet, the field weakening region can be reduced (the amount of field weakening can be reduced), and the spatial harmonics that cause magnetostriction can be reduced. For this reason, generation | occurrence | production of an eddy current in a permanent magnet can be restrict | limited, heat_generation | fever can be suppressed, demagnetization by the temperature change of a permanent magnet can be suppressed, a heat-resistant grade can be lowered | hung and cost can be reduced.

In addition, the center adjustment groove can be adjusted to increase the magnetic resistance in the vicinity of the d-axis between the rotor and the stator side teeth. By forming the gap, the magnetic flux in the vicinity of the d-axis can be adjusted. Along with the decrease, an increase in the armature magnetic flux interlinking can be suppressed. Therefore, it is possible to prevent the drive efficiency from being lowered due to an increase in torque ripple or iron loss.
Further, the side adjustment groove can increase the magnetic resistance in the vicinity of both outer end portions of the V-shaped permanent magnet of the rotor, and can suppress harmonics that are to be superimposed on the interlinked magnetic flux waveform. Therefore, it is possible to suppress the cogging torque and prevent the drive efficiency from being lowered due to an increase in torque ripple and iron loss.
In addition, in a flux barrier having a structure in which the number of slots per phase per pole is two and a side bridge is formed between the outer peripheral surface side of the rotor, the gap between the midpoint of the inner surface on the outer end side and the d-axis is The angle θ8 is set to 64.7 ° to 74.2 ° (electrical angle), and the d-axis side inner surface side extends from the middle point in a direction parallel to the outer peripheral surface of the rotor so as to function as a bridge portion, By making the included angle θ9 between the q-axis side inner surface refracted at the midpoint and the q-axis side extended surface of the d-axis side inner surface 0 ° to 37 ° (mechanical angle), the torque is hardly reduced, Cogging torque can be reduced.
As a result, it is possible to realize a low-cost electric rotating machine that rotates with high energy density and high quality.

According to the second aspect of the present invention, the included angle θ8 between the midpoint of the inner surface on the outer end side of the flux barrier and the d-axis is further reduced to 64.9 ° to 74.2 ° (electrical angle). Thus, the torque ripple can be further reduced together with the cogging torque, the electromagnetic vibration of the stator (stator) core generated due to the torque ripple can be reduced, and the electromagnetic noise associated therewith can also be reduced.
Furthermore, in addition to the specific matter of the above-described embodiment, the included angle θ8 (electrical angle) between the midpoint of the inner surface on the outer end side of the flux barrier and the d-axis is 66 ° to 68 ° or 70 ° to 72 °. By making the included angle θ9 (mechanical angle) between the q-axis side inner surface and the d-axis side inner surface the q-axis side extension surface is 10 ° to 27 °, the torque can be reduced more effectively. In addition, cogging torque and torque ripple can be reduced.

Further, the gap is formed so that the extension space to the d-axis side expands toward the axis of the rotor, so that the armature magnetic flux entering the rotor from the q-axis side on one side of the magnetic pole can be reduced. It is possible to detour around the outer peripheral surface side of the permanent magnet and to be detoured toward the q-axis side on the other side, and to be saturated together with the magnetic flux toward the outer peripheral surface side of the permanent magnet. It can be avoided. Therefore, the reluctance torque due to the armature magnetic flux can be used more effectively, and the total torque can be increased.
This gap can be effectively synthesized even if the armature magnetic flux is not canceled on the d-axis side by forming the extension space to the d-axis side so as to expand toward the outer peripheral surface side of the rotor. The direction of the magnetic flux that cannot be made can be made appropriate. Therefore, the combined magnetic flux of the armature magnetic flux and the magnet magnetic flux can be passed through a path that effectively contributes to the generation of torque, and the total torque can be further increased.
In addition, the fifth and seventh spatial harmonics are suppressed by setting the included angle θ6 (electrical angle) between the outer ends of the flux barriers on both ends of the permanent magnet to 144 ° to 154.3 °. Can do. Also, the included angle θ2 (mechanical angle) from the d-axis to the outer peripheral surface side of the permanent magnet is 27.5 ° to 72.5 °, 37.5 ° to 82.5 °, and 37.5 By setting the angle to 72.5 °, the torque at the maximum load or low load can be increased. At this time, the torque ripple and the 6th and 12th harmonic torque are suppressed to reduce electromagnetic vibration and noise. Can be reduced.

FIG. 1 is a diagram showing an embodiment of an IPM type electric rotating machine (motor) according to the present invention, and is a plan view showing a schematic overall configuration thereof. FIG. 2 is a magnetic flux diagram of the armature magnetic flux at the time of low load driving in the structure of the embodiment. FIG. 3 is a magnetic flux diagram of magnet magnetic flux at the time of low load driving in the structure of the embodiment. FIG. 4 is a graph showing torque characteristics with respect to the current phase of a V-shaped IPM motor having no large gap on the d-axis side. FIG. 5A is a magnetic flux diagram of magnet magnetic flux of a V-shaped IPM motor without a large gap on the d-axis side. FIG. 5B is a vector diagram of the magnetic flux in the vicinity of the d-axis of a V-shaped IPM motor without a large gap on the d-axis side. FIG. 6A is a magnetic flux diagram of armature magnetic flux at the time of maximum load driving of a V-shaped IPM motor without a large gap on the d-axis side. FIG. 6B is a vector diagram of armature magnetic flux in the vicinity of the d-axis at the time of maximum load driving of the V-shaped IPM motor without a large gap on the d-axis side. FIG. 7 is a model diagram showing the relative relationship between the magnetic flux vector on the outer peripheral side of the magnetic pole (permanent magnet) and the armature magnetic flux vector when the V-shaped IPM motor without a large gap on the d-axis side is driven at the maximum load. FIG. 8 is a graph showing the correspondence (characteristic) between the current phase and the output torque with respect to the input current of the IPM type motor. FIG. 9 is a magnetic flux diagram of the armature magnetic flux when the V-shaped IPM motor without a large gap on the d-axis side is driven at a low load. FIG. 10 is a path diagram showing a path taken by the combined magnetic flux together with a magnetic flux diagram of the combined magnetic flux of the magnetic flux and the armature magnetic flux when the V-shaped IPM motor without a large gap on the d-axis side is driven at a low load. FIG. 11 is a graph showing a change in generated torque and a reduction rate of torque ripple when the embedded permanent magnet of the V-shaped IPM motor with a d-axis side gap is shortened. FIG. 12 is a graph showing changes in the fifth-order spatial harmonics superimposed when the embedded permanent magnet of the V-shaped IPM motor with a d-axis side gap is shortened. FIG. 13 is a graph showing a torque generation ratio in a low load driving region of a V-shaped IPM motor without a large gap on the d-axis side and a V-shaped IPM motor with a d-axis side gap. FIG. 14 is a graph showing a torque generation ratio in a maximum load drive region of a V-shaped IPM motor without a large gap on the d-axis side and a V-shaped IPM motor with a d-axis side gap. FIG. 15 is a magnetic flux diagram showing an armature magnetic flux at the time of maximum load driving of a V-shaped IPM motor with a d-axis side gap. FIG. 16 is a magnetic flux diagram showing a combined magnetic flux of a magnetic flux and an armature magnetic flux when a V-shaped IPM motor with a d-axis side gap is driven at a low load. FIG. 17 is a magnetic flux diagram showing a combined magnetic flux of the magnet magnetic flux and the armature magnetic flux when the V-shaped IPM motor with the d-axis side gap is driven at the maximum load. FIG. 18 includes a magnetic flux diagram showing a combined magnetic flux of the magnetic flux and the armature magnetic flux at the time of maximum load driving of the V-shaped IPM motor with a d-axis side gap, and a structure to be compared with the structure of the present embodiment in FIG. FIG. FIG. 19 is a graph showing the instantaneous torque in the average torque generated in the structure A of the present embodiment in FIG. 17 and the comparative structure B in FIG. FIG. 20 is a graph showing the ratio of the harmonic torque superimposed on the instantaneous torque waveform of FIG. 19 generated in the present embodiment structure A of FIG. 17 and the comparative structure B of FIG. FIG. 21 is a graph showing the content ratio of the spatial harmonic component included in the one inter-linkage magnetic flux waveform via the gap G in the present embodiment structure A in FIG. 17 and the comparison structure B in FIG. FIG. 22 is a graph showing a change in torque when the distance R2 / distance R1 of the rotor from the axial center of the end wall surface of the flux barrier 17c on the axial center side is used as a parameter. FIG. 23 is a graph showing changes in torque when the outer radius R1 / rotor distance R2 from the axial center of the end wall surface on the axial center side of the flux barrier 17c is used as a parameter. FIG. 24 shows the magnetic flux vector and armature near the corner of the d-axis side of the permanent magnet when the maximum load is driven in the V-shaped IPM motor that forms a large gap on the d-axis side but is not enlarged on the outer peripheral surface side. It is a model figure which shows the relative relationship of a magnetic flux vector. FIG. 25 shows the magnetic flux vector and armature in the vicinity of the corner of the d-axis side of the permanent magnet when the V-shaped IPM motor that has a large gap on the d-axis side and expands to the outer peripheral surface is driven at the maximum load. It is a model figure which shows the relative relationship of a magnetic flux vector. FIG. 26 is a structural diagram in which one magnetic pole of a rotor showing parameters used when determining the size and shape of the enlarged gap shown in FIG. 25 is enlarged. FIG. 27 is a structural diagram showing a model example of the shape when the parameter DLd shown in FIG. 26 is changed. FIG. 28 is a graph showing changes in torque and harmonic torque when the ratio of DLd to outer radius R1 shown in FIG. 26 is changed as a parameter. FIG. 29 is a graph showing changes in torque ripple when the ratio of DLd to the outer radius R1 shown in FIG. 26 is changed as a parameter. FIG. 30 is a graph showing changes in torque and harmonic torque when the ratio of θ1 to magnet opening degree θ2 shown in FIG. 26 is changed as a parameter. FIG. 31 is a graph showing changes in torque ripple when the ratio of θ1 to magnet opening degree θ2 shown in FIG. 26 is changed as a parameter. FIG. 32 is a graph showing the instantaneous torque in the average torque for comparing the case where the flux barrier that is the enlarged gap is provided with the case where it is not enlarged. FIG. 33 is a graph showing the ratio of the harmonic torque superimposed on the waveform of the instantaneous torque in the average torque of FIG. FIG. 34A is a magnetic flux diagram of magnet magnetic flux of a V-shaped IPM motor having no large gap on the d-axis side and having no center groove formed thereon. FIG. 34B is a vector diagram of a combined magnetic flux of an armature magnetic flux and a magnet magnetic flux in the vicinity of the d-axis at the maximum load of a V-shaped IPM motor without a center gap formed on the d-axis side and having no center gap. FIG. 35A is a magnetic flux diagram of the magnetic flux of a V-shaped IPM motor with no center groove formed with a large gap on the d-axis side. FIG. 35B is a vector diagram of the combined magnetic flux of the armature magnetic flux and the magnet magnetic flux in the vicinity of the d-axis at the maximum load of the V-shaped IPM motor having no center groove formed with a large gap on the d-axis side. FIG. 36 shows a one-tooth linkage comparing the structure without a large groove on the d-axis side shown in FIG. 34A and the structure without a center groove formed with a large gap on the d-axis side shown in FIG. 35A. It is a graph which shows a magnetic flux waveform. FIG. 37 is a graph showing the content of spatial harmonics superimposed on one inter-linkage magnetic flux waveform by expanding the magnetic flux waveform shown in FIG. 36 by Fourier series. FIG. 38 is a vector diagram of a combined magnetic flux of an armature magnetic flux and a magnet magnetic flux in the vicinity of the d-axis at the maximum load of a V-shaped IPM motor having a center groove formed with a large gap formed on the d-axis side. FIG. 39 is a graph showing a torque waveform at the maximum load comparing the present embodiment with the structure without the center groove shown in FIG. 35A. FIG. 40 is a graph comparing the degree of superposition of harmonic torque superimposed on the torque waveform by expanding the torque waveform shown in FIG. 39 by Fourier series. FIG. 41 is an enlarged structural view of one magnetic pole of the rotor showing parameters used when determining the dimension and shape of the center groove. FIG. 42 is a graph showing changes in torque ripple when the ratio of R4 to the outer radius R1 in the dimensional shape of the center groove shown in FIG. 41 is changed as a parameter. FIG. 43 is a graph showing a phase voltage waveform and a line voltage waveform when the outer opening angle θa in the dimension shape of the center groove shown in FIG. 41 is changed as a parameter. FIG. 44 is a graph showing a torque waveform at a low load comparing the present embodiment with the structure having no center groove shown in FIG. 35A. FIG. 45 is a graph comparing the degree of superposition of harmonic torque superimposed on the torque waveform by expanding the torque waveform shown in FIG. 44 by Fourier series. FIG. 46 is a structural diagram showing the positional relationship of stator teeth in one magnetic pole in a structure in which side grooves are not formed. 47 is a graph showing a gap magnetic flux waveform at the time of no load of the structure in which the side groove is not formed shown in FIG. FIG. 48 is a graph showing a gap magnetic flux waveform at the maximum load of the structure in which the side groove is not formed shown in FIG. FIG. 49 is an enlarged structural diagram of one magnetic pole of the rotor showing parameters used when determining the size and shape of the side grooves formed on the outer peripheral surface of the rotor. FIG. 50 shows changes in torque, harmonic torque, and torque ripple when the inner sandwich angle θ5 / outer sandwich angle θ4 from the d-axis in the dimension shape of the side groove shown in FIG. 49 is changed as a parameter at the maximum load. It is a graph. FIG. 51 is a graph showing changes in torque and torque ripple when the inside groove angle θ5 / outer angle θ4 from the d-axis in the dimension shape of the side groove shown in FIG. 49 is changed as a parameter at the time of low load. FIG. 52 is a graph showing changes in torque and torque ripple when the groove depth RG / air gap width AG in the dimension shape of the side groove shown in FIG. 49 is changed as a parameter at the maximum load. FIG. 53 is a graph comparing the magnitudes of harmonics superimposed on the gap magnetic flux waveform with and without the side grooves when there is no load. FIG. 54 is a graph comparing the magnitude of torque ripple based on torque waveforms with and without side grooves at the maximum load. FIG. 55 is a graph comparing the magnitude of torque ripple based on torque waveforms with and without side grooves at low loads. FIG. 56 is a graph for confirming the reduction rate of the cogging torque from the cogging torque waveform with and without the side groove at the time of no load. FIG. 57 is an enlarged structural view of one magnetic pole of the rotor showing the magnetic pole opening degree θ6 and the magnet opening degree θ2. FIG. 58 is a graph showing an approximate waveform of a gap magnetic flux linked to one tooth. FIG. 59 is a conceptual explanatory diagram showing the relationship between the approximate waveform of the gap magnetic flux interlinking one tooth, the magnetic pole opening degree, and the magnet opening degree. FIG. 60 is a graph showing the theoretical waveform (rectangular wave) of the gap magnetic flux linked to one tooth and the actual waveform (trapezoidal wave). FIG. 61 is a graph showing changes in torque, harmonic torque, and torque ripple when the magnet opening degree θ6 is changed as a parameter at the maximum load. FIG. 62 is a graph showing changes in torque, harmonic torque, and torque ripple when the magnet opening degree θ6 is changed as a parameter at low load. FIG. 63 is an enlarged structural view of one magnetic pole of the rotor showing the shape of the side bridge. FIG. 64 is a vector diagram of the magnetic flux near the side bridge when the V-shaped IPM motor is not loaded. FIG. 65A is a graph showing a magnetic flux density waveform in the air gap between the rotor and the stator when there is no load. FIG. 65B is an enlarged graph of the rising region of the magnetic flux density waveform in FIG. 65A. FIG. 66 is a vector diagram of armature magnetic flux in the vicinity of the side bridge at the maximum load of the V-shaped IPM motor. FIG. 67 is a contour diagram of the analysis result of the mechanical strength showing the location where a large Mises stress occurs during high-speed rotation of the V-shaped IPM motor. FIG. 68 is a graph showing a change in cogging torque when the refraction point on the inner surface on the outer end side of the outer flux barrier is changed. FIG. 69 is a graph showing changes in torque and torque ripple when the refraction location on the inner surface of the outer flux barrier on the outer end side is changed. FIG. 70 is a graph showing changes in torque and torque ripple when the amount of refraction on the outer end side inner surface of the outer flux barrier is changed.

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. 1 to 70 are views showing an embodiment of an IPM type electric rotating machine according to the present invention. Here, in the description of the present embodiment, the rotation direction is illustrated by taking as an example a case where the rotor is rotated counterclockwise (CCW) with respect to the stator.
In FIG. 1, an electric rotating machine (motor) 10 includes a stator (stator) 11 formed in a substantially cylindrical shape, and a rotary drive shaft 13 that is rotatably housed in the stator 11 and coincides with an axis. And a fixed rotor (rotor) 12. The electric rotating machine 10 has a performance suitable for mounting in a wheel or wheel as a drive source similar to that of an internal combustion engine, for example, in a hybrid vehicle (HEV) or an electric vehicle (EV).

The stator 11 is formed with a plurality of stator teeth 15 extending in the normal direction of the axial center so that the outer peripheral surface 12a of the rotor 12 faces the inner peripheral surface 15a via the gap G. . A three-phase winding (not shown) constituting a coil for generating a magnetic flux for rotationally driving the rotor 12 accommodated inside is wound around the stator teeth 15 by distributed winding.
The rotor 12 is manufactured to have an IPM (Interior Permanent Magnet) structure in which a pair of permanent magnets 16 are embedded as one magnetic pole so as to be V-shaped to open toward the outer peripheral surface 12a. The rotor 12 is formed such that a V-shaped space 17 that is fitted in a corner portion 16a of a plate-like permanent magnet 16 extending in the front and back direction of the drawing and is housed in a stationary state faces the outer peripheral surface 12a. .
The V-shaped space 17 is a space 17a in which the permanent magnet 16 is fitted and accommodated, and spaces 17b and 17c (hereinafter referred to as flux) that are located on both sides in the width direction of the permanent magnet 16 and function as flux barriers that restrict the wraparound of the magnetic flux. Barriers 17b and 17c). In the V-shaped space 17, the permanent magnet 16 is extended in the normal direction between the spaces 17 c so that the permanent magnet 16 can be positioned and held against the centrifugal force during high-speed rotation. A center bridge 20 for connecting and supporting is formed. The side bridge 30 having a similar function will be described later.

In the electric rotating machine 10, the space between the stator teeth 15 on the stator 11 side constitutes a slot 18 for forming a coil by being wound through a winding. On the other hand, in the rotor 12, six stator teeth 15 on the stator 11 side face each of the eight sets of permanent magnets 16. In short, the electric rotating machine 10 is constructed such that the six slots 18 on the stator 11 side correspond to one magnetic pole formed on the pair of permanent magnets 16 side on the rotor 12 side. That is, the electric rotating machine 10 is wound with 8 poles (4 pole pairs), 48 slots, single-phase distributed winding 5 pitches, with the N and S poles of the permanent magnet 16 alternately arranged for each adjacent magnetic pole. It is made to a wired three-phase IPM motor. In other words, the electric rotating machine 10 is manufactured in an IPM type structure in which the number of slots per phase per pole q = (number of slots / number of poles) / number of phases = 2.
Thus, the electric rotating machine 10 can be driven to rotate by energizing the coil in the slot 18 of the stator 11 and passing the magnetic flux through the rotor 12 facing the stator teeth 15. At this time, the electric rotating machine 10 (the stator 11 and the rotor 12) tries to minimize the magnetic path through which the magnetic flux passes, in addition to the magnet torque caused by the attractive force and the repulsive force generated between the permanent magnets 16. It can be rotationally driven by the total torque including the reluctance torque. Therefore, the electric rotating machine 10 can output electrical energy that is energized and input as mechanical energy from the rotary drive shaft 13 that rotates integrally with the rotor 12 with respect to the stator 11.
In addition, the stator 11 and the rotor 12 are laminated in the axial direction so that a thin plate made of electromagnetic steel plate material such as silicon steel has a thickness corresponding to a desired output torque, and is caulked so as to maintain the laminated state. 19 or the like.

Here, as shown in FIG. 2 as a magnetic flux diagram, the electric rotating machine 10 includes a plurality of stator teeth 15 corresponding to a pair of permanent magnets 16 constituting one magnetic pole, and the outer peripheral side of the stator 11 ( A winding coil is distributedly wound in the slot 18 so as to form a magnetic path (armature magnetic flux) that passes through the rotor 12 from the back side of the stator teeth 15. The permanent magnet 16 is accommodated in the fitting space 17a of the formed V-shaped space 17 so as to follow the magnetic path of the armature magnetic flux Ψr, in other words, so as not to prevent the formation of the armature magnetic flux Ψr. Has been.
The magnetic path (magnet magnetic flux Ψm) of the permanent magnet 16 extends vertically from the N and S poles on the front and back surfaces of the pair of permanent magnets 16 constituting one magnetic pole as shown in FIG. 3 as a magnetic flux diagram. In particular, on the stator 11 side, the corresponding stator teeth 15 pass through the back side.

In the IPM structure in which the permanent magnet 16 is embedded in the V-shape in the rotor 12, the direction of the magnetic flux generated by the magnetic pole, that is, the central axis between the V-shaped permanent magnets 16 is defined as the d-axis. The central axis between the permanent magnets 16 between adjacent magnetic poles that are orthogonal to each other electrically and magnetically is defined as the q axis. The rotor 12 is formed so that the inner space 17c located on the d-axis side of the V-shaped space 17 is enlarged to a large gap toward the axis and functions as a flux barrier 17c. The optimum size and shape of the flux barrier 17c in the V-shaped space 17 will be described later.
As a result, in the electric rotating machine 10, as shown in FIG. 2, the armature magnetic flux Ψr entering the rotor 12 from the stator teeth 15 is greatly increased so as not to go around the outer periphery of the V-shaped space 17 ( It is formed so as to take a path detouring toward the axis) and returning to the stator teeth 15. In short, in the electric rotating machine 10, the rotor 12 is constructed as a V-shaped IPM motor with a d-axis gap.
In addition, the electric rotating machine 10 is arranged so that the armature magnetic flux Ψr entering from the stator teeth 15 corresponding to the d-axis does not overlap with many fifth-order and seventh-order spatial harmonics that cause an increase in torque ripple. A center groove (center adjusting groove) 21 extending in a direction parallel to the inner peripheral surface 15a of the stator teeth 15 (axial direction) is formed on the outer peripheral surface of the stator teeth 15. The optimum dimensional shape of the center groove 21 will be described later.
Further, the electric rotating machine 10 has a side groove that suppresses torque pulsation in the entire drive region by reducing cogging torque at no load and torque ripple at low load and maximum load while minimizing torque reduction. (Side adjustment grooves) 22 are formed on the outer peripheral surfaces of the pair of permanent magnets 16 forming the magnetic poles. The optimum dimensional shape of the side groove 22 will be described later.

Ｐｐ：極対数、Ψｍ：電機子（ステータティース１５）鎖交磁石磁束、
ｉｄ：線電流のｄ軸成分、ｉｑ：線電流のｑ軸成分、
Ｌｄ：ｄ軸インダクタンス、Ｌｑ：ｑ軸インダクタンス As described above, in the case of the electric rotating machine 10 having the IPM structure in which the permanent magnet 16 is embedded in the V shape in the rotor 12, the torque T can be expressed by the following equation (1), as shown in FIG. In addition, high torque and high efficiency operation is realized by driving at a current phase that maximizes the sum of the magnet torque Tm and the reluctance torque Tr.

Pp: number of pole pairs, Ψm: armature (stator teeth 15) interlinkage magnet magnetic flux,
id: d-axis component of line current, iq: q-axis component of line current,
Ld: d-axis inductance, Lq: q-axis inductance

By the way, in the case of the rotor 12A of related technology provided with the flux barrier 17d equivalent to the flux barrier 17b outside the V-shaped space 17 instead of the flux barrier 17c of the d-axis side gap, the magnetic flux diagram of FIG. A magnetic path of the illustrated permanent magnet 16 is formed, and the magnet magnetic flux Ψm is a vector Vm in the direction illustrated in the magnetic flux vector diagram of FIG. 5B. Further, the armature magnetic flux Ψr generated by energizing the coil accommodated in the slot 18 is formed in the magnetic path shown in the magnetic flux diagram of FIG. 6A, and becomes the vector Vr in the direction shown in the magnetic flux vector diagram of FIG. 6B. It has become.
In this type of electric rotating machine, at the time of maximum load driving, the current phase angle is advanced to achieve high torque and high efficiency driving. In the related art rotor 12A, as shown in the magnetic flux vector diagrams of FIGS. 5B and 6B, in the small region A1 near the d axis located on the outer peripheral side of the V-shaped space 17 (magnetic pole), the magnetic flux Ψm and the armature The reluctance torque Tr is in a state of being driven while canceling out (cancelling) the magnet torque Tm because the magnetic flux Ψr has a reverse magnetic field relationship. In short, as shown in FIG. 7, the magnetic pole outer peripheral side small area A1 is an interference area where the magnet magnetic flux Ψm and the armature magnetic flux Ψr face each other in a reverse positional relationship at an included angle of 90 degrees or more. The armature magnetic flux Ψr is wasted to suppress (cancel) the magnetic flux Ψm generated in the range B on the d-axis side of the permanent magnet 16 adjacent to the small side area A1.
From this, it can be said that the d-axis side range B of the permanent magnet 16 corresponding to the magnetic pole outer peripheral side small area A1 does not actively contribute to the torque T. By using a magnetic circuit that maintains the same salient pole ratio while reducing the portion B, the magnet amount of the permanent magnet 16 itself can be reduced.
Here, since the torque T is expressed by the above equation (1), when the magnet amount of the permanent magnet 16 is reduced, the reluctance torque Tr is increased to be equivalent to the case where the magnet amount of the permanent magnet 16 is not reduced. can do. The reluctance torque Tr can be increased by increasing the difference between the d-axis inductance Ld and the q-axis inductance Lq, that is, the salient pole ratio.
Therefore, in the rotor 12 of the present embodiment, the salient pole ratio is increased while the amount of the permanent magnet 16 is reduced by replacing the d-axis side range B of the permanent magnet 16 with a gap (restricted region) having a small magnetic permeability. Thus, a torque T equal to or higher than that before replacement can be obtained. In other words, the reluctance torque Tr can be increased by effectively using the armature magnetic flux Ψr that has been wasted to suppress the magnetic flux Ψm generated in the d-axis side range B of the permanent magnet 16. Even if the magnet amount of the magnet 16 is reduced, an equivalent torque T can be obtained.

β：電流位相角度、Ｉａ：相電流値 The torque T can also be expressed as in the following formula (2). In the low load region where the current value Ia is small, the ratio of the magnet torque Tm is high, and the current value Ia is low as shown in FIG. The current phase β at the maximum torque becomes closer to zero. Waveforms i to v in FIG. 8 indicate current phase-torque characteristics at the respective current values Ia (i) to Ia (v), and the magnitude of the current value Ia is i <ii <iii <iv <iv <. The relationship is v. Therefore, the ratio (dependence) of the magnet torque Tm naturally increases during low-load driving, but a magnetic circuit that effectively uses the magnet torque Tm to the maximum is desirable.

β: current phase angle, Ia: phase current value

In the related art rotor 12A, as shown in FIG. 9, in the low load region with a low current value, the current phase β is driven under a condition close to zero, so that the magnetic flux amount of the armature magnetic flux Ψr becomes the q axis. (Between adjacent permanent magnets 16 of different magnetic poles). For this reason, as a path of the magnetic flux Ψs obtained by synthesizing the magnetic flux Ψm with the armature magnetic flux Ψr, it is preferable to use a magnetic circuit that passes through the magnetic paths MP1 and MP2 shown in FIG. As a result, the synthesized magnetic flux Ψs can increase the q-axis inductance Lq by dispersing the q-axis magnetic path (magnetic flux) (avoid saturation), and can actively use the reluctance torque Tr. can do.
The magnetic path MP1 is connected to the rotor teeth 12A via the air gap G from the stator teeth 15 on the stator 11 side and enters between the magnetic poles, and then forms a magnetic pole on the rotation direction traveling side (left side in the figure). A path is taken through the permanent magnet 16 on the side from the inner peripheral side. Further, the magnetic path MP1 takes a path that passes through the outer peripheral area A2 of the magnetic pole and returns to the stator teeth 15 through the air gap G again.
The magnetic path MP2 enters between the magnetic poles in the same manner as the magnetic path MP1, and then passes through the outer peripheral side area A2 of the magnetic pole through the separation-side permanent magnet 16 that forms the magnetic pole on the rotation direction traveling side from the inner peripheral side. Then, a path to return to the stator teeth 15 again through the air gap G is taken.

For example, in the magnetic paths MP1 and MP2, when both end sides (magnetic pole outer end portions) of the pair of permanent magnets 16 are cut and moved to the inside, a large flux barrier exists on both end sides and the vicinity of the center of the magnetic pole In particular, it becomes difficult to take a path on the right side of the magnetic pole outer peripheral side area A2, and the entire area A2 cannot be used effectively.
On the other hand, when the center side (the inner end of the magnetic pole) of the pair of permanent magnets 16 is cut and moved to the outside, a large flux barrier exists on the center side, and the magnetic flux path can be dispersed on both sides of the magnetic pole. In addition, the magnetic flux can pass through the region A2 evenly by actively and effectively including the path on the right side of the magnetic pole outer region A2. In the case of this structure, the magnetic path that connects the N pole and the S pole of the adjacent permanent magnet 16 of the magnetic pole after the permanent magnet 16 of the magnetic pole on the reverse side in the rotation direction has passed through from the outer periphery to the inner periphery. MP3 can also be taken. In this magnetic path MP3, it is possible to pass through the outer periphery side area A2 of the magnetic pole on the traveling side in the rotation direction through the same path as the magnetic path MP1, and the dispersion efficiency of the magnetic flux is high.
Therefore, the rotor 12 has an embedded structure of a pair of permanent magnets 16 that form magnetic poles, while maintaining a V shape so as not to disturb the armature magnetic flux Ψr that generates the reluctance torque Tr, the rotor 12 It is preferable to adopt a shape that approaches the outer end portion. Further, it is preferable to adopt a structure in which a flux barrier 17c that restricts the magnetic flux from taking a short circuit path is formed between the pair of permanent magnets 16 (the inner end of the magnetic pole). Further, a center groove 21 is formed on the outer peripheral surface of the rotor 12 on the d-axis to limit the saturation of the armature magnetic flux Ψr entering from the stator teeth 15 on the stator 11 side, in other words, to distribute the magnetic flux Ψr. It is preferable to adopt a structure that By adopting such a structure, the rotor 12 can disperse the q-axis magnetic path (magnetic flux), increase the q-axis inductance Lq, and actively use the reluctance torque Tr.

The permanent magnet 16 compares and determines the optimum value of the length (width) Wpm in the longitudinal direction in the drawing with reference to the case where the length Wpm is not shortened.
Specifically, with the number of poles P and the outer radius R1 from the axis of the rotor 12 to the outer peripheral surface being fixed values, the length Wpm of the permanent magnet 16 installed at the outer end of the magnetic pole is a variable (inner end side). It is determined by changing the ratio δ calculated by the following equation (3), with the position of the end side as displacement). As a determining factor, the change in the torque T at the maximum load per unit unit and the change in the reduction rate of the torque ripple that is the fluctuation range of the torque T with respect to the ratio δ are displayed as a graph by magnetic field analysis. Then, as shown in FIG. In per unit unit, for example, 1.0 [p. u. ] Means the same.
δ = (P × Wpm) / R1 (3)
In FIG. 11, the ratio δ = 1.84 is the case of the permanent magnet 16 having a shape dimension (magnet reduction amount 0%) that does not shorten the length Wpm, and the ratio shape δ = 1.38 (magnet reduction amount 24. 7%), it can be seen that a torque T equivalent to 1.0 (p.u.) can be obtained. The permanent magnet 16 can obtain an equivalent torque T by setting the ratio δ = 1.38 even during normal low-speed rotational load.
Here, in FIG. 11, a rotor 12A of related technology including flux barriers 17b and 17d having the same size on the inner and outer ends of the V-shaped space 17 is set as a comparison target. On the other hand, in the case of the rotor 12 of the present embodiment, the armature magnetic flux Ψr can be effectively divided and distributed by providing the flux barrier 17c and the center groove 21. Therefore, in the rotor 12, the reluctance torque Tr can be generated effectively, and the torque T is improved and the torque ripple is reduced even at the ratio δ = 1.84 where the permanent magnet 16 has the same length Wpm. Yes. That is, in FIG. 11, the change of the torque T and the torque ripple with respect to the ratio δ is illustrated by shortening the length Wpm of the permanent magnet 16 with the structure of the rotor 12. When the length Wpm of the permanent magnet 16 is shortened with the structure of the rotor 12A of the related art, there is no significant change in the torque T from the ratio δ = 1.84 to the ratio δ = 1.38 (1 .0 [pu]]).

  Moreover, in the electric rotating machine, an induced voltage (counterelectromotive voltage) corresponding to the amount of the permanent magnet to be embedded is generated with the rotation of the rotor, and spatial harmonics of magnetostriction caused by the field weakening are superimposed. become. In this spatial harmonic, the fifth, seventh, eleventh, and thirteenth components cause torque ripple and cause an increase in iron loss. From this, for example, when the generation of fifth-order spatial harmonics with respect to the ratio δ is graphed in units of per unit, the result is as shown in FIG. 12, and the fifth-order space becomes smaller as the ratio δ = 1.75 or less. It can be seen that the generation of harmonics can be suppressed. In this case, the magnet amount of the permanent magnet 16 can be reduced by 4.7% or more, and the internal efficiency of the permanent magnet 16 can be improved while reducing the iron loss by reducing the spatial harmonics of the magnetostriction and improving the driving efficiency. The generation of eddy currents at this point can be limited to suppress heat generation.

From this, in the rotor 12 of the present embodiment, in order to reduce the amount of the permanent magnet 16 used while obtaining the torque T equivalent to the rotor 12A of the related art, the length Wpm of the permanent magnet 16 is shortened. (The magnet amount is reduced by 24.7%) and the ratio δ is preferably about 1.38, and the torque ripple can be reduced. In short, the permanent magnet 16 has a ratio δ = 1.38 (magnet reduction amount 24.7%) to 1.75 (magnet reduction amount 4.7%) in accordance with desired characteristics such as torque T and torque ripple. The size and shape may be appropriately selected.
Therefore, the electric rotating machine 10 is a V-shaped IPM motor with a d-axis gap formed by shortening the length Wpm of the permanent magnet 16 so as to have the equivalent torque T and forming the dimensional shape with the ratio δ = 1.38. When the magnetic field analysis is performed in the case of a V-shaped IPM motor that does not shorten the permanent magnet 16, the ratio of the magnet torque Tm and the reluctance torque Tr changes as shown in FIGS. It turns out that output is possible. The V-shaped IPM motor with a d-axis gap has a structure with a large gap flux barrier 17c on the d-axis side, and the simple V-shaped IPM motor has a structure with a small flux barrier 17d on the d-axis side. It is.
FIG. 13 illustrates the ratios of torques Tm and Tr in the low load region, and FIG. 14 illustrates the ratios of torques Tm and Tr in the maximum load region. In any case, in the case of a V-shaped IPM motor with a d-axis gap, it can be seen that the reluctance torque Tr is increased instead of decreasing the magnet torque Tm in order to shorten the permanent magnet 16. That is, the electric rotating machine 10 replaces the permanent magnet 16 near the d-axis to form a large gap space flux barrier 17c and a center groove 21, so that the magnetic pole outer peripheral small area A1 shown in FIG. 6B and FIG. The magnet magnetic flux Ψm that cancels the armature magnetic flux Ψr can be reduced. As a result, the electric rotating machine 10 can increase the q-axis inductance Lq so that the difference (saliency ratio) from the d-axis inductance Ld is larger than that of the non-shortened V-shaped IPM motor, and the reluctance torque Tr can be increased. The same torque T can be secured by making effective use.

With this structure, as shown in FIG. 15 as a magnetic flux diagram, the electric rotating machine 10 causes the armature magnetic flux Ψr concentrated in the small area A1 on the outer peripheral side of the pair of permanent magnets 16 forming the magnetic poles to The magnetic path Mr1 that passes through the magnetic pole outer peripheral side small area A1 can also be effectively divided (divided) into a magnetic path Mr2 that bypasses the inner peripheral side of the d-axis side space 17c of the V-shaped space 17. As a result, the electric rotating machine 10 reduces the magnetic interference between the magnet magnetic flux Ψm and the armature magnetic flux Ψr (d-axis / q-axis), and on the traveling side (left side in the drawing) of the magnetic pole outer peripheral small area A1. It is possible to effectively contribute to the generation of the torque T by avoiding local magnetic saturation.

Therefore, as shown in the magnetic flux diagram of FIG. 16, the electric rotating machine 10 passes through a magnetic path MP0 through which the combined magnetic flux Ψs of the magnet magnetic flux Ψm and the armature magnetic flux Ψr mainly passes through the permanent magnet 16 during low load driving. On the other hand, when the maximum load is driven, the combined magnetic flux Ψs can be divided into the magnetic path MP1 and the magnetic path MP2 as shown in the magnetic flux diagram of FIG. As a result, the magnetic interference can be reduced and the local magnetic saturation state can be avoided, and the torque T equal to or higher than that of the permanent magnet 16 can be efficiently generated while the amount of the permanent magnet 16 is reduced. Note that the ratio of the magnetic flux Ψm to the combined magnetic flux Ψs during low load driving is larger than the armature flux Ψr.
Further, the electric rotating machine 10 replaces the permanent magnet 16 with, for example, a ratio δ = 1.44 and replaces it with a low-permeability flux barrier 17c (reduces the magnetic flux Ψm) to reduce the magnet amount by 23%. As the inertia (inertial force) is reduced, the induced voltage constant can be reduced by about 13.4%, and the output on the high-speed rotation side can be increased. Furthermore, in the electric rotating machine 10, the heat generation due to the eddy current generated in the permanent magnet 16, iron loss, and electromagnetic noise can be suppressed by reducing the spatial harmonics that become magnetic distortion.

On the other hand, for example, as shown in the magnetic flux diagram of FIG. 18, in the case of the flux barrier 17 e that is not expanded toward the axial center of the rotor 12, the combined magnetic flux Ψs cannot be sufficiently divided. Furthermore, local magnetic saturation on the rotation direction traveling side (left side in the figure) of the magnetic pole outer peripheral small area A1 cannot be avoided.
In this embodiment structure A of the flux barrier 17c shown in FIG. 17 and the comparative structure B of the flux barrier 17e shown in FIG. 18, the magnitude of the torque and its magnitude are shown in FIG. Comparing with fluctuation (torque ripple), it can be seen that the torque of the structure A is increased by about 6%, and at the same time, the torque ripple is reduced and can be rotated with high quality. In FIG. 19, the average torque is calculated based on the structure B of FIG. 18, and the instantaneous torque corresponding to the rotation angle (electrical angle) is expressed in units of per unit together with the structure B of FIG. 17. This case is illustrated.

In the structures A and B, when the waveform shown in FIG. 19 is expanded by Fourier series, the harmonic torque superimposed on the torque can be compared as shown in FIG. It can be seen that the 12th and 24th harmonic torque can be greatly reduced. Thereby, in the structure A of the present embodiment, in particular, the 12th harmonic torque can be greatly reduced to suppress the generation of judder during climbing acceleration, and the electromagnetic noise can also be greatly reduced. FIG. 20 shows the ratio (%) of harmonic torque included in the torques of structures A and B.
Furthermore, in the structures A and B, when the magnetic flux waveform interlinked through the gap G is expanded in one stator tooth 15 by Fourier series expansion, and the contents of the 11th and 13th spatial harmonic components are compared, FIG. As can be seen, the structure A can be reduced more than the structure B. In FIG. 21, the basic waveform components of the single interlinkage magnetic fluxes of the structures A and B are normalized and shown in units of per unit.

By the way, in the case of three phases, the torque ripple of the electric rotating machine 10 has a 6f-order component (in terms of electrical angle) due to the spatial harmonics superimposed on the magnetic flux waveform for each pole of one phase and the time harmonics included in the phase current ( f = 1, 2, 3,...: natural number).
The cause of torque ripple will be described below. The three-phase output (electric power) P (t) and the torque τ (t) have an angular velocity of ωm, an induced electromotive force of each phase Eu (t), Ev (t), If Ew (t) and the current of each phase are Iu (t), Iv (t), and Iw (t), they can be obtained by the following equations (4) and (5).
P (t) = _Eu (t) _Iu (t) + _Ev (t) _Iv (t) + _Ew (t) _Iw (t) (4)
τ (t) = P (t) / ω _m
= [E _u (t) I _u (t) + E _v (t) I _v (t) + E _w (t) I _w (t)] (5)
The three-phase torque is the sum of the torques of the U-phase, V-phase, and W-phase, where m represents the harmonic component of the current, n represents the harmonic component of the voltage, and the U-phase current I _u (t) Is represented by the following equation (6), the U-phase torque τ _u (t) can be expressed by the following equation (7).

また、この誘起電圧は、磁束を時間微分して求めることができることから、各誘起電圧に含まれる高調波の次数と１相１極磁束に含まれる高調波も同じ次数成分が発生することになる。その結果、３相交流モータにおいては、磁束（誘起電圧）に含まれる空間高調波次数ｎと相電流に含まれる時間高調波次数ｍとの組み合わせが６ｆになるときに、その６ｆ次成分のトルクリプルが発生していることになる。
よって、３相モータのトルクリプルは、上述するように、１相１極における磁束波形における空間高調波ｎと相電流の時間高調波ｍにおいては、ｎ±ｍ＝６ｆ（ｆ：自然数）のときに発生することから、例えば、１１次と１３次の空間高調波（ｎ＝１１、１３）が重畳していると相電流の基本波（ｍ＝１）との合わせにより１２次の高調波トルクが発生することが分かる。 Since the phase current I (t) and the phase voltage E (t) are both symmetrical waves, “n” and “m” are only odd numbers. V-phase torque and W-phase torque other than the U phase, respectively U-phase induced voltage _E u (t), with respect to U phase current _I u (t) "+ 2π / 3 (rad)", "- 2π / 3 ( rad) "phase difference, the overall torque is canceled (offset) so that only the coefficient term of" 6 "remains,
6f = n ± m (f: natural number), s = nα _n + mβ _m , t = nα _n −mβ _m
And can be expressed as the following equation (8).

In addition, since this induced voltage can be obtained by time differentiation of the magnetic flux, the same order component is generated in the harmonic order contained in each induced voltage and the harmonic contained in the 1-phase 1-pole magnetic flux. . As a result, in the three-phase AC motor, when the combination of the spatial harmonic order n included in the magnetic flux (induced voltage) and the time harmonic order m included in the phase current becomes 6f, the torque ripple of the 6f-order component Will occur.
Therefore, the torque ripple of the three-phase motor is, as described above, when n ± m = 6f (f: natural number) in the space harmonic n in the magnetic flux waveform in one phase and one pole and the time harmonic m in the phase current. For example, if the 11th and 13th spatial harmonics (n = 11, 13) are superposed, the 12th harmonic torque is combined with the fundamental wave of the phase current (m = 1). It can be seen that it occurs.

And in this electric rotating machine 10, as the flux barrier 17c of the V-shaped space 17 in the rotor 12, the size of the permanent magnet 16 is optimized at the ratio δ = 1.44 and the enlargement size toward the axis is optimized. Therefore, the end wall surface position on the axial center side is determined.
First, returning to FIG. 1, the structure of the rotor 12 is such that the distance R2 in the normal direction from the axial center of the end wall surface on the axial center side of the flux barrier 17c is changed to the outer peripheral surface. This is determined by the torque characteristics shown in FIGS. 22 and 23, which are obtained when the ratios R2 / R1 and R3 / R2 with respect to the inner radius R3 from the outer radius R1 to the inner peripheral surface are used as parameters. Here, the size and shape of the rotor 12 is such that the permeability (ease of magnetic flux) deteriorates due to the Mises stress caused by the compressive stress applied to the electromagnetic steel sheet when the rotary drive shaft 13 is press-fitted. It is determined by taking into account numerical values. 22 and 23 show the torque obtained at the maximum load in units of per unit with reference to the comparative structure B of FIG.

First, it can be seen from FIG. 22 that a torque equal to or greater than the structure B can be obtained within a range A in which R2 / R1 is 0.56 to 0.84. Preferably, 0.565 to 0 near the position where the tendency changes. The separation distance R2 of the end portion on the axial center side of the flux barrier 17c is set so as to be within a range B of .75, more preferably within a range C of about 0.59 to 0.63 where the torque increases by about 5%. decide.
Furthermore, it can be seen from FIG. 23 that a torque equal to or greater than the structure B can be obtained within a range A in which R3 / R2 is 0.54 to 0.82, preferably 0.60 to 0 near the position where the tendency changes. , And more preferably within a range C of about 0.70 to 0.77 where the torque is increased by about 5%. decide.
Thereby, the magnetic path width of the magnetic path MP2 in FIG. 17 can be sufficiently secured, and the size of the flux barrier 17c can be determined so that magnetic saturation does not occur in the magnetic path MP2.

In the rotor 12B shown in FIG. 24, as described above, even when the length (width) in the longitudinal direction of the permanent magnet 16 is set to the optimum value Wpm, the magnet is formed near the corner 16a approaching the d axis. There is a vector Vr of the armature magnetic flux Ψr that opposes the vector Vm of the magnetic flux Ψm. Specifically, in the vicinity of the corner portion 16a approaching the d-axis, the vector Vr of the armature magnetic flux Ψr passing through the magnetic path toward the axial center side deepest part of the magnetic pole outer peripheral small region A1 is the vector Vm of the magnet magnetic flux Ψm. However, there remains a state of a reverse magnetic field that opposes (interferes) and cancels (cancels) in the opposite direction exceeding 90 °. For this reason, in the structure of the rotor 12B, the armature magnetic flux Ψr passing near the d-axis side corner 16a of the permanent magnet 16 is wasted to suppress (cancel) the magnetic flux Ψm.
From this, in this electric rotating machine 10 (rotor 12), as shown in FIG. 25, the flux barrier 17c is formed in the space | gap shape which expands toward the outer peripheral surface 12a also on the d-axis side. Thereby, in this rotor 12, the magnetic path where the vector Vr of the armature magnetic flux Ψr near the corner 16a of the permanent magnet 16 approaching the d-axis is less than 90 ° with respect to the vector Vm of the magnet magnetic flux Ψm. The armature magnetic flux Ψr and the magnet magnetic flux Ψm can be effectively used so that the armature magnetic flux Ψr passes.

Specifically, in the electric rotating machine 10, as the flux barrier 17c of the V-shaped space 17 in the rotor 12, the enlarged gap toward the outer peripheral surface 12a side while the permanent magnet 16 is dimensioned with a ratio δ = 1.44. In order to optimize the above, the dimension shapes 1 and 2 are determined.
First, as the dimension shape 1 of the flux barrier 17c of the rotor 12, as shown in FIG. 26, the outer periphery from the intersection point Y between the extended surface of the outer peripheral surface side end surface (planar shape) 17cu of the flux barrier 17c and the d axis. A separation distance DLd to the surface 12a (intersection point X) is determined. For example, the separation distance DLd is determined by the average torque, harmonic torque, and torque ripple obtained when the ratio DLd / R1 to the outer radius R1 of the rotor 12 is used as a parameter. In other words, the size and shape 1 of the flux barrier 17c is such that the outer peripheral surface 12a has an outer periphery so that optimum characteristics can be obtained, such as not saturating the magnetic flux density of the magnetic path MP1 passing through the magnetic pole outer peripheral region A2 in the rotor 12. The distance (separation distance) DLd from the surface side end surface 17cu to the d-axis side end is determined.
For example, from the outer peripheral surface 12a of the rotor 12 to the outer peripheral surface side end surface 17cu of the flux barrier 17c, the outer peripheral surface side wall surface (outer surface of the permanent magnet 16) 17au of the accommodating space 17a of the V-shaped space 17 is shown in FIG. Is expanded from the DLd / R1 = 0.194 coincident with the extended surface to DLd / R1 = 0.086 on the outer peripheral surface 12a side. In this case, it can be seen that the torque characteristics change as shown in the graphs of FIGS. In FIG. 28, the average torque obtained at the maximum load with DLd / R1 = 0.194 as a reference is shown in units of per unit. Further, the high-frequency torque in FIG. 28 illustrates the superposition ratio of the sixth-order and twelfth-order components (electrical angles), and the torque ripple in FIG. 29 illustrates the torque fluctuation rate.

  According to FIG. 28, the dimension shape 1 of the flux barrier 17c of the rotor 12 is simply set within the range A of DLd / R1 = 0.098 to 0.194, so that the accommodation space 17a of the V-shaped space 17 is simply set. It can be seen that a larger torque can be obtained than a structure in which the outer peripheral surface side wall surface 17au is simply extended. As this dimension shape 1, preferably, the 12th-order harmonic torque can be reduced by setting within the range B of DLd / R1 = 0.11 to 0.194, and more preferably DLd / R1 = 0.11 to 0.194. The maximum torque can be obtained by setting within the range C of about /R1=0.12 to 0.14. Further, from FIG. 29, the torque ripple can be minimized by setting the best shape BP1 of DLd / R1 = 0.139 as the dimension shape 1.

Further, as the dimension shape 2 of the flux barrier 17c of the rotor 12, as shown in FIG. 26, the outer peripheral surface side end surface 17cu of the flux barrier 17c is opposed to the outer peripheral surface side wall surface 17au of the accommodating space 17a of the V-shaped space 17. To determine the tilt angle α.
For example, the inclination angle α is based on DLd / R1 = 0.139, the included angle θ1 between the outer peripheral surface side end face 17cu of the flux barrier 17c and the d axis, and the accommodation space 17a of the V-shaped space 17. A ratio θ1 / θ2 of the included angle θ2 between the outer peripheral surface side wall surface 17au and the d-axis is determined. This ratio θ1 / θ2 is determined by the average torque, the harmonic torque, and the torque ripple shown in FIGS. 30 and 31 obtained when changing as a parameter. In other words, as the dimension shape 2 of the flux barrier 17c, the armature magnetic flux Ψr suppresses the magnetic flux Ψm near the corner 16a of the permanent magnet 16 approaching the d-axis of the magnetic pole outer peripheral small area A1 of the rotor 12. The inclination angle α is determined so that an optimum characteristic can be obtained by forming no magnetic path. In FIG. 30, the average torque obtained at the maximum load with θ1 / θ2 = 1.7 as a reference is shown in units of per unit. Further, the high-frequency torque in FIG. 30 illustrates the superposition rate of the sixth-order and twelfth-order components, and the torque ripple in FIG. 31 illustrates the torque fluctuation rate. This θ2 is sometimes referred to as a magnet opening degree of the permanent magnet 16, and from this, θ1 can also be referred to as a flux barrier opening degree.

  According to FIG. 30, as the dimension shape 2 of the flux barrier 17c of the rotor 12, a large torque can be obtained and a 12th-order harmonic can be obtained by setting the range D in the range of θ1 / θ2 = 1.2 to 1.7. It can be seen that the torque can be reduced. Further, as shown in FIG. 31, the dimensional shape 2 can be set to the maximum torque / minimum torque ripple by preferably using the best point shape BP2 of θ1 / θ2 = 1.52.

  Here, considering both the dimensional shape 1 and 2 of the flux barrier 17c, when DLd / R1 is in the range A of 0.098 to 0.194, θ1 under this condition is divided by θ2 to be displaced. Thus, a suitable torque characteristic can be obtained by setting θ1 / θ2 = 1.0 to 2.13. Further, when the range B is in the range of DLd / R1 = 0.11 to 0.194, similarly, more preferable torque characteristics can be obtained by setting θ1 / θ2 = 1.0 to 2.02. Can do.

  In addition, when optimized with DLd / R1 = 0.139 and θ1 / θ2 = 1.5 considering both the dimensional shape 1 and 2 of the flux barrier 17c, as shown in FIG. 32, as shown in FIG. Compared to the comparative structure example, the torque ripple can be kept small while increasing the average torque by about 1.8%. Further, in the dimension shapes 1 and 2, as shown in FIG. 33, the 12th and 24th harmonic torques can be greatly reduced as compared with the comparative structure example shown in FIG. As a result, in these dimensional shapes 1 and 2, particularly, the 12th harmonic torque can be greatly reduced to suppress the generation of judder when accelerating uphill, and the electromagnetic noise can also be greatly reduced.

Further, in the rotor 12A shown in FIG. 34A, since the permanent magnet 16 exists up to the vicinity of the d-axis, a large amount of magnet magnetic flux Ψm is generated in the magnetic pole outer peripheral region A2. On the other hand, in the rotor 12C not provided with the center groove 21 shown in FIG. 35A, a gap flux barrier 17c is formed in the vicinity of the d-axis, so that the magnetic flux Ψm generated from the permanent magnet 16 is reduced. The orthogonality decreases, in other words, the magnetic flux density of the magnet magnetic flux Ψm near the d-axis decreases. For this reason, for the q-axis magnetic path Ψq, the inductance increases as the magnetic resistance near the d-axis decreases. As a result, in the rotor 12C, due to the difference in density of the magnetic flux interlinking with the outer peripheral surface 12a, harmonics are superimposed on the magnetic flux, and the efficiency is reduced due to an increase in torque ripple and iron loss. .
For example, in the vicinity of the d-axis of the rotor 12A, as shown in the magnetic flux vector diagram at the maximum load in FIG. 34B, the magnetic flux density linked from the facing stator teeth 15D corresponding to the magnetic path loop of the armature magnetic flux Ψr. Is not expensive. On the other hand, in the vicinity of the d-axis of the rotor 12C, as shown in the magnetic flux vector diagram at the maximum load in FIG. 35B, the interlinkage magnetic flux density becomes higher than the magnetic flux in the stator teeth 15D in FIG. Magnetic flux to be increased.
This means that one tooth chain that passes through the gap G between the rotor 12A (no flux barrier 17d and the center groove 21) and the rotor 12B (no flux barrier 17c and the center groove 21) and one stator tooth 15. Comparing the alternating magnetic flux waveforms, as shown in the graph of FIG. 36, the rotor 12B is more likely to flow the magnetic flux at the location indicated by “P” in the drawing where the vicinity of the d-axis is affected, and the harmonics are more easily superimposed. It has become. For example, when the magnetic flux waveform shown in FIG. 36 is expanded in the Fourier series, as shown in FIG. 37, the magnetic flux waveform of the rotor 12B has a higher content ratio of the fifth and seventh spatial harmonics than the rotor 12A. It can also be seen from the overlapping.

Therefore, the electric rotating machine 10 forms a center groove 21 that adjusts so as to increase the magnetic resistance in the gap G with the inner peripheral surface 15a of the stator teeth 15 on the d-axis of the outer peripheral surface 12a of the rotor 12. doing. In the rotor 12 in which the center groove 21 is formed, as shown in the magnetic flux vector diagram at the maximum load in FIG. 38, it is possible to suppress an increase in magnetic flux entering from the stator teeth 15 facing near the d-axis of the rotor 12. is made of.
Further, when the torque waveforms of the rotor 12 (with the center groove 21) and the rotor 12C (without the center groove 21) are compared, as shown in the graph of FIG. [pu]]), the torque waveform of the rotor 12 with the center groove 21 can reduce the amplitude, and torque ripple can be suppressed. Further, when the torque waveform shown in FIG. 39 is expanded by Fourier series, as shown in FIG. 40, the torque waveform of the rotor 12 with the center groove 21 is higher in the 6th, 12th, 18th, and 24th harmonics. Wave torque can be greatly reduced. Note that FIG. 39 shows a torque waveform of the instantaneous torque based on the average torque of the rotor 12C (1.0 [pu]].

In the electric rotating machine 10, the optimum dimensional shape of the center groove 21 in the rotor 12 is determined based on torque characteristics such as the torque ripple.
As shown in FIG. 41, the center groove 21 has a ratio R4 / ratio to the outer radius R1 to the outer peripheral surface 12a of the rotor 12 by changing the separation distance R4 from the axial center to the groove bottom 21a in the normal direction. The dimension and shape are determined by the torque ripple shown in FIG. 42 obtained when R1 is used as a parameter.
First, the depth of the center groove 21 is formed in the following dimension shape so that the torque ripple generated at the maximum load can be reduced on the basis of the dimension shape without the center groove 21 (R4 / R1 = 1.0).
0.98 ≦ R4 / R1 <1.0

Further, the center groove 21 of the rotor 12 needs to have a dimensional shape determined from a relative relationship to the stator teeth 15 on the stator 11 side, and as shown in FIG. The outer opening angle θa of the outer peripheral surface 12a and the inner opening angle θb of the groove bottom 21a inside the outer peripheral surface 12a can be defined.
When the rotor 12 is changed by using the outer opening angle θa of the center groove 21 as a parameter, as shown in the graph in FIG. 43 in which the phase voltage and the line voltage are associated with each other, the peak F and the top W in the figure are obtained. It is affected at the points indicated by.
Specifically, for example, the width of G1 to G3 of the U-phase voltage waveform in FIG. 43 depends on the width of the outer opening angle θa of the center groove 21 from the relative positional relationship between the stator 11 and the rotor 12. Change. When the outer opening angle θa is narrowed, the U-phase voltage waveform becomes narrower between G1 and G3 and becomes a sharp waveform with the top W being the highest vertex, and the line voltage waveform has a peak F approaching the top W. Thus, the waveform approximates a triangular wave. On the contrary, the U-phase voltage waveform becomes a waveform in which the top W between G1 and G3 becomes flat when the outer opening angle θa of the center groove 21 is increased, and the line voltage waveform has a peak F from the top W. It becomes a waveform that approximates a trapezoidal wave that spreads apart and becomes easier to superimpose the fifth and seventh spatial harmonics.

Here, as described above, the center groove 21 needs to increase the magnetic resistance in the gap G between the rotor 12 and the stator teeth 15 (decrease the magnetic permeability), while making the outer opening angle θa too wide. In addition, since the fifth and seventh spatial harmonics are easily superimposed, it is necessary to form the minimum necessary size and shape.
As shown in FIG. 41, the structure of the rotor 12 and the stator 11 includes an opening width SO of the slot 18 on the rotor 12 side, a facing width TB of the inner peripheral surface 15a of the stator teeth 15, and an inner periphery of the stator teeth 15. Assuming that the tip end portion width TW inside the surface 15a and the air gap width AG of the gap G between the rotor 12 and the stator teeth 15 are as follows.
First, since it is necessary to increase the magnetic resistance in the gap G, the center groove 21 needs to be equal to or larger than the facing width TB of the stator teeth 15. As a lower limit value of the outer opening angle θa, the shape surrounded by the facing width TB and the axis of the rotor 12 approximates to an isosceles triangle (2 × right triangle).
2 × tan ⁻¹ ((TB / 2) / (R1 + AG)) ≦ θa
It can be.
Further, the slot 18 needs to satisfy an opening width SO> air gap width AG of the slot 18 in consideration of automatic coil insertion and a necessary energy density. From this relationship, the magnetic resistance in the gap G is lower than the opening space of the slot 18, and it is necessary to reduce the amount of magnetic flux interlinked from the tip corner K of the stator teeth 15 (see FIG. 36) to the rotor 12 side. For this reason, the center groove 21 needs to be equal to or smaller than the width to the inner peripheral surface 15a of the adjacent stator teeth 15, and as an upper limit value of the outer opening angle θa, similarly,
θa ≦ 2 × tan ⁻¹ ((SO + (TB / 2)) / (R1 + AG))
It can be.

Next, the inner opening angle θb of the groove bottom 21a of the center groove 21 is set to an outer opening angle θa equal to or smaller than the width to the inner peripheral surface 15a of the adjacent stator teeth 15 as an upper limit, similarly to the outer opening angle θa.
θb ≦ 2 × tan ⁻¹ ((SO + (TB / 2)) / (R1 + AG))
It can be.
On the other hand, the lower limit value of the inner opening angle θb of the groove bottom 21 a of the center groove 21 is adjusted so that the lower limit value of the outer opening angle θa is the facing width TB of the stator teeth 15 and the magnetic resistance in the gap G is increased. Therefore, 0 ° ≦ θb without groove bottom 21a
It is good.
The facing width TB and the tip width TW of the stator teeth 15 are not satisfied when the tip of the stator teeth 15 is pointed.
TW ≦ TB
It becomes.

Here, in this rotor 12, when the torque waveform is compared with the rotor 12C without the center groove 21 similarly at the time of low load, as shown in the graph of FIG. 0.0 [pu]]), the torque waveform of the rotor 12 with the center groove 21 has a smaller amplitude, and torque ripple can be suppressed. When the torque waveform shown in FIG. 44 is expanded in the Fourier series, the torque waveform of the rotor 12 with the center groove 21 can reduce the sixth harmonic torque as shown in FIG.
In the above, the influence of the center groove 21 on the torque characteristics will be mainly described. However, the center groove 21 is useful because it can be used as a mark at the time of manufacturing such as assembly. For example, when the so-called skew is applied in a state where the positional relationship of the permanent magnet 16 in the axial direction is twisted, the presence or absence of the skew can be confirmed from the linearity of the center groove 21 in the axial direction. .

In addition, in the rotor 12D without the side groove 22 shown in FIG. 46, as shown in FIG. 47, the magnetic flux density waveform in the gap G at no load is deformed from a fundamental wave to a waveform close to a trapezoidal wave. I understand. In this gap G, the gap magnetic flux waveform corresponding to the structure of the stator teeth 15 on the stator 11 side, the V-shaped permanent magnet 16 on the rotor 12 side, and the flux barriers 17b and 17c of the V-shaped space 17 is further increased to a spatial harmonic. Superposition of the waves causes an increase in torque ripple, electromagnetic noise, and iron loss.
The gap magnetic flux waveform has an electrical angle of 90 ° corresponding to the d-axis, an electrical angle of 0 ° and 180 ° corresponding to the q-axis, and an electrical angle of 30 for each of the stator teeth 15a to 15g in one magnetic pole of the rotor 12D. Regions A to G corresponding to each ° correspond to each other. This gap magnetic flux waveform is recessed before and after the region A corresponding to the d-axis side flux barrier 17c (gap), and the magnetic flux density is too high between the regions B and C and between the regions E and F compared to the basic waveform. I understand that. That is, in the rotor 12D, the third stator teeth 15c from the second stator teeth 15b toward the traveling direction side from the d-axis and the third stator teeth 15e from the second stator teeth 15e toward the backward direction side from the d-axis. It can be seen that superposition of spatial harmonics increases with the stator teeth 15f.
Therefore, in the rotor 12D, the chain is within two ranges (d axis ± 30 ° to 60 °) of the outer peripheral surface 12a corresponding to between the stator teeth 15b and 15c and between the stator teeth 15e and 15f. It is effective to form a pair of side grooves 22 for reducing the crossing magnetic flux density.

By the way, in an IPM type motor, a so-called step skew is applied between permanent magnets in the axial direction by twisting the rotor, so that a torque ripple of a specific order can be canceled. For example, in the case of a three-phase motor, a 12th-order torque ripple can be completely canceled by applying a step skew of an electrical angle of 15 °.
Specifically, when the 12th harmonic superimposed on the magnetic flux is expressed as a function,
F (θ) = sin12θ
The waveform with an electrical angle of 15 ° is
F (θ + 15 °) = sin12 (θ + 15 °) = − sin12θ
Theoretically, it can be canceled by canceling with the 11th and 13th spatial harmonics, and as a result, the 12th torque ripple can be reduced.
From this, not only when there is no load, but when the gap magnetic flux waveform where the harmonics at the time of loading are superimposed is confirmed, the waveform is as shown in FIG. In FIG. 48, both cases where there is a step skew without the side groove 22 are shown.
In this gap magnetic flux waveform, it can be confirmed that the spatial harmonics to be superimposed are suppressed by applying a step skew. However, as in the case of no load, the gap between the regions B and C and the region E are compared with the basic waveform. It can be seen that the magnetic flux density between F and F is too high.

And in this electric rotating machine 10, the optimal dimension shape of the side groove | channel 22 in the rotor 12 is determined based on such torque characteristics, such as a torque and a torque ripple.
As shown in FIG. 49 (FIG. 26), the side groove 22 has a narrow angle between the extended surface of the outer peripheral surface 12a side wall surface (outer peripheral side wall surface 17au) of the permanent magnet 16 and the d-axis, so-called magnet opening degree. The angle between θ2 and the extension line connecting the outer peripheral surface 12a side corner 16b of the permanent magnet 16 from the axial center and the d axis, so-called magnet end opening angle θ3, the outer end side 22o and the d axis The formation position can be defined by the outer sandwiching angle θ4 between them and the inner sandwiching angle θ5 between the inner side edge 22i and the d-axis.
First, when the side groove 22 is positioned outside the magnet end opening angle θ3 and the magnet opening degree θ2, the side groove 22 corresponds to between the regions C and D and between the regions F and G in the gap magnetic flux waveform shown in FIG. Thus, the magnetic flux density is deviated from the reduced position. Further, in the rotor 12, Mises stress caused by the centrifugal force of the permanent magnet 16 when rotating at a high speed is concentrated on a side bridge 30 described later that connects and supports the magnetic pole between the outer peripheral surface 12a and the flux barrier 17b. For this reason, a certain amount of width is required to prevent breakage due to the stress concentration. From this, as the formation position of the side groove 22,
Inner included angle θ5 <Outer included angle θ4 ≦ Magnet end opening angle θ3
It becomes.

Further, the side groove 22 has a dimension and shape determined by the torque, harmonic torque, and torque ripple torque characteristics shown in FIGS. 50 and 51, which are obtained when the ratio of the inner sandwich angle θ5 / the outer sandwich angle θ4 is used as a parameter. .
First, according to the torque characteristic at the maximum load in FIG. 50, the side groove 22 is based on the rotor 12D (θ5 / θ4 = 1.0) without the side groove 22 (1.0 [pu]]. ),
0.945 ≦ θ5 / θ4 ≦ 0.98
By using the size and shape, torque ripple can be effectively reduced while obtaining a certain amount of torque. In particular, the side groove 22 can minimize torque ripple by setting θ5 / θ4 = 0.97.
Further, the side groove 22 is also obtained from the torque characteristics at the time of low load in FIG.
θ5 / θ4 ≦ 0.98
By using the size and shape, torque ripple can be effectively reduced while obtaining a certain amount of torque.

Further, as shown in FIG. 49, the side groove 22 has a dimension and shape determined by the torque shown in FIG. 52 and the torque characteristics of the torque ripple obtained when the ratio of the groove depth RG / air gap width AG is used as a parameter. To do.
First, according to the torque characteristics at the maximum load in FIG. 52, the side groove 22 is based on the rotor 12D without the side groove 22 (RG / AG = 0.0) (1.0 [pu]]. ),
0.00 <RG / AG ≦ 0.73
By using the size and shape, torque ripple can be effectively reduced while obtaining a certain amount of torque. In particular, the side grooves 22 can minimize torque ripple by setting about 0.30 ≦ RG / AG ≦ 0.45.

Thereby, the electric rotating machine 10 forms the side groove 22 at the optimum position of the outer peripheral surface 12a of the rotor 12 as shown in the graph of the gap magnetic flux waveform in FIG. , C and the magnetic flux density between regions E and F can be reduced.
Further, as shown in the graph of the torque waveform at the maximum load in FIG. 54 and the torque waveform at the low load in FIG. 55, the electric rotary machine 10 forms the side groove 22 at the optimum position on the outer peripheral surface 12 a of the rotor 12. As a result, torque ripple can be reduced in any case.
Furthermore, the electric rotating machine 10 can reduce the cogging torque by 50% or more by forming the side groove 22 at the optimum position of the outer peripheral surface 12a of the rotor 12 as shown in the graph of the cogging torque waveform in FIG. Is done.

  Incidentally, when the electric rotating machine 10 has an IPM structure in which the permanent magnet 16 is embedded in the rotor 12 so as to have the positional relationship shown in FIG. 57, the change in magnetic flux in one tooth of the stator teeth 15 of the stator 11 is shown in FIG. As shown at 58, a square wave can be approximated. By superimposing low-order spatial harmonics such as the fifth and seventh orders on this magnetic flux waveform, iron loss and torque ripple, which is the fluctuation range of torque, increase, resulting in a decrease in efficiency due to waste as thermal energy. This is a cause of vibration and noise. Iron loss can be divided into hysteresis loss and eddy current loss. Hysteresis loss is the product of frequency and magnetic flux density, and eddy current loss is the product of the square of frequency and magnetic flux density, so the loss can be reduced by suppressing spatial harmonics, and the input of electrical energy Drive efficiency can be improved. In FIG. 58, the vertical axis is the field magnetic flux and the horizontal axis is the time, with respect to one stator tooth 15, there is no flux linkage between L1, and the flux is linked forward and reverse between L2. , An approximate rectangular wave of the magnetic flux waveform in one electrical angle period T (4L1 + 2L2) is illustrated.

よって、空間高調波を考慮して磁束密度Ｂを次式（１１）のように表したときには、径方向電磁力ｆｒは磁束密度Ｂの２乗を含むことから、空間高調波の重畳は径方向電磁力ｆｒの増加の要因となる。すなわち、空間高調波を低減することは、トルクリプルの低減、引いては、モータ電磁騒音の低減と共に駆動効率の向上を実現できる。

The electromagnetic noise of the motor (electric rotating machine) is generated when the stator vibrates due to the electromagnetic force acting on the starter (stator) side. The electromagnetic force acting on the stator is the same as that of the rotor (rotor). There are radial electromagnetic force due to the magnetic coupling of the stator and circumferential electromagnetic force due to torque. When the motor is approximated by a linear magnetic circuit for each stator tooth 15, the radial electromagnetic force is considered to be magnetic flux φ, magnetic energy W, radial electromagnetic force fr, magnetic resistance Rg, magnetic flux density B, magnetic flux. Assuming that the interlinkage area S, the distance x between the air gaps G, and the magnetic path permeability μ, the magnetic energy W and the radial electromagnetic force fr can be expressed by the following equations (9) and (10).

Therefore, when the magnetic flux density B is expressed as in the following equation (11) in consideration of the spatial harmonics, the radial electromagnetic force fr includes the square of the magnetic flux density B, so that the superposition of the spatial harmonics is the radial direction. This increases the electromagnetic force fr. In other words, reducing the spatial harmonics can reduce torque ripple and, in turn, reduce motor electromagnetic noise and improve drive efficiency.

In the case of the electric rotating machine 10 which is a distributed winding type three-phase IPM motor in which the number of slots per phase per pole = 2, since 12 slots 18 correspond to one magnetic pole pair, the electrical angle of 1 Within the period, there are 12 slots 18 in which the magnetic resistance becomes large, and the 11th and 13th spatial harmonics n are superimposed on the magnetic flux waveform by the magnetic resistance of the corresponding slot 18. The eleventh and thirteenth spatial harmonics n are generally called slot harmonics and are easily provided with a skew angle twisted about the axis according to the installation position of the permanent magnet 16 in the axial direction. Can be reduced.
However, in the case of the three-phase IPM structure, as shown in FIG. 58, the magnetic flux waveform in which the field magnetic flux interlinks with one stator tooth 15 becomes a substantially rectangular wave. The seventh-order spatial harmonic n (6f-order = sixth-order harmonic) is easily superimposed and difficult to reduce.
For this reason, in order to reduce torque ripple, it is necessary to employ a structure that reduces the fifth and seventh spatial harmonics.

A Fourier transform equation f (t) obtained by approximating a rectangular waveform of the magnetic flux waveform in one stator tooth 15 having the three-phase IPM structure is expressed as the following equation (12), and the magnetic flux waveform F illustrated in FIG. (T) can be expressed as the following formula (13). If this magnetic flux waveform F (t) is an approximate expression including spatial harmonics up to the 7th order, it is expressed as the following expression (14). 15), it can be seen from this equation that the following condition 1 or condition 2 must be satisfied in order to reduce the fifth or seventh harmonic.
Condition 1: “cos5ω · L1 = 0”
Condition 2: “cos7ω · L1 = 0”

By the way, referring to the magnetic flux waveform of FIG. 58, the following equation (16) is obtained. Here, since “L1, L2> 0”, it can be understood that the fifth-order spatial harmonics can be reduced to zero by satisfying the following condition 1A when this is arranged.
Angular frequency (angular velocity) ω = 2π / T = 2π / (4L1 + 2L2) (16)
Condition 1: 5ωL1 = 5 · 2πL1 / (4L1 + 2L2) = ± π / 2 (17)
Condition 1A: L1 = L2 / 8
Similarly, the modified expression of condition 2 is as shown in the following expression (18), and “L1, L2> 0”. Therefore, when this is rearranged, the following condition 2A is satisfied to satisfy the seventh spatial harmonic. It can be seen that the wave can be reduced to zero.
Condition 2: 7ωL1 = 7 · 2πL1 / (4L1 + 2L2) = ± π / 2 (18)
Condition 2A: L1 = L2 / 12

In the electric rotating machine 10 with the number of slots per pole per phase = 2, since the outer radius R1 of the rotor 12 is used and the following relationship is established, the following equations (19) and ( 20).
Mechanical angle 45 degrees = electrical angle period T / 2
V (m / sec) = 2πR1 · (45 ° / 360 °) / (T / 2)
= 2πR1 · (45 ° / 360 °) / ((4L1 + 2L2) / 2)
= R1 (m) ・ ω (rad / sec) (19)
2L1 + L2 = π / 4ω (20)
By substituting condition 1A and condition 2A for this, the following condition can be derived.
5th-order spatial harmonics = 0 ⇒ (L2, L1) = (π / 5ω, π / 40ω)
7th spatial harmonic = 0 ⇒ (L2, L1) = (3π / 14ω, π / 56ω)
From this, in the electric rotating machine 10, by laying out so as to satisfy the following relational expression (21), it is possible to reduce the fifth and seventh spatial harmonics and suppress the torque ripple.
π / 5ω ≦ L2 ≦ 3π / 14ω (sec) (21)

Here, “L2” in the relational expression (21) corresponds to a region forming a magnetic path on the rotor 12 side facing the stator teeth 15 in the magnetic flux waveform of FIG. 58, and the flux barriers on both sides of the permanent magnet 16. The expansion angle θ6 about the axis center in a range including the region up to the outer end portion of 17b, in other words, the magnetic pole opening degree θ6 can be obtained.
Referring to the magnetic flux waveform in FIG. 58, the relational expression “θ = ωt” is established.
It can be replaced with “θ1 = ωL2” and can be expressed as follows in various display formats. For example, in an electric rotating machine 10 in which the number of slots per phase per pole is 2 in an 8-pole 48-slot motor structure (a structure in which 6 slots correspond to one magnetic pole), one of two poles in eight poles is one cycle. Therefore, the 360 ° rotation of the rotor 12 with one mechanical angle period corresponds to four electrical angles, and the following relational expression is established.
π / 5 (rad) ≦ θ6 (mechanical angle) ≦ 3π / 14 (rad)
36 (degree) ≦ θ6 (mechanical angle) ≦ 270/7 (degree)
θ6 (mechanical angle) = (8 poles / 2 poles) · θ6 (electrical angle)
144 (degree) ≦ θ6 (electrical angle) ≦ 154.3 (degree)
From this, in the electric rotating machine 10, as shown in FIG. 59, the magnetic pole opening degree θ6 of one magnetic pole including the permanent magnet 16 and the outer end portions of the both end side flux barriers 17b is arranged as follows. Is installed in the rotor 12. Note that θ7 in FIG. 59 corresponds to the aperture between the q axes.
36 ° ≦ θ6 (mechanical angle) ≦ 38.6 °
144 ° ≦ θ6 (electrical angle) ≦ 154.3 °

  Incidentally, at this time, the magnetic pole opening degree θ6 of one magnetic pole in the rotor 12 corresponds to the period L2 in which the magnetic flux is linked to the stator teeth 15 in the approximate waveform of the magnetic flux waveform as shown in FIG. As shown, the interlinkage period L2 is located at the center of the inter-q axis θ7, and has a magnetic flux waveform at a timing at which the d-axis coincides with the center line of the interlinkage period L2. The angle θ7 in FIG. 57 is a mechanical angle of 45 ° corresponding to the angle between the q axes, and is a half-cycle electrical angle θ in the magnetic flux waveform.

  Therefore, the electric rotating machine 10 has a torque ripple when the magnetic pole opening degree θ6 including the flux barrier 17b of the permanent magnet 16 in the rotor 12 is m = 1, which is a basic waveform of the time harmonic m of the phase current. An angular range (144 ° ≦ θ6 (electrical angle) ≦ 154.3 °) that suppresses the fifth and seventh orders of the spatial harmonics n of the phase voltage to the 6f order (n = 5, 7), which is a specific order effective for reduction. ), The torque ripple can be reduced to reduce the vibration and noise, and the rotary shaft 13 can be driven to rotate with high quality. At the same time, it is possible to suppress the iron loss such as hysteresis loss and eddy current loss as well as heat loss by reducing vibration by reducing the torque ripple, and it is possible to drive the rotation with high efficiency with little loss.

Actually, as shown in FIG. 60, a leakage magnetic flux is generated at both shoulders with respect to a magnetic flux waveform approximated to a rectangular wave, and thus a slight deviation occurs from a theoretical value (waveform). This slight deviation can be adjusted by magnetic field analysis or the like within a range of 144 ° ≦ magnetic pole opening degree θ6 (electrical angle) ≦ 154.3 °.
The electric rotating machine 10 has a tendency that the armature magnetic flux Ψr flows near the q-axis (q-axis magnetic path) less affected by the magnetic flux Ψm than the d-axis side at the maximum load, and the magnetic flux density tends to increase. Therefore, when the q-axis magnetic path becomes close to magnetic saturation, the magnetic permeability decreases and the torque decreases. Therefore, the magnetic pole opening degree θ6 is advantageously smaller (narrower) in order to secure the q-axis magnetic path as much as possible and increase the torque (magnetic flux passage efficiency), and has a value close to 144 ° (electrical angle). . This magnetic pole opening degree θ6 is subjected to magnetic field analysis from the correlation with the facing width TB of the stator teeth 15 of the stator 11, the opening width SO of the slots 18, the air gap width AG between the rotor 12 and the stator teeth 15, and the like. Thus, 146.8 ° (electrical angle) is determined as an optimum value that can reduce fifth-order and seventh-order spatial harmonics and also reduce cogging torque.

Further, the electric rotating machine 10 is determined by the torque shown in FIGS. 61 and 62, the sixth-order and twelfth-order harmonic torque, and the torque characteristics of the torque ripple, with the magnet opening degree θ2 as a parameter. 61 and 62, these torque characteristics are illustrated with θ2 = 90 ° (electrical angle) as a reference (1.0 [pu]].
First, as shown in FIG. 61, when the magnet opening degree θ2 (mechanical angle) is less than 27.5 ° when the maximum load is applied, the torque is greatly reduced, and when it exceeds 72.5 °, torque ripple and harmonics are exceeded. It is preferable to be within the range E of 27.5 ° to 72.5 ° because the torque becomes large, and more preferably within the range F of about 37.5 ° to 67.5 ° from the viewpoint of the torque.
As shown in FIG. 62, when the magnet opening degree θ2 (mechanical angle) is low, when the torque is less than 37.5 °, the torque sharply decreases, and when it exceeds 82.5 °, the torque drops sharply. At the same time, it is preferable that the torque ripple and the harmonic torque become large, so that it falls within the range G of 37.5 ° to 82.5 °. Is more preferable.
From these maximum loads and low loads, the magnet opening θ2 (mechanical angle) is preferably within 37.5 ° to 72.5 °, and from torque, it is about 42.5 ° to 67.5 °. More preferably, the angle is 52.5 ° because the torque can be maximized while suppressing torque ripple and harmonic torque.

  Incidentally, as shown in FIG. 63, the electric rotating machine 10 employs an IPM structure in which the permanent magnet 16 is embedded in the rotor 12 in a V shape, so that in addition to the center bridge 20 described above, the flux barrier 17b By providing the side bridge 30 on the outer end side, the magnetic poles including the pair of permanent magnets 16 are connected and supported so as to maintain the shape against the Mises stress caused by the centrifugal force when rotating at high speed. The center bridge 20 extends in a normal direction that coincides with the d-axis from the axis of the rotor 12 and supports the magnetic poles. The side bridge 30 is formed between the outer peripheral surface 12a of the rotor 12 and the outer end side inner surface 17b1 of the flux barrier 17b, and the outer (outer peripheral surface) of the pair of permanent magnets 16 forming one magnetic pole in the rotor 12. 12a side) and the adjoining magnetic pole side q-axis side are connected and supported.

As shown in the no-load magnetic flux vector diagram of FIG. 64, the side bridge 30 is located between the d-axis side and the q-axis side of the magnetic pole (ideally, it is desired to suppress the magnetic flux wraparound as much as possible. ), And also functions as a wraparound magnetic path for the magnetic flux Ψm (shown as a vector Vm in the drawing) of the permanent magnet 16. Further, the side bridge 30 is in a state in which the region where the magnetic flux Ψm is linked to the stator teeth 15 via the air gap G is switched between the q-axis side and the d-axis side as the rotor 12 rotates. It also functions as a magnetic path. The side bridge 30 can adjust the magnetic resistance according to the shape of the outer end side inner surface 17b1 of the flux barrier 17b located on the back side of the outer peripheral surface 12a of the rotor 12, and with the rotation of the rotor 12, Thus, the magnetic flux density of the magnet magnetic flux Ψm passing through the chain can be changed.
When no load is applied, the magnetic flux Ψm changes with a waveform in which the magnetic flux density in the air gap G between the stator 11 (stator teeth 15) and the rotor 12 is close to a rectangular wave as shown in FIG. 65A. The cogging torque is generated by the change in the magnetic flux density of the magnet magnetic flux Ψm. The magnetic flux density of the magnetic flux Ψm is ideal because smooth drive can be realized by changing it with a waveform that approximates a sine wave. However, since it is difficult to achieve, the temporal change of the magnetic flux (dΨ / dt) It is effective to reduce the cogging torque. In particular, as shown in FIG. 65B, it is effective to moderate the temporal change in the rising region and convergence region of the magnetic flux (illustrated by the density waveform). From this, it is conceivable to optimize the shape of the outer end side inner surface 17b1 of the flux barrier 17b forming the side bridge 30 in order to reduce the cogging torque.

  Further, as shown in the magnetic flux vector diagram at the maximum load in FIG. 66, the side bridge 30 similarly has an armature magnetic flux between the stator teeth 15 via the air gap G as the rotor 12 rotates. It also functions as a magnetic path while a region where Ψr (shown as a vector Vr in the figure) is linked is switched between the q-axis side and the d-axis side. The armature magnetic flux Ψr becomes a magnetic flux waveform that approximates a rectangular wave, similarly to the magnet magnetic flux Ψm, and as described above, the (6f ± 1) orders such as the fifth, seventh, eleventh, and thirteenth orders. Therefore, torque ripples are generated. Accordingly, the side bridge 30 similarly has an outer end side inner surface 17b1 of the flux barrier 17b so as to moderate the temporal change (dΨ / dt) of the armature magnetic flux Ψr, particularly in the magnetic flux rising region and the convergence region. It is effective that the torque ripple can be reduced.

Therefore, in the electric rotating machine 10, the thickness (width in the figure) of the side bridge 30 is adjusted by gently changing the shape of the outer end side inner surface 17b1 of the flux barrier 17b with respect to the outer peripheral surface 12a of the rotor 12. To adjust the magnetic resistance in the air gap G.
As shown in FIG. 67, the side bridge 30 connects and supports the region located on the outer side (outer peripheral surface 12a side) of the pair of permanent magnets 16 of the rotor 12 together with the center bridge 20. The stress is concentrated on the d-axis side region MS1 of the outer peripheral surface 12a of the rotor 12 and the q-axis side region MS2 of the outer end side inner surface 17b1 of the flux barrier 17b. On the center bridge 20 side, Mises stress is concentrated in the outer peripheral surface side region MS3 of the rotor 12.
Accordingly, returning to FIG. 63, the side bridge 30 refracts the outer end side inner surface 17b1 at the intermediate point 17b1m of the both end side corners 17b1c of the outer end side inner surface 17b1 of the flux barrier 17b, and moves to the q-axis side. The thickness (width in the drawing) is increased to form a so-called fillet shape. As a result, the side bridge 30 makes the q-axis region MS2 in a shape advantageous to the Mises stress, and adjusts so that the magnetic resistance in the air gap G gradually decreases. Adjustment is made so that Ψm and armature magnetic flux Ψr change gently.

  Specifically, the outer end side inner surface 17b1 of the flux barrier 17b inside the rotor 12 of the side bridge 30 has a d-axis side inner surface 17b1d and a q-axis side inner surface 17b1q on both sides of the intermediate point 17b1m. The outer end-side inner surface 17b1 includes an angle θ8 between a straight line passing through the axis of the rotor 12 and the intermediate point 17b1m and the d-axis, an extension surface of the d-axis side inner surface 17b1d toward the q-axis side, and a q-axis It is determined by the characteristics of torque, cogging torque, and torque ripple obtained when the included angle θ9 with the side inner surface 17b1q is changed as a parameter. In this characteristic comparison, the unit is illustrated in units of units based on the included angle θ8 = 74.2 ° (the included angle θ9 = 0, no refraction) in the above-described structure in which the magnetic pole opening degree θ6 is optimized. In addition, at both end corners 17b1c and intermediate point 17b1m of the outer end side inner surface 17b1 of the flux barrier 17b, a curved shape is formed so that both end sides of the d axis side inner surface 17b1d and the q axis side inner surface 17b1q are smoothly continuous. Thus, it is formed in a so-called chamfered shape.

First, the outer end side inner surface 17b1 of the flux barrier 17b of the side bridge 30 has an included angle θ8 (electrical angle) of the intermediate point 17b1m of 64.7 ° or more and less than 74.2 ° as shown in FIG. It can be seen that setting the range I can reduce the cogging torque at no load. It is understood that the cogging torque can be more effectively reduced by setting the included angle θ8 to a range J of 66 ° to 72 ° more preferably.
As shown in FIG. 69, the inner surface 17b1 of the flux barrier 17b of the side bridge 30 has an included angle θ8 (electrical angle) of the intermediate point 17b1m of 64.9 ° or more and less than 74.2 °. It can be seen that by setting the range K, the torque ripple can be reduced while minimizing the decrease in torque at the maximum load. More preferably, by setting the included angle θ8 in the range L of 66 ° to 78 °, the torque ripple can be reduced more effectively, and in the range M of 70 ° to 72 °, it is closer to 72 °. It can be seen that the torque ripple can be effectively reduced while further suppressing the decrease in torque.
On the other hand, the outer end side inner surface 17b1 of the flux barrier 17b of the side bridge 30 is, as shown in FIG. 70, an included angle θ9 (mechanical angle) between the extended surface of the d-axis side inner surface 17b1d and the q-axis side inner surface 17b1q. The bending angle θ9 (mechanical angle) of the q-axis side inner surface 17b1q with respect to the d-axis side inner surface 17b1d is set to a range N that exceeds 0 ° and is equal to or less than 37 °, thereby minimizing a decrease in torque at the maximum load. It can be seen that torque ripple can be more effectively reduced while suppressing. It can be seen that the included angle θ9 is more preferably 10 ° in the range P of 10 ° to 27 °, so that the torque ripple can be effectively reduced while further suppressing the decrease in torque.

As described above, in the present embodiment, the d-axis side range B of the permanent magnet 16 is reduced and replaced with the large flux barrier 17c, so that the magnet magnetic flux Ψm in the direction that cancels the armature magnetic flux Ψr is eliminated to interfere (cancel) each other. And the passage of the armature magnetic flux Ψr in the range B can be restricted.
Therefore, a large magnet torque Tm and a reluctance torque Tr can be obtained by effectively using the armature magnetic flux Ψr and the magnet magnetic flux Ψm on the d-axis side while reducing the amount of permanent magnets 16 used. In addition, the output on the high speed rotation side can be increased by reducing the induced voltage constant, and the heat generation due to the eddy current of the permanent magnet 16 can be suppressed to suppress the demagnetization due to the temperature change, thereby lowering the heat resistant grade. Can reduce costs.

Further, the separation distance R2 to the axial center side end portion of the flux barrier 17c is such that the relationship (size and shape) between the outer radius R1 and the inner radius R3 of the rotor 12 is 0.56 ≦ R2 / R1 ≦ 0.84, and By satisfying 0.54 ≦ R3 / R2 ≦ 0.82, a large torque T can be generated efficiently.
Further, the flux barrier 17c is configured such that the separation distance DLd to the outer peripheral surface of the rotor 12 is 0.098 ≦ DLd / R1 <0.194 with respect to the outer radius R1 of the rotor 12, thereby efficiently increasing a large torque. Can be generated well. Further, the flux barrier 17c preferably further has DLd / L so that 0.12 ≦ DLd / R1 ≦ 0.14 and 1.2 ≦ flux barrier opening angle θ1 / magnet opening angle θ2 ≦ 1.7. By setting R1 = 0.139 and θ1 / θ2 = 1.52, larger torque can be generated efficiently.

Further, the center groove 21 of the rotor 12 has a harmonic torque of 0.98 ≦ R4 / R1 <1.0 with respect to the outer radius R1 of the rotor 12 by setting the length R4 to the groove bottom 21a. And torque ripple can be effectively reduced.
Further, the center groove 21 has 2 × tan ⁻¹ ((tooth-to-face width TB / 2) / (rotor outer radius R1 + air gap width AG)) ≦ outer opening angle θa ≦ 2 × tan−1 ((slot opening Width SO + (tooth-to-face width TB / 2)) / (rotor outer radius R1 + air gap width AG)), 0 ° ≦ inner opening angle θb ≦ 2 × tan ⁻¹ ((slot opening width SO + (tooth-to-face width TB / 2)) / (Rotor outer radius R1 + air gap width AG)), and the shape of the tooth tip width TW ≦ tooth-to-face width TB to further suppress harmonic torque and further reduce torque ripple. Can do.

  Further, the side groove 22 of the rotor 12 has an outer clamping angle θ4 ≦ magnet end opening angle θ3, 0.945 ≦ inner clamping angle θ5 / outer clamping angle θ4 ≦ 0.98, 0.00 <groove depth RG / By setting the air gap width AG ≦ 0.73, it is possible to suppress the spatial harmonics that are to be superimposed on the gap magnetic flux waveform and prevent the drive efficiency from being reduced due to the increase of cogging torque, torque ripple, and iron loss. can do.

  Further, as the structure of the pair of permanent magnets 16 embedded in the V shape, 144 ° ≦ magnetic pole opening degree θ6 (electrical angle) ≦ 154.3 ° and 27.5 ° to 37.5 ° ≦ magnet opening degree θ2 ( (Mechanical angle) ≦ 72.5 ° to 82.5 °, more preferably 37.5 ° ≦ θ2 (mechanical angle) ≦ 72.5 ° to increase the torque at maximum load or low load. In this case, the torque ripple and the 6th and 12th harmonic torque can be suppressed to reduce electromagnetic vibration and electromagnetic noise.

In addition to the above structure, the included angle θ8 between the d-axis and the intermediate point 17b1m between the d-axis side inner surface 17b1d and the q-axis side inner surface 17b1q of the side bridge 30 is 64.9 ° to 74.2 ° (electrical angle ), And the included angle θ9 between the extended surface of the d-axis side inner surface 17b1d and the q-axis side inner surface 17b1q is set to 0 ° to 37 ° (mechanical angle). Torque ripple can be reduced. For this reason, electromagnetic vibration of the stator (stator) core generated due to torque ripple can be reduced, and electromagnetic noise accompanying this can be reduced. When the purpose is to reduce cogging torque, the included angle θ8 may be relaxed to 64.7 ° or more.
Furthermore, the included angle θ8 is set to 66 ° to 68 ° or 70 ° to 72 °, and the included angle θ9 is set to 10 ° to 27 °, so that the torque can be reduced more effectively and hardly. Cogging torque and torque ripple can be reduced.
As a result, the rotor 12 in the stator 11 can be produced at low cost and can be driven to rotate with high energy density and high quality.

Here, in this embodiment, the electric rotating machine 10 having a configuration of an 8-pole 48-slot motor will be described as an example. However, the present invention is not limited to this. If the structure has a number of slots per phase per pole q = 2, The present invention can be preferably applied as it is. For example, it can also be applied to a motor structure of 6 poles, 36 slots, 4 poles, 24 slots, and 10 poles and 60 slots.
The scope of the present invention is not limited to the illustrated and described exemplary embodiments, but includes all embodiments that provide the same effects as those intended by the present invention. Further, the scope of the invention is not limited to the combinations of features of the invention defined by the claims, but may be defined by any desired combination of particular features among all the disclosed features. .

10 Electric rotating machine (IPM type)
11 Stator 12 Rotor 12a Outer peripheral surface 13 Rotating drive shaft 15 Stator teeth 16 Permanent magnet 17 V-shaped space 17b, 17c Flux barrier 17b1 Outer end inner surface 17b1d d-axis inner surface 17b1m Intermediate point 17b1q q-axis inner surface 18 Slot 20 Center Bridge 21 Center groove 22 Side groove 30 Side bridge B d-axis side range G Air gap θ8 Angle between the d-axis and the intermediate point θ9 Angle between the inner surface of the d-axis and the inner surface of the q-axis

Claims (2)

A rotor that is embedded with a permanent magnet and rotates integrally with the drive shaft, and that the rotor is housed in a relatively rotatable manner, and a coil is housed in a slot between a plurality of teeth facing the rotor to function as an armature. An electric rotating machine including a stator and configured so that the number of slots per phase per pole is 2.
The permanent magnets are arranged in a V shape that opens toward the outer peripheral surface of the rotor,
The armature magnetic flux generated by the armature on the d-axis side when the permanent magnet exists up to the d-axis side in the magnetic flux direction coinciding with the central axis of the permanent magnet for each magnetic pole formed by the permanent magnet. Replacing the permanent magnet in the range that generates magnet magnetic flux in the direction of cancellation with a small gap of magnetic permeability,
A center adjustment groove parallel to the axis is formed on the d-axis on the outer peripheral surface of the rotor, and a pair of side adjustment grooves parallel to the axis is formed on both outer end sides of the permanent magnet on the outer peripheral surface. And
A flux barrier projecting from both outer end sides of the permanent magnet toward the outer peripheral surface of the rotor;
A side bridge that connects and supports the q-axis side and the d-axis side in the magnetic flux direction between the magnetic poles of the rotor is formed between the inner surface on the outer end side of the flux barrier and the outer peripheral surface of the rotor. And
The outer surface on the outer end side of the flux barrier has a d-axis side inner surface and a q-axis side inner surface on both sides of an intermediate point between both corners located on the back side of the outer peripheral surface of the rotor,
When the angle between the straight line passing through the center of the rotor and the intermediate point of the inner surface on the outer end side of the flux barrier and the d axis is θ8,
64.7 ° ≦ θ8 (electrical angle) ≦ 74.2 °
Satisfy the relationship
The d-axis side inner surface is extended in a direction parallel to the outer peripheral surface of the rotor from an intermediate point of the outer end side inner surface of the flux barrier,
The q-axis side inner surface, when the included angle between the d-axis side inner surface and the extension surface to the q-axis side is θ9,
0 ° <θ9 (mechanical angle) ≦ 37 °
An IPM type electric rotating machine characterized by satisfying the above relationship. The included angle θ8 is:
64.9 ° ≦ θ8 (electrical angle) ≦ 74.2 °
The IPM type electric rotating machine according to claim 1, wherein the relationship is satisfied.

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