JP2013135570A - Dc-dc converter - Google Patents

Abstract

PROBLEM TO BE SOLVED: To suppress switching loss, voltage stress, and EMI of a semiconductor element.SOLUTION: When a MOSFET Qm is turned on, a current flows from a DC power supply 2 via a first resonance path passing through the MOSFET Qm, a capacitor C2, a diode D3, an inductor Ls, and a capacitor C1. If the capacitance value of the capacitor C1 is smaller than that of the capacitor C2, a voltage of the capacitor C1 reaches an input voltage Vs prior to a voltage of the capacitor C2, and the energy of the inductor Ls is moved to the capacitor C2 via a second resonance path passing through the inductor Ls, a diode D1, the capacitor C2, and the diode D3. When a resonance current becomes zero, the voltage of the capacitor C2 reaches near the input voltage Vs. When the MOSFET Qm is turned off, storage charges of the capacitors C1 and C2 flow via the diode D1, a diode D2, and an inductor Lm. Then, a reflux current flows via a diode Df.

Description

  The present invention relates to a DC-DC converter including a resonance circuit.

  Various types of circuit topology have been devised to improve power density and / or efficiency of power converters. Passive elements such as inductors, capacitors, transformers, etc. can be reduced in size by increasing the frequency of signals passing through these elements. However, as the switching frequency increases, the performance of the power converter is limited by the loss of the switching element. Switching loss of a semiconductor element occurs during a turn-on period or a turn-off period.

  For example, the DC-DC converter described in Patent Document 1 increases conversion efficiency by reducing switching loss by a resonance technique. This DC-DC converter enables soft switching and reduces switching loss by using a resonant circuit including two switching elements, an inductor, a transformer, two capacitors, and two diodes.

However, this circuit configuration has the following disadvantages.
(1) It has a dissipative current circulation path in which energy in the resonance circuit is not regenerated to the input power source or the load.
(2) High voltage stress is applied to the switching element by the resonance operation.
(3) Two switching elements are required. For this reason, a special gate driver for adding dead time is required so that both switching elements are not turned on at the same time. Since power cannot be sent to the output side within the dead time period, power loss (dead time loss) occurs. Furthermore, since the number of switching elements increases, the size and cost of the DC-DC converter increase.
(4) A large transformer increases the size and cost of the system.
(5) The semiconductor element (diode) on the secondary side becomes hard switching. The hard switching of the diode is non-ideal switching in which the diode is energized in the reverse direction during the recovery time due to the minority carrier accumulation effect, and loss occurs.

  In addition, the resonant power converter described in Patent Document 2 is configured to operate in a wide input voltage range by using PWM and variable frequency modulation. The half-bridge resonant converter described in Patent Document 3 performs constant frequency operation and ZVS operation by controlling the duty cycle. The resonant power converter described in Patent Document 4 uses a burst mode trigger unit that outputs a burst width signal with a duty cycle of 50% to the bridge switch circuit, thereby increasing the converter efficiency at no load or light load. ing. However, the converters disclosed in Patent Documents 2 to 4 also have the same or similar disadvantages as the above (1) to (5).

US Pat. No. 7,952,334 US Pat. No. 7,990,739 US Pat. No. 7,948,775 US Patent Application Publication No. 2011/0085354 Specification

  The present invention has been made in view of the above circumstances, and an object of the present invention is to provide a DC-DC converter that can suppress switching loss, voltage stress, and EMI of a semiconductor element, and can suppress increase in size and cost.

  According to a first aspect of the present invention, there is provided a DC-DC converter including a main switching element and a main inductor interposed in series with a first power supply line extending from one end of an input terminal to one end of an output terminal with an intermediate node interposed therebetween, Resonant circuit and freewheeling diode connected in parallel between the intermediate node and the second power line extending from the other end of the input terminal to the other end of the output terminal, and main switching based on the output voltage or output current from the output terminal And a control circuit for controlling the element.

  The circuit part excluding the resonant circuit in the above configuration constitutes a step-down chopper type DC-DC converter. However, in the conventional step-down chopper circuit, the main switching element and the free wheel diode are hard-switched. That is, ZVS (Zero-Voltage Switching) and ZCS (Zero-Current Switching) do not occur when the main switching element is turned off, and ZVS and ZCS do not occur when the main switching element is turned on. Further, when the main switching element is turned on, the freewheeling diode causes a recovery current to flow in the reverse direction due to the minority carrier accumulation effect. Such hard switching results in a decrease in conversion efficiency.

  Therefore, a resonant circuit is added to realize soft switching of the semiconductor element (main switching element, freewheeling diode). The resonant circuit includes a first diode and a first capacitor connected in series with a first node between a first power supply line and a second power supply line, and between the first power supply line and the second power supply line. And a second capacitor and a second diode connected in series across the second node, and a resonant inductor and a third diode connected in series between the first node and the second node. The control circuit drives the main switching element on to charge the first capacitor and the second capacitor, and then drives the main switching element off to discharge the first capacitor and the second capacitor. The operations and effects of this configuration are as follows. However, since the forward voltage of the diode is low, it is assumed that it is approximately zero. Further, the current flowing through the resonance circuit is referred to as resonance current.

  When the control circuit drives the main switching element ON, a resonance current flows from the input terminal via the first resonance path passing through the main switching element, the second capacitor, the third diode, the resonance inductor, and the first capacitor. The resonance current gradually increases and charges the first and second capacitors. Eventually, when the sum of the voltage of the first capacitor and the voltage of the second capacitor exceeds the voltage applied to the input terminal (hereinafter referred to as input voltage), the resonance current turns from increasing to decreasing.

  As an example, when the capacitance value of the first capacitor is smaller than the capacitance value of the second capacitor and the ON drive period has a sufficient width for charging the first and second capacitors, the voltage of the first capacitor is the second voltage. The input voltage is reached before the capacitor voltage and the first diode is energized. Thereafter, the resonance current flows through the second resonance path passing through the resonance inductor, the first diode, the second capacitor, and the third diode, and the energy of the resonance inductor is transferred to the second capacitor. When the resonance current eventually becomes zero, the voltage of the second capacitor reaches the vicinity of the input voltage, and the first diode and the third diode are disconnected.

  Conversely, when the capacitance value of the first capacitor is larger than the capacitance value of the second capacitor, the voltage of the second capacitor reaches the input voltage before the voltage of the first capacitor, and the second diode is energized. Thereafter, the resonance current flows through the third resonance path passing through the resonance inductor, the first capacitor, the second diode, and the third diode, and the energy of the resonance inductor is transferred to the first capacitor. When the resonance current eventually becomes zero, the voltage of the first capacitor reaches the vicinity of the input voltage, and the second diode and the third diode are disconnected. That is, energy corresponding to the input voltage (or a voltage close thereto) can be stored in the first and second capacitors by the resonance operation.

  In many cases, the inductance value of the main inductor is set larger than the inductance value of the resonant inductor. For this reason, a change in current flowing from the input terminal to the main inductor through the main switching element during the above-described resonance operation is reduced. Further, since energy is transmitted to the output side via the resonance circuit, the current flowing through the main inductor via the main switching element can be reduced accordingly.

  When the control circuit drives off the main switching element, the charges accumulated in the first and second capacitors are discharged through the first and second diodes, respectively, and the discharge current flows to the main inductor. Eventually, when the voltages of the first and second capacitors become zero, the freewheeling diode is energized and the freewheeling current flows. That is, the energy accumulated in the first and second capacitors can be regenerated to the load by the discharge operation and the reflux operation.

  Since the main switching element is turned off when the charging voltage of the first and second capacitors is equal to or close to the input voltage, the turn-off becomes almost ideal ZVS. Also, when the main switching element is driven off, the resonance current has not already flowed through the main switching element or the resonance current is small, so that the current cut off by the main switching element is reduced accordingly, and the turn-off approaches ZCS.

  Similarly, when the main switching element is turned on, the current that the main switching element should initially flow is zero or small, and accordingly, the current flowing through the main switching element at the time of turn-on decreases and approaches the ZCS. Further, since the freewheeling diode and the first, second, and third diodes do not change from the energized state to the reverse voltage applied state, soft switching is achieved in which no recovery current flows.

  According to this means, by providing the resonance circuit, soft switching of the semiconductor element can be realized and EMI can be reduced. The resonant circuit can regenerate the input energy to the load, and the voltage stress on the semiconductor element due to the resonant operation does not increase. Also, there is only one main switching element, and no dead time loss occurs even when the diode is replaced with a switching element as will be described later. Furthermore, since a transformer is unnecessary and can be constituted by at least one switching element, an increase in the size and cost of the DC-DC converter can be suppressed.

  According to the means described in claim 2, after the main switching element is driven on, the control circuit drives off the main switching element after the current flowing through the resonant inductor becomes zero. As a result, energy corresponding to the input voltage can be stored in the first and second capacitors of the resonance circuit.

  According to the means described in claim 3, after the main switching element is driven off, the control circuit drives the main switching element on after the current flowing through the first capacitor and the second capacitor becomes zero. Thereby, all the energy accumulated in the first and second capacitors can be regenerated to the load side.

  According to the means described in claim 4, the control circuit drives the main switching element on after the current flowing through the first capacitor and the second capacitor becomes zero and further after the current flowing through the freewheeling diode becomes zero. To do. Thereby, the turn-on of the main switching element can be an ideal ZCS.

  According to the means described in claim 5, some or all of the free wheel diode and the first to third diodes are replaced with transistors. The control circuit drives the transistor on during the energization period of the diode replaced with the transistor. By this synchronous switch operation, it is possible to reduce a loss caused by energization of the diode.

  According to the means described in claim 6, a plurality of sets of circuits each including a main switching element, a main inductor, a resonance circuit, and a free-wheeling diode are provided in parallel between the input terminal and the output terminal. The control circuit drives each main switching element with a different phase angle. With this multi-phase DC-DC converter, inductor ripple is reduced and transient response is increased. In addition, the current flowing through the main inductor can be reduced to 1 / (number of parallel connections).

The block diagram of the DC-DC converter which shows the 1st Embodiment of this invention Configuration diagram of control circuit The figure which shows the voltage waveform and current waveform of each part Explanatory diagram showing the actual current path Explanatory drawing which shows an ideal electric current path about Drawing 4 (a) and (b) FIG. 1 equivalent diagram showing a second embodiment of the present invention FIG. 1 equivalent view showing a third embodiment of the present invention FIG. 1 equivalent view showing a fourth embodiment of the present invention FIG. 3 equivalent diagram in the second to fourth embodiments FIG. 1 equivalent view showing a fifth embodiment of the present invention Waveform diagram of gate-source voltage

In each embodiment, substantially the same parts are denoted by the same reference numerals and description thereof is omitted.
(First embodiment)
Hereinafter, the first embodiment will be described with reference to FIGS. 1 to 5. FIG. 1 shows the configuration of a step-down DC-DC converter. The DC-DC converter 1 receives a DC voltage Vs from a DC power source 2 through input terminals 1a and 1b, and outputs a DC voltage Vo to a load 3 (indicated by a resistor Ro) through output terminals 1c and 1d. To do. Hereinafter, the DC voltages Vs and Vo are referred to as an input voltage Vs and an output voltage Vo, respectively.

  In the first power supply line 4 from the input terminal 1a to the output terminal 1c, an N-channel MOSFET Qm (main switching element) and a main inductor Lm are interposed in series with the intermediate node na interposed therebetween. A parasitic capacitor and a parasitic diode are formed in the MOSFET Qm. A capacitor Co that performs a filter action is connected between the output terminals 1c and 1d. The second power supply line 5 extending from the input terminal 1b to the output terminal 1d is a ground line of this circuit.

  A resonance circuit 6 and a free wheel diode Df are connected in parallel between the intermediate node na of the power supply line 4 and the power supply line 5. The resonance circuit 6 is connected in series with the first node D1 and the first capacitor C1 connected in series with the first node n1 between the power supply lines 4 and 5 and the second node n2 between the power supply lines 4 and 5. The second capacitor C2 and the second diode D2 are connected, and the resonance inductor Ls and the third diode D3 are connected in series between the first node n1 and the second node n2. In the present embodiment, the capacitance value of the first capacitor C1 is set smaller than the capacitance value of the second capacitor C2.

  The voltage detection circuit 7 includes, for example, a resistance voltage dividing circuit, and outputs a detection output voltage Vo (det) obtained by dividing the output voltage Vo. As shown in FIG. 2, the control circuit 8 includes a comparator 9, a reference voltage generation circuit 10, resistors 11 to 14, and a driver 15. The control circuit 8 compares the detected output voltage Vo (det) with the reference voltage Vref to obtain hysteresis. Control is performed to output a gate signal to the MOSFET Qm.

  Although not shown, the DC-DC converter 1 includes a current detection circuit that detects a resonance current flowing through the resonance circuit 6 (for example, a current that flows through the resonance inductor Ls), a current detection circuit that detects a current flowing through the main inductor Lm, and the like. Also good.

  Next, the operation of the present embodiment will be described with reference to FIGS. Since the forward voltage of the diode is low, it is approximately zero. FIG. 3 shows the gate-source voltage VGS (Qm) of the MOSFET Qm, the drain current ID (Qm) of the MOSFET Qm, the current ILs of the resonant inductor Ls, the voltage VC1 of the first capacitor C1, and the voltage VC2 of the second capacitor C2 from the top. The waveforms of the current IDf of the free wheel diode Df and the current ILm of the main inductor Lm are shown. FIGS. 4A to 4D show actual current paths that change in this order as time passes. FIGS. 5A and 5B show current paths when an ideal operation is assumed with respect to FIGS. 4A and 4B as described later.

  When the control circuit 8 turns on the MOSFET Qm at time t1, the input voltage Vs is applied to the series resonant circuit including the second capacitor C2, the resonant inductor Ls, and the first capacitor C1. At the same time, the input voltage Vs is also applied to the output circuit composed of the main inductor Lm and the capacitor Co (or the load 3).

  As a general configuration, the inductance of the main inductor Lm is larger than the inductance of the resonant inductor Ls. Therefore, when the MOSFET Qm is on, the impedance of the series resonant circuit is low, and the impedance of the main inductor Lm is relatively high. . For this reason, the main inductor Lm during the resonance period is regarded as open under ideal conditions where the impedance of the main inductor Lm is sufficiently large. FIG. 5 shows a current path in such an ideal case.

  When the MOSFET Qm is turned on, the current ILs of the resonant inductor Ls, the voltage VC1 of the first capacitor C1, and the voltage VC2 of the second capacitor C2 are zero. On the other hand, the current ILm flowing through the main inductor Lm when the MOSFET Qm is turned on is not zero in the current continuous mode and is zero in the current discontinuous mode. Changes in the drain current ID (Qm) flowing through the MOSFET Qm before and after the MOSFET Qm is turned on are limited by the resonant inductor Ls and the main inductor Lm.

  Therefore, in the current discontinuous mode, the drain current ID (Qm) becomes zero before and after the turn-on, so that the MOSFET Qm becomes an ideal ZCS. On the other hand, in the continuous current mode, the drain current ID (Qm) equal to the current ILm flows simultaneously with the turn-on, so that the ideal ZCS is not achieved. However, since the DC-DC converter 1 sends energy to the output side via the resonance circuit 6, the drain current ID (Qm) at the time of turn-on becomes smaller than that of the step-down chopper circuit of the conventional configuration, and can approach ZCS.

  Here, ZVS (Zero-Voltage Switching) of the MOSFET Qm refers to a switching state in which the drain-source voltage is maintained at zero during a period in which the drain current increases due to turn-on or a period in which the drain current decreases due to turn-off. Also, ZCS (Zero-Current Switching) of the MOSFET Qm refers to a switching state in which the drain current is maintained at zero during a period in which the drain-source voltage decreases due to turn-on or a period in which the drain-source voltage increases due to turn-off. . Under these switching states, the product of the drain-source voltage and the drain current becomes zero, and the switching loss is reduced. This is called soft switching of the switching element.

  Further, when a reverse voltage is applied from a state in which a forward current is flowing through the diode, a reverse recovery current flows through the diode during the recovery time due to a minority carrier accumulation effect. This is called hard switching of the diode following the switching element. On the other hand, no recovery current flows when a reverse voltage is applied from a state in which no forward current flows through the diode. This is called soft switching of the diode.

  When the MOSFET Qm is turned on, since no current flows through the first, second, and third diodes D1, D2, and D3, each diode is soft-switched. On the other hand, the freewheeling diode Df performs soft switching in the current discontinuous mode and performs hard switching in the current continuous mode. However, since the DC-DC converter 1 sends energy to the output side via the resonance circuit 6, the current flowing through the freewheeling diode Df at the time of turn-on becomes smaller than that of the step-down chopper circuit of the conventional configuration, and approaches soft switching.

  In the turn-on period, two different resonance paths are formed. The first resonance path is a path that passes from the DC power source 2 through the MOSFET Qm, the second capacitor C2, the third diode D3, the resonance inductor Ls, and the first capacitor C1, as shown in FIGS. 4 (a) and 5 (a). . In the first resonance stage, the resonance current gradually increases and charges the first capacitor C1 and the second capacitor C2. At this time, the resonance current ILs flowing through the resonance inductor Ls changes as shown by the equations (1) to (4).

The peak value of the current ILs flowing through the resonant inductor Ls is expressed by equation (5).

  Eventually, when the sum of the voltage VC1 of the first capacitor C1 and the voltage VC2 of the second capacitor C2 exceeds the input voltage Vs, the applied voltage to the resonant inductor Ls changes from positive to negative. Turn (time t2). Since the capacitance value of the first capacitor C1 is smaller than the capacitance value of the second capacitor C2, the voltage VC1 of the first capacitor C1 reaches the input voltage Vs before the voltage VC2 of the second capacitor C2 (time t3). As a result, the diode D1 starts energization, and the first resonance stage ends.

  In the subsequent second resonance stage, a second resonance path passing through the resonance inductor Ls, the first diode D1, the second capacitor C2, and the third diode D3 is formed as shown in FIGS. The energy of the resonant inductor Ls is transferred to the second capacitor C2. At this time, the resonance current ILs flowing through the resonance inductor Ls changes as shown by the equations (6) to (8). The resonance angular frequency ω2 at the second resonance stage expressed by the equation (8) is lower than the resonance angular frequency ω1 at the first resonance stage expressed by the equation (4).

  Eventually, when the resonance current becomes zero at time t4 and the voltage VC2 of the second capacitor C2 reaches the input voltage Vs, the currents of the first diode D1 and the third diode D3 become zero. When the energy stored in the resonant inductor Ls becomes excessive, a path from the resonant inductor Ls to the first diode D1, the main inductor Lm, the capacitor Co (or load 3), the second diode D2, and the third diode D3. Resonance current flows, and excess energy is sent to the output side. When the resonance current becomes zero, the second resonance phase ends.

  If the capacitance value of the first capacitor C1 is larger than the capacitance value of the second capacitor C2, the voltage VC2 of the second capacitor C2 reaches the input voltage Vs before the voltage VC1 of the first capacitor C1. In the second resonance stage in this case, a second resonance path passing through the resonance inductor Ls, the first capacitor C1, the second diode D2, and the third diode D3 is formed, and the energy of the resonance inductor Ls is transferred to the first capacitor C1. .

  When the detected output voltage Vo (det) becomes higher than the reference voltage Vref and the output signal of the comparator 9 becomes L level at time t5, the control circuit 8 drives the MOSFET Qm off. As a result, the process proceeds to the discharge stage shown in FIG. When the MOSFET Qm is turned off, since the first capacitor C1 and the second capacitor C2 are charged up to the input voltage Vs, the drain-source voltage becomes zero and becomes ZVS. Further, under the ideal conditions shown in FIG. 5, since the current flowing through the main inductor Lm is zero, the drain current ID (Qm) is also zero and becomes ZCS. On the other hand, as shown in FIG. 4, even when a current flows through the main inductor Lm, the current flowing through the main inductor Lm at the time of turn-off becomes smaller than the step-down chopper circuit of the conventional configuration and approaches ZCS.

  When the MOSFET Qm is turned off, the charge accumulated in the first capacitor C1 and the charge accumulated in the second capacitor C2 are discharged through the first diode D1 and the second diode D2, respectively, and the discharge current is supplied to the main inductor Lm. Flowing. At this time, the voltage VC1 of the first capacitor C1 and the voltage VC2 of the second capacitor C2 start drooping toward zero at a moderate rate according to the equation (9). Io is the output current, and Δt is the elapsed time from time t5.

  Eventually, when the voltage VC1 of the first capacitor C1 and the voltage VC2 of the second capacitor C2 become zero at time t7, as shown in FIG. 4D, the freewheeling diode Df is energized and the freewheeling current flows. Transition. Through the discharge stage and the free wheel stage, the energy stored in the first capacitor C1 and the second capacitor C2 can be supplied (regenerated) to the load 3. Thereafter, when the detection output voltage Vo (det) falls below the reference voltage Vref and the output signal of the comparator 9 becomes H level, the control circuit 8 drives the MOSFET Qm on again.

  According to the present embodiment described above, the high-side first power supply line 4 includes the MOSFET Qm and the main inductor Lm in series, and converts the input voltage Vs to a desired output voltage Vo lower than that. 1 is obtained. Since the resonance circuit 6 is provided between the intermediate node na of the first power supply line 4 and the second power supply line 5, the MOSFET Qm can be turned off by ideal ZVS and ideal or approximate ZCS. . Further, the MOSFET Qm can be turned on with an ideal or approximate ZCS. Furthermore, the free-wheeling diode Df can be soft-switched so that a recovery current does not flow ideally or approximately. The DC-DC converter 1 can soft-switch these semiconductor elements in a very wide input voltage range and a very wide load range.

  As a result, the switching loss is reduced, so that the efficiency of the DC-DC converter 1 can be increased. Further, since generation of a surge voltage or a recovery current having a steep waveform can be suppressed, EMI can be reduced. Further, by increasing the switching frequency, passive components such as the inductors Ls, Lm, and the capacitor Co can be reduced while suppressing the switching loss, and the size and cost of the DC-DC converter 1 can be reduced. In addition, output voltage ripple can be reduced.

  Since the resonance circuit 6 can regenerate all the energy accumulated in the first capacitor C1 and the second capacitor C2 to the load side by the resonance operation, there is no energy dissipation due to the resonance operation. Further, energy is stored in the resonance circuit 6 during the ON period of the MOSFET Qm, and the stored energy is sent to the load side during the OFF period, so that the energy transfer efficiency can be increased. Furthermore, the voltage applied to the semiconductor elements (MOSFET Qm, diodes D1 to D3, Df) and the passive elements is not more than the input voltage Vs in spite of the resonance operation, and voltage stress does not increase. The diodes D1 to D3 of the resonance circuit 6 perform soft switching.

  Since the main switching element is only one MOSFET Qm and does not have a configuration of upper and lower arms, one driver circuit is sufficient for the switching operation, and no dead time loss occurs. Furthermore, the transformer which each converter of prior art literature had was unnecessary, and the increase in the size and cost of the DC-DC converter 1 can be suppressed.

(Second to fourth embodiments)
6 to 8 show the configurations of the DC-DC converters of the second to fourth embodiments, respectively. FIG. 9 is a waveform diagram commonly used in describing these embodiments. IDf shown in FIG. 9 is a current flowing in a MOSFET Qf described later. These second to fourth embodiments have a configuration in which each diode is sequentially replaced with a synchronous switch made of a MOSFET, as compared with the first embodiment. In this case, in order to accurately control the on / off drive timing of the synchronous switch, it is preferable to include a current detection circuit that detects a resonance current flowing through the resonance inductor Ls, a voltage detection circuit that detects each voltage of the nodes n1 and n2, and the like. .

  A DC-DC converter 21 shown in FIG. 6 is obtained by replacing the free-wheeling diode Df with a MOSFET Qf, and the MOSFET Qm and Qf are driven by the control circuit 22. The gate-source voltage VGS (Qf) of the MOSFET Qf is turned on during the period from time t7 to t8.

  The DC-DC converter 31 shown in FIG. 7 is obtained by replacing the first diode D1 and the second diode D2 of the resonance circuit 32 with MOSFETs Q1 and Q2, respectively. The control circuit 33 drives the MOSFETs Qm, Qf, Q1, and Q2. To do. The gate-source voltage VGS (Q1) of the MOSFET Q1 becomes the on drive level in the period from time t3 to t7. However, the period from time t4 to t5 may be an off drive level. The gate-source voltage VGS (Q2) of the MOSFET Q2 becomes the on drive level during the period from time t5 to t7. The MOSFETs Q1 and Q2 are turned on at ZVS and turned off at ZVS and ZCS.

  A DC-DC converter 41 shown in FIG. 8 is obtained by replacing the third diode D3 of the resonance circuit 42 with a MOSFET Q3. The control circuit 43 drives the MOSFETs Qm, Qf, Q1, Q2, and Q3. The gate-source voltage VGS (Q3) of the MOSFET Q3 is turned on during the period from time t1 to t4. MOSFET Q3 is turned on and off at ZCS. According to the second to fourth embodiments described above, the same operation and effect as those of the first embodiment can be obtained, and loss generated in the diode can be reduced.

(Fifth embodiment)
Next, a fifth embodiment will be described with reference to FIGS. 10 and 11. A DC-DC converter 51 shown in FIG. 10 has a multi-phase configuration in which n (n = 2, 3,...) DC-DC converters 1 shown in FIG. 1 are connected in parallel. FIG. 11 shows the waveforms of the gate-source voltages VGS (Qm1), VGS (Qm2), and VGS (Qm3) of the MOSFETs Qm1, Qm2, and Qm3 when n = 3. In this embodiment, PWM control is performed, and the control circuit 52 drives the MOSFETs Qm1, Qm2, and Qm3 with a phase difference of 120 °. FIGS. 11A and 11B show waveforms when the duty ratio is 33% and 50%, respectively. According to the present embodiment, inductor ripple and output voltage ripple are reduced, and transient response is improved. Further, the current flowing through the main inductor Lm can be reduced to 1 / n.

(Other embodiments)
As mentioned above, although preferred embodiment of this invention was described, this invention is not limited to embodiment mentioned above, A various deformation | transformation and expansion | extension can be performed within the range which does not deviate from the summary of invention.

The capacitance value of the first capacitor C1 and the capacitance value of the second capacitor C2 may be equal.
The control circuits 8, 22, 33 and 43 may perform PWM control. The control circuit 52 may perform hysteresis control. Further, the control circuits 8, 22, 33, 43, and 52 may perform output current control for controlling the output current Io instead of the output voltage control for controlling the output voltage Vo.

  In each of the above embodiments, the MOSFET Qm is turned off after the current flowing through the resonant inductor Ls becomes zero (after time t4) after the MOSFET Qm is turned on. However, when the efficiency is allowed to decrease, the MOSFET Qm may be driven off before the current flowing through the resonant inductor Ls becomes zero.

  In each of the above embodiments, the MOSFET Qm is turned on after the current flowing through the first capacitor C1 and the second capacitor C2 becomes zero after the MOSFET Qm is turned off. However, when the efficiency is allowed to decrease, the MOSFET Qm may be turned on before the current flowing through the first capacitor C1 and the second capacitor C2 becomes zero.

  Not limited to the second to fourth embodiments, in general, some or all of the free wheel diode Df and the first to third diodes D1, D2, and D3 can be replaced with transistors. In this case, at least in the energization period of the diode replaced with the transistor, the replaced transistor may be driven on. These embodiments may have a multi-phase configuration as in the fifth embodiment.

  The main switching element and the transistor are not limited to MOSFETs but may be semiconductor elements such as IGBTs and bipolar transistors.

  In the drawings, 1, 21, 31, 41, 51 are DC-DC converters, 1a, 1b, 21a, 21b, 31a, 31b, 41a, 41b, 51a are input terminals, 1c, 1d, 21c, 21d, 31c, 31d. , 41c, 41d and 51c are output terminals, 4 is a power supply line (first power supply line), 5 is a power supply line (second power supply line), 6, 32 and 42 are resonance circuits, and 8, 22, 33, 43 and 52 are provided. Are control circuits, Qm, Qm1, Qm2,..., Qmn are MOSFETs (main switching elements), Qf, Q1, Q2, and Q3 are MOSFETs (transistors), D1 is a first diode, D2 is a second diode, and D3 is a third diode. Diode, Df is freewheeling diode, C1 is first capacitor, C2 is second capacitor, Lm is main inductor, Ls is resonant inductor, n1 is first node, n2 is second node, and na is intermediate It is over de.

Claims (6)

A main switching element and a main inductor interposed in series with a first power line extending from one end of the input terminal to one end of the output terminal with an intermediate node interposed therebetween;
A resonant circuit and a free wheel diode connected in parallel between the intermediate node of the first power line and the second power line extending from the other end of the input terminal to the other end of the output terminal;
A control circuit for controlling the main switching element based on an output voltage or an output current from the output terminal,
The resonant circuit is:
A first diode and a first capacitor connected in series with a first node between the first power line and the second power line;
A second capacitor and a second diode connected in series with a second node between the first power line and the second power line;
A resonance inductor and a third diode connected in series between the first node and the second node;
The control circuit drives the main switching element on to charge the first capacitor and the second capacitor, and then drives the main switching element off to discharge the first capacitor and the second capacitor. DC-DC converter characterized by this.   2. The DC-DC converter according to claim 1, wherein the control circuit drives the main switching element off after the current flowing in the resonant inductor becomes zero after the main switching element is turned on.   2. The control circuit according to claim 1, wherein after the main switching element is turned off, the main switching element is turned on after the currents flowing through the first capacitor and the second capacitor become zero. 2. The DC-DC converter according to 2.   The control circuit is configured to turn on the main switching element after the current flowing through the first capacitor and the second capacitor becomes zero and further after the current flowing through the freewheeling diode becomes zero. The DC-DC converter according to claim 3. A part or all of the free wheel diode and the first to third diodes are replaced with transistors,
5. The DC-DC converter according to claim 1, wherein the control circuit drives the transistor on during a current-carrying period of a diode replaced with the transistor. Between the input terminal and the output terminal, a plurality of sets of circuits each including the main switching element, the main inductor, the resonance circuit, and the reflux diode are provided in parallel,
6. The DC-DC converter according to claim 1, wherein the control circuit drives the main switching elements at different phase angles.

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