JP2013021787A - Synchronous motor control apparatus - Google Patents

Abstract

PROBLEM TO BE SOLVED: To reduce effects due to a current ripple, etc. at inversion of a torque command, for example, thereby ensuring smooth drive.SOLUTION: Presented here is a control apparatus for a synchronous motor including a stator which has a salient pole structure so that magnetic resistance has a level difference with respect to the needle advance direction and a needle made of a soft magnetic material, whose permanent magnet is disposed in the advance direction so as to have a magnetic polarity periodically changing in the advance direction on a plane opposed to the stator, the needle having a coil wound round it so that the magnetic polarity on the needle surface changes when current is applied to the coil. In the control apparatus, the polarity of a permanent magnet flux component on the needle surface is made to change according to the polarity of a torque command, or a weight arithmetic unit is provided in which magnet flux of the permanent magnet of the needle is calculated for a first pole pair and a second pole pair portion differing by πrad in electric angle independently of each other, and then the weights of first and second magnetic flux is calculated according to the relative positions of the stator salient and the needle and the polarity of the torque command.

Description

  The present invention relates to a control device for a synchronous motor used in a machine tool or the like, and more particularly, a stator having a magnetic salient pole, and a synchronous motor called an inductor type in which a permanent magnet and a winding are provided on a mover. The present invention relates to a control apparatus for a synchronous motor that can perform a stable and smooth drive without being affected by a current ripple due to an inductance change caused by the structure of the motor by performing current control and voltage control with high accuracy.

  FIG. 9 shows an example of a block diagram of a general synchronous motor control device.

  Since the known vector control is based on the current control method, for the convenience of explanation, the expression of the field component is d-axis and the torque (armature) current component is q-axis.

  In FIG. 9, STQC is a torque command value, SIQC is a q-axis current command value, SVU, SVV, and SVW are U-phase, V-phase, and W-phase voltage command values, SPHS is a phase value, and SPD is a mover position detection value. , SVD is a mover speed detection value.

  Reference numeral 1 denotes a q-axis current command calculation unit that converts a torque command value STQC from the host controller into a q-axis current command value SIQC.

  The q-axis current command value SIQC is calculated by the 2 / 3-phase converter 2 as a 3-phase voltage command, that is, current command values for the U phase, V phase, and W phase each having a phase difference of 2π / 3. The d-axis current command value SIDC in vector calculation is treated as zero.

  Although not shown in particular, each phase current detection value obtained by the current detection means provided for detecting the current applied to the electric motor 9 is fed back, and the current command value is compared with the current command value by a calculator such as a PI calculator. By calculating the deviation, the voltage command values SVU, SVV, SVW of each phase are obtained.

  Reference numeral 4 denotes a power converter, which applies a voltage based on the voltage command values SVU, SVV, SVW to the electric motor 9 to cause a current to flow through each phase winding of the electric motor 9.

  Reference numeral 10 denotes a mover position detecting means.

  Normally, the mover position detection means 10 is directly connected to the mover of the electric motor 9 and outputs the mover position detection value SPD to the phase calculation unit 7 and the speed calculation unit 5.

  The speed calculation unit 5 performs a calculation process such as differentiation on the mover position detection value SPD to obtain the mover speed detection value SVD.

  The phase calculation unit 7 outputs the U-phase, V-phase, and W-phase q-axis current command value phase values SPHS to the voltage command calculation unit 3 based on the mover position detection value SPD obtained from the mover position detection means 10.

  Representative examples of the synchronous motor controlled by the control device include a permanent magnet (SPM) type in which a permanent magnet is attached to the surface of the mover, a permanent magnet insertion (IPM) type in which the permanent magnet is covered with a soft magnetic material, and A reluctance (RM) type in which the mover is composed only of a soft magnetic material can be used.

  Both synchronous motors obtain the thrust of the motor by controlling the q-axis (armature) current that is a torque component.

  A reluctance type or permanent magnet interpolated synchronous motor that uses reluctance force has a permeance change or inductance due to the shape of the mover structure and stator teeth in the moving direction as seen from the stator winding on the surface of the mover. Changes occur, which change the energy stored between the stator and the mover and cause torque ripple.

  Therefore, when it is used for a feed axis of a machine tool, torque ripple appears as a streak on the workpiece, which causes problems such as machining defects and noise and vibration when the motor is driven.

JP2001-45736

  When controlling a conventional electric motor, for example, a permanent magnet (SPM) type in which a permanent magnet is attached to the surface of a mover, or a permanent magnet insertion (IPM) type synchronous motor in which a permanent magnet is covered with a soft magnetic material, the permanent magnet is used as a stator. The permanent magnet or winding is independently provided for each stator, such as a winding (or vice versa) on the mover, and the polarity of the magnetic flux of the permanent magnet when current / voltage control is performed. Is always fixed to either positive or negative and control is performed.

  When the electric motor to which the present invention is applied uses a reluctance force, the stator has a magnetic salient pole, and the movable motor includes a permanent magnet and a winding, when performing voltage control, Since the magnetic flux polarity of the permanent magnet changes in accordance with the polarity of the thrust, the conventional voltage control method for d-axis voltage control in vector control has a control voltage error in a steep operation such as high-speed operation or direction reversal. Becomes larger, an overcurrent flows through the motor winding, and the control becomes unstable, or a change in energy stored in the winding inductance increases, causing torque ripple.

  The object of the present invention is to realize efficient high-bandwidth driving in a synchronous motor, particularly an inductor type synchronous motor using reluctance force and permanent magnet force, and stable control with low torque ripple at low speed rotation or high torque generation It is an object of the present invention to provide a control device for a synchronous motor that can improve efficiency and improve efficiency.

  The above object is achieved by the following means.

  The present invention comprises a stator having a salient pole structure so that the magnetic resistance periodically has a height difference with respect to the moving direction of the mover, and a soft teeth material surface of the mover facing the stator. In addition, a permanent magnet having a different polarity is arranged adjacent to the moving direction so as to have a magnetic polarity that periodically changes in the moving direction of the moving element, and the moving element is wound with a winding. , A control device for a synchronous motor in which a magnetic pole property of a tooth surface of the mover changes by applying a current to a winding, and a voltage command is calculated according to a polarity of a torque command corresponding to a moving direction of the mover The polarity of the permanent magnet magnetic flux component on the surface of the mover used in the above is changed.

  In addition, the magnetic flux of the permanent magnet of the mover is independently dq-axis vector-calculated for the second pole pair portion that differs by π rad in electrical angle from the first pole pair. It is preferable to include a weight calculation unit that performs weighting calculation of the first and second magnetic fluxes according to the relative position.

  As a result, the d-axis (permanent magnet) magnetic flux component changes according to the polarity of the thrust and is used for voltage control. Therefore, according to the present invention, efficient and stable control during high-speed driving can be obtained, and Control to reduce torque ripple of the electric motor becomes possible.

  As described above, according to the present invention, in the inductor type synchronous motor using the reluctance force and the permanent magnet force, it is possible to realize a high bandwidth capable of stable driving even at high speed operation, and problems such as overcurrent. As a result, it is possible to obtain a control device for a synchronous motor that can achieve stable controllability with low torque ripple.

It is explanatory drawing which shows 1st Embodiment of the control apparatus of the synchronous motor of this invention. It is the figure which showed the mode of the magnetic flux at the time of the coil non-energization of the synchronous motor to which the control apparatus of the synchronous motor of this invention is applied. It is an operation principle diagram of the synchronous motor to which the control device for the synchronous motor of the present invention is applied. It is a mode and operation example of the magnetic flux at the time of coil energization of a synchronous motor to which a control device of a synchronous motor of the present invention is applied. It is a mode and operation example of the magnetic flux at the time of coil energization of a synchronous motor to which a control device of a synchronous motor of the present invention is applied. It is explanatory drawing which shows the winding linkage magnetic flux of the synchronous motor of this invention. It is explanatory drawing which shows 2nd Embodiment of the control apparatus of the synchronous motor of this invention. It is supplementary explanatory drawing in 2nd Embodiment of the control apparatus of the synchronous motor of this invention. It is explanatory drawing which shows the control apparatus form of the conventional synchronous motor.

  Embodiments of the present invention will be described below with reference to the accompanying drawings. Unless otherwise noted, the same symbols, number elements, signals, etc. have the same functions and performance.

  In this description, an inductor type linear synchronous motor using a reluctance force and a permanent magnet force is used, and the description will be made with a two-phase model for convenience of description.

  FIG. 1 is an example of an embodiment of the present invention and an example of a control block diagram.

  STQC is the torque command value, SIQC is the q-axis current command value, SVU, SVV, and SVW are the U-phase, V-phase, and W-phase voltage command values, SPHS is the phase value, SPD is the mover position detection value, and SVD is movable This is the child speed detection value.

  Reference numeral 1 denotes a q-axis current command calculation unit that converts a torque command value STQC from the host controller into a q-axis current command value SIQC.

  The q-axis current command value SIQC is supplied to the 2 / 3-phase conversion unit 2, and the 2/3 conversion unit 2 converts the q-axis current command value into a three-phase voltage command, that is, a U phase having a phase difference of 2π / 3. , V-phase and W-phase current command values are calculated.

  Although not shown in particular, each phase current detection value obtained by the current detection means provided for detecting the current applied to the electric motor 9 is fed back, and the current command value is compared with the current command value by a calculator such as a PI calculator. A deviation is calculated, and voltage command values SVU, SVV, SVW for each phase are calculated based on the deviation.

  Reference numeral 4 denotes a power converter, which applies a voltage based on the voltage command values SVU, SVV, SVW to the electric motor 9 to cause a current to flow through each phase winding of the electric motor 9.

  Reference numeral 10 denotes a mover position detecting means.

  Normally, the mover position detection means 10 is directly connected to the mover of the electric motor 9 and outputs the mover position detection value SPD to the phase calculation unit 7 and the speed calculation unit 5.

  The speed calculation unit 5 calculates a mover speed detection value SVD from the mover position detection value SPD by performing a calculation process such as differentiation on the mover position detection value SPD.

  The phase calculation unit 7 calculates the U-phase, V-phase, and W-phase q-axis current command value phase values SPHS based on the mover position detection value SPD obtained from the mover position detection means 10, and uses this to calculate the voltage command calculation unit. 3 is output.

  The feature of this embodiment is that the polarity of the torque command STQC, which is the thrust polarity of the mover, is determined by the polarity determination unit 8, and the determination result is reflected in the polarity of the magnetic flux value SPHIM by the d-axis magnetic flux calculation unit 6. Thus, the voltage command calculation unit 3 performs the voltage command calculation with the magnetic flux value and polarity according to the thrust polarity.

  Without the above processing, in the inductor type electric motor treated in this description, the magnetic flux due to the winding, the magnetic flux of the permanent magnet, and the inductance change due to the structure of the stator and the mover of the electric motor are not taken into consideration. Since the calculation is performed, the harmonics are superimposed on the current flowing through the motor windings, and thus ripples are generated in the currents. Therefore, the resultant thrust is also rippled.

  FIGS. 2A to 2C are examples of an electric motor to which the synchronous motor control device of the present invention is applied, in which no current is applied to the windings C1 and C2.

  The electric motor of this example is an inductor type electric motor that obtains a magnetic attractive force (reluctance force) by the magnetic salient pole of the stator 21 and uses the permanent magnets M1 and M2 of the mover 22.

  The stator 21 has only a magnetic salient pole that protrudes upward in the figure, and has no permanent magnet or winding. The stator 21 is composed of a belt-like yoke extending linearly in the horizontal direction in the figure and a trapezoidal salient pole projecting upward from the yoke, and the salient poles are formed at regular intervals in the longitudinal direction of the yoke. The shape is as follows. Accordingly, the stator 21 has salient pole structures in which the concave portions and the salient poles are alternately positioned, and the magnetic resistance periodically has a difference in height with respect to the moving direction of the mover described later.

  The mover 22 is made of a soft magnetic material, and includes a tooth extending downward in the drawing and a yoke connecting the teeth. In this example, the mover 22 has two teeth, and has a U-shape as a whole. Windings C1 and C2 are wound around the two teeth of the mover 22, respectively. In addition, a pair of permanent magnets M1 and M2 are provided at the tips of the teeth of the mover 22 facing the recesses or salient poles of the stator 21 respectively. In this example, a permanent magnet M1 on the left side and a permanent magnet M2 on the right side are provided at the tip of the left tooth in the figure, and a permanent magnet M2 on the left side and a permanent magnet M1 on the right side at the tip of the right tooth in the figure. Is provided. In this example, the width of the teeth of the mover 22 is twice the width of the salient poles of the stator 21, and the width of the recess in the middle of the mover 22 corresponds to the width of the salient poles of the stator 21.

  The permanent magnet M1 generates a magnetic flux φm1 and the permanent magnet M2 generates a magnetic flux φm2. The right permanent magnet M1 in the figure has an N pole on the lower side, an S pole on the upper side, and a permanent magnet M1 on the left side. In the figure, the lower side is the S pole and the upper side is the N pole. The right side permanent magnet M2 in the figure is the lower side in the figure, the upper side is the N pole, and the left side permanent magnet M2 is the lower side in the figure. The upper side is the S pole. The magnetic flux φm1 and the magnetic flux φm2 are preferably the same.

  Thus, the mover 22 has permanent magnets M1 and M2 of different polarities adjacent to each other on the surface of the teeth facing the stator 21 so as to have a magnetic polarity that periodically changes in the travel direction of the mover 22. Are arranged.

  FIG. 2A shows the state of the magnetic flux when the permanent magnet M1 of the mover 22 is opposed to the two salient poles of the stator 21 and the two permanent magnets M2 are opposed to the recesses of the stator. Show. In this case, the magnetic flux φm1 generated by the two permanent magnets M1 is dominant in the magnetic flux interlinking the windings C1 and C2. However, since the permanent magnet M1 forms a magnetic loop with the magnetic flux φ2 with the adjacent permanent magnet M2 on the same tooth, the magnetic flux interlinking the windings C1, C2 is all the magnetic flux φm1 of the permanent magnet M1. Are not interlinked.

  Hereinafter, in the description, the magnetic flux in the mover 22 is expressed as φa, and the magnetic flux in the stator 21 is expressed as φs. In the case of FIG. 2A, a magnetic loop that rotates around the mover 22 and the stator 21 is formed by the magnetic fluxes φa and φs.

  In FIG. 2B, the permanent magnet M1 and the permanent magnet M2 of the movable element 22 are opposed to the salient pole of the stator 21 by 1/2, and the concave portion of the stator 21 is also opposed by 1/2. The state of the magnetic flux in the case is shown. In this case, since a magnetic loop is formed between the magnetic flux φm1 of the permanent magnet M1 and the magnetic flux φ2 of the adjacent permanent magnet M2, there is almost no magnetic flux interlinking the windings C1 and C2, and therefore, in the mover 22 The magnetic flux φa and the magnetic flux φs in the stator 21 are almost zero.

  2C shows a state in which the position of the mover is deviated by π rad in electrical angle with respect to FIG. 2B, and the permanent magnet M1 of the mover 22 is arranged on the salient pole of the stator 21 as in FIG. And the permanent magnet M2 are opposed to each other by 1/2, and the state of the magnetic flux in the case where the concave portions are also opposed to each other by 1/2 is shown. Since a magnetic loop is formed between the magnetic flux φm1 of the permanent magnet M1 and the magnetic flux φ2 of the permanent magnet M2, there is almost no magnetic flux interlinking the windings C1, C2, and therefore the magnetic flux φa in the mover 22 and the stator Similarly, the magnetic flux φs in 21 is almost zero.

  FIG. 3 is an example in the case of applying the general concept of vector control to the inductor type synchronous motor as shown in FIG. 2 applied in the present invention, and is a principle diagram of the synchronous motor control method of the present invention.

  FIG. 3A shows a method for setting the d and q axes between the mover 22 and the stator 21. The + d axis and the -d axis are set at the magnetic pole center according to the polarity of the mover permanent magnet. The boundary between the permanent magnets M1 and M2 (the magnetic pole polarity reversal part) is set as the q axis. With the center of the magnetic salient pole of the stator 21 as a reference, by setting the electrical angle to 0 or 2π at intervals of the period λ, the d-axis and q-axis of the mover 22 with respect to the reference 0 (or 2πrad) The phase of the current applied to the windings C1 and C2 is determined according to the relative position.

  3 (b) and 3 (c) show the state of FIG. 3 (a) as a vector diagram, and FIG. 3 (b) shows the case of FIG. 3 (c) when the thrust polarity is positive (F> 0). Indicates the case where the thrust polarity is negative (F <0). What is characteristic here is that the polarity of the d-axis magnetic flux component of the permanent magnet changes according to the polarity of the thrust (whether the thrust is positive or negative).

  When vector control is performed with a motor using a general permanent magnet, the polarity of this d-axis magnetic flux or d-axis current component is constant regardless of the polarity of the thrust, and is fixed to either positive or negative polarity. Therefore, the control method of the present invention is greatly different.

  As can be seen from FIGS. 3B and 3C, another feature of this control method is that the vehicle is decelerating from the state of acceleration (FIG. (B)) in the same direction (arrow Vel) (FIG. (C)). Even in the case of performing, it is a point where the regenerative operation is hardly performed and the power running state is obtained. For this reason, there are two sets of inductor motors handled in the present invention so that the magnetic properties of the pole pairs of the permanent magnets on the surface of the mover are different, and the set of permanent magnets is switched according to the polarity of thrust. It is good to consider that. That is, there are a case where a pair of permanent magnets M1 is used and a case where a pair of permanent magnets M2 is used.

  FIGS. 4 and 5 are diagrams illustrating the case where current is applied to the windings C1 and C2, and shows that the generated thrust polarity differs depending on the direction of the magnetic flux generated by the windings C1 and C2. .

  4A and 5A are in a non-energized state, the relative positions of the stator 21 and the mover 22 are the same, and the magnetic fluxes φa and φb are also the same. FIG. 2A is the same state.

  4 (b) and 4 (c), magnetic fluxes φc1 and φc2 (counterclockwise in the drawing from the right tooth through the yoke to the left tooth in the mover 22 in the direction of the drawing due to the current flowing in the windings C1 and C2. (B) and (c) differ in the direction of the generated thrust due to the difference in the relative positions of the mover 22 and the stator 21 even if the winding magnetic flux is the same. The generated force is a magnetic attractive force (reluctance force) F0 acting between the permanent magnets M1 and M2, the winding magnetic fluxes φc1 and φc2, and the stator 22 salient pole, and is decomposed into attractive forces in the x and y directions in the drawing. The thrust used as an electric motor is a force Fx in the x direction. In FIG. 4A, the magnetic flux φm1 is in the same direction as the magnetic fluxes φc1 and φc2, and in FIG. 4B, the magnetic flux φm2 is in the same direction as the magnetic fluxes φc1 and φc2.

  In FIGS. 5B and 5C, magnetic fluxes φc1 and φc2 are generated in the clockwise direction in the mover 22 toward the drawing by the current flowing through the windings C1 and C2, as opposed to FIG. 4, even if the winding magnetic flux is the same, the direction of thrust generated differs between (b) and (c) due to the difference in relative position between the mover 22 and the stator 21.

  FIG. 6 shows the change of the interlinkage magnetic flux with respect to the moving direction of the mover 22 of the winding C1 (or winding C2). (A) is a positive thrust (F> 0), (b) is This is a magnetic flux change in the case of negative thrust (F <0). The magnetic flux generated by applying a current to the winding C1 or C2 is φc, and the magnetic flux φ increases in proportion to the magnitude of the current. The curved lines shown by broken lines indicate the magnetic fluxes generated by the permanent magnets M1 and M2, and the magnetic flux φm1 generated by the permanent magnet M1 and the magnetic flux φm2 generated by the permanent magnet M2 are dominant depending on the relative positions of the mover 22 and the stator 21. Become.

  Thus, in the present embodiment, even if the relative positions of the mover 22 and the stator 21 are the same according to the acceleration direction (thrust direction) of the mover 22, the polarity of the d axis in the vector control is changed. The desired thrust can be obtained. For example, the movement of the mover 22 in the positive direction controls the magnetic field by the windings C1 and C2 as shown in FIG. 6 (a) in the state of FIG. 5 (b) and FIG. 4 (c). Do that. When the mover 22 is moved in the negative direction, the state shown in FIG. 5C and the state shown in FIG. 4B are changed to the magnetic field generated by the windings C1 and C2, as shown in FIG. This is done by controlling

  In the current control of the windings C1 and C2, vector control for controlling the d-axis and q-axis magnetic flux is performed. In this case, as described above, the polarity of the d-axis is switched depending on the direction of thrust at that time. Thereby, appropriate thrust control can be achieved.

  FIG. 7 shows another example of the embodiment of the present invention. The difference from FIG. 1 is that a first dq vector calculation unit 71 and a second dq vector calculation unit 72 are provided. The first dq vector calculation unit 71 and the second dq vector calculation unit 72 calculate the magnetic fluxes of the permanent magnet M1 and the permanent magnet M2 independently, respectively, and the weight function calculation unit 73 calculates the phase information (fixed with the movable element 22). The weighting is performed according to the relative position of the child 21).

  It should be noted that the vectors handled by the first dq vector computing unit 71 and the second dq vector computing unit 72 are characterized in that the π rad phase is shifted by an electrical angle. That is, in FIGS. 6A and 6B, in the range of −π / 2 to π / 2, the magnetic flux φm1 by the permanent magnet M1 is used, and the first dq vector calculation unit 71 performs vector calculation. . At this time, the calculation is performed by changing the polarity of the d-axis according to the polarity of the thrust. In the range of π / 2 to −π / 2, the magnetic flux φm2 by the permanent magnet M2 is used, and the second dq vector calculation unit 72 performs vector calculation. Also in this case, the calculation is performed by changing the polarity of the d-axis according to the polarity of the thrust.

  FIG. 8 shows an example of the weighting function calculation unit 73. From the calculation result of the phase calculation unit 7, the weighting function 81 for the first dq vector calculation unit 71 and the weighting function 82 for the second dq vector calculation unit 72 are shown. Is calculated. The weighting function 81 and the weighting function 82 intersect with each other only in a minute section of 2Δθ in phase, and are set according to a change in the permanent magnet magnetic flux that affects the structure such as a permanent magnet mounting interval on the surface of the mover.

  Although not particularly illustrated, the weighting function 81 and the weighting function 82 also change the polarity of the weight KPHI according to the polarity of the torque command value STQC.

  By such multiplication of weights in the weight function calculation unit 73, the first dq vector calculation unit 71 and the second dq vector calculation unit 72 can always calculate, and the output can be selected and used for each π.

  The following changes may be made without departing from the spirit of the invention.

  * In the above description, the linear motor has been described. However, the technology of the present invention may be applied to a rotary synchronous motor. Further, structurally, the mover (rotor) and the stator may be interchanged.

  1 q-axis current command calculation unit, 2 2/3 phase conversion unit, 3 voltage command calculation unit, 4 power converter, 5 speed calculation unit, 6 d-axis magnetic flux calculation unit, 7 phase calculation unit, 8 polarity determination unit, 9 Electric motor, 10 mover position detecting means, 71 1st dq vector computing unit, 72 2nd dq vector computing unit, 73 weight function computing unit, 74 polarity determiner, STQC torque command, SIQC q-axis current command, SPD mover position detection value .

Claims (2)

A stator having a salient pole structure so that the magnetic resistance periodically has a height difference with respect to the moving direction of the mover;
Made of soft magnetic material, permanent magnets of different polarities are arranged adjacent to each other in the traveling direction so as to have a magnetic polarity that periodically changes in the traveling direction of the movable member, on the tooth surface of the movable member facing the stator. With a mover,
The movable element has a winding wound thereon, and is a control device for a synchronous motor in which a magnetic property of a tooth surface of the movable element is changed by applying a current to the winding,
A control apparatus for a synchronous motor, wherein a polarity of a permanent magnet magnetic flux component on a surface of a mover used for calculation of a voltage command is changed according to a polarity of a torque command corresponding to a traveling direction of the mover. In the synchronous motor control device according to claim 1,
The magnetic flux of the permanent magnet of the mover is dq-axis vector-calculated independently for the second pole pair portion that differs by π rad in electrical angle from the first pole pair, and the relative position between the stator salient pole and the mover. A control apparatus for a synchronous motor, comprising a weight calculation unit that performs weighting calculation of the first and second magnetic fluxes according to the frequency.

Images