JP2012154887A - Clutter eliminator, radar device, clutter elimination method and clutter elimination program - Google Patents

Abstract

PROBLEM TO BE SOLVED: To sufficiently suppress clutter even when relative speed difference between a target and a clutter source is small.SOLUTION: A Doppler shift frequency estimation part 331a estimates a Doppler shift frequency of a reflector based on complex received signals I, Q from the reflector and makes the estimated value a Doppler estimated value ω. A pseudo spectrum generating part 331b generates a pseudo spectrum U by giving signal strength corresponding to the Doppler estimated value ω based on the complex received signals I, Q. A CFAR processing 332 performs suppression processing of clutter using the pseudo spectrum U.

Description

  The present invention relates to a clutter remover for removing clutter in a radar, a radar apparatus including the clutter remover, a clutter removal method for removing clutter in a radar, and a program for removing clutter.

  In a radar apparatus, generally, a target (such as an aircraft or a ship) is detected by capturing a reflected wave of emitted radio waves, and the detected target is displayed on a display. Since there may be a reflector that generates clutter (unnecessary reflected waves) such as waves and rain around the target to be displayed on the display, Patent Document 1 (Patent No. 3006815) is used to suppress clutter. No. 1) and Non-Patent Document 1 (supervised by Takashi Yoshida, revised radar technology, Institute of Electronics, Information and Communication Engineers, 1996, p. 217) are used. In the pulse Doppler processing, the target signal and the clutter are separated based on the difference between the Doppler shift frequencies of the two, and the clutter is selectively suppressed.

  However, even when trying to suppress clutter by pulse Doppler processing described in Patent Document 1 or the like when detecting a ship or a buoy, the relative speed of the target with respect to the radar antenna and the relative speed of the clutter source with respect to the radar antenna Since the difference is small, the difference in Doppler shift frequency between the target signal and the clutter is also small, and the clutter may not be sufficiently suppressed.

  An object of the present invention is to sufficiently suppress clutter even when a relative speed difference between a target and a clutter source is small.

  A clutter remover for solving the above-described problem is a Doppler shift frequency estimation unit that estimates a Doppler shift frequency of a reflector based on a complex reception signal from the reflector, and uses the estimated value as a Doppler estimation value; A pseudo spectrum generation unit that generates a pseudo spectrum by giving a signal intensity corresponding to the Doppler estimation value based on the complex received signal, and a clutter that performs processing for suppressing the clutter component included in the complex received signal using the pseudo spectrum. A suppression processing unit.

  According to this clutter remover, a spectrum having a narrower bandwidth than the Doppler spectrum targeted by the conventional pulse Doppler processing can be obtained as a pseudo spectrum. In the clutter suppression processing unit, by suppressing the signal intensity using such a pseudo spectrum, even if the relative speed difference between the target and the clutter source is small, these are separated and the clutter is selectively suppressed. be able to.

  A clutter removal method for solving the above-described problem is a Doppler shift frequency estimation step in which a Doppler shift frequency of a reflector is estimated based on a complex reception signal from the reflector, and the estimated value is a Doppler estimated value; A pseudo spectrum generating step for generating a pseudo spectrum by giving a signal intensity corresponding to the Doppler estimation value based on the complex received signal, and a process for suppressing a clutter component included in the complex received signal using the pseudo spectrum. A suppression processing step.

  According to this clutter removal method, a spectrum having a narrower bandwidth than the Doppler spectrum targeted by the conventional pulse Doppler processing can be obtained as a pseudo spectrum. In the clutter suppression processing step, by suppressing the signal strength of the received data based on such a pseudo spectrum, even if the relative speed difference between the target and the clutter source is small, the clutter is selectively separated. Can be suppressed.

  The clutter removal program for solving the above problem is a function of estimating the Doppler shift frequency of the reflector based on the complex reception signal from the reflector, and using the estimated value as the Doppler estimated value, and the Doppler estimated value. The computer has the function of generating a pseudo spectrum by giving the signal strength corresponding to the signal based on the complex received signal and the function of using the pseudo spectrum to suppress the clutter component contained in the complex received signal. Let

  According to the clutter removal program, a spectrum having a narrower bandwidth than the Doppler spectrum targeted by the conventional pulse Doppler processing is obtained as the pseudo spectrum. By suppressing the signal intensity of the received data based on such a pseudo spectrum, even when the relative speed difference between the target and the clutter source is small, these can be separated and the clutter can be selectively suppressed.

  According to the present invention, even when the relative speed difference between the target and the clutter source is small, the clutter can be selectively suppressed without damaging the target signal by using the pseudo spectrum having a narrow bandwidth. .

1 is a block diagram showing a schematic configuration of a radar apparatus according to a first embodiment of the present invention. The figure for demonstrating the arrangement | sequence structure of a complex received signal. The block diagram for demonstrating schematic structure of the clutter removal device which concerns on 1st Embodiment. The conceptual diagram about the amplitude of the complex received signal which concerns on a target. The conceptual diagram about the phase of the complex received signal which concerns on a target. The schematic diagram of the Doppler spectrum obtained by discrete Fourier transform. The figure which shows notionally the Doppler spectrum obtained from one reflector in a Doppler spectrum production | generation part. The figure which shows notionally the pseudo spectrum obtained from one reflector in a pseudo spectrum production | generation part. The conceptual diagram for demonstrating the effect of the CFAR process using a pseudo spectrum. FIG. 5 is a block diagram showing another configuration of the clutter remover according to the first embodiment. The flowchart for demonstrating the method of a clutter removal. The block diagram for demonstrating schematic structure of the signal processing part which concerns on 2nd Embodiment. The block diagram for demonstrating schematic structure of the clutter removal device which concerns on 3rd Embodiment. The block diagram for demonstrating schematic structure of the clutter removal device which concerns on 4th Embodiment.

<First Embodiment>
Hereinafter, a radar apparatus according to a first embodiment of the present invention will be described with reference to the drawings. FIG. 1 is a block diagram showing a schematic configuration of this marine radar apparatus. A radar apparatus 100 illustrated in FIG. 1 is a ship radar apparatus that is provided in, for example, a ship and detects a target such as another ship on the sea, a buoy, or land. As shown in FIG. 1, the radar apparatus 100 includes an antenna 1, a transmission / reception unit 2, a signal processing unit 3, and a display device 4. Hereinafter, each element constituting the radar apparatus 100 will be described in detail. Here, a marine radar apparatus will be described as an example of a radar apparatus. However, the present invention can be applied to a radar apparatus that can obtain received signals having the same phase by continuous transmission and reception, for example, weather radar, harbor monitoring, and the like. The present invention can also be applied to radar devices for other uses such as radar. Such a radar apparatus includes not only a solid-state radar apparatus using a semiconductor amplifier as a transmitter but also a magnetron radar apparatus.

[Configuration of antenna 1]
In the radar apparatus 100, the antenna 1 transmits a beam of a pulsed radio wave (radar transmission signal) having a sharp directivity and receives a reflected wave from a target around it. The beam width is set to 2 degrees, for example. The antenna 1 repeats the above transmission and reception while rotating in a horizontal plane. The number of rotations is, for example, 24 rpm. A unit of processing performed while the antenna 1 rotates once is called one scan. In addition, transmission and reception operations in a period from when a radar transmission signal is transmitted to immediately before the next radar transmission signal is transmitted are called sweeps. The time for one sweep, that is, the transmission cycle is, for example, 1 ms. The number of received data per sweep is called the number of sample points.

  The antenna 1 receives a radar reception signal including a reflected wave (target signal) from a target by concentrating and emitting radar transmission signals in a certain direction. In addition to the target signal component, the radar reception signal may include components such as radio wave interference waves and receiver noise from clutter and other radar devices.

  The distance from the antenna 1 to the target is obtained from the time difference between the reception time of the radar reception signal including the target signal and the transmission time of the radar transmission signal corresponding to the radar reception signal. Further, the direction of the target is obtained from the direction of the antenna 1 when transmitting the corresponding radar transmission signal.

[Configuration of the transceiver 2]
The transmission / reception unit 2 generates a radar transmission signal and sends it to the antenna 1. Further, the transmission / reception unit 2 takes in the radar reception signal from the antenna 1 and converts the frequency of the radar reception signal. Therefore, in this embodiment, the transmission / reception unit 2 includes a signal generator 21, a frequency converter 22 (first frequency converter), a local oscillator 23, a transmission / reception switch 24, and a frequency converter 25 (second frequency converter). Is provided.

  The signal generator 21 generates a radar transmission signal having an intermediate frequency at the same time interval or at different time intervals, and outputs it to the frequency converter 22. In the description of this embodiment, generating the above-described intermediate frequency radar transmission signal at the same time interval means that the transmission cycle of the radar transmission signal is constant. Also, generating the above-described intermediate frequency radar transmission signal at different time intervals means that the transmission cycle of the radar transmission signal changes.

  In this embodiment, the radar transmission signal generated by the signal generator 21 is, for example, a frequency modulation signal known as a chirp signal, but the signal generator 21 generates a phase modulation signal or an unmodulated pulse. In addition, the radar apparatus 100 can have the same configuration. Note that the bandwidth or pulse width of the pulse generated by the signal generator 21 can be changed according to the display distance of the radar image set in the display device 4.

  The frequency converter 22 mixes the output signal of the signal generator 21 with the local signal output from the local oscillator 23, converts the output signal of the signal generator 21 to a frequency, and outputs it to the transmission / reception switch 24. The frequency band of the output signal of the frequency converter 22 is, for example, a 3 GHz band or a 9 GHz band.

  The duplexer 24 is configured to be connectable to the antenna 1. The transmission / reception switch 24 switches signals between the antenna 1 and the transmission / reception unit 2. That is, the transmission / reception switch 24 prevents the radar transmission signal from entering the reception circuit (ie, the frequency converter 25) during transmission, and prevents the radar reception signal from entering the transmission circuit (ie, the frequency converter 22) during reception. Like that. As the transmission / reception switching device 24, for example, an electronic component such as a circulator is used.

  The frequency converter 25 takes in a radar reception signal output from the antenna 1 via the transmission / reception switch 24. Then, the frequency converter 25 mixes the radar reception signal with the local signal output from the local oscillator 23, converts the output signal of the transmission / reception switch 24 into an intermediate frequency, and outputs it to the signal processing unit 3 at the subsequent stage.

  In the transmission / reception unit 2 in FIG. 1, illustration of an amplifier and a filter is omitted.

[Configuration of Signal Processing Unit 3]
The signal processing unit 3 converts the radar reception signal into a digital signal and performs signal processing. Therefore, in this embodiment, the signal processing unit 3 includes an A / D (Analog to Digital) converter 31, a quadrature detector 32, and a clutter remover 33. The signal processing unit 3 can be realized by a digital circuit such as an ASIC (Application Specific Integrated Circuit).

  The A / D converter 31 converts an analog intermediate frequency signal output from the frequency converter 25 (transmission / reception unit 2) into a digital signal.

  The quadrature detector 32 performs quadrature detection of a digital intermediate frequency signal (hereinafter referred to as a radar reception signal S) output from the A / D converter 31.

Specifically, in this embodiment, the quadrature detector 32 determines an I (In-Phase) signal and a Q (with a phase difference of π / 2 from the I (In-Phase) signal from the radar reception signal S output from the A / D converter 31. Quadrature) signal. Here, the I signal and the Q signal (hereinafter abbreviated as “I” and “Q” as appropriate) are a real part and an imaginary part of the complex envelope signal Z of the radar reception signal S, respectively. Hereinafter, the complex envelope signal Z is simply referred to as a complex reception signal Z. The amplitude of the complex received signal Z is represented by (I ² + Q ² ) ^1/2 and the phase of the complex received signal Z is represented by tan ⁻¹ (Q / I).

  The clutter remover 33 removes receiver noise and clutter from the output signal (I, Q) of the quadrature detector 32 and outputs received data from which the clutter has been removed to the display device 4. The reception data output from the clutter remover 33 is the amplitude value (band synthesis reception data W) of the radar reception signal S at each position specified by the azimuth and distance. The removal processing of the clutter and the like and generation of the band synthesis reception data W will be described in detail later.

[Configuration of Display Device 4]
The display device 4 includes devices such as a CPU, a memory, and an input device (not shown). In this display device 4, the received data (band synthesis received data W) obtained in each sweep is stored in an image display memory, and the stored data is read out from this memory in a predetermined order, and is displayed as an LCD (Liquid Display on a Crystal Display).

  Usually, the radar image is displayed in a bird's eye view centering on the position of the radar device (antenna). The origin of display corresponds to the position of the radar device. The operator of the radar apparatus 100 can recognize the azimuth and distance of the target from the position where the amplitude of the reflected wave (target signal) from the target is displayed on the radar image.

  As described above, the radar reception signal obtained by one scan has information on the azimuth and the distance. The radar reception signal obtained in the k-th sweep is (k / K) × 360 (deg) (0 ≦≦ K) where the number of sweeps per scan is K and the first sweep is taken as a reference (0 deg). It is obtained from a reflector in the direction of k ≦ K−1). Data obtained from the same direction is associated with the same direction number. The reception data obtained in the k-th sweep is given an azimuth number k.

  The received data obtained by the n-th sampling of each sweep is (n / N) × L, where N is the number of sample points per sweep and L is the range (maximum display distance with the antenna 1 as the origin). This is data obtained from a reflector located at a distance of (0 ≦ n ≦ N−1). The reception data obtained by the nth sampling is given a distance number n.

Hereinafter, the complex received signal Z is treated as a two-dimensional array with respect to the bearing number k and the distance number n, and this array is represented by Z [k, n]. The configuration of the array is shown in FIG. The horizontal direction corresponds to the direction, and the vertical direction corresponds to the distance. The element indicated by hatching in FIG. 2 is Z [k ₀ , n ₀ ]. In the following, for signals other than the complex reception signal Z, when an individual array element is designated, a number indicating the element is written together as Z [k, n], and the type of the signal is indicated. In the case of designation, the number indicating the element is omitted and indicated as Z.

[Configuration of Clutter Remover 33]
Next, the configuration of the clutter remover 33 of the signal processing unit 3 will be described with reference to FIG. FIG. 3 is a block diagram for explaining the configuration of the clutter remover 33. As shown in FIG. 3, the clutter remover 33 includes a Doppler spectrum generation unit 331, a CFAR processing unit 332, and a band synthesis unit 333.

  Here, in order to clarify the difference between the conventional pulse Doppler processing and the clutter removal method according to the present invention, the Doppler spectrum targeted by the conventional pulse Doppler processing will be described. The Doppler spectrum is defined as a set of values indicating the intensity of each component when the received signal is decomposed into components having a plurality of different Doppler shift frequencies.

  4 and 5 conceptually show graphs in which the amplitude and phase of the complex received signal Z from the target approaching the antenna are plotted while the distance number n is fixed and the azimuth number k is changed. FIG. Since the antenna 1 is rotating, the amplitude of the complex reception signal Z is large when the center of the beam is directed toward the target as shown in FIG. 4, and is small when the end of the beam is directed toward the target. .

The amount of change in phase per sweep is proportional to the relative speed and therefore proportional to the Doppler shift. When the distance number corresponding to the position of a target is n ₀ and the azimuth number is k ₀ , a plurality of complex receptions arranged in the azimuth direction (lateral direction) with [k ₀ , n ₀ ] as the center in the array shown in FIG. A Doppler spectrum can be obtained by extracting the signal Z and subjecting the data string to discrete Fourier transform. Since the radar repeats transmission and reception while rotating the antenna, the number of sweeps that can continuously obtain received signals from the same target is limited by the antenna rotation speed, beam width, and transmission cycle. Therefore, the Doppler spectrum obtained by the discrete Fourier transform has a spread corresponding to the number of sweeps even if the relative velocity of the target does not change. In addition, in the Doppler spectrum of rain and waves, the unevenness of the movement of raindrops and waves in the region irradiated with the beam also contributes to the spread.

  FIG. 6 schematically shows a graph in which the Doppler spectra obtained by the discrete Fourier transform are plotted at the same coordinates on the assumption that a white wave with a relative speed of 3 knots exists around a target with a relative speed of 8 knots. FIG. The abscissa represents the Doppler shift frequency converted into speed by using the relationship of relative speed v = wavelength λ of transmission pulse × Doppler shift frequency fd / 2. In FIG. 6, a curve DT1 shows the Doppler spectrum of the target, and a curve DC1 shows the Doppler spectrum of the white wave. For the reasons described above, the spectra spread and overlap each other.

[Configuration of Doppler Spectrum Generation Unit 331]
The Doppler spectrum generation unit 331 illustrated in FIG. 3 includes a Doppler shift frequency estimation unit 331a and a pseudo spectrum generation unit 331b. 3 generates a pseudo spectrum U instead of the conventional Doppler spectrum as shown in FIG. 6 and outputs it.

  First, an outline of processing for generating the pseudo spectrum U performed by the Doppler shift frequency estimation unit 331a and the pseudo spectrum generation unit 331b will be described. The Doppler shift frequency estimation unit 331a estimates one Doppler shift frequency of a reflector located at a specific distance and a specific orientation. Hereinafter, the Doppler shift frequency is defined as “amount of phase change per sweep of the complex reception signal Z”. This estimated Doppler shift frequency is referred to as a Doppler estimated value and is represented using the symbol ω. According to the above definition, ω takes a value from 0 to 2π (0 ≦ ω <2π). The pseudo spectrum generation unit 331b determines the signal strength of the Doppler estimated value ω based on the complex received signal, and the signal strength at a frequency other than the Doppler estimated value ω or a frequency other than the Doppler estimated value ω and its vicinity has a very small value. A pseudo spectrum (pseudo spectrum) is generated.

  FIG. 7 shows a conventional Doppler spectrum obtained from a complex received signal of one reflector, and FIG. 8 shows a pseudo spectrum obtained from one reflector. The pseudo spectrum illustrated in FIG. 8 is a line spectrum in which the signal intensity of the frequencies other than the Doppler estimation value ω and the vicinity thereof is zero. As can be seen from a comparison between FIG. 7 and FIG. 8, the pseudo spectrum has a narrower spectrum width than the conventional Doppler spectrum. The pseudo spectrum PT1 of the Doppler spectrum (curve DT1) of the target shown in FIG. 6 and the pseudo spectrum PC1 of the white wave Doppler spectrum (curve DC1) are generated, and the pseudo spectra of both are the same as in FIG. When plotted at the coordinates, the result is as shown in FIG. 9, and the spectra do not overlap. For this reason, in the CFAR processing described later, even if clutter is suppressed, the target signal is not suppressed, so that a sufficient clutter removal effect can be obtained.

[Calculation of Doppler estimated value ω]
Next, a method of calculating the Doppler estimated value ω in the Doppler shift frequency estimation unit 331a will be described. In the Doppler shift frequency estimation unit 331a, a plurality of (M) Doppler estimation values ω at a specific distance n = n ₀ and a specific direction k = k ₀ are at the same distance n ₀ with the direction k ₀ as the center. It is calculated using the complex received signal Z. These M complex received signals Z are called orientation data series. For example, when M = 5 is set, the Doppler estimation value ω of the complex reception signal Z [k ₀ , n ₀ ] is, for example, Z [k ₀ −2, n ₀ ], Z [k ₀ −1, n _0], Z _[k 0, _{n 0],} Z _[k 0 + 1, _{n 0],} is estimated based on the orientation data series of five elements of Z _[k 0 + 2, _{n 0].}

Here, a method of calculating the Doppler estimated value ω by pulse pair processing will be described. For easy understanding, the five complex received signals Z [k ₀ -2, n ₀ ], Z [k ₀ -1, n ₀ ], Z [k ₀ , n ₀ ], Z [k ₀ +1, n ₀ ], Z [k ₀ +2, n ₀ ] are simply written as z [0], z [1], z [2], z [3], z [4] To do. These conjugate complex numbers are represented by *, for example, the complex number of z [0] is represented as z ^* [0].

The sum of products of each element of the azimuth data series and the conjugate complex number of the element immediately before that element, that is, z [1] · z ^* [0] + z [2] · z ^* [1] + z [3 ] ^* Z ^* [2] + z [4] ^* z ^* [3] is R, the Doppler estimated value ω is given by arg [R]. Here, arg [R] represents the angle of deviation of R. This R is called a complex correlation coefficient.

  When the number of sample points per sweep is N, the Doppler shift frequency estimation unit 331a calculates N Doppler estimated values ω while sequentially changing the distance number n from 0 to N-1. In the above example, the sum of the products of each element of the azimuth data series and the conjugate complex number of the element immediately preceding that element is the complex correlation coefficient R. Not limited to those. For example, it may be the previous one, and can generally be estimated using the previous (M-1).

[Generation of pseudo spectrum U]
The pseudo spectrum generation unit 331b generates the pseudo spectrum U as illustrated in FIG. 8 based on the complex reception signal Z [k ₀ , n ₀ ] and the Doppler estimation value ω. The pseudo spectrum generation unit 331b defines J frequency sections defined by dividing the frequency range from 0 to 2π into J equal parts. When the Doppler estimation value ω is input from the Doppler shift frequency estimation unit 331a, the pseudo spectrum generation unit 331b calculates which section of the J frequency sections the Doppler estimation value ω belongs to. Here, for example, it is assumed that the Doppler estimation value ω belongs to the j _0th frequency section.

Next, the pseudo spectrum generation unit 331b sets the absolute value of the complex reception signal Z [k ₀ , n ₀ ] to the signal intensity of the j _0th frequency section, sets the signal intensity of other frequency sections to 0, and sets the pseudo spectrum U Is generated. By doing so, a pseudo spectrum U as shown in FIG. 8 is obtained.

The pseudo spectrum generation unit 331b generates N pseudo spectra while sequentially changing the distance number n from 0 to N-1 where N is the number of sample points per sweep. The N pseudo spectra are represented by a two-dimensional array U [j, n]. For example, the pseudo spectrum of the distance number n ₀ is represented by U [j, n ₀ ]. However, 0 ≦ j ≦ J−1.

[CFAR processing unit 332]
Distance and pseudo spectrum U [j, n] arranged in two-dimensional frequency for the signal strength, if a fixed frequency interval to j _0, distance number n of the data sequence {U [j _0, n] | 0 ≦ n ≦ N−1} is defined. This series has N elements. In the following description, this series, that is, a data series {U [j, n] | 0 ≦ 0 in the distance (n) direction obtained by fixing the frequency section (j) of the pseudo-spectrum array U [j, n]. n ≦ N−1} is referred to as band division reception data.

  The CFAR processing unit 332 performs a CFAR (Constant False Alarm Rate) process on the band division reception data corresponding to each frequency section. In the CFAR processing, the intensity of clutter or receiver noise is estimated based on the intensity data series in the distance direction including the data of interest, and the estimated value is subtracted from the data of interest. By this processing, even if the clutter strength fluctuates temporally and spatially, the clutter strength after subtraction takes a substantially constant value. Therefore, if the intensity of the target echo is sufficiently larger than the intensity of the clutter, it is possible to easily remove only the clutter by comparing the intensity data after subtraction with an appropriate constant threshold value. For this CFAR processing, either a parametric CFAR that estimates the intensity of clutter by assuming a probability density function of clutter or noise amplitude, or a nonparametric CFAR that does not assume a specific probability density function may be used. For example, parametric CFAR cell-average log CFAR processing is performed by the CFAR processing unit 332 to suppress noise and clutter.

[Bandwidth synthesis unit 333]
The band synthesizer 333 receives from the CFAR processing unit 332 a band division reception data sequence that has been subjected to CFAR processing for each frequency section (j). This data series is represented by {V [j, n] | 0 ≦ n ≦ N−1}. The band synthesizing unit 333 selects the maximum value among the J band division reception data {V [j, n] | 0 ≦ j ≦ J−1} regarding each distance number n. This maximum value is represented by band synthesis reception data and W. That is, the band synthesis unit 333 outputs a band synthesis reception data sequence {W [n] | 0 ≦ n ≦ N−1} in each sweep. Further, the clutter remover 33 repeats all the above processes for each sweep. Therefore, the clutter remover 33 performs band synthesis reception data {W [k, n] | 0 ≦ n ≦ N−1 for all azimuth numbers (k) and all distance numbers (n) during one scan. 0 ≦ K−1} is generated.

  The display device 4 converts the value of the band synthesis reception data W generated by the band synthesizing unit 333 into a color or luminance and displays an image.

<Modification 1-1>
In the radar apparatus 100 of the above embodiment, the case where the Doppler shift frequency estimation unit 331a calculates the Doppler estimated value by the pulse pair processing has been described, but the Doppler estimated value ω can also be obtained by other methods. For example, the Doppler estimation value can also be obtained using discrete Fourier transform. First, discrete Fourier transform is performed to generate a Doppler spectrum as shown in FIG. Next, the frequency at which the signal intensity is maximum in the generated Doppler spectrum is set as the Doppler estimated value ω.

<Modification 1-2>
In the Doppler spectrum generation unit 331 of the above embodiment, the case where the pseudo spectrum is generated by the Doppler shift frequency estimation unit 331a and the pseudo spectrum generation unit 331b has been described. However, the Doppler spectrum generation unit can also be configured by other devices. .

  FIG. 10 is a block diagram showing a configuration of a clutter remover 33A having a Doppler spectrum generation unit 331A of another configuration. The Doppler spectrum generation unit 331A includes J filters f1, f2, f3... FJ having different center frequencies, a maximum value selector 331c, and a pseudo spectrum generation unit 331d. For example, the center frequency of each of the filters f1, f2, f3... FJ is set to the center frequency of each of the J frequency intervals defined by dividing the frequency range from 0 to 2π into J equal parts. The maximum value selector 331c compares the absolute values of the outputs from the plurality of filters f1, f2, f3... FJ, and selects the one indicating the maximum value. The center frequency of the selected filter is the Doppler estimation value ω of the above embodiment. The pseudo spectrum generation unit 331d generates the pseudo spectrum U using the Doppler estimation value ω and the complex reception signal Z as described in the above embodiment. Instead of using the complex received signal Z, the output from the selected filter may be used.

  Accordingly, as in the above embodiment, when the output of the Doppler spectrum generation unit 331A is processed by the CFAR processing unit 332, and the band synthesis unit 333 performs band synthesis, the relative speed difference between the target and the clutter source is small. In addition, it is possible to sufficiently suppress the clutter by distinguishing from the signal component of the target by using the Doppler estimated value ω.

<Modification 1-3>
In the radar apparatus of the above embodiment, an example in which the signal processing unit 3 (clutter removal unit 33) is configured by hardware has been described. However, the function of the signal processing unit 3 (clutter removal unit 33) is realized by software. You may do it. In this case, a control unit such as a CPU that reads a program from a recording medium such as a ROM performs functions (for example, a series of processes shown in FIG. 11) after the A / D converter 31 of the signal processing unit 3 of the embodiment. Etc.) can be realized.

  Explaining along FIG. 11, first, quadrature detection of the radar reception signal S is performed in step S10. Next, the Doppler estimation value ω of the reflector is extracted based on the complex reception signal Z obtained by the quadrature detection (step S11). The Doppler estimation value ω is extracted for the positions specified by all the azimuth numbers k and all the distance numbers n, as performed by the Doppler shift frequency estimation unit 331a of the first embodiment.

  Next, in step S12, a Doppler spectrum is generated based on the Doppler estimation value ω. The generation of the Doppler spectrum is performed by using the absolute value of the complex reception signal Z as the signal intensity in the frequency section to which the Doppler estimation value ω belongs, and the signal intensity in other frequency sections is set to 0, as the pseudo spectrum generation unit 331b performs. By doing so, the pseudo spectrum U is generated as a Doppler spectrum.

  Next, in step S13, a data series {U [j, n] | 0 ≦ n ≦ N in the distance (n) direction obtained by fixing the frequency section (j) of the pseudo-spectrum array U [j, n]. −1}, that is, CFAR processing is performed on the band division reception data, and clutter suppression processing is performed. Band synthesis reception data W is generated based on the band division reception data subjected to the CFAR process in step S13 (step S14). In order to perform display on the display device 4, the band synthesized reception data is converted into video display data (step S15).

  FIG. 11 shows a case where a pseudo spectrum is generated as a Doppler spectrum. However, as shown in FIG. 10, a configuration using a plurality of filters f1 to fJ and a maximum value selector 331c is also used. It can be configured by software.

<Modification 1-4>
In the pseudo spectrum generation, the pseudo spectrum generation units 331b and 331d of the above embodiment set all signal intensities in the frequency sections other than the frequency section (referred to as “main section”) corresponding to the Doppler estimation value ω to zero. However, the pseudo spectrum generation unit may make the signal intensity in the frequency section near the main section take a value corresponding to the signal intensity of the main section in generating the pseudo spectrum. For example, in the frequency section in the vicinity of the main section, the signal intensity is α times (0 <α ≦ 1) the signal intensity of the main section, and the value of α is increased in the section closer to the main section. Good. In this way, even if the main section related to the clutter varies slightly according to the distance (n), the pseudo spectrum has a “non-zero value” in the distance direction in the main section and its neighboring frequency sections. Therefore, the clutter is more reliably suppressed by the CFAR processing unit.

<Features>
(1) The Doppler spectrum generation units 331 and 331A respectively estimate the Doppler shift frequencies of reflectors such as targets and clutter sources from the complex reception signal Z for a plurality of positions specified by the distance number n and the azimuth number k. Thus, the estimated value is set as a Doppler estimated value ω.

  In the Doppler spectrum generation units 331 and 331A, the Doppler shift frequency estimation unit 331a and the maximum value selector 331c output the Doppler estimation value ω. Then, the pseudo spectrum generation units 331b and 331d generate the pseudo spectrum U using the output Doppler estimation value ω. This pseudo spectrum U is a spectrum having a signal intensity of 0 other than the Doppler estimated value ω or other than the Doppler estimated value ω and its vicinity. When CFAR processing (clutter suppression processing) is performed with such a narrow pseudo spectrum in the CFAR processing unit 332 (clutter suppression processing unit), even if the relative speed difference between the target and the clutter source is small, Since the target signal and the clutter belong to different frequency sections, it is possible to suppress only the clutter without impairing the target signal.

  (2) The pseudo-spectrum generation unit 331b determines whether the Doppler estimated value ω belongs to one of the frequency sections defined by dividing the frequency range from 0 to 2π into J equal parts, and other than the frequency section to which the Doppler estimated value ω belongs. By making the signal strength of 0 a zero, it is possible to easily generate a pseudo spectrum U having a signal strength of 0 outside the vicinity of the Doppler estimation value ω. When such a pseudo spectrum U is used, the target and the clutter source can be separated with a resolution corresponding to the width of the frequency section.

Second Embodiment
Next, a radar apparatus according to a second embodiment of the present invention will be described with reference to FIG. The configuration of the radar apparatus according to the second embodiment is the same as the radar apparatus 100 of the first embodiment, except that the configuration of the signal processing unit 3B is different from the configuration of the signal processing unit 3 of the first embodiment. Therefore, the description of the radar apparatus according to the second embodiment is limited to the configuration of the signal processing unit 3B.

  As shown in FIG. 12, the signal processing unit 3 </ b> B includes an A / D converter 31, a quadrature detector 32, and a clutter remover 33, similarly to the radar apparatus 100. The processes performed in the A / D converter 31, the quadrature detector 32, and the clutter remover 33 are as described in the first embodiment performed for the signal processing unit 3.

  In addition to them, the signal processing unit 3B includes a pulse Doppler processor 33B and a signal synthesis unit 34. The pulse Doppler processor 33 </ b> B is provided in parallel to the clutter remover 33. The pulse Doppler processor 33B performs pulse Doppler processing on the complex reception signal Z. The pulse Doppler processing performed by the pulse Doppler processor 33B is the same pulse Doppler processing as described in Patent Document 1 and Non-Patent Document 1. Therefore, the process performed by the pulse Doppler processor 33B is different from the process performed by the clutter remover 33.

  The signal synthesizer 34 is configured to output an average value of the output of the clutter remover 33 and the output of the pulse Doppler processor 33B. Alternatively, the signal synthesizer 34 is configured to compare the output of the clutter remover 33 and the output of the pulse Doppler processor 33B and output the larger one.

<Modification 2-1>
In the description of the second embodiment, the case where the pulse Doppler processor 33B and the signal synthesis unit 34 are added to the signal processing unit 3 of the first embodiment shown in FIG. 1 has been described. However, the clutter remover shown in FIG. A pulse Doppler processor 33B and a signal synthesis unit 34 can be added to the signal processing unit of the radar apparatus having 33A.

<Modification 2-2>
In the radar apparatus according to the second embodiment, the signal processing unit 3B (clutter removal unit 33, pulse Doppler processing unit 33B, and signal synthesis unit 34) is shown as an example of hardware, but the signal processing unit 3B ( The functions of the clutter remover 33, the pulse Doppler processor 33B, and the signal synthesis unit 34) may be realized by software.

<Features>
In the radar apparatus 100 according to the first embodiment, when each complex reception signal Z includes both a target signal component and a clutter component, and the clutter component is larger than the target signal component, the clutter and the target signal are detected. May both be suppressed. Even in such a case, if the relative speed difference between the target and the clutter source is larger than the bandwidth of the Doppler spectrum, the pulse Doppler processor 33B can suppress the clutter without suppressing the target signal. .

  Since the signal synthesizer 34 outputs the average or larger one of the output of the clutter remover 33 and the output of the pulse Doppler processor 33B, even if the target signal is suppressed by the clutter remover 33, the pulse Doppler processor 33B. Complemented by the output of. As a result, even at a position where each complex reception signal Z includes both the target signal component and the clutter component, the clutter can be suppressed to the same level or more as in the conventional pulse Doppler processing.

<Third Embodiment>
Next, a radar apparatus according to a third embodiment of the present invention will be described with reference to FIG. The configuration of the radar apparatus according to the third embodiment is the same as that of the first embodiment except that the configuration of the Doppler spectrum generation unit 331C of the signal processing unit 3C is different from the Doppler spectrum generation unit 331 of the signal processing unit 3 of the first embodiment. It becomes the same as the radar apparatus 100 of the form. Therefore, the description of the radar apparatus according to the third embodiment is limited to the configuration of the Doppler spectrum generation unit 331C.

  As illustrated in FIG. 13, the Doppler spectrum generation unit 331C includes a Doppler shift frequency estimation unit 331a and a pseudo spectrum generation unit 331b, similarly to the Doppler spectrum generation unit 331 illustrated in FIG. Since the processes performed by the Doppler shift frequency estimation unit 331a and the pseudo spectrum generation unit 331b are the same as those described for the Doppler spectrum generation unit 331, description thereof is omitted.

  In the Doppler spectrum generation unit 331C, a filter unit 331e that is not provided in the Doppler spectrum generation unit 331 is added between the quadrature detector 32 and the pseudo spectrum generation unit 331b. The filter unit 331e performs a filtering process on the complex reception signal Z output from the quadrature detector 32 according to the Doppler estimation value ω provided from the Doppler shift frequency estimation unit 331a.

  The filter unit 331e performs a filtering process that is easier to pass through the direction data series as it is closer to the Doppler estimation value ω.

A specific example of the filter processing of the filter unit 331e will be described as follows. The filter coefficient h (m) of the filter unit 331e is defined as e ^−jωm (m = 0, 1, 2,..., M−1). Here, j is an imaginary unit, and ω is a Doppler estimated value. Then, the filter unit 331e performs a product-sum operation on the filter coefficient h and the element of the azimuth data series. That is, the complex reception signal Zf after filtering is given by the sum of products of the filter coefficient h and the elements of the azimuth data series. For example, the elements of the azimuth data sequence of the complex received signal Z [k ₀ , n ₀ ] are Z [k ₀ −2, n ₀ ], Z [k ₀ −1, n ₀ ], Z [k ₀ , n ₀ ]. , Z [k ₀ +1, n ₀ ], Z [k ₀ +2, n ₀ ], the filtered complex received signal Zf [k ₀ , n ₀ ] is Z [k ₀ −2, n _0]. ] + Z [k ₀ -1, n ₀ ] · e ^−jω + Z [k ₀ , n ₀ ] · e ^−j2ω + Z [k ₀ + 1, n ₀ ] · e ^−j3ω + Z [k ₀ + 2, n ₀ ]. e ^−j4ω .

<Modification 3-1>
In the description of the third embodiment, the case where the Doppler spectrum generation unit 331 of the first embodiment is replaced with the Doppler spectrum generation unit 331C has been described. However, the clutter remover 33 of the second embodiment is the same as that of the third embodiment. It can also be replaced with a clutter remover 33C.

<Modification 3-2>
In the radar apparatus of the third embodiment, an example in which the signal processing unit 3C (clutter removal unit 33C) is configured by hardware has been described. However, the function of the signal processing unit 3C (clutter removal unit 33C) is implemented by software. It may be realized.

<Features>
As can be seen from the equation for calculating the complex received signal Zf [k ₀ , n ₀ ] in the filter unit 331e, the phase change amount between adjacent elements of the azimuth data series is the Doppler estimated value ω for the target signal. Therefore, the azimuth data is added in the same phase by filtering. That is, the azimuth data series of the target signal strengthens each other. On the other hand, since receiver noise fluctuates randomly, it is not added in the same phase. Therefore, when the filter processing according to the Doppler estimation value ω by the filter unit 331e is performed, the signal-to-noise ratio (S / N) of the target signal with respect to the receiver noise is improved. That is, the receiver noise can be suppressed simultaneously with suppressing the clutter.

<Fourth embodiment>
Next, a radar apparatus according to a fourth embodiment of the invention will be described with reference to FIG. The configuration of the radar apparatus according to the fourth embodiment is the same as that of the first embodiment except that the configuration of the Doppler spectrum generation unit 331D of the signal processing unit 3D is different from the Doppler spectrum generation unit 331 of the signal processing unit 3 of the first embodiment. This is the same as the radar device 100 of the embodiment. Therefore, the description of the radar apparatus according to the fourth embodiment is limited to the configuration of the Doppler spectrum generation unit 331D.

  As illustrated in FIG. 14, the Doppler spectrum generation unit 331D includes a Doppler shift frequency estimation unit 331a, similar to the Doppler spectrum generation unit 331 illustrated in FIG. Since the processing performed by the Doppler shift frequency estimation unit 331a is the same as that described for the Doppler spectrum generation unit 331, description thereof is omitted.

  In the Doppler spectrum generation unit 331D, a bandwidth estimation unit 331f that is not provided in the Doppler spectrum generation unit 331 is added between the quadrature detector 32 and the pseudo spectrum generation unit 331g. The bandwidth estimation unit 331f calculates a bandwidth index σ indicating the spread of the Doppler spectrum for the complex reception signal Z output from the quadrature detector 32.

The bandwidth index σ is defined as σ = 1− | R | / R ₀ using the complex correlation coefficient R, and R ₀ is given by the following equation. That is, R ₀ = | Z [k ₀ -2, n ₀ ] · Z [k ₀ -1, n ₀ ] | + | Z [k ₀ -1, n ₀ ] · Z [k ₀ , n ₀ ] | + | Z [k ₀ , n ₀ ] · Z [k ₀ + 1, n ₀ ] | + | Z [k ₀ + 1, n ₀ ] · Z [k ₀ + 2, n ₀ ] |. Therefore, the bandwidth index σ always takes a value of 0 or more and 1 or less. The bandwidth index σ estimated by the bandwidth estimation unit 331f is output to the pseudo spectrum generation unit 331g.

  The pseudo spectrum generation unit 331g has the same function as the pseudo spectrum generation unit 331b described in the first embodiment. Furthermore, in addition to the function of the pseudo spectrum generation unit 331b, the pseudo spectrum generation unit 331g has a function of setting the signal intensity to 0 over all frequencies of the pseudo spectrum according to the value of the bandwidth index σ. That is, if the bandwidth index σ is larger than a predetermined value set in advance, the pseudo spectrum generation unit 331g sets all elements of the corresponding pseudo spectrum to zero.

<Modification 4-1>
In the description of the fourth embodiment, the case where the Doppler spectrum generation unit 331 of the first embodiment is replaced with the Doppler spectrum generation unit 331D has been described. However, the clutter remover 33A shown as a modification of the first embodiment has been described. A bandwidth estimation unit 331f can also be added. In this case, the maximum value selector 331c has a function of setting all elements of the corresponding Doppler spectrum to 0 if the bandwidth index σ is larger than a predetermined value set in advance.

<Modification 4-2>
In the description of the fourth embodiment, the case where the Doppler spectrum generation unit 331 of the first embodiment is replaced with the Doppler spectrum generation unit 331D has been described. However, the clutter removal unit 33 of the second embodiment is replaced by the fourth embodiment. It can also be replaced with a clutter remover 33D.

<Modification 4-3>
In the description of the fourth embodiment, the case where the Doppler spectrum generation unit 331 of the first embodiment is replaced with the Doppler spectrum generation unit 331D has been described. However, the bandwidth estimation unit 331f is added to the clutter removal unit 33C of the third embodiment. It is also possible to replace the pseudo spectrum generation unit 331b of the third embodiment with the pseudo spectrum generation unit 331g of the fourth embodiment.

<Modification 4-4>
In the radar apparatus of the fourth embodiment, an example in which the signal processing unit 3D (clutter removal unit 33D) is configured by hardware has been described, but the function of the signal processing unit 3D (clutter removal unit 33D) is implemented by software. It may be realized.

<Modification 4-5>
In the fourth embodiment, the bandwidth index σ is used as an index indicating the spread of the Doppler spectrum of the azimuth data series. However, another index may be used as an index indicating the spread of the Doppler spectrum of the azimuth data series. For example, the spread index may be calculated based on the Doppler spectrum obtained by the discrete Fourier transform.

<Features>
When only the receiver noise is included in the azimuth data series, each element of the azimuth data series is independent of each other and takes a random value, so that the correlation coefficient R becomes small. Therefore, the bandwidth index σ takes a relatively large value. . Further, when the azimuth data series includes interference from other radars, only a specific element of the azimuth data series takes a remarkably large value, so that R ₀ takes a large value with respect to the correlation coefficient R. The index σ takes a relatively large value.

  In this way, all elements of the Doppler spectrum can be set to 0 when the azimuth data series includes only receiver noise or when interference from other radars is included, so that receiver noise and interference can be suppressed. And the target can be detected more stably.

DESCRIPTION OF SYMBOLS 1 Antenna 2 Transmission / reception part 3 Signal processing part 4 Display device 33,33A, 33C, 33D Clutter remover 34 Signal composition part 331,331A, 331C, 331D Doppler spectrum generation part 331a Doppler shift frequency estimation part 331b, 331d, 331g pseudo spectrum Generation unit 331c Maximum value selector 331e Filter unit 331f Bandwidth estimation unit 332 CFAR processing unit 333 Band synthesis unit

Japanese Patent No. 3006815 Supervised by Takashi Yoshida, revised radar technology, The Institute of Electronics, Information and Communication Engineers, 1996, p. 217

Claims (14)

Estimating a Doppler shift frequency of the reflector based on a complex reception signal from the reflector, and a Doppler shift frequency estimation unit using the estimated value as a Doppler estimation value;
A pseudo spectrum generation unit that generates a pseudo spectrum by giving a signal intensity corresponding to the Doppler estimation value based on the complex reception signal;
A clutter suppression processing unit that performs suppression processing of clutter included in the complex reception signal using the pseudo spectrum; and
A clutter remover. The Doppler shift frequency estimation unit divides a frequency range from 0 to 2π to define a plurality of frequency sections, extracts a frequency section to which the Doppler estimation value belongs from among the plurality of frequency sections,
The pseudo spectrum generation unit gives a signal strength in a frequency section to which the Doppler estimation value belongs based on the complex reception signal, and sets a signal strength in all frequency sections to which the Doppler estimation value belongs to 0 except for a frequency section to which the Doppler estimation value belongs. Generating the pseudo spectrum by:
The clutter remover according to claim 1. The Doppler shift frequency estimation unit divides a frequency range from 0 to 2π to define a plurality of frequency sections, extracts a frequency section to which the Doppler estimation value belongs from the plurality of frequency sections as a main section,
The pseudo spectrum generation unit gives the signal strength in the main section based on the complex received signal, and the frequency section near the main section is smaller than the signal intensity in the main section and closer to the main section, the main section. Generating the pseudo spectrum by giving a signal strength close to the signal strength of the interval;
The clutter remover according to claim 1. The Doppler shift frequency estimation unit sets a frequency that gives a maximum value of signal strength in a Doppler spectrum obtained from the complex reception signal as a Doppler estimation value.
The clutter remover according to any one of claims 1 to 3. The Doppler shift frequency estimation unit calculates a Doppler estimation value by pulse pair processing using the complex reception signal.
The clutter remover according to claim 4. The Doppler shift frequency estimation unit calculates a Doppler estimation value by a discrete Fourier transform process using the complex reception signal.
The clutter remover according to claim 4. The Doppler shift frequency estimator is
A plurality of filters having different center frequencies,
A maximum value selector that sets the Doppler estimation value as the center frequency of the plurality of filters that output the maximum value;
The clutter remover of claim 1, comprising: A filter unit that performs filtering on the complex reception signal according to the Doppler estimation value is further provided in the previous stage of the pseudo spectrum generation unit,
The clutter remover according to any one of claims 1 to 7. A bandwidth estimation unit for calculating an index indicating the spread of the Doppler spectrum;
The pseudo spectrum generation unit reduces the signal intensity over the entire frequency of the pseudo spectrum when the index is larger than a predetermined value.
The clutter remover according to any one of claims 1 to 8. The clutter suppression processing unit includes a constant false alarm rate processing unit that performs a constant false alarm rate process on a series whose signal intensity is the pseudo spectrum.
The clutter remover according to any one of claims 1 to 9. A band synthesizing unit that synthesizes received data from the output of the clutter suppression processing unit;
A Doppler processing unit that performs processing different from processing performed by the Doppler shift frequency estimation unit, the pseudo spectrum generation unit, the clutter suppression processing unit, and the band synthesis unit, and suppresses clutter components included in the complex reception signal. When,
A signal synthesizing unit that outputs an average value or a larger value of the output of the band synthesizing unit and the output of the Doppler processing unit;
Further comprising
The clutter remover according to any one of claims 1 to 10. A clutter remover according to any one of claims 1 to 11,
A transmission / reception unit that outputs a radar reception signal obtained by radiating a radar transmission signal to the clutter remover;
A radar apparatus comprising: Estimating a Doppler shift frequency of the reflector based on a complex received signal from the reflector, and a Doppler shift frequency estimating step using the estimated value as a Doppler estimated value;
Generating a pseudo spectrum by giving a signal strength corresponding to the Doppler estimate value based on the complex received signal; and
A clutter suppression processing step of performing suppression processing of clutter included in the complex reception signal using the pseudo spectrum; and
A clutter removal method comprising: A function of estimating the Doppler shift frequency of the reflector based on a complex reception signal from the reflector, and setting the estimated value as a Doppler estimated value;
A function of generating a pseudo spectrum by giving a signal intensity corresponding to the Doppler estimation value based on the complex received signal;
A function of performing suppression processing of clutter included in the complex reception signal using the pseudo spectrum;
A program for removing clutter to make a computer realize.

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