JP2011234089A - Nonlinear waveguide-waveguide converter, and communication device using nonlinear waveguide-waveguide converter - Google Patents

Abstract

PROBLEM TO BE SOLVED: To achieve size and cost reduction even when a filter is inserted into a waveguide circuit side, to sufficiently suppress an unwanted signal, and to attenuate even harmonics in a wide frequency band.SOLUTION: A converter comprises: a second waveguide 10B provided with a short-circuit plane S; a probe 14 inserted into a waveguide 10 from an H plane thereof at a tip of a microwave strip line 13 formed on a dielectric substrate 12; and a columnar conductor 16 which is arranged while projecting from a short-circuit plane S to nearby the probe 14, and adapted to form a resonance circuit resonating to a predetermined frequency. The waveguide 10 (10A and 10B) has a width of a rectangle in a magnetic field direction set to a value such that the circumference of a reception frequency band is equal to or less than a cutoff frequency of the waveguide, and the probe 14 is provided with an open stub. Further, a waveguide step impedance converter is provided between the waveguide of the converter and a transmitter-side waveguide.

Description

  The present invention relates to a non-waveguide line-waveguide converter and a communication device, particularly a converter between a non-waveguide line and a waveguide, such as a distributed constant line or a coaxial line, used in a microwave circuit. Thus, the present invention relates to a non-waveguide line-waveguide converter having a filter function for suppressing a desired unnecessary frequency component and a configuration of a communication device.

  Conventionally, in a converter between a non-waveguide line such as a microstrip line and a waveguide, when a filter is inserted in order to suppress a specific frequency before and after the conversion unit, a BRF using a distributed constant circuit (Band rejection filter), LPF (low pass filter), HPF (high pass filter), BPF (band pass filter) are used. In the waveguide portion, a BPF or BRF formed by coupling a plurality of resonators is used. When the suppression band is relatively narrow, a conductor plate attached in a direction orthogonal to the waveguide transmission direction A single resonance circuit that suppresses a predetermined frequency is inserted depending on the shape (Patent Documents 1 and 2 below).

  FIGS. 12A and 12B show the configuration of the non-waveguide line-waveguide converter. In this converter, the inside of the metal waveguide 1 in which a rectangular cavity is formed. Then, the tip of the microstrip line 3 formed on the dielectric substrate 2 is inserted as the probe 4. That is, the probe 4 is projected from the H surface (upper surface) of the waveguide 1 to the inside. Reference numeral 5 denotes a through hole.

  On the other hand, in two-way communication using satellites, there are strict requirements for unwanted signals sent from transmitters due to interference with receivers in communication devices and legal regulations, and products that are also intended for general consumer use Therefore, the demand for reduction in price and power consumption is also increasing. For this reason, it is necessary to suppress these unnecessary signals by a simple method without incurring costs as much as possible, and since the suppression mechanism is also required after the final amplifier, the loss of transmission signals by these circuits can be large. As a result, the power consumption is increased.

Japanese Patent No. 4301722 JP 2007-180655 A Japanese Patent No. 4262192

  By the way, when a filter is inserted on the non-waveguide line side, since a circuit is generally formed on a dielectric substrate, a high Q value filter cannot be realized due to loss of the dielectric. That is, for example, when suppressing the image signal and the reception frequency band noise, it is not possible to adopt when the detuning frequency between the passband and the suppression frequency is small, and even if it is a condition that can be adopted without any problem in these frequency relationships, There are problems such as large insertion loss and large occupation area.

  On the other hand, when a filter is inserted on the waveguide circuit side, it is easy to realize a high Q value / low loss circuit. However, a filter circuit formed by a waveguide is generally used as a waffle iron type filter, for example. Due to the three-dimensional shape, the occupied volume is large, and depending on the shape and required characteristics, precise machining is required, which is often accompanied by an increase in cost.

  In order to eliminate these problems and suppress unnecessary image signals and reception frequency band noise, for example, in Patent Documents 1 and 2, a combination of a tapered waveguide and a metal plate filter is used. This metal plate filter is difficult to make an electrically stable mounting state with respect to the waveguide. That is, when mechanical caulking or the like is used as an inexpensive mounting method for the metal plate filter, an increase in insertion loss of the transmission band due to a poor electrical (microwave) contact or within the intended suppression frequency band. It is necessary to carry out production management such as screening tests to eliminate these problems.

  In addition, the metal plate filter itself requires a small mechanically occupied volume. However, when other waveguide circuits (filters and duplexers) are connected and mounted before and after the metal plate filter, the metal plate is structurally An appropriate length is required between the filter and another waveguide circuit, which prevents miniaturization.

  Furthermore, since the filter material requires some rust prevention treatment, plating treatment may cause an increase in skin resistance depending on the type of plating treatment, resulting in an increase in insertion loss in the transmission band. It becomes.

  On the other hand, the second harmonic, which is an unnecessary signal, has an adverse effect due to higher-order modes when a suppression circuit is installed in the waveguide. Therefore, as described above, a microstrip circuit or the like formed on a dielectric substrate is used. Since the signal suppression filter is formed, a loss is also generated in the transmission band, and these addition effects cause a decrease in transmission power, resulting in a decrease in transmitter power efficiency, resulting in an increase in power consumption. It was generated.

  As a technique for suppressing such second harmonics, there is Patent Document 3 described above. As shown in FIG. 12 (A), Patent Document 1 discloses that λ / 2 (λ: suppression frequency) The metal piece 6 is provided at a position separated by (wavelength). Further, this metal piece 6 has a length in the longitudinal direction (insertion direction of the probe 4) of λ / 2, and is configured such that the tip of the probe 4 is pseudo-grounded at a target frequency, for example, twice. Harmonics can be suppressed.

  However, since this suppression circuit has a high Q value due to the circuit configuration, it is difficult to ensure sufficient attenuation when the suppression frequency band is wide.

  The present invention has been made in view of the above-described problems, and the object of the present invention is to achieve downsizing and cost reduction even when a filter is inserted on the waveguide circuit side. A non-waveguide line-waveguide converter and a non-waveguide line-waveguide converter that can be suppressed, can attenuate a wide frequency band even in harmonics, and do not cause a decrease in transmission power. It is to provide a communication device used.

In order to achieve the above object, the invention according to claim 1 is a non-waveguide in which a waveguide provided with a short-circuited surface is inserted into the waveguide from its H-plane (plane parallel to the magnetic field). A non-waveguide line-waveguide converter comprising a probe formed at the tip of the line, and a resonance that projects from the short-circuit surface of the waveguide and is close to the probe and resonates at a predetermined frequency A columnar conductor for forming a circuit is provided.
That is, by projecting the columnar conductor from the short-circuited surface of the waveguide to form a coaxial structure, a parallel resonant circuit composed of an inductor (L) and a capacitor (C) is formed, and the columnar conductor is brought close to the probe. By capacitive coupling, a series resonant circuit composed of an inductor and a capacitor is configured.

The invention of claim 2 is characterized in that an open stub having a predetermined frequency as a grounding condition is provided at a position near the waveguide H surface of the probe.
The invention of claim 3 is a rectangular waveguide provided with a short-circuited surface, and is inserted into the waveguide from its H surface (a surface parallel to the magnetic field) and formed at the tip of the non-waveguide line. In a communication device using a non-waveguide line-waveguide converter comprising a probe, a resonance circuit that protrudes from the short-circuited surface of the waveguide and is arranged close to the probe and resonates at a predetermined frequency And the width of the rectangular waveguide in the magnetic field direction is set to a value in the vicinity of the reception frequency band equal to or lower than the cutoff frequency of the waveguide.

When the converter is connected to an output side (transmitter side) waveguide having a size different from that of the waveguide defined by the transmission frequency, the magnetic field of the rectangular waveguide is interposed between them. A waveguide step impedance converter having the same direction width as the magnetic field direction width of the converter and a different electric field direction width can be provided.
Further, the distributed constant line end probe and the open stub can be formed so as to leave only a portion of the dielectric substrate where the probe and the open stub are formed.

  According to the first aspect of the present invention, by providing the columnar conductor, the waveguide has a coaxial structure, and is equivalent to the impedance characteristics resulting from the length of the columnar conductor and the appropriate coupling capacitance between the columnar conductor and the probe. Thus, a resonant circuit that appears to be a short circuit at the probe end is formed, and this resonant circuit can suppress a predetermined frequency band in the image signal and the reception frequency band noise.

  According to the configuration of the second aspect, the open stub is provided in the probe in the waveguide without the ground conductor even on the dielectric substrate, and the connection point of the open stub in the probe is grounded at a predetermined suppression frequency. By being equivalent, a predetermined frequency band, for example, the second harmonic, is suppressed under good attenuation characteristics. Further, since the probe is disposed in the vicinity of the H plane, a transmission wave in a predetermined frequency band can be efficiently attenuated before switching to the waveguide mode.

  According to the configuration of the third aspect, in the communication apparatus used for the transmitter or the like, the waveguide is provided between the waveguide of the non-waveguide line-waveguide converter and the output-side waveguide. Since the step impedance converter is provided, the resonance circuit having the columnar conductor could not be suppressed while maintaining the suppression characteristics of the waveguide set to a value in which the vicinity of the reception frequency band is equal to or lower than the cutoff frequency of the waveguide. Unnecessary frequencies can be suppressed.

  According to the non-waveguide line-waveguide converter of the present invention, it is possible to achieve desired suppression characteristics in image signals and reception frequency band noise without increasing the size and volume of the converter. . In addition, since a resonance circuit for suppression is built into the converter, it is possible to avoid interference with other waveguide circuits connected to the converter, and structurally realized by changing the shape of the parts Therefore, there is an effect that the addition of parts such as a metal plate filter, that is, an increase in cost is not caused, and the overall size can be reduced and the cost can be reduced. Further, there is an advantage that there is no variation in characteristics during production, no special rust prevention treatment is necessary, and no increase in insertion loss due to this rust prevention treatment is caused.

According to the second aspect of the present invention, since the filter function is realized by the probe portion in the waveguide even though the circuit is formed on the substrate, for example, substantial dielectric loss is reduced in suppressing the second harmonic. In addition, since the filter circuit (open stub) is directly connected to the probe, the Q value can be appropriately lowered while being a circuit formed in the waveguide, and the second in a wide frequency band. It becomes possible to suppress harmonics with a sufficient amount of attenuation.
In addition, since both the filter according to the first and second embodiments form the filter in the waveguide, the insertion loss in the transmission frequency band can be minimized, and the decrease in transmission power and the increase in power consumption can be suppressed. There is an effect that it becomes possible.

  According to the invention of claim 3, in the transmitter, for example, an image signal and a part of the noise in the reception frequency band are suppressed by the resonance circuit using the columnar conductor, and the second harmonic is suppressed by the open stub. Due to the cutoff effect of the waveguide, it is possible to satisfactorily suppress reception band noise and image signals that cannot be suppressed by a resonance circuit using columnar conductors.

BRIEF DESCRIPTION OF THE DRAWINGS The structure of the non-waveguide line-waveguide converter based on 1st Example of this invention is shown, A figure (A) is a front view, A figure (B) is sectional drawing of the center part of a figure (A). . It is a circuit diagram which shows the equivalent circuit of the non-waveguide line-waveguide converter of 1st Example. It is a graph which shows the characteristic when not having a columnar conductor in the non-waveguide line-waveguide converter of 1st Example. It is a graph which shows the characteristic of the non-waveguide line-waveguide converter of 1st Example. The structure of the non-waveguide line-waveguide converter of 2nd Example is shown, A figure (A) is a front view, A figure (B) is sectional drawing of the center part of a figure (A), A figure (C) is It is the figure which looked at the probe direction from the bottom face side of a waveguide. The structure of the transmitter using the non-waveguide line-waveguide converter of 3rd Example is shown, A figure (A) is a front view, A figure (B) is sectional drawing of the center part of a figure (A), FIG. (C) is a front view of a portion taken along line II in FIG. (B). It is sectional drawing which shows the other structure of the 1st waveguide of 3rd Example. It is a graph which shows the characteristic of the transmitter using the non-waveguide line-waveguide converter of 3rd Example. It is a graph which shows the characteristic of the 2nd harmonic frequency vicinity of the transmitter using the non-waveguide line-waveguide converter of 3rd Example. The structure of the non-waveguide line-waveguide converter of 4th Example is shown, A figure (A) is a front view, A figure (B) is sectional drawing of the center part of a figure (A). The structure of the transmitter using the non-waveguide line-waveguide converter of 5th Example is shown, A figure (A) is a front view, A figure (B) is sectional drawing of the center part of a figure (A), FIG. (C) is a front view of a portion taken along line II-II in FIG. (B). The structure of the conventional non-waveguide line-waveguide converter is shown, FIG. (A) is a front view, FIG. (B) is sectional drawing of the center part of FIG. (A).

  1A and 1B show a non-waveguide line-waveguide converter according to a first embodiment of the present invention. As shown in FIG. Consists of a first waveguide 10A and a second waveguide 10B provided with a short surface S. The dielectric substrate 12 is sandwiched between the first waveguide 10A and the second waveguide 10B. Provided. In this dielectric substrate 12, a probe 14 extending from the tip of a microstrip line (distributed constant circuit) 13 is inserted into the inside from the upper H surface (surface parallel to the magnetic field) of the waveguide 10. In addition, a through hole 5 for grounding is provided so as to contact each wall surface of the portion where the first waveguide 10A and the second waveguide 10B are combined.

Then, in the first embodiment, the second substantially lambda _0/4 from the short side S of the waveguide 10B probes _14: While in the position of the length of (lambda ₀ the wavelength of a pass frequency), the short side S A prismatic (rectangular) columnar conductor 16 is formed which protrudes from the center position toward the probe 14 and is close to the probe 14. The columnar conductor 16 is approximately a quarter of the wavelength lambda ₁ of the suppression frequency from the short plane S, that is, is set to a length of lambda _1/4, the tip of the columnar conductor 16 is the relationship approaching the probe 14 Become. By providing the columnar conductor 16, the second waveguide 10B has a coaxial structure.

  When an electromagnetic field analysis is performed on a circuit provided with such a columnar conductor 16, an electromagnetic field distribution close to the TEM mode is recognized in the columnar conductor 16 at a certain specific frequency. That is, this TEM mode is generated when the columnar conductor 16 is equivalent to a quarter wavelength in the mode, and a strong TEM mode is not excited at other frequencies, and the inside of the waveguide is almost normal. It becomes tube mode.

  In the first embodiment, since the tip of the columnar conductor 16 in which the TEM mode is excited is disposed at a relatively close distance from the probe 14, it is considered that it is directly electrically coupled to the probe 14. Therefore, if expressed in a simple equivalent circuit based on these, it can be expressed as shown in FIG.

  In FIG. 2, 18 is a waveguide input terminal, 19 is an input waveguide, 20 is a probe, 21 is a non-waveguide line output terminal, 22 is a waveguide line up to a short plane, 23 is a short plane, 24 Represents a resonance circuit generated in the TEM mode of the columnar conductor 16, and 25 represents a coupling capacitance parasitic between the columnar conductor 16 and the probe 14. That is, the coaxial structure in which the columnar conductor 16 is provided in the second waveguide 10B constitutes a parallel resonant circuit 24 of L (inductor) C (capacitance), and the columnar conductor 16 is brought close to the probe 14 and capacitively coupled. Thus, for example, an LC series resonance circuit for suppressing an image signal and reception frequency band noise is configured.

The first embodiment has the above configuration, and the operation thereof will be described next.
As shown in FIG. 2, the coaxial structure including a substantially lambda _1/4 of the length of the columnar conductor 16 which projects into the center of the short side S, will be parallel resonant circuit 24 is formed, the resonance At the resonance frequency f ₀₁ of the circuit 24, the circuit end is equivalent to an open circuit, but the effective impedance of the resonance circuit 24 is inductive on the lower side than this f ₀₁ , and is formed by this equivalent inductor and the coupling capacitor 25. A series resonance circuit is generated, and at the resonance frequency _f02 , the tip of the probe 14 is equivalent to a short circuit.

These can be expressed by the following formulas 1 and 2. However, the equivalent inductor in the resonance circuit 15 is L1, the equivalent capacitor is C1, and the coupling capacitance 25 is C2.

That is, at the resonance frequency _f02 , the end of the probe 14 appears to be short-circuited, so that attenuation characteristics can be obtained. In the pass band, the characteristics are almost the same as those of a normal non-waveguide line-waveguide converter. It will be.
Further, since this resonant circuit has a very low loss and a high Q value, if the coupling of the columnar conductor 16 to the probe 14 is set sparse, even if the detuning frequency between the pass frequency and the suppression frequency is small, the resonance circuit passes. It is possible to minimize the adverse effect on the bandwidth.

  The resonance circuit 24 is generated in a coaxial structure, and its characteristics change depending on the size (thickness etc.) and length of the columnar conductor 16 with respect to the second waveguide 10B, and the coupling capacitance. 25 is mainly determined by the distance from the probe 14. Accordingly, desired characteristics can be obtained by appropriately combining these parameters, and actual design is performed using an electromagnetic field simulator or the like.

  FIG. 3 and FIG. 4 show characteristic examples by simulation of the first embodiment. FIG. 3 shows a case where there is no resonance circuit, and FIG. 4 shows a case where the columnar conductor 16 is provided. This is a result when the suppression frequency is set in the vicinity of 12.75 GHz with .75 to 14.5 GHz. As shown in FIG. 4, when the columnar conductor 16 is provided, a large amount of attenuation is obtained in the vicinity of 12.75 GHz. With such characteristics, for example, a part of the noise in the reception frequency band is suppressed satisfactorily. can do.

  In the first embodiment, the conversion from the microstrip line (distributed constant line) to the waveguide line has been described. However, the present invention is not limited to the conversion from the coaxial line to the waveguide line, or other various transmission lines. This can also be applied to the conversion.

5A to 5C show the configuration of the second embodiment applied to a coaxial line. In this second embodiment, the tip of the central conductor of the coaxial line 28 is used as the probe 29 in the waveguide 27 from the H surface (surface parallel to the magnetic field) of the upper surface of the waveguide 27 provided with the short surface S. Protruding and arranged. The protruding length of approximately lambda _1/4 from the center of the short side S of the waveguide 27, a prismatic columnar conductor 30 so as to be close to the probe 29 is placed.

  Even with the configuration of the second embodiment, the same effects as those of the first embodiment can be obtained. In the first and second embodiments, the columnar conductors 16 and 30 are formed in a rectangular column shape, but the columnar conductor may be formed in other shapes such as a columnar shape. Moreover, although the rectangular thing was used as the waveguides 10 and 27, a circular waveguide can also be used.

  FIGS. 6A to 6C show the configuration of the third embodiment. In the third embodiment, the second harmonic is suppressed in addition to the suppression of the image signal and the reception frequency band noise. It is an apparatus that can be applied to a transmitter that is a communication device. Specifically, it is applied to a satellite communication system or the like in the Ku-Band band (transmission frequency: 13.75 to 14.5 GHz, reception frequency: 10.95 to 12.75 GHz). Among the transmitters of this system, what is called “Universal Band” normally has an IF frequency of 0.95 to 1.7 GHz and a local oscillator frequency of 12.8 GHz, so that the frequency of the image signal is 11.10 to 11. .85 GHz.

In the third embodiment, the rectangular second waveguide 10B in which the columnar conductor 16 described in the first embodiment is formed and the size (the magnetic field direction and the electric field) of the second waveguide 10B will be described in detail later. The first waveguide 10 </ b> C has a rectangular waveguide portion 32 having the same length as the direction). Further, as shown in FIG. 6B, a dielectric substrate 12 is provided so as to be sandwiched between the first waveguide 10C and the second waveguide 10B, and this dielectric substrate 12 is provided with a microstrip. The line 13 and the probe 14 are formed, and open stubs 33a and 33b are formed in a form connected to the probe 14 in the vicinity of the H surface in the waveguide 10B and substantially parallel to the H surface. The Open stub 33a of the third embodiment, 33b is provided for suppressing the second harmonic, D ≒ λ _2/4 from the center of the probe _14: The length of the (lambda ₂ wavelength of a predetermined suppression frequency) It is held and the tip is opened.

That is, approximately lambda _2/4 of the length of the open stub 33a, by providing the 33b, at a predetermined suppression frequency, the open stub 33a, 33b becomes the connection point grounding equivalent of the probe 14 connected, third In the embodiment, good attenuation characteristics can be obtained in the frequency band of the second harmonic. The open stubs 33a and 33b are arranged not near the center position in the waveguide section 32 or the second waveguide 10B but in the vicinity of the H plane, so that the transmission wave in the target frequency band is changed to the waveguide mode. It can be attenuated efficiently before switching.

  Furthermore, in the third embodiment, the width (long side width) d of the rectangular waveguide direction 32 of the first waveguide 10C and the second waveguide 10B is set to 11.6 mm, for example. The cutoff frequency of the wave tube (32, 10B) is set to 12.75 GHz or more which is the upper limit of the reception frequency band and less than 13.75 GHz (12.75 to 13.75 GHz) which is the lower limit of the transmission band. An attenuation amount proportional to the length in the propagation direction of the waveguide portion 32 of the waveguide 10C is obtained. That is, the single resonance suppression characteristic obtained by the columnar conductor 16 generates a steep attenuation characteristic (FIG. 4) at a frequency immediately below the transmission band, and below the suppression frequency is the above waveguide. By generating an attenuation characteristic in which the attenuation amount increases as the frequency decreases due to the cut-off, a characteristic (FIG. 10 described later) obtained by adding these attenuation characteristics can be obtained.

  By the way, normally, there are standardized dimensions recommended for the frequency used in the waveguide used for connection between devices. For example, WR75 (R120 in IEC) of the EIA standard is used at the frequency of the embodiment. Therefore, the irregular-sized waveguide (32, 10B) as described above is inconvenient. Therefore, in the third embodiment, the WR75 is used as the transmitter-side (output-side) waveguide section (final output terminal) 35 of the first waveguide 10C, and the microstrip line with a reduced size in the magnetic field direction- Between the waveguide converter section (10B, 32) and the transmitter side waveguide section 35, a step impedance converter 36 for impedance matching as well as physical size conversion is provided.

  In this step impedance converter 36, waveguide portions 36a and 36b are linearly connected in a step shape. As this step impedance converter, a non-waveguide line-waveguide converter portion (32 , 10B), the impedance becomes too high if only the magnetic field direction (H surface) is limited. In view of this, the step impedance converter 36 of the embodiment has a waveguide section 36a in which both the magnetic field direction and the electric field direction are narrowed, and a waveguide section 32 in which the width in the magnetic field direction is the same and the width in the electric field direction is reduced. The wave tube is composed of 36b. That is, the impedance of the portion is made close to the impedance of the size of the transmitter-side waveguide portion 35 by the waveguide portion 36a, and the reception frequency band based on the above-described waveguide cutoff frequency is set by the waveguide portion 36b. Is effectively obtained.

  FIG. 7 shows another configuration of the step impedance converter (impedance matching circuit) of the first waveguide. The first waveguide 10D of this example is a waveguide for bending the waveguide. A tube portion (bend waveguide) 36d is provided, and a waveguide portion 36c is disposed between the tube portion and the transmitter-side waveguide portion 35. The waveguide portion 36c and the waveguide portion 36d of the step impedance converter 36 have the same rectangular size, that is, the width of the magnetic field direction and the electric field direction as those of the waveguides 36a and 36b. The center of (E surface) is offset (shifted downward). Similar characteristics can be obtained by the step impedance converter 36 having such a configuration.

  According to the third embodiment, the equivalent circuit shown in FIG. 2 is formed by the configuration of the waveguide portion 32 as the non-waveguide line-waveguide converter and the second waveguide 10B. The LC parallel resonance circuit 24 is configured by the coaxial structure provided with the columnar conductors 16, and the LC series resonance circuit is configured by capacitively coupling the columnar conductors 16 to the probe 14.

  As described above, when the impedance matching circuit is designed on the condition that the step impedance converter 36 having a constant width in the magnetic field direction is used, the characteristics shown in FIG. 8 can be obtained. is there. In other words, a characteristic that is attenuated steeply from a slightly lower frequency of 13.75 GHz, which is the lower limit of the transmission frequency band, to about 12.75 GHz, and a greatly attenuated characteristic is also obtained at the lower frequency. Image signals, reception frequency band noise, and the like can be satisfactorily suppressed.

  In the third embodiment, since the open stubs 33a and 33b are provided on the probe 14, the second harmonic can be suppressed. FIG. 9 shows an example of characteristics obtained when the suppression frequency is 27.5 to 29 GHz, which is the second harmonic band of the pass frequency, in the third embodiment, as shown in FIG. It can be seen that attenuation of about 43 dB is obtained in the suppression frequency band of 27.5 to 29 GHz, and that the band width of 30 dB suppression is as wide as 5 GHz.

  FIGS. 10A and 10B show the configuration of the fourth embodiment in which the area of the dielectric substrate in the waveguide of the above embodiment is reduced. The configuration of the probe 14 and the open stubs 33a and 33b of the fourth embodiment is the same as that of the third embodiment, but the portion of the dielectric substrate 38 where the probe 14 and the open stubs 33a and 33b are formed. The dielectric substrate in the other region 50 is cut and removed in accordance with the internal space of the waveguide 10 except for the portion where the through hole 5 is formed.

  According to the configuration of the fourth embodiment, since the dielectric substrate in the region 50 is removed, there is an effect that the insertion loss due to the presence of the dielectric substrate 38 can be further reduced.

FIGS. 11A to 11C show the configuration of a communication device (transmitter) according to a fifth embodiment in which the features of the third embodiment are applied to a coaxial line. In the fifth embodiment, the tip of the central conductor of the coaxial line 28 is used as the probe 29 in the waveguide 40 from the H surface (surface parallel to the magnetic field) of the upper surface of the waveguide 40 provided with the short surface S. The open stubs 41a and 41b are connected to and formed on the probe 29 at a position near the H surface in the waveguide 40 and substantially parallel to the H surface. The open stub 41a, 41b, similar to the third embodiment, approximately from the center of the probe 29 λ _2/4: is the length of (lambda ₂ wavelength of a predetermined suppression frequency).

  The waveguide 40 is provided with a waveguide portion 42a having a rectangular magnetic field direction width of, for example, 11.6 mm, as in the third embodiment, and the cutoff frequency of the waveguide portion 42a. Is set to 12.75 GHz or more, which is the upper limit of the reception frequency band, and less than 13.75 GHz, which is the lower limit of the transmission band, so that an amount of attenuation proportional to the length of the waveguide section 42a can be obtained. A step impedance converter 43 including waveguide portions 43a and 43b is provided between the transmitter-side waveguide 42b of the waveguide 40 and the waveguide portion 42a. As in the third embodiment, the device is similar to the waveguide portion 43a in which both the magnetic field direction and the electric field direction are narrowed, and the waveguide portion 42a has the same width in the magnetic field direction as that of the waveguide portion 42a. The wave tube is composed of 43b.

  According to the fifth embodiment, as in the third embodiment, unnecessary frequencies such as image signals and reception frequency band noise can be suppressed and the second harmonic can be suppressed with good attenuation characteristics. It becomes possible.

  In the third to fifth embodiments, two open stubs 33a, 33b, 41a and 41b are provided. However, one open stub may be provided, and three or more open stubs may be provided in all directions. It is also possible.

Furthermore, in the third embodiment, the effect obtained by installing the open stubs 33a, 33b, 41a, 41b is as follows. That is,
a. Since there is no GND on the back surface of the dielectric substrate (12, 38) in the open stub portion and the electromagnetic field distribution is dispersed, the influence of the dielectric substrate is reduced, and the loss is less than that of the distributed constant line portion.
b. The open stub is provided in the probe (14, 29) in the vicinity of the wall surface of the waveguide H, and the transmission mode of this portion has not yet been converted to the waveguide mode. Therefore, it is possible to avoid the problem of a significant decrease in the amount of suppression due to the influence of the waveguide-side load condition.
c. The open stub is a circuit formed in the waveguide circuit, but the Q value is not higher than necessary because it is formed on the dielectric substrate and directly connected to the probe. Compared with the converter of Patent Document 3 described in (4), a broadband suppression characteristic can be obtained.
d. An open stub is equivalent to a case where a capacitive stub is added in the middle of the line at the passing frequency. Therefore, if the circuit design taking this into account is performed, it is possible to achieve the desired characteristics at the passing frequency. is there.

The present invention can be applied to a converter between a non-waveguide line and a waveguide, such as a microstrip line and a coaxial line, and a communication device equipped with a non-waveguide line-waveguide converter.
In addition, in the satellite communication system currently in operation, the reception frequency is lower than the transmission frequency, and the up-conversion of the transmitter is performed by mixing with a local oscillator signal having a frequency lower than the transmission frequency. The present invention can be applied to a transmitter of a satellite communication system in any frequency band. Further, if the relationship between the pass band and the suppression frequency is the same (the suppression frequency is lower than the pass band), the present invention can be applied to a communication device other than the satellite communication transmitter.

10, 10A-10C, 27, 40 ... waveguide,
2, 12, 38 ... dielectric substrate,
3, 13 ... microstrip line,
4, 14, 29 ... probe, 16, 30 ... columnar conductor,
24 ... Parallel resonant circuit, 25 ... Coupling capacitance,
28 ... Coaxial line, 32, 36a to 36d, 42a, 43a, 43b ... Waveguide section,
33a, 33b, 41a, 41b ... open stub,
35, 42b ... Transmitter side waveguide section,
36, 43 ... Step impedance converter.

Claims (3)

A waveguide provided with a short-circuit surface;
A non-waveguide line-waveguide converter comprising a probe inserted into the waveguide from its H-plane and formed at the tip of the non-waveguide line;
A non-waveguide line-waveguide characterized in that a columnar conductor is provided to project from the short-circuit surface of the waveguide and close to the probe and to form a resonant circuit that resonates at a predetermined frequency. converter.   2. The non-waveguide line-waveguide converter according to claim 1, wherein an open stub having a predetermined frequency as a grounding condition is provided at a position near the waveguide H surface of the probe. A rectangular waveguide provided with a short-circuit surface;
In a communication apparatus using a non-waveguide line-waveguide converter, comprising a probe inserted into the waveguide from its H-plane and formed at the tip of a non-waveguide line,
Protruding from the short-circuited surface of the waveguide and close to the probe, providing a columnar conductor for configuring a resonance circuit that resonates at a predetermined frequency,
A non-waveguide line-waveguide converter characterized in that the width of the rectangular magnetic field direction of the waveguide is set to a value in which the vicinity of the reception frequency band is equal to or lower than the cutoff frequency of the waveguide. Communication device used.

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