JP2006191428A - Microstrip line waveguide converter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a microstrip line waveguide converter capable of reducing reflection losses at an input end and an output end, even if a coupling hole is not formed in a dielectric layer. <P>SOLUTION: The microstrip line waveguide converter 1 leads an electromagnetic wave propagating on a microstrip line 10 via a coupling hole 70 to a waveguide to be connected 90. The reflection losses at the input end I and the output end O can be reduced by adjusting the length or width of a stub provided at the tip of a line conductor 12, the length of a projection of the line conductor 12 projecting into the coupling hole 70, the width of the stop of a reactance stop formed in a patterned layer 32, the distance d from the patterned layer 32 up to a short-circuiting board 64, and so on. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a microstrip line waveguide converter that guides an electromagnetic wave propagating through a microstrip line to a waveguide and guides an electromagnetic wave propagating through the waveguide to a microstrip line.

  In a wireless communication device, a microstrip line waveguide converter that guides an electromagnetic wave propagating through a microstrip line to a waveguide and guides an electromagnetic wave propagating through the waveguide to the microstrip line is widely used.

  This is because signal generators, frequency converters, amplifiers, etc. are configured as planar circuits, whereas antenna feed systems connected to planar circuits require low-loss transmission characteristics and waveguides are used. It is.

  FIG. 12 shows a microstrip line waveguide converter 5 having a general conventional configuration. FIG. 11 is a perspective view showing the appearance of the microstrip line waveguide converter 5. A connected waveguide 90 is connected to the ground conductor layer 36 side of the multilayer substrate 30 on which the microstrip line 10 is formed, and the pattern layer 32 of the portion of the multilayer substrate 30 to which the connected waveguide 90 is connected is connected. On the side, a shielding housing 60 is provided to prevent loss due to radiation of electromagnetic waves. FIG. 12A is a cross-sectional view taken along a plane AA′BB ′ (hereinafter referred to as a propagation axis cross section) perpendicular to the ground conductor including the propagation axis Z of the microstrip line 10 in FIG. 11, and FIG. FIG. 11 is a cross-sectional view taken along a plane CC′DD ′ (hereinafter referred to as a propagation cross section) parallel to the propagation cross section including the stub 14 provided at the open end of the line conductor 12 of the microstrip line 10 in FIG. FIG. 12C shows the configuration of the pattern layer 32, and FIG. 12D shows the configuration of the ground conductor layer 36. As can be seen from FIGS. 12A and 12B, the multilayer substrate 30 includes a pattern layer 32, a dielectric layer 34, and a ground conductor layer 36. The line conductor 12 is provided on the pattern layer 32, and the microstrip line 10 is configured by the dielectric layer 34 and the ground conductor layer 36. The pattern layer 32 is provided with a ground pattern 38 so as to surround the line conductor 12. Here, a region surrounding the line conductor 12 shown in FIG. The ground pattern 38 is electrically connected to the ground conductor layer 36 through a plurality of through holes 40 penetrating the dielectric layer 34, and the same potential as the ground conductor layer 36 is achieved.

  The ground conductor layer 36 and the dielectric layer 34 are provided with a coupling hole 70 having the same shape and size as the cross section of the connected waveguide 90, and electromagnetic waves propagating through the microstrip line 10 are guided. It leads to the output end O of the tube 90. The line conductor 12 is opened at a position protruding above the coupling hole 70, and a stub 14 for matching the microstrip line 10 and the connected waveguide 90 is provided at the tip.

  The shielding housing 60 provided on the pattern layer 32 is in electrical contact with the ground pattern 38 so that the ground pattern 38 and the ground conductor layer 36 have the same potential. This shielding housing 60 is composed of a wall portion 62 and a short-circuit plate 64, forms a waveguide whose terminal is short-circuited, and loss due to radiation at the coupling portion between the microstrip line 10 and the connected waveguide 90. Reduce.

  In addition, since the shape and size of the edge surrounding the conductor line by the ground pattern 38 match the shape and size of the cross section of the connected waveguide 90, the wall 62 of the shielding housing 60, the ground pattern 38, The through hole 40 constitutes a waveguide having a short-circuit plate 64 connected to the end.

  Here, considering the case where an electromagnetic wave is excited from the microstrip line 10 side and a load is connected to the connected waveguide 90 side, the input end I is connected to the microstrip line 10 side, and the connected waveguide is connected. An output terminal O is defined on the 90 side.

  At the input end I and the output end O of the microstrip line waveguide converter 5, it is preferable that the reflection loss is small. This is because a large reflection loss at the input terminal I and the output terminal O may cause problems such as an increase in transmission loss due to the generation of a reflected wave and abnormal oscillation of the active circuit. In general, the high frequency circuit connected to the microstrip line waveguide converter 5 is designed on the assumption that impedance matching is achieved at the connection end. If the reflection loss at the input terminal I or the output terminal O is not sufficiently reduced, it means that impedance matching with the adjacent circuit connected thereto is not sufficiently achieved, and this sufficiently improves the design performance of the adjacent circuit. It can also cause problems that cannot be demonstrated.

Therefore, the microstrip line waveguide converter 5 having a conventional structure, the length l _s or width of the stub 14 provided at the tip of the line conductor 12 w, protruding length l _c of the line conductor 12, or short-circuiting plate 64 By adjusting the distance d to the pattern layer 32, the reflection loss at the input / output end I and the output end O is reduced.

  Here, a case has been described in which electromagnetic waves are excited from the microstrip line 10 side and a load is connected to the connected waveguide 90 side, but on the contrary, the connected waveguide 90 side The same argument holds for the case where electromagnetic waves are excited from and the load is connected to the microstrip line 10 side. That is, reducing the reflection loss at the input / output terminals is an important matter in determining the configuration of the microstrip line waveguide converter 5. Since the microstrip line waveguide converter 5 can be considered as a lossless passive circuit, an electromagnetic wave is excited from the microstrip line 10 side, and a load is connected to the connected waveguide 90 side. The insertion loss has the same value when the electromagnetic wave is excited from the connected waveguide 90 side and the load is connected to the microstrip line 10 side.

  The configuration for guiding the electromagnetic wave propagating to the microstrip line as described here to the waveguide provided on the back side thereof by a coupling means such as a coupling hole is disclosed in the following document.

Japanese Patent Laid-Open No. 2003-037406

  In the microstrip line waveguide converter 5 shown in FIG. 12, a coupling hole 70 having the same shape and size as the cross section of the connected waveguide 90 is provided in the ground conductor layer 36 and the dielectric layer 34. Here, the reason why the coupling hole 70 is provided in the dielectric layer 34 is as follows.

Assuming that the coupling hole 70 is not provided in the dielectric layer 34 as in the microstrip line waveguide converter 7 shown in FIG. 13, the region r ₁ of the pattern layer coupling portion 50 formed in the pattern layer 32. And the boundary surface b ₁ between the upper layer region r ₂ of the coupling hole 70 provided in the ground conductor layer 36 of the dielectric layer 34 and the coupling provided in the region r ₂ and the ground conductor layer 36. The medium constant of the electromagnetic wave becomes discontinuous at the boundary surface b ₂ between the region r ₃ formed in the hole 70 and the reflection loss at the input end I and the output end O increases. The increase in the reflection loss varies depending on the relative dielectric constant of the dielectric applied to the dielectric layer 34. However, in the range of the relative dielectric constant of the general dielectric applied to the dielectric layer 34, the reflection loss increases as described above. By adjusting the length l _s or width w of the stub 14 provided at the tip of the line conductor 12, the protruding length l _c of the line conductor 12, or the distance d between the short-circuit plate 64 and the pattern layer 32, as shown in FIG. It is impossible to avoid.

Therefore, in the conventional configuration, not only the ground conductor layer 36 but also the dielectric layer 34 is provided with a coupling hole 70 having the same shape and size as the cross-section of the connected waveguide 90, and at the tip of the line conductor 12. the length of the provided stubs 14 l _s or width w, since a possible to reduce the reflection loss by adjusting the distance d protruding length l _c of the line conductor 12 or to short-circuiting plate 64 and the pattern layer 32, is there.

However, when a coupling hole having the same shape and size as the cross-section of the connected waveguide 90 is provided in the dielectric layer 34, the shape and size of the coupling hole 70 provided in the ground conductor layer 36, etc. Sufficient manufacturing accuracy is required. As a result, man-hours and costs in production are increased. Therefore, as a microstrip line waveguide converter, it is preferable that the dielectric layer 34 is not provided with a coupling hole, and the region r _{2 in} FIG. 13 is filled with a dielectric.

  The present invention has been made to solve such a problem, and provides a microstrip line waveguide converter capable of reducing reflection loss at an input / output end without providing a coupling hole in a dielectric layer. .

  The present invention guides an electromagnetic wave propagating through a microstrip line to the waveguide by connecting the waveguide to a coupling hole provided in the ground conductor of the microstrip line, and converts the electromagnetic wave propagating through the waveguide into the microstrip. A microstrip line waveguide converter that leads to a line, the microstrip line having a ground region that has the same potential as the ground conductor on a surface where the line conductor of the microstrip line exists, and the ground region is connected to the waveguide A reactance restriction is formed.

  Further, the present invention guides the electromagnetic wave propagating through the microstrip line to the waveguide by connecting the waveguide to the coupling hole provided in the ground conductor of the microstrip line, and the electromagnetic wave propagating through the waveguide. A microstrip line waveguide converter that leads to a microstrip line, provided on a surface of the microstrip line where a line conductor is present, and is in contact with the grounding area that is at the same potential as the grounding conductor. And a matching stub formed on a line conductor of the microstrip line, wherein the ground region forms a reactance stop for the waveguide.

  Further, the present invention guides the electromagnetic wave propagating through the microstrip line to the waveguide by connecting the waveguide to the coupling hole provided in the ground conductor of the microstrip line, and the electromagnetic wave propagating through the waveguide. A microstrip line waveguide converter that leads to a microstrip line, wherein the coupling hole is provided with a reactance stop for the waveguide.

  Further, the present invention guides the electromagnetic wave propagating through the microstrip line to the waveguide by connecting the waveguide to the coupling hole provided in the ground conductor of the microstrip line, and the electromagnetic wave propagating through the waveguide. A microstrip line waveguide converter that leads to a microstrip line, and is provided in a region opposite to the region where the ground conductor of the microstrip line exists across the surface where the line conductor of the microstrip line exists. And a matching stub formed on the line conductor of the microstrip line, and the coupling hole is provided with a reactance restrictor for the waveguide.

  In addition, the present invention is preferably applied to a configuration in which a dielectric exists between the coupling hole and the line conductor of the microstrip line.

  According to the present invention, the design parameter of the microstrip line waveguide converter in which the waveguide is connected to the coupling hole provided in the ground conductor of the microstrip line is added. Therefore, it is possible to realize a microstrip line waveguide converter with sufficiently reduced reflection loss at the input / output end without providing a coupling hole in the dielectric layer between the line conductor and the ground conductor of the microstrip line. it can.

  FIG. 1 shows a propagation section of the microstrip line waveguide converter 1 according to the first embodiment of the present invention, and FIG. 2 shows a propagation axis section thereof. The external shape is the same as in FIG. As can be seen from FIGS. 1 and 2, the multilayer substrate 30 includes a pattern layer 32, a dielectric layer 34, and a ground conductor layer 36.

  FIG. 3A shows the configuration of the pattern layer 32, and FIG. 3B shows the configuration of the ground conductor layer 36. The line conductor 12 is provided on the pattern layer 32, and the microstrip line 10 is configured by the dielectric layer 34 and the ground conductor layer 36. The pattern layer 32 is provided with a ground pattern 38 so as to surround the line conductor 12. Here, a region formed by providing the reactance diaphragm 52 in the region of the pattern layer coupling portion 50 shown in FIG. The ground pattern 38 is electrically connected to the ground conductor layer 36 through a plurality of through holes 40 penetrating the dielectric layer 34, and the same potential as the ground conductor layer 36 is achieved. It is preferable that the interval between the plurality of through holes 40 be sufficiently shorter than the free space wavelength of the electromagnetic wave propagating through the microstrip line 10 and the connected waveguide 90.

  The ground conductor layer 36 is provided with a coupling hole 70 having the same shape and size as the section of the connected waveguide 90, and guides the electromagnetic wave propagating through the microstrip line 10 to the connected waveguide 90. The electromagnetic wave propagating through the connected waveguide 90 is guided to the microstrip line 10. Unlike the microstrip line waveguide converter 5 shown in FIG. 12, the dielectric layer 34 is not provided with a coupling hole, and the upper layer portion of the ground conductor layer 36 where the coupling hole 70 is provided is It is blocked by a dielectric. The line conductor 12 is opened at a position protruding above the coupling hole 70, and a stub 14 for matching the microstrip line 10 and the connected waveguide 90 is provided at the tip.

  The shielding housing 60 provided on the pattern layer 32 is in electrical contact with the ground pattern 38 so that the ground pattern 38 and the ground conductor layer 36 have the same potential. This shielding housing 60 is composed of a wall portion 62 and a short-circuit plate 64, forms a waveguide whose end is short-circuited, and reduces loss due to radiation at the coupling portion between the microstrip line 10 and the connected waveguide 90. To reduce.

  In the microstrip line waveguide converter 5 shown in FIG. 12, the shape and size of the pattern layer coupling portion 50 are matched with the shape and size of the cross section of the connected waveguide 90. In the present embodiment, as a result of configuring the aperture pattern layer coupling portion 54 formed by providing the reactance aperture 52 in the pattern layer coupling portion 50 that matches the cross-sectional shape and size of the connected waveguide 90, the pattern The area contributing to electromagnetic field coupling in the layer 32 is narrowed only in the portion where the reactance restrictor 52 is provided.

  Here, the reactance restriction means a restriction valve that acts as a reactance circuit with respect to a fundamental propagation mode electromagnetic field (hereinafter simply referred to as a fundamental propagation mode) of a waveguide. That is, in the configuration of FIG. 1 and FIG. 2, the basic propagation mode is in the plane in which the waveguide formed by the wall 62 of the shielding housing 60, the ground pattern 38, and the through hole 40 is cut by the pattern layer 32. It can be understood that a throttle valve acting as reactance, that is, a reactance throttle 52 is provided.

Here, the general property of the reactance restriction will be described. FIGS. 4A and 4B show the configuration of a reactance diaphragm that acts as an inductance on the cross section of the waveguide 92. 4A is a view of the reactance stop 94 viewed from a cross section in which the propagation direction of the waveguide 92 is a normal line, and FIG. 4B is a view of the reactance stop 94 viewed from the H plane cross section of the waveguide 92. It is a figure. The arrows in FIG. 4 indicate the electric field direction in the basic propagation mode. When the waveguide 92 is represented by the transmission line 80 for the basic propagation mode, the reactance restriction 94 of FIGS. 4A and 4B is inserted into the transmission line 80 in a shunt as shown in FIG. 4C. It can be expressed as equivalent to inductance. The diaphragm widths d _L1 and d _L2 may be different. In general, as the values of d _L1 and d _L2 are larger, the inductance value in FIG. 4C tends to be larger. 4D and 4E show the configuration of the reactance stop 96 that acts as a capacitance on the cross section of the waveguide 92. FIG. 4D is a view of the reactance stop 96 viewed from a cross section in which the propagation direction of the waveguide 92 is a normal line, and FIG. 4E is a view of the reactance stop 96 viewed from the E plane cross section of the waveguide 92. It is a figure. The arrows in FIG. 4 indicate the electric field direction in the basic propagation mode. When the waveguide 92 is represented by the transmission line 80 for the fundamental propagation mode, the reactance stop 96 of FIGS. 4D and 4E is equivalent to the capacitance inserted in the shunt as shown in FIG. Can be expressed as a thing. The diaphragm widths d _C1 and d _C2 may be different. In general, the larger the values of d _C1 and d _C2 , the larger the capacitance value shown in FIG. More generally, as shown in FIG. 4G, it is possible to configure a reactance stop 98 that combines the properties of both inductance and capacitance.

  Next, the operation principle of the microstrip line waveguide converter 1 of FIGS. 1 and 2 will be described with reference to FIG. In the state where the electromagnetic wave propagates to the microstrip line 20 in FIG. 5A, an electromagnetic field having a distribution as shown in FIG. In FIG. 5A, a solid line represents an electric field, and a broken line represents a magnetic field. From this, it can be seen that the electromagnetic field distribution in the cross section of the microstrip line 20 is composed of an electric field connecting the ground conductor layer 26 and the line conductor 22 and a magnetic field surrounding the line conductor 22. On the other hand, the basic propagation mode propagating through the waveguide is composed of an electric field connecting the H-plane tube walls as shown in FIG. 5B and an annular magnetic field perpendicular to the surface where the electric field distribution is zero.

  As can be seen from FIGS. 5A and 5B, the distribution of the magnetic field surrounding the line conductor 22 of the microstrip line 20 and the annular magnetic field perpendicular to the plane where the electric field distribution in the waveguide 92 is zero. The distribution of is approximate. Therefore, the cross section of the microstrip line 10 and the H surface of the connected waveguide 90 are joined, and the magnetic field surrounding the line conductor 12 of the microstrip line 10 excites the magnetic field in the basic propagation mode as shown in FIG. By doing so, it can be seen that a configuration for guiding the electromagnetic field propagating through the microstrip line 10 to the connected waveguide 90 is possible.

  In general, when different types of lines are connected, it is known that the reflection loss and the insertion loss can be reduced by coupling the lines on a plane having an approximate electromagnetic field distribution. Therefore, when connecting the microstrip line and the waveguide, the distribution of the magnetic field surrounding the line conductor of the microstrip line and the annular magnetic field perpendicular to the plane where the electric field distribution in the waveguide is zero. Therefore, it can be said that joining the cross section of the microstrip line and the H-plane of the waveguide is advantageous from the viewpoint of reducing reflection loss and insertion loss.

  The above has focused on the point that the electromagnetic field propagating through the microstrip line 10 is guided to the connected waveguide 90 by magnetic coupling. However, the electromagnetic field propagating through the microstrip line 10 is also affected by electrical coupling. Guided to the connecting waveguide 90. When electromagnetic waves are incident from the microstrip line 10, an electric field is generated that connects the tip of the line conductor 12 and the shielding housing 60 as shown in FIG. Since this electric field has a component in the same direction as the electric field direction of the fundamental propagation mode, the fundamental propagation mode is excited and the electromagnetic field propagating through the microstrip line 10 is guided to the connected waveguide 90.

  Next, the reflection loss at the input end I of the microstrip line 10 will be described with reference to FIG. When an electromagnetic wave enters from the input end I, the fundamental propagation mode is excited in the connected waveguide 90 by the electromagnetic wave. However, the microstrip line waveguide converter 1 has innumerable configurations that generate reflected waves at the input end I, and the reflected waves generated in each of these configurations are added, and this is ultimately input. It becomes a reflected wave with respect to the electromagnetic wave incident on the microstrip line waveguide converter 1 from the end I. Hereinafter, the reflected wave generated by each configuration that generates the reflected wave in the microstrip line waveguide converter 1 is referred to as a local reflected wave, and is incident from the input terminal I, which is a result of adding and summing the local reflected waves. It will be distinguished from the final “reflected wave” for electromagnetic waves. The state in which the reflection loss is sufficiently small is a state in which the reflected wave with respect to the incident electromagnetic wave is sufficiently small as a result of addition and summing in relation to amplitude and phase so that the locally reflected wave is canceled. .

One of the configurations that cause a local reflected wave at the input end I is a discontinuous surface S ₁ . Since the conductors of the ground conductor layer 36 forming a discontinuous surface S ₁ in the line conductor 12 pair is interrupted, electromagnetic field boundary condition is satisfied of the electromagnetic wave incident here a discontinuous surface S _1, from a discontinuous surface S ₁ A transmitted wave is generated on the connected waveguide 90 side, and a locally reflected wave is generated on the microstrip line 10 side.

Further, in the portion of the line conductor 12 protruding above the coupling hole 70, it can be considered that the current is distributed so that the electric field component becomes zero on the surface of the line conductor 12. Then, the equivalent circuit connected by the relationship between the voltage which is excited in the line conductor 12 in the current and discontinuous surface S ₁ on the discontinuous surface S ₁ distributed as such is determined. This equivalent circuit is a coupling circuit to the connected waveguide 90 connected to the discontinuous surface S ₁ , and its input impedance is adjusted by changing the shape of the portion protruding above the coupling hole 70. Can do. This is from the standpoint of electromagnetic theory, nothing but that it is possible to adjust the transmission coefficient and reflection coefficient at the discontinuity surface S _1, by adjusting the reflection coefficient at the discontinuity surface S ₁ is where occurs locally reflected waves It is none other than adjusting the amplitude and phase of the signal. Therefore, it is possible to adjust the local reflected wave amplitude and phase caused by the discontinuous surface S ₁ by changing the portion having a shape protruding on the coupling holes 70 of the line conductor 12, microstrip line waveguide In the tube converter 1, this can be done by changing the length l _s or width w of the stub 14 provided at the tip or the protruding length l _c of the line conductor 12.

  Furthermore, the waveguide formed by the wall portion 62 constituting the shielding housing 60 and the waveguide formed by the arrangement of the plurality of through holes 40 have a fundamental propagation mode due to electromagnetic waves incident from the input end I. Excited.

As in FIG. 11, the region of the aperture pattern layer coupling portion 54 formed in the pattern layer 32 is r ₄ , and the region of the upper layer portion of the coupling hole 70 provided in the ground conductor layer 36 in the dielectric layer 34. R ₅ , and r ₆ is a region formed in the coupling hole 70 provided in the ground conductor layer 36. The boundary surface between the region r ₄ and the region r ₅ is b ₃ , and the boundary surface between the region r ₅ and the region r ₆ is b ₄ .

The basic propagation mode excited at the boundary surface b ₃ in FIG. 2 and propagating in the direction of the short-circuit plate 64 is reflected at the short-circuit plate 64 and arrives again at the boundary surface b _3, and is reflected again there. Multiple reflection is performed between ₃ and the short-circuit plate 64. In the basic propagation mode, a local reflected wave is excited with a certain amplitude and phase in the microstrip line 10 while being subjected to multiple reflections between the boundary surface b ₃ and the short-circuit plate 64.

Fundamental propagation modes propagating in the direction of the excited boundary b ₄ at the interface b ₃ in FIG. 2 are reflected at the boundary surface b _4, repeated that again arrived at the boundary surface b _3, where it reflected again, Multiple reflection is performed between the boundary surface b ₃ and the boundary surface b ₄ . In the basic propagation mode, a local reflected wave is excited in the microstrip line 10 with a certain amplitude and phase while performing multiple reflections between the boundary surface b ₃ and the boundary surface b ₄ .

Incidentally, a fundamental propagation mode of multiple reflection between the short-circuiting plate 64 and the boundary surface b _3, a boundary surface b ₃ and the boundary surface b ₃ between the fundamental propagation mode of multiple reflection between the boundary surface b ₄ The fundamental propagation mode in which multiple reflections are performed contributes to the excitation of electromagnetic waves to the output end O of the connected waveguide 90. Here, reflection and transmission of the fundamental propagation mode at the boundary surfaces b ₃ and b ₄ are considered, but the boundary surface b which is a boundary surface between the region r _M and the region r ₄ inside the shielding housing 60. _M, and also the reflection and transmission of the fundamental propagation mode of the boundary surface at a boundary surface b _W between the interior of the region r _W and the region r ₆ of the connection waveguide 90 is of course contemplated. However, in the present embodiment, since the thickness t of the pattern layer 32 and the ground conductor layer 36 is made sufficiently thinner than the wavelength, the influence can be ignored.

In the microstrip line waveguide converter 1 of FIGS. 1 and 2, a reactance stop 52 is provided in the region r ₄ . As described above, the reactance stop 52 can be regarded as the reactance inserted in the shunt when the waveguide is regarded as a transmission line, and the basic propagation reflected at the boundary surface b ₃ by changing the reactance value. The mode amplitude and phase can be adjusted. The reactance value can be adjusted by changing the diaphragm widths d _L1 and d _L2 in FIG. Therefore, by changing the diaphragm widths d _L1 and d _L2 of the reactance diaphragm 52, the amplitude and phase of the fundamental propagation mode reflected on the boundary surface b ₃ are adjusted, and the locally reflected wave excited on the microstrip line 10 is adjusted. Can be adjusted in amplitude and phase.

Further, when the distance d from the pattern layer 32 to the short-circuit plate 64 is changed, the phase of the fundamental propagation mode arriving at the boundary surface b3 is changed. A local reflected wave excited by the microstrip line 10 while being subjected to multiple reflections between the boundary surface b ₃ and the short-circuit plate 64 or between the boundary surface b ₃ and the boundary surface b ₄ arrives at the boundary surface b ₃ . Therefore, the amplitude and phase of the excited local reflected wave are determined according to the phase relationship of the individual fundamental propagation modes. Therefore, if the phase of the fundamental propagation mode arriving at the boundary surface b3 is changed by changing the distance d from the pattern layer 32 to the short-circuit plate 64, the amplitude of the local reflected wave excited on the microstrip line 10 and It can be said that the phase can be adjusted.

From the above, in the present embodiment, there are mainly the following four design parameters for adjusting the amplitude and phase of the locally reflected wave. (1) Length l _s or width w of the stub 14 (2) Projection length l _c of the line conductor 12 (3) Diaphragm widths d _L1 and d _L2 of the reactance diaphragm 52 in the diaphragm pattern layer coupling portion 54 (4) Distance d from pattern layer 32 to short-circuit plate 64. By adjusting the design parameters of (1) to (4), all the local waves generated on the microstrip line 10 side when the electromagnetic wave is input from the input terminal I to the microstrip line waveguide converter 1 are obtained. If the reflected wave is canceled, the reflection loss at the input terminal I can be reduced.

  Here, the case where the microstrip line 10 is the input side has been considered. The microstrip line waveguide converter 1 according to the present embodiment can be considered as a lossless transmission circuit. Therefore, under the condition that a matching load is connected to the connected waveguide 90, if the reflection loss at the input end I is sufficiently small, it can be considered that the insertion loss is also sufficiently small. Further, due to the reciprocity of the lossless transmission circuit, when the connected waveguide 90 is the input side, the insertion loss when the electromagnetic wave propagates from the connected waveguide 90 to the microstrip line 10 and the connected conductor The reflection loss seen from the wave tube 90 toward the microstrip line waveguide converter 1 can also be said to be sufficiently small.

Next, a design method of the microstrip line waveguide converter 1 of FIGS. 1 and 2 will be described. In designing, the length l _s or width w of the stub 14, the protruding length l _c of the line conductor 12, the widths d _L1 and d _L2 of the diaphragm of the reactance diaphragm 52, the distance d from the surface to the short-circuit plate 64, etc. It is preferable to use main design parameters.

First, as the initial values of the design parameters, the length of the stub 14 at the tip of the line conductor 12 is set to zero, and the distance d from the pattern layer 32 to the short-circuit plate 64 is set to the in-tube wavelength of the waveguide formed by the wall 62. The diaphragm widths d _L1 and d _L2 of the reactance diaphragm 52 are set to zero. In this initial state, the reflection loss and insertion loss of the microstrip line waveguide converter 1 are obtained by electromagnetic field simulation or experiment.

Next, the design parameters are optimized so that the reflection loss and insertion loss obtained by electromagnetic field simulation or experiment have desired values. In the microstrip line waveguide converter 5 having a conventional structure shown in FIG. 12, since the coupling hole 70 is provided on the dielectric layer 34, the length l _s or width w of the stub 14, protruding length of the line conductor 12 By adjusting l _c and the distance d from the pattern layer 32 to the short-circuit plate 64 as design parameters, the reflection loss and the insertion loss can be set to desired values. However, in this embodiment, the dielectric layer 34 is not provided with a coupling hole, and the upper layer portion of the ground conductor layer 36 where the coupling hole 70 is provided is blocked by the dielectric. Therefore, the length l _s or width w of the stub 14, protruding length l _c of the line conductor 12, the only distance d from the pattern layer 32 to short circuit plate 64 becomes insufficient design parameters. Therefore, in addition to these design parameters, the diaphragm widths d _L1 and d _L2 of the reactance diaphragm 52 are used as design parameters.

  Here, a case where a reactance restrictor in the pattern layer 32 that acts as an inductance is taken up, but one that acts as a capacitance or a combination of the properties of both inductance and capacitance is also applicable. FIG. 6 shows a configuration of the pattern layer 32 to which the reactance stop 54 acting as a capacitance is applied.

The length l _s or width w of the stub 14, the width d _L1, d _L2 of the aperture of the reactance diaphragm 52, the distance from the surface to the short-circuiting plate 64 d, have been a major design parameters etc., in addition to these Needless to say, various elements that determine the configurations of FIGS. 1, 2, 3 and 6 can be selected as design parameters.

  Next, a microstrip line waveguide converter according to a second embodiment of the present invention will be described. FIG. 7 shows a propagation cross section of the microstrip line waveguide converter 3 according to the second embodiment, and FIG. 8 shows a propagation axis cross section thereof. The external shape is the same as in FIG. In the present embodiment, the diaphragm reactance is not provided in the pattern layer coupling portion 50 as in the microstrip line waveguide converter 1 according to the first embodiment, but the diaphragm reactance is provided in the coupling hole 70 in the ground conductor layer 36. It is provided.

  As can be seen from FIGS. 7 and 8, the multilayer substrate 30 includes a pattern layer 32, a dielectric layer 34, and a ground conductor layer 36. FIG. 9A shows the configuration of the pattern layer 32, and FIG. 9B shows the configuration of the ground conductor layer 36. The line conductor 12 is provided on the pattern layer 32, and the microstrip line 10 is configured by the dielectric layer 34 and the ground conductor layer 36. The pattern layer 32 is provided with a ground pattern 38 so as to surround the line conductor 12. Here, the holed region shown in FIG. The ground pattern 38 is electrically connected to the ground conductor layer 36 through a plurality of through holes 40 penetrating the dielectric layer 34, and the same potential as the ground conductor layer 36 is achieved. It is preferable that the interval between the plurality of through holes 40 be sufficiently shorter than the free space wavelength of the electromagnetic wave propagating through the microstrip line 10 and the connected waveguide 90.

  The shielding housing 60 provided on the pattern layer 32 is in electrical contact with the ground pattern 38 so that the ground pattern 38 and the ground conductor layer 36 have the same potential. This shielding housing 60 is composed of a wall portion 62 and a short-circuit plate 64, forms a waveguide whose end is short-circuited, and reduces loss due to radiation at the coupling portion between the microstrip line 10 and the connected waveguide 90. To reduce.

  The ground conductor layer 36 is provided with an aperture coupling hole 74 formed by providing a reactance aperture 72 in the coupling hole 70 matched with the cross-sectional shape and size of the connected waveguide 90, and is provided with a microstrip line. The electromagnetic wave propagating through 10 is guided to the connected waveguide 90. Unlike the configuration of the microstrip line waveguide converter 1 shown in FIGS. 1 and 2, the dielectric layer 34 is not provided with the coupling hole 70, and the ground conductor layer 36 is provided with the aperture coupling hole 74. The upper layer portion of the portion is filled with a dielectric. The line conductor 12 is opened at a position protruding above the aperture coupling hole 74, and a stub 14 for matching the microstrip line 10 and the connected waveguide 90 is provided at the tip. .

  FIG. 9B shows the configuration of the ground conductor layer 36 of the microstrip line waveguide converter 3. In this configuration, a reactance stop 72 is provided in a coupling hole 70 having the same shape and size as the cross-section of the connected waveguide 90. The general nature of the reactance restriction is as described above.

  When the microstrip line and the waveguide are connected, the distribution of the magnetic field surrounding the line conductor of the microstrip line and the distribution of the magnetic field in the H plane of the waveguide are approximated. It has been mentioned above that joining the cross section and the H-plane of the waveguide is advantageous from the viewpoint of reducing reflection loss and insertion loss. Since this embodiment uses this principle as in the first embodiment, the details of the operation principle are omitted.

Next, the reflection loss at the input end I of the microstrip line 10 will be described with reference to FIGS. The reflected wave with respect to the electromagnetic wave incident on the microstrip line waveguide converter 3 according to the present embodiment from the input end I is described as the sum of the local reflected waves as in the first embodiment. Can do. One of the configurations that cause local reflected waves is the discontinuous surface S _2. By changing the shape of the portion of the line conductor 12 that protrudes above the aperture coupling hole 74, the discontinuous surface S ₂ is used. The same applies to the point that the amplitude and phase of the generated local reflected wave can be adjusted. Therefore, also in the microstrip line waveguide converter 3 according to the present embodiment, by changing the length l _s or width w or line protrusion of conductor 12 length l _c, stub 14 provided at the distal end , it is possible to perform amplitude and phase adjustment of the local reflection wave generated at the discontinuous plane S _2.

Here, as in FIG. 13, the region of the pattern layer coupling portion 50 formed in the pattern layer 32 is r ₇ , and the upper layer portion of the aperture coupling hole 74 provided in the ground conductor layer 36 of the dielectric layer 34. The region is r ₈ , and the region formed in the aperture coupling hole 74 provided in the ground conductor layer 36 is r ₉ . The boundary surface between the region r ₇ and the region r ₈ is b ₅ , and the boundary surface between the region r ₈ and the region r ₉ is b ₆ .

The basic propagation that multi-reflects between the boundary surface b ₅ and the short-circuit plate 64 is reflected on the waveguide constituted by the wall 62 by the electromagnetic wave incident on the microstrip line waveguide converter 3 from the input end I. In the waveguide formed of the mode and the plurality of through holes 40, a fundamental propagation mode in which multiple reflection is performed between the boundary surface b ₅ and the boundary surface b ₆ is excited, and the microstrip line 10 has a certain amplitude. And a local reflected wave is excited with a phase.

In microstrip waveguide converter 3 of FIG. 7 and FIG. 8, the reactance aperture 72 is provided in the region r _9. As described above, the reactance stop 72 can be regarded as a reactance inserted in the shunt when the waveguide is regarded as a transmission line. By changing the reactance value, the basic value reflected at the interface b ₆ is reflected. The amplitude and phase of the propagation mode can be adjusted. The reactance value can be adjusted by changing the diaphragm widths d _L1 and d _L2 in FIG. Accordingly, by changing the diaphragm widths d _L1 and d _L2 of the reactance diaphragm 72, the amplitude and phase of the fundamental propagation mode reflected at the boundary surface b ₆ and the fundamental propagation mode arriving at the boundary surface b ₅ are adjusted. The amplitude and phase of the locally reflected wave excited on the strip line 10 can be adjusted.

Further, when the distance d from the pattern layer 32 to the short-circuit plate 64 is changed, the phase of the fundamental propagation mode arriving at the boundary surface b ₅ is changed. A local reflected wave that is excited by the microstrip line 10 while being subjected to multiple reflections between the pattern layer 32 and the short-circuit plate 64 or between the boundary surface b ₅ and the boundary surface b ₆ arrives at the boundary surface b ₅ . Since the electromagnetic waves are excited by the individual fundamental propagation modes, the amplitude and phase of the excited local reflected wave are determined according to the phase relationship of the individual fundamental propagation modes. Therefore, if the phase of the fundamental propagation mode arriving at the boundary surface b ₅ is changed by changing the distance d from the pattern layer 32 to the short-circuit plate 64, the amplitude of the locally reflected wave excited on the microstrip line 10. It can be said that the phase can be adjusted.

Here, reflection and transmission of the fundamental propagation mode at the boundary surfaces b ₅ and b ₆ are considered, but the boundary surface b which is a boundary surface between the region r _M and the region r ₅ inside the shielding housing 60. _M, and also the reflection and transmission of the fundamental propagation mode of the boundary surface at a boundary surface b _W between the interior of the region r _W and the region r ₉ of the connecting waveguides 90 are of course contemplated. However, in the present embodiment, since the thickness t of the pattern layer 32 and the ground conductor layer 36 is made sufficiently thinner than the wavelength, the influence can be ignored.

From the above, in the present embodiment, there are mainly the following four parameters for adjusting the amplitude and phase of the locally reflected wave. (1) Length l _s or width w of the stub 14 (2) Projection length l _c of the line conductor 12 (3) (4) Pattern of the diaphragm widths d _L1 and d _L2 of the reactance diaphragm 72 in the diaphragm coupling hole 74 The distance d from the layer 32 to the shorting plate 64; By adjusting the design parameters (1) to (4), when an electromagnetic wave is input to the microstrip line waveguide converter 3, local reflected waves generated on the microstrip line 10 side are canceled out. If it is in a state, reflection loss can be reduced.

  Here, the case where the microstrip line 10 is the input side has been considered. The microstrip line waveguide converter 3 according to the present embodiment can be considered as a lossless transmission circuit. Therefore, if the reflection loss at the input end I in the microstrip line 10 is sufficiently small, the insertion loss when the electromagnetic wave propagates from the connected waveguide 90 to the microstrip line 10 and the microstrip line conduction from the connected waveguide 90. It can be said that the reflection loss seen from the wave tube converter 3 side is also sufficiently small.

The design of the microstrip line waveguide converter 3 of FIGS. 7 and 8 can be performed in the same manner as the design of the microstrip line waveguide converter 1 according to the first embodiment. That is, the design parameters are adjusted so that the reflection loss and the insertion loss have desired values. In this embodiment, the width d _L1, d _L2 length l _s or width w, protruding length l _c of the line conductor 12, the reactance aperture 72 in the diaphragm coupling hole 74 aperture stub 14, a short circuit from the pattern layer 32 The distance d to the plate 64 is preferably a design parameter. In the first embodiment, the aperture width of the reactance aperture 52 in the aperture pattern layer coupling portion 54 is a design parameter. In the present embodiment, the aperture width of the reactance aperture 72 in the aperture coupling hole 74 is the design parameter. Is different.

  Here, the case where a reactance diaphragm acting as an inductance is applied as the reactance diaphragm in the diaphragm coupling hole 74 has been taken up. Is also applicable. FIG. 10 shows the configuration of the ground conductor layer 36 in the case where a reactance stop 76 acting as a capacitance is applied.

The main design includes the length l _s or width w of the stub 14, the protruding length l _c of the line conductor 12, the widths d _L1 and d _L2 of the diaphragm of the reactance diaphragm 72, the distance d from the surface to the short-circuit plate 64, and the like. Although parameters are used, it goes without saying that in addition to these, various elements shown in FIGS. 7 to 10 can be selected as design parameters.

  The embodiment of the present invention has been described above. It goes without saying that the present invention is not limited to these embodiments, and various embodiments are possible within the scope of the gist of the present invention.

It is a figure which shows the structure in the propagation cross section of the microstrip line waveguide converter which concerns on 1st Embodiment. It is a figure which shows the structure in the propagation axis cross section of the microstrip line waveguide converter which concerns on 1st Embodiment. It is a figure which shows the structure of the pattern layer and ground conductor layer of the microstrip line waveguide converter which concerns on 1st Embodiment. It is a figure explaining the reactance stop of a waveguide. It is a figure which shows the principle of operation of the microstrip line waveguide converter which concerns on 1st Embodiment. It is a figure which shows the structure of the pattern layer to which the reactance stop which acts as a capacitance is applied. It is a figure which shows the structure in the propagation cross section of the microstrip line waveguide converter which concerns on 2nd Embodiment. It is a figure which shows the structure in the propagation axis cross section of the microstrip line waveguide converter which concerns on 2nd Embodiment. It is a figure which shows the structure of the pattern layer and ground conductor layer of the microstrip line waveguide converter which concerns on 2nd Embodiment. It is a figure which shows the structure of the grounding conductor layer to which the reactance restriction | limiting which acts as a capacitance is applied. It is a figure which shows the external appearance of a microstrip line waveguide converter. It is a figure which shows the structure of the microstrip line waveguide converter of a conventional structure. In the microstrip line waveguide converter of the conventional structure, it is a figure which shows the structure which blocked the upper layer part of the coupling hole provided in the grounding conductor layer with the dielectric material.

Explanation of symbols

  1, 3, 5, 7 Microstrip line waveguide converter, 10 Microstrip line, 12, 22 Line conductor, 14 Stub, 30 Multilayer substrate, 32 Pattern layer, 34 Dielectric layer, 26, 36 Ground conductor layer, 38 ground pattern, 40 through hole, 50 pattern layer joint, 52, 56, 72, 76, 94, 96, 98 reactance diaphragm, 54 diaphragm pattern layer joint, 60 shielding housing, 62 wall, 64 short circuit board, 70 coupling hole, 74 aperture coupling hole, 80 transmission line, 90 connected waveguide, 92 waveguide.

Claims (5)

By connecting the waveguide to the coupling hole provided in the ground conductor of the microstrip line, the electromagnetic wave propagating through the microstrip line is guided to the waveguide, and the electromagnetic wave propagating through the waveguide is guided to the microstrip line. A stripline waveguide converter comprising:
Provided with a ground region having the same potential as the ground conductor on the surface where the line conductor of the microstrip line exists,
The microstrip line waveguide converter, wherein the ground region forms a reactance restriction for the waveguide. By connecting the waveguide to the coupling hole provided in the ground conductor of the microstrip line, the electromagnetic wave propagating through the microstrip line is guided to the waveguide, and the electromagnetic wave propagating through the waveguide is guided to the microstrip line. A stripline waveguide converter comprising:
A ground region provided on the surface where the line conductor of the microstrip line is present, and the same potential as the ground conductor;
A shielding housing provided in contact with the grounding area;
A matching stub formed on the line conductor of the microstrip line;
With
The microstrip line waveguide converter, wherein the ground region forms a reactance restriction for the waveguide. By connecting the waveguide to the coupling hole provided in the ground conductor of the microstrip line, the electromagnetic wave propagating through the microstrip line is guided to the waveguide, and the electromagnetic wave propagating through the waveguide is guided to the microstrip line. A stripline waveguide converter comprising:
A microstrip line waveguide converter, wherein the coupling hole is provided with a reactance stop for the waveguide. By connecting the waveguide to the coupling hole provided in the ground conductor of the microstrip line, the electromagnetic wave propagating through the microstrip line is guided to the waveguide, and the electromagnetic wave propagating through the waveguide is guided to the microstrip line. A stripline waveguide converter comprising:
A shielding housing provided in a region opposite to the region where the ground conductor of the microstrip line exists across the surface where the line conductor of the microstrip line exists,
A matching stub formed on the line conductor of the microstrip line;
With
A microstrip line waveguide converter, wherein the coupling hole is provided with a reactance stop for the waveguide. The microstrip line waveguide converter according to claim 1, wherein:
A microstrip line waveguide converter, wherein a dielectric is present between the coupling hole and the line conductor of the microstrip line.

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