JP2011130154A - Optical reception device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To increase phase control speed by simplifying a configuration for phase state control in an optical reception device. <P>SOLUTION: The optical reception device includes: a demodulator 1 which includes a delay interferometer 2 including optical circuits 21 and 22 being first and second optical circuits that belong to first and second sequences respectively and twin photodiodes 31 and 32 for receiving output light of the optical circuits 21 and 22 respectively and demodulates differentially modulated optical signals PS1 and PS2 independently of each other using the first and second sequences to output electric signals ES1 and ES2 being first and second demodulated signals with a prescribed phase shifted from each other; a control part 50 being a first control part which controls the phase state of the optical signal PS1 in the optical circuit 21 using the electric signal ES1; an adder 75 being an arithmetic operation part which arithmetically operates the electric signals ES1 and ES2 to output an arithmetic operation signal; and a control part 70 being a second control part which controls the phase state of the optical signal PS2 in the optical circuit 22 using the electric signal ES2 and the arithmetic operation signal. <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to an optical receiving apparatus, and more particularly to an optical receiving apparatus that receives an optical signal modulated by a differential M-phase phase shift modulation method such as a DQPSK (Differential Quadrature Phase Shift Keying) method.

Conventionally, a differential M-phase phase shift modulation method (M = 2 ^N , where N is a natural number of 2 or more) is known as a modulation method used in optical communication. A typical example of this modulation method is the DQPSK method.

  In optical communication using the DQPSK scheme, 0, 0.5π, π, and 1 between time-sequential symbols corresponding to four types of data 0, 1, 2, and 3 in a transmission apparatus. An optical signal to which four types of phase differences of .5π are given is generated, and the optical signal is transmitted from the optical transmission device, whereby 2-bit information is transmitted. This optical signal propagates through the optical fiber and is input to the optical receiver. In the optical receiver, the optical signal is demodulated by the demodulator, and four types of data 0, 1, 2, and 3 are reproduced from the output of the demodulator.

  Patent Document 1 discloses a technique for demodulating an optical signal modulated by the DQPSK method. As described in Patent Document 1, the demodulator of the optical reception device in the DQPSK system is configured by a splitter, two asymmetric Mach-Zehnder interferometers (MZI), and a twin photodiode (PD). There are many.

  The splitter bifurcates the received optical signal, and the bifurcated optical signal is sent to two MZIs, respectively. In each MZI, an optical path length difference is provided between two optical paths by increasing the physical length of the other optical path with respect to one optical path. A delay time equal to the symbol period is given.

  Furthermore, a phase shift amount of + 0.25π is given to the signal propagating through the shorter optical path of one MZI so that the two systems of electrical signals output from the twin photodiodes are symmetric, and the other MZI A signal propagating through the shorter optical path is given a phase shift amount of −0.25π. The twin photodiode is composed of two photodiodes that convert output light from each MZI into an electric signal, and a transimpedance amplifier that converts a current into a voltage.

  Generally, in order to demodulate received light modulated by the DQPSK method while suppressing the bit error rate, it is desirable to suppress the difference between the current value of the phase shift amount and the target value within each MZI within 1 °. .

  In this regard, Non-Patent Document 1 discloses a technique for controlling the phase shift amount of MZI in a DBPSK (Differential Binary Phase Shift Keying) method. The target value of the MZI phase shift amount in the DBPSK system is 0 or π. When the low frequency dither signal generated by the low frequency oscillator is superimposed on the heater voltage of the long arm, the phase shift amount of the MZI vibrates accordingly. The envelope of the output signal of the photodiode vibrates. Furthermore, since the oscillation frequency, phase, and amplitude of the light intensity of the output light of the MZI change depending on the center value of the phase shift amount, the oscillation frequency, phase, and amplitude of the output envelope of the twin photodiode are also shifted by the phase shift of the MZI. It depends on the central value of the quantity.

  For example, when the dither frequency is 10 Hz and the center value of the phase shift amount of MZI is 0 or π, the output envelope of the twin photodiode has a vibration frequency of 20 Hz and a phase of 0.5π. Have. Therefore, when the dither signal and the envelope are synchronously detected, 0 is obtained. Further, when the central value of the phase shift amount of MZI is 0 or a deviation δ (0 <δ <0.5π) that is more positive than π, the vibration frequency is 10 Hz and the phase is 0. Therefore, when the dither signal and the envelope are synchronously detected, a positive value is obtained. On the other hand, when the center value of the phase shift amount of MZI is 0 or has a deviation δ (−0.5π <δ <0) that is more negative than π, it has a vibration frequency of 10 Hz and a phase of π. Therefore, when the dither signal and the envelope are detected synchronously, a negative value is obtained.

  As described above, the phase shift amount is changed by changing the heater voltage of the long arm so that the phase shift amount is reduced if the synchronous detection result is a positive value and the phase shift amount is increased if the result is negative. It can be seen that the value should be adjusted. This is a control method of the phase shift amount in the DBPSK system disclosed in Non-Patent Document 1.

  Also in the DQPSK system, a method of applying a low-frequency dither signal can be used as in the DBPSK system. That is, by individually superimposing a dither signal on each MZI and performing synchronous detection, phase shift amounts of 0.25π, 0.75π, 1.25π, 1.75π are given to one MZI, and the other The MZI can be adjusted so as to give a difference of 0.5π or 1.5π from the phase shift amount of the MZI. However, the difference from the DBPSK method is that the above-described phase shift amount may not be reached only by the above method. That is, the convergence points in the method of superimposing the dither signal are 0.25π, 0.75π, 1.25π, and 1.75π, and the convergence point is determined only by the initial value of the phase shift amount. . Here, considering the periodicity of 2π, 1.75π is equivalent to −0.25π. Since 0.25π and 1.25π differ by π, in this case, only the sign of the output signal of the twin photodiode is inverted, which represents essentially the same data. Therefore, if one MZI converges to 0.25π, the other MZI must converge to 0.75π or 1.75π corresponding to the phase shift difference of 0.5π or 1.5π. . However, when synchronous detection is performed by individually superimposing dither signals on each MZI, both MZIs have the same phase and anti-phase characteristics, such as a combination of 0.25π or a combination of 0.25π and 1.25π depending on the initial value. May converge to phase.

  In order to solve this technical problem, Patent Document 1 proposes a technique for controlling the phase shift amount of each MZI in the DQPSK system. An optical receiving apparatus using the DQPSK system in Patent Document 1 includes a demodulator, a CDR, a MUX (multiplexer), a paralleling unit, a reception processing unit, a control unit, an interferometer control unit, a clock recovery control unit, and a multiplexing control unit. Is done.

  The CDR has a function of reproducing the clock signal and the data signal from the output signal of the demodulator, and the MUX has a function of multiplexing the data signal in synchronization with the clock signal output from the CDR. The paralleling unit has a function of branching the clock signal and data signal output from the MUX into 16 branches and outputting a low frequency data signal, and the reception processing unit is controlled so that frame synchronization is established. And a function of transmitting a control signal to the interferometer control unit, the clock recovery control unit, and the multiplexing control unit.

  If the CDR logic inversion process is executed through the clock recovery control unit and the MUX bit swap process is executed through the multiplexing control unit, if the frame synchronization is not successfully acquired, the MZIs are in phase or antiphase. Therefore, a process for controlling the phase shift amount of each MZI in the demodulator is executed through the interferometer control unit. The logical inversion process, the bit swap process, and the MZI phase shift amount control process are repeated until the frame synchronization is successfully acquired. As a result, each MZI converges to an optimum phase shift amount, so that frame synchronization is established and normal communication is possible.

JP 2007-288702 A (FIGS. 1 and 3)

Becoarn, L.M. Boissau, T. Julien, B.B. Grandpierre, G.G. Dupont, S.M. Plantady, P.M. Marcelou, J.M. -F. "Investigation on a stable DPSK receiver at 10Gb / s with a 5111 km-long transmission," 31st European Conference on Optical Communication 2005. 1, pp. 9-10, 2005. (Figure 3)

  By the technique of the above-mentioned patent document 1, each MZI can be converged to an optimum phase shift amount. However, the sequence to which one MZI belongs has four reception states of 0.25π, 0.75π, 1.25π, and 1.75π, and the sequence to which the other MZI belongs also has 0.25π, 0.75π, 1.. Since four reception states of 25π and 1.75π can be taken independently, all the reception states are 16 ways. In order to determine that one of them is applicable, according to the technique of Patent Document 1 described above, the control of the optical receiving apparatus covers each MZI, CDR, and MUX, so that the control of the optical receiving apparatus is complicated. . Moreover, since such complicated control is realized, the configuration becomes complicated. Further, because of these, there is a problem that the time for establishing normal communication is delayed.

  The present invention has been made in view of the above-described problems, and receives an optical signal modulated by a differential modulation method. The optical receiver has a simplified configuration and can perform normal communication at high speed. The purpose is to provide a device.

  An optical receiver according to a first aspect of the present invention receives a delay interferometer including first and second optical circuits belonging to the first and second series, and output light of the first and second optical circuits, respectively. The first and second twin photodiodes are provided, and the differentially modulated optical signals are separately demodulated using the first and second series, and the first and second demodulated signals that are shifted from each other by a predetermined phase are output. Using the first demodulated signal, a first control unit for controlling the phase state of the optical signal in the first optical circuit, and calculating the first and second demodulated signals, And a second control unit that controls a phase state of the optical signal in the second optical circuit using the second demodulated signal and the arithmetic signal.

  An optical receiver according to the second aspect of the present invention includes a delay interferometer including first and second optical circuits belonging to the first and second series, and output light from the first and second optical circuits. First and second twin photodiodes that receive light respectively, and first and second demodulated signals that are differentially demodulated using the first and second series and deviated from each other by a predetermined phase. A first demodulator that outputs the first demodulated signal, a first control unit that controls a phase state of the optical signal in the first optical circuit, and a multiplier that multiplies the first and second demodulated signals. A multiplier that outputs a signal; and a second controller that controls a phase state of the optical signal in the second optical circuit by using the multiplied signal.

  According to the optical receiver of the first aspect of the present invention, the delay interferometer including the first and second optical circuits belonging to the first and second series, and the output light of the first and second optical circuits are used. First and second twin photodiodes that receive light respectively, and first and second demodulated signals that are differentially demodulated using the first and second series and deviated from each other by a predetermined phase. Using the first demodulated signal, a first control unit for controlling the phase state of the optical signal in the first optical circuit, and calculating the first and second demodulated signals. By providing a calculation unit that outputs a signal, and a second control unit that controls a phase state of the optical signal in the second optical circuit using the second demodulated signal and the calculation signal, Since the phase difference with the second series can be determined, the configuration for phase difference control It was iodinated, and it is possible to perform high-speed normal communication.

  According to the optical receiver of the second aspect of the present invention, the delay interferometer including the first and second optical circuits belonging to the first and second series and the outputs of the first and second optical circuits are provided. First and second twin photodiodes for receiving light respectively, and first and second differentially modulated optical signals are separately demodulated using the first and second series and shifted by a predetermined phase from each other. A demodulator that outputs a demodulated signal, a first control unit that controls a phase state of the optical signal in the first optical circuit using the first demodulated signal, and the first and second demodulated signals are multiplied. The first sequence and the second sequence by including a multiplication unit that outputs a multiplication signal and a second control unit that controls a phase state of the optical signal in the second optical circuit using the multiplication signal. Phase difference can be discriminated, so the configuration for phase difference control is simplified and correct. It is possible to perform Do high-speed communication.

1 is a diagram illustrating a configuration of an optical receiving device according to Embodiment 1. FIG. 2 is a cross-sectional view showing a structure of a delay interferometer 2. FIG. It is a graph of the relationship between the phase θi1 and the magnitude in the electrical signal ES1P. It is a figure which shows the low frequency signal LFS1. It is a figure which shows the envelope of electrical signal ES1P. It is a graph of the relationship between the phase θi1 and the magnitude in the electrical signal ES1P. It is a figure which shows the envelope of electrical signal ES1P. It is a graph of the relationship between the phase θi1 and the magnitude in the electrical signal ES1P. It is a figure which shows the envelope of electrical signal ES1P. It is a graph which shows the relationship between the amplitude and phase of detection signal DS1. It is a graph which shows the relationship between the amplitude of detection signal DS2b, and average phase error (delta) 2. It is a graph which shows the relationship between detection signal DS2a and average phase error (delta) 2. It is a graph which shows the relationship between average signal A2a and average phase error (delta) 2. It is a graph which shows an example of the time change of average signal A2a, detection signal DS2a, and average phase error (delta) 2. 6 is a diagram illustrating a configuration of an optical receiving apparatus according to Embodiment 2. FIG. It is a graph which shows the relationship between the average value signal by the average value calculation part 731b, and average phase error (delta) 2. FIG. 10 is a diagram illustrating a configuration of an optical receiving apparatus according to a third embodiment. FIG. 10 is a diagram illustrating a configuration of an optical receiving device according to a fourth embodiment. It is a graph which shows the relationship between the amplitude of detection signal DS2b, and average phase error (delta) 2. FIG. 10 is a diagram illustrating a configuration of an optical receiving device according to a fifth embodiment. It is a graph which shows the relationship between the average value which average value calculation part 731b outputs, and average phase error (delta) 2. It is a figure which shows the structure of the optical receiver which concerns on the modification 1. FIG. It is a figure which shows the structure of the optical receiver which concerns on the modification 2.

<A. Embodiment 1>
<A-1. Configuration>
FIG. 1 is a diagram showing a configuration of an optical receiving apparatus according to Embodiment 1 of the present invention. The optical receiving apparatus according to the first embodiment is an optical signal modulated with N-bit transmission data using a differential M-phase phase shift modulation method (M = ^2N , where N is a natural number of 2 or more). It is an optical receiver that receives PS. In the following, for example, in the case of M = 4 (N = 2), that is, in the case of receiving an optical signal PS modulated with 2-bit transmission data using the DQPSK scheme, the optical receiver according to the first embodiment Will be described. Note that the present invention is not limited to DQPSK but can be applied to further multiple values such as D8PSK.

  Here, 0, 0.5π, π, and 1.5π are respectively assigned to four types of values that can be taken by the 2-bit optical signal PS transmitted to the optical receiver. Therefore, in the optical signal PS input to the present optical receiver, the phase difference Δφi given between two consecutive symbols in time series is 0, 0.5π, π, and 1.5π. Either one.

  As shown in FIG. 1, an optical receiving apparatus according to the present invention uses a demodulator 1 that demodulates an optical signal modulated by the DQPSK method, and electric signals ES1N and ES2P that are demodulated signals. A data reproducing unit 60 that reproduces transmission data, a control unit 50 that is a first control unit that controls a phase state of an optical signal in the demodulator 1 using an electric signal ES1P that is a demodulated signal, and an electric that is a demodulated signal. An adder 75, which is a calculation unit that calculates signals ES1P and ES2N and outputs a calculation signal, and a second control unit that controls the phase state of the optical signal in the demodulator 1 using the electrical signal ES2N and the calculation signal. And a certain control unit 70.

<A-1-1. Configuration of Demodulator 1>
The demodulator 1 includes a delay interferometer 2 to which an optical signal PS is input, and a photoelectric conversion device 3 that converts output light output from the delay interferometer 2 into current and voltage. The delay interferometer 2 includes a splitter 20 that branches the optical signal PS into two optical signals PS1 and PS2, and two optical circuits 21 and 22 (first and second optical circuits) to which the respective optical signals PS1 and PS2 are input. ). The photoelectric conversion device 3 includes two twin photodiodes 31 and 32 (first and second twin photodiodes), and the two twin photodiodes 31 and 32 correspond to the two light receiving elements 4 and 5, respectively. Two transimpedance amplifiers 6 and 7 are provided. A series including the optical circuit 21 and the twin photodiode 31 is a first series, and a series including the optical circuit 22 and the twin photodiode 32 is a second series.

  The optical circuit 21 is an asymmetric Mach-Zehnder interferometer (MZI) having two optical paths 210 and 211 through which the optical signal PS1 propagates. In the optical circuit 21, the optical path 210 is set to have a longer physical length than the optical path 211, and an optical path length difference is provided between the two optical paths 210 and 211. Thus, a delay time equal to the symbol period is given to the optical signal PS1 propagating through the optical path 210.

  Similarly, the optical circuit 22 is an asymmetric Mach-Zehnder interferometer (MZI) having two optical paths 220 and 221 through which the optical signal PS2 propagates. In the optical circuit 22, the optical path 220 has a longer physical length than the optical path 221, and an optical path length difference is provided between the two optical paths 220 and 221. Thereby, a delay time equal to the symbol period is given to the optical signal PS2 propagating through the optical path 220.

  The optical circuit 21 includes a heater 213 that functions as a phase shift unit that gives a phase shift to the optical signal PS1 propagating through the optical path 211. The phase shift amount α1 that the heater 213 gives to the optical signal PS1 is controlled by the control unit 50 to be any one of 0.25π, 0.75π, 1.25π, and 1.75π.

  Similarly, the optical circuit 22 includes a heater 223 that functions as a phase shift unit that gives a phase shift to the optical signal PS2 propagating through the optical path 221. The phase shift amount α2 that the heater 223 gives to the optical signal PS2 is controlled by the control unit 70 to be either α1 + 0.5π or α1 + 1.5π.

  FIG. 2 is a sectional view showing the structure of the delay interferometer 2. The delay interferometer 2 according to the first embodiment is formed as a quartz-based planar optical circuit on a silicon substrate.

  When forming the delay interferometer 2, for example, as shown in FIG. 2, first, a glass of 50 μm thickness is first deposited on a silicon substrate 230 having a thickness of 1 mm by using a flame deposition method to form an under cladding layer 231. Form. Next, a glass whose relative refractive index is about 0.75% higher than that of the under cladding layer 231 is deposited on the under cladding layer 231 by using a flame deposition method, and then reactive ion etching is performed on the glass. The glass is patterned to form optical paths 210, 211, 220, and 221 as cores.

  Then, a glass having a thickness of 20 μm is deposited on the under cladding layer 231 so as to cover the optical paths 210, 211, 220, and 221 by using a flame deposition method, thereby forming an over cladding layer 232. Thereafter, a thin film made of tantalum nitride (TaN) is deposited on the overcladding layer 232 using a sputtering method, and the thin film is patterned using a photoengraving technique and an ion etching method.

  Thus, heaters 212, 213, 222, and 223 made of tantalum nitride are formed in the vicinity of the optical paths 210, 211, 220, and 221 respectively.

The optical circuits 21 and 22 are not limited to quartz-based waveguide devices, but may be devices such as semiconductors such as InP, ferroelectrics such as LiNbO ₃ , and spatially coupled devices.

<A-1-2. Configuration of Data Reproducing Unit 60>
The data reproducing unit 60 includes two CDR circuits 61 and 62 and a reproducing unit 63. The CDR circuit 61 extracts a clock signal from the electrical signal ES1N output from the demodulator 1, and identifies the electrical signal ES1N as “1” or “0” in synchronization with the clock signal. The CDR circuit 62 extracts a clock signal from the electrical signal ES2P output from the demodulator 1, and identifies the electrical signal ES2P as “1” or “0” in synchronization with the clock signal. The reproduction unit 63 reproduces and outputs 2-bit transmission data based on the identification results in the CDR circuits 61 and 62.

<A-1-3. Configuration of Control Unit 50>
The control unit 50 outputs an envelope detector 51 that is a first detector for detecting the electrical signal ES1P that is the first demodulated signal, a signal generator 52 that generates a low-frequency signal LFS1, and an envelope detector 51 that outputs the signal. An operation amount determination unit 53 to which the detection signal DS1 and the low frequency signal LFS1 as the first detection signal to be input are input, and an adder for adding the control voltage VT1 and the low frequency signal LFS1 output from the operation amount determination unit 53 54.

  The envelope detector 51 detects the envelope of the electric signal ES1P output from the demodulator 1 and outputs a detection signal DS1 indicating the envelope. The signal generator 52 generates and outputs a low frequency signal LFS1 having a period longer than the symbol period in the optical signal PS.

  The operation amount determination unit 53 is a multiplier 530 that multiplies the low-frequency signal LFS1 and the detection signal DS1, and a first average value calculation unit that calculates an average value of the output signal (first detection signal) from the multiplier 530. An average value calculation unit 531 and a determination unit 532 that is a first determination unit that determines the control voltage VT1 using the average signal A1 that is the first average value output from the average value calculation unit 531. Yes. The control voltage VT1 is an operation amount for the heaters 212 and 213 of the demodulator 1, that is, a voltage supplied to the heaters 212 and 213. The average value calculation unit 531 calculates the average value of the output signal of the multiplier 530 and thereby performs synchronous detection.

  The adder 54 adds the low frequency signal LFS1 to the control voltage VT1 and supplies the obtained signal to the heaters 212 and 213 as the control signal CS1. When the control signal CS1 is supplied to the heaters 212 and 213, the temperatures of the optical paths 210 and 211 change according to the control signal CS1. Then, the phase shift amount α1 given to the optical signal PS1 transmitted through the optical paths 210 and 211 changes. As described above, the heaters 212 and 213 change the phase shift amount α1 applied to the optical signal PS1 transmitted through the optical paths 210 and 211 in accordance with the control signal CS1. Since the control signal CS1 changes according to the change of the low frequency signal LFS1, the phase shift amount α1 changes slowly according to the frequency of the low frequency signal LFS1.

<A-1-4. Configuration of Control Unit 70>
The control unit 70 is an envelope detector 71a that is a second detector that detects the electrical signal ES2N that is the second demodulated signal, and a third detector that detects the signal (calculation signal) added by the adder 75. Envelope detector 71b, signal generator 72 for generating low-frequency signal LFS2, detection signals DS2a (second detection signal), DS2b (third detection signal) and low-frequency output from envelope detectors 71a and 71b An operation amount determination unit 73 to which the signal LFS2 is input, an adder 76 that adds a low-frequency signal LFS2 and a sweep voltage (described later) from the operation amount determination unit 73, an output of the adder 76, and an operation amount determination And an adder 74 for adding the control voltage VT2 from the unit 73.

  The envelope detector 71a detects the envelope of the electric signal ES2N output from the demodulator 1 and outputs a detection signal DS2a indicating the envelope.

  The adder 75 outputs the result of adding the electrical signal ES1P output from the demodulator 1 and the electrical signal ES2N.

  The envelope detector 71b detects the envelope of the output result of the adder 75 and outputs a detection signal DS2b indicating the envelope.

  The signal generator 72 generates and outputs a low frequency signal LFS2 having a period longer than the symbol period in the optical signal PS.

  The operation amount determination unit 73 multiplies the low-frequency signal LFS2 and the detection signal DS2a (second detection signal), and an average value (second average) of the output signal (second detection signal) from the multiplier 730a. Value), a second average value calculation unit 731a that is a second average value calculation unit, a detection monitor unit 734 that measures a detection signal DS2b that is a third detection signal, and a second average that is output from the average value calculation unit 731a. Using the average signal A2a that is a value and the output from the detection monitor unit 734, a control unit VT2, its initial value, a determination unit 732 that determines a sweep voltage, and a sweep to which the sweep voltage is input Part 733. The control voltage VT2 is an operation amount for the heaters 222 and 223 of the demodulator 1, that is, a voltage supplied to the heaters 222 and 223. The average value calculator 731a calculates the average value of the output signal of the multiplier 730a, and thereby performs synchronous detection.

  The adder 74 adds the output signal of the adder 76 to the control voltage VT2, and supplies the obtained signal to the heater 222 as the control signal CS2. When the control signal CS2 is supplied to the heater 222, the temperature of the optical path 221 changes according to the control signal CS2. Then, the phase shift amount α2 given to the optical signal PS2 transmitted through the optical path 221 changes.

  The envelope detectors 51, 71a, 71b are configured by, for example, a half-wave rectifier circuit or a full-wave rectifier circuit using a Schottky barrier diode. The average value calculation units 531 and 731a are constituted by, for example, a low-pass filter, an arithmetic circuit that calculates a direct current component by integration calculation executed at a cycle of the low-frequency signals LFS1 and LFS2, or Fourier transform calculation.

<A-2. Operation>
<A-2-1. Operation of Delay Interferometer 2>
The operation of the delay interferometer 2 in the demodulator 1 will be described. In the optical circuit 21, the input optical signal PS1 and the optical signal PS1 delayed by one symbol period interfere with the optical signal PS1, and the light intensity is increased for each symbol period according to the phase difference between the optical signals PS1. The changing optical signal is output from the two output ports of the optical circuit 21. The phase difference at a certain timing between the optical signal PS1 input to the optical circuit 21 and the optical signal PS1 delayed by one symbol period is the timing in the optical signal PS input to the delay interferometer 2. Therefore, the optical signal output from the optical circuit 21 has four kinds of values of the phase difference Δφi in the optical signal PS. Depending on (0, 0.5π, π, 1.5π), it has four or less kinds of light intensity. The optical circuit 21 outputs two complementary optical signals having different light intensities.

  Similarly, in the optical circuit 22, the input optical signal PS2 and the optical signal PS2 delayed by one symbol period interfere with each other, and each symbol period is determined according to the phase difference between the optical signals PS2. An optical signal whose light intensity changes is output from the two output ports of the optical circuit 22. The optical circuit 22 outputs two complementary optical signals having different light intensities.

  In the delay interferometer 2, for example, when the voltage supplied to the heater 213 changes, the temperature of the optical path 211 underneath changes, and the refractive index in the optical path 211 changes. As a result, the phase of the optical signal PS1 propagating through the optical path 211 changes. The control unit 50 controls the voltage supplied to the heaters 212 and 213, thereby controlling the phase shift amount α1 given to the optical signal PS1 propagating through the optical paths 210 and 211. Similarly, the control unit 70 controls the voltage supplied to the heaters 222 and 223, thereby controlling the phase shift amount α2 given to the optical signal PS2 propagating through the optical paths 220 and 221.

  In the above example, the heaters are provided in both the optical paths 210 and 211, but the heater may be provided only in the optical path 211, or the heater may be provided only in the optical path 210. That is, a heater is provided in at least one of the optical paths 210 and 211, and the heater is controlled so that the phase of the optical signal PS1 propagating in the optical path 211 is advanced by 0.25π than the optical signal PS1 propagating in the optical path 210. do it. Similarly, in the optical circuit 22, a heater is provided in at least one of the optical paths 220 and 221 so that the phase of the optical signal PS2 propagating through the optical path 221 is 0.25π more than that of the optical signal PS2 propagating through the optical path 220. The heater may be controlled so as to be delayed.

<A-2-2. Operation of Photoelectric Conversion Device 3>
In the photoelectric conversion device 3, the light receiving elements 4 and 5 convert the optical signals output from the optical circuits 21 and 22, respectively, into electrical signals. The light receiving element 4 includes two photodiodes 4a and 4b connected in series, and the two optical signals output from the optical circuit 21 are irradiated to the photodiodes 4a and 4b, respectively. From the light receiving element 4, the difference between the current generated by the photodiode 4a and the current generated by the photodiode 4b is output as a differential current signal.

  Similarly, the light receiving element 5 includes two photodiodes 5a and 5b connected in series, and the two optical signals output from the optical circuit 22 are irradiated to the photodiodes 5a and 5b, respectively. From the light receiving element 5, the difference between the current generated by the photodiode 5a and the current generated by the photodiode 5b is output as a differential current signal.

  As the photodiodes 4a, 4b, 5a, and 5b, for example, InP-based photodiodes are employed.

  The transimpedance amplifier 6 converts the differential current signal output from the light receiving element 4 into a voltage signal, and outputs the voltage signal as electrical signals ES1P and ES1N. Here, the electrical signal ES1P and the electrical signal ES1N are signals having opposite phases. When the envelope waveform of the optical signal PS in one symbol period is represented by A (t), the electric signal ES1P is represented by the following formula (1). Here, t represents time.

A ² (t) cos (Δφi + α1) (1)
As can be understood from the equation (1), the electrical signal ES1P depends on a value obtained by adding the phase difference Δφi between the optical signal PS and a signal obtained by delaying the optical signal PS by one symbol period and the phase shift amount α1. This is a signal whose intensity changes.

  When the difference with respect to + 0.25π of the phase shift amount α1 is Δ1, Equation (1) can be rewritten as the following Equation (2).

A ² (t) cos (Δφi + 0.25π + Δ1) (2)
The transimpedance amplifier 7 converts the differential current signal output from the light receiving element 5 into a voltage signal, and outputs the voltage signal as electrical signals ES2P and ES2N. Here, the electrical signal ES2P and the electrical signal ES2N are signals having opposite phases. The electrical signal ES2P is expressed by the following equation (3), similarly to the electrical signal ES1P.

A ² (t) cos (Δφi + α2) (3)
As can be understood from the equation (3), the electrical signal ES2P depends on the value obtained by adding the phase difference Δφi between the optical signal PS and the signal obtained by delaying the optical signal PS by one symbol period and the phase shift amount α2. This is a signal whose intensity changes.

  When the difference with respect to −0.25π of the phase shift amount α2 is Δ2, Equation (3) can be rewritten as the following Equation (4).

A ² (t) cos (Δφi−0.25π + Δ2) (4)

<A-2-3. Operation of control unit 50>
Next, the operation of the control unit 50 will be described. As described above, the control unit 50 controls the voltage supplied to the heaters 212 and 213, thereby changing the refractive index in the optical paths 210 and 211 and controlling the phase shift amount α1.

  In order to reduce the bit error of the transmission data reproduced in the data reproducing unit 60, the phase shift amount α1 is controlled to any one of the target values 0.25π, 0.75π, 1.25π, and 1.75π. In addition, the control unit 70 needs to control α2 to either the target value α1 + 0.5π or α1 + 1.5π.

  This is achieved by controlling Δ1 to be any of 0, 0.5π, π, and 1.5π and controlling Δ2 to be either Δ1 or Δ1 + π from the equations (1) to (4). Is equivalent to

  When the frequency of the low frequency signal LFS1 generated by the signal generator 52 is ω, the electrical signal ES1P output from the demodulator 1 can be expressed by the following equation (5) from the equation (2).

A ² (t) cos (Δφi + 0.25π + β1 · sin (ωt) + δ1) (5)
Here, δ1 represents the time average value of the difference Δ1 between the phase shift amount α1 and 0.25π. Hereinafter, δ1 is referred to as “average phase error δ1”. Β1 (> 0) represents the degree to which the phase shift amount α1 is modulated by the low frequency signal LFS1. Hereinafter, β1 is referred to as “modulation degree β1”. If the degree of modulation β1 is too large, the phase shift amount α1 will fluctuate greatly, and the bit error rate at the data reproducing unit 60 will increase. .

  Since the target values 0.25π, 0.75π, 1.25π, and 1.75π of the phase shift amount α1 correspond to the average phase error δ1 of 0, 0.5π, π, and 1.5π, respectively, the control unit 50 The phase shift amount α1 is controlled so that the average phase error δ1 becomes one of the above values.

<A-2-3-1. When δ1 = 0>
FIG. 3 is a graph showing the relationship between the phase θi1 and the amplitude of the electrical signal ES1P when δ1 = 0. In FIG. 3, A ² (t) = 1 is set for convenience of explanation. Here, the phase θi1 is expressed by the following equation (6).

θi1 = Δφi + 0.25π + β1 · sin (ωt) + δ1 (6)
In FIG. 3, the horizontal axis indicates the value obtained by dividing the phase θi1 by π, and the vertical axis indicates the magnitude of the electric signal ES1P. The four white circles in FIG. 3 indicate the relationship between the phase θi1 and the magnitude of the electrical signal ES1P when t = 0, and the four white circles (white circle 101, white circle 102, white circle 103, white circle 104) are From left to right, the value when white circle 101 is Δφi = 0π, the value when white circle 102 is Δφi = 0.5π, the value when white circle 103 is Δφi = π, and the value when white circle 104 is Δφi = 1.5π. The value is shown. Also, the graph 100 in FIG. 3 shows how the phase θi1 at the white circle changes when the time t changes. In the graph 100, the time t becomes lower (to the right in the drawing). Has passed.

  As indicated by white circle 101, white circle 102, white circle 103, and white circle 104 in FIG. 3, at t = 0, θi1 = 0.25π, 0.75π, 1.25π, and 1.75π. The magnitude of the electrical signal ES1P at θi1 = 0.25π is the same as the magnitude of the electrical signal ES1P at θi1 = 1.75π, and the magnitude of the electrical signal ES1P at θi1 = 0.75π, The magnitude of the electric signal ES1P at θi1 = 1.25π is the same value. The electrical signal ES1P at θi1 = 0.25π, 1.75π is in an antiphase relationship with the electrical signal ES1P at θi1 = 0.75π, 1.25π.

  As shown in FIG. 3, the phase θi1 increases from the initial state when 0 <t <π / ω and then decreases, and returns to the initial state when t = π / ω. The phase θi1 increases after decreasing from the initial state when π / ω <t <2π / ω, and returns to the initial state at t = 2π / ω.

  The magnitude of the electrical signal ES1P changes according to such a change in the phase θi1. As shown in FIG. 3, when 0 <t <π / ω, the magnitude of the electrical signal ES1P at Δφi = 0 is smaller than that at t = 0, and the magnitude of the electrical signal ES1P at Δφi = 1.5π. Increases from t = 0. Therefore, when 0 <t <π / ω, the magnitude of the electrical signal ES1P is larger in the case of Δφi = 1.5π than in the case of Δφi = 0. Since the phase θi1 returns to the initial state at t = π / ω, the magnitude of the electric signal ES1P is the same between Δφi = 1.5π and Δφi = 0. When π / ω <t <2π / ω, the magnitude of the electrical signal ES1P at Δφi = 0 is greater than t = 0, and the magnitude of the electrical signal ES1P at Δφi = 1.5π is greater than t = 0. Decrease. Therefore, when π / ω <t <2π / ω, the magnitude of the electrical signal ES1P is larger in the case of Δφi = 0 than in the case of Δφi = 1.5π.

  From the above, in the case of δ1 = 0, the magnitude of the electrical signal ES1P at Δφi = 1.5π is the largest when 0 <t <π / ω, and Δφi = 0 at π / ω <t <2π / ω. The magnitude of the electrical signal ES1P at the maximum is the largest. Therefore, when the envelope detector 51 detects the envelope of the electric signal ES1P with a time constant sufficiently longer than the symbol period of the optical signal PS and shorter than the period of the low frequency signal LFS1, 0 <t <π / ω. Then, the same result as when detecting the envelope of the electric signal ES1P where Δφi = 1.5π is always obtained. When π / ω <t <2π / ω, the envelope of the electric signal ES1P where Δφi = 0 is always detected. The same result as when you do.

  Here, considering the waveform as shown in FIG. 4 as the low frequency signal LFS1, the envelope of the electric signal ES1P detected by the envelope detector 51 is as shown in FIG. Here, the horizontal axis in FIGS. 4 and 5 indicates time t [s] normalized by the symbol period, and the vertical axis in FIG. 4 indicates the low-frequency signal LFS1 normalized so that the maximum value is 1. The vertical axis of FIG. 5 indicates the magnitude of the detection signal DS1 output from the envelope detector 51.

The period of the low-frequency signal LFS1 shown in FIG. 4 is set to 20 × 10 ⁷ times one symbol period. For example, if one symbol period is 50 ps, the period and frequency of the low-frequency signal LFS1 are 10 ms and 100 Hz, respectively. When the low frequency signal LFS1 is as shown in FIG. 4, the time constant in the envelope detector 51 is set to a value close to 10 ms.

From the above, when δ1 = 0, the magnitude of the detection signal DS1 output from the envelope detector 51 is 0 <t <π / ω.
cos (1.75π + β1 · sin (ωt)) (7)
And π / ω <t <2π / ω,
cos (0.25π + β1 · sin (ωt)) (8)
It can be said that it is proportional to.

  The detection signal DS1 has a periodicity of a period π / ω. Therefore, when δ1 = 0, the average value of the output signal of the multiplier 530 is

Therefore, it becomes zero.

<A-2-3-2. When β1 <δ1 <0.25π>
FIG. 6 is a graph showing the relationship between the phase θi1 and the magnitude of the electrical signal ES1P when β1 <δ1 <0.25π, similarly to FIG. As shown in FIG. 6, when β1 <δ1 <0.25π, the magnitude of the electric signal ES1P (white circle 104) at Δφi = 1.5π is always the largest. Here, since β1 is set to a very small value, when β1 is considered as 0, when 0 <δ1 <0.25π, the electric signal ES1P (white circle 104) always with Δφi = 1.5π. It can be said that the size of is the largest. In this case, considering the waveform as shown in FIG. 4 as the low frequency signal LFS1, the envelope of the electric signal ES1P detected by the envelope detector 51 is as shown in FIG. Here, the horizontal axis in FIG. 7 indicates time t [s] normalized by the symbol period, and the vertical axis indicates the magnitude of the detection signal DS1 output from the envelope detector 51.

  From the above, when 0 <δ1 <0.25π, the detection signal DS1 can be expressed as cos (β1 · sin (ωt) + 1.75π + δ1). When this equation is Taylor-expanded to the first order term with respect to β1, the detection signal D1 is expressed by the following equation (11).

cos (β1 · sin (ωt) + 1.75π + δ1)
→ cos (1.75π + δ1) −sin (1.75π + δ1) β1 · sin (ωt) (11)
Therefore, the output signal of multiplier 530 is expressed by the following equation (12).

Bcos (1.75π + β1 · sin (ωt) + δ1) sin (ωt)
→ B (−β1 · sin (1.75π + δ1) / 2 + cos (1.75π + δ1) sin (ωt) + β1 · sin (1.75π + δ1) cos (2ωt) / 2) (12)
When 0 <δ1 <0.25π, the average value calculation unit 531 calculates the average value of the output signal of the multiplier 530 represented by Expression (12). Since B> 0 and β1> 0, from equation (12), when 0 <δ1 <0.25π, the average value of the output signal of the multiplier 530 is positive.

<A-2-3-3. −0.25π <δ1 <-β1>
FIG. 8 is a graph showing the relationship between the phase θi1 and the magnitude of the electric signal ES1P when −0.25π <δ1 <−β1 as in FIGS. As shown in FIG. 8, in the case of −0.25π <δ1 <−β1, the magnitude of the electric signal ES1P (white circle 101) at Δφi = 0 is always the largest. Here, since β1 is set to a very small value, when β1 is considered as 0, when −0.25π <δ1 <0, the electric signal ES1P (white circle 101) of Δφi = 0 is always obtained. It can be said that the size is the largest. In this case, considering the waveform as shown in FIG. 4 as the low frequency signal LFS1, the envelope of the electric signal ES1P detected by the envelope detector 51 is as shown in FIG.

  From the above, when −0.25π <δ1 <0, the detection signal DS1 can be expressed as cos (β1 · sin (ωt) + 0.25π + δ1). When the output signal of the multiplier 530 is obtained in the same manner as in the case of 0 <δ1 <0.25π, the output signal is expressed by the following equation (13).

Bcos (0.25π + β1 · sin (ωt) + δ1) sin (ωt)
→ B (−β1 · sin (0.25π + δ1) / 2 + cos (0.25π + δ1) sin (ωt) + β1 · sin (0.25π + δ1) cos (2ωt) / 2) (13)
In the case of −0.25π <δ1 <0, the average value calculation unit 531 calculates the average value of the output signal of the multiplier 530 represented by Expression (13). Since B> 0 and β1> 0, from Equation (13), when −0.25π <δ1 <0, the average value of the output signal of the multiplier 530 is negative.

  The determination unit 532 determines the control voltage VT1 for the heater 212 based on the average value calculated by the average value calculation unit 531. For example, when performing PID control, the control voltage VT1 is set to a linear sum of the average value calculated by the average value calculation unit 531, the differential value of the average value, and the integral value of the average value. The manipulated variable determination unit 53 is controlled so that the average phase error δ1 converges to a desired value by continuing to execute the above processing for each cycle of the low frequency signal LFS1.

  FIG. 10 is a graph showing the relationship between the amplitude of the detection signal DS1 and the average phase error δ1. Again, the average phase error δ1 of 0, 0.5π, π, and 1.5π is a normal operating point. This period is 0.5π, and has a maximum value when the phase is 0.25π, 0.75π, 1.25π, and 1.75π, and when the phase is 0, 0.5π, π, and 1.5π. Has a local minimum. Since the period is 0.5π, the average phase errors δ1 of 0, 0.5π, π, and 1.5π are all equal and cannot be distinguished. Therefore, when PID control is performed, the phase converges to the phase closest to the initial condition. That is, whether the average phase error δ1 converges to 0, 0.5π, π, or 1.5π depends on the initial phase.

<A-2-4. Operation of control unit 70>
Next, the operation of the control unit 70 will be described. As described above, the control unit 70 controls the voltage supplied to the heaters 222 and 223, thereby changing the refractive index in the optical paths 220 and 221 and controlling the phase shift amount α2. The control by the controller 70 starts after the average phase error δ1 has converged. Here, since the same result is obtained when the average phase error δ1 converges to 0, 0.5π, π, or 1.5π, the case where the average phase error δ1 converges to 0 will be described.

First, the adder 75 outputs the result of adding the electrical signal ES1P output from the demodulator 1 and the electrical signal ES2N. Assuming that the frequency of the low-frequency signal LFS2 generated by the signal generator 72 is ω, the electrical signal ES2N output from the demodulator 1 is similar to the equation (5) related to the electrical signal ES1P.
−A ² (t) cos (Δφi−0.25π + β2 · sin (ωt) + δ2) (14)
It is expressed. Here, the average phase error δ2 represents the time average value of the difference between the phase shift amount α2 and 0.25π. The modulation degree β2 (> 0) represents the degree to which the phase shift amount α2 is modulated by the low frequency signal LFS2. Here, for simplification, β2 = β1 = β. Therefore, the output signal of the adder 75 is
A ² (t) {cos (Δφi + 0.25π + β · sin (ωt)) − cos (Δφi−0.25π + β · sin (ωt) + δ2)} (15)
It is expressed. FIG. 11 shows the relationship between the amplitude of the detection signal DS2b output from the envelope detector 71b and the average phase error δ2. When δ2 = 0.5π, the amplitude of the detection signal DS2b has a minimum value, and when δ2 = 1.25π and 1.75π, the amplitude of the detection signal DS2b has a maximum value.

  On the other hand, the envelope detector 71a detects the envelope of the electric signal ES2N and outputs a detection signal DS2a. FIG. 12 is a graph showing the relationship between the detection signal DS2a and the average phase error δ2. Similar to the detection signal DS1 in the control unit 50, it has a periodicity of 0.5π, and the detection signal DS2a is minimal when the average phase error δ2 is 0, 0.5π, π, and 1.5π. . However, unlike the detection signal DS1, when the average phase error δ1 converges to 0, the average phase error δ2 of 0 and π is a normal operating point, but the average phase error δ2 of 0.5π and 1.5π is false. This is the operating point.

  Next, the detection signal DS2a and the low frequency signal LFS2 are multiplied by the multiplier 730a, and the average signal A2a is calculated by the average value calculator 731a. FIG. 13 is a graph showing the relationship between the average signal A2a and the average phase error δ2. Similar to the detection signal DS2a, the average signal A2a has a periodicity of 0.5π, and becomes 0 when the average phase error δ2 is 0, 0.5π, π, and 1.5π.

  Further, since the average phase error δ2 depends on the wavelength, it is difficult to calculate what value the average phase error δ2 takes when the control voltage VT2 is given. The delay interferometer 2 can be manufactured by controlling the rate of change with respect to the control voltage VT2 with high accuracy. This is because the rate of change of the average phase error δ2 with respect to the control voltage VT2 is determined by the resistance value of the heater 222 and the temperature dependence of the refractive index of the optical paths 210, 211, 220, 221 and these can be controlled with high accuracy. Because. In general, since the average phase error δ2 has a characteristic proportional to the power of the control voltage VT2, the average phase error δ2 is a quadratic function of the control voltage VT2. Therefore, if the average phase error δ2 at a certain control voltage VT2 can be calculated, the average phase error δ2 at other control voltages VT2 can be calculated with high accuracy.

  Using these characteristics, convergence is made to a normal operating point having an average phase error δ2. FIG. 14 shows an example of temporal changes in the average signal A2a, the detection signal DS2a, and the average phase error δ2 from the start of control until the average phase error δ2 is drawn to a normal operating point and the control is stabilized. The determining unit 732 causes the sweep unit 733 to sweep the voltage by an amount by which the average phase error δ2 changes by 2π or more, thereby obtaining the control voltage VT2 that minimizes the amplitude of the detection signal DS2b. Since this corresponds to δ2 = 0.5π, a control voltage VT2 corresponding to an average phase error δ2 of 0 or π is calculated from this result. Finally, the operation amount for the heater 222 corresponding to the average phase error δ2 of 0 or π, that is, the control voltage VT2 supplied to the heater 222 is set to its initial value. Here, the sweep is not limited to monotonous change.

  Thereafter, the voltage sweep by the sweep unit 733 is stopped, the output of the sweep unit 733 is set to 0, and then the same control as the control unit 50 is performed. That is, the initial value of the control voltage VT 2 and the low frequency signal LFS 2 are added by the adder 74 and applied to the heaters 222 and 223. If the average signal A2a is positive, the average phase error δ2 is deviated positively. Therefore, the control voltage VT2 for the heaters 222, 223 is increased so as to be zero, and conversely if the average signal A2a is negative. Since the average phase error δ2 is negatively shifted, the control voltage VT2 for the heaters 222 and 223 is reduced so that the average phase error δ2 becomes zero. The manipulated variable determining unit 73 is controlled so that the average phase error δ2 converges to a desired value by continuing to execute the above processing for each cycle of the low frequency signal LFS2. Here, since the initial value of the control voltage VT2 is set to an operation amount corresponding to the average phase error δ2 of 0 or π, the average phase error δ2 does not converge to 0.5π or 1.5π. As described above, the phase shift amount α2 given to the optical signal PS2 propagating through the optical path 221 is controlled so as to coincide with α1 + 0.5π or α1 + 1.5π which is a target value.

  Further, for example, not only the amplitude of the detection signal DS2b but also the average signal A2a may be measured at the time of sweeping by the sweep unit 733. That is, only when the average signal A2a becomes 0, the detection signal DS2b is measured, and the control voltage VT2 that minimizes the detection signal DS2b is calculated. Similar to the above example, based on this result, the control voltage VT2 corresponding to the average phase error δ2 of 0 or π is set to the initial value, and automatic control is performed. Alternatively, the detection signal DS2b may be measured only when the average signal A2a changes from negative to positive, except when the average signal A2a changes from positive to negative, when the average signal A2a becomes zero.

  Further, a mechanism may be added in which not only the minimum value of the detection signal DS2b but also the maximum value is measured, thereby increasing the number of measurement points and checking whether the calculation result is valid.

  In the first embodiment, the example in which the electrical signal ES1P and the electrical signal ES2N are added is shown. However, either the electrical signal ES1P or the electrical signal ES1N may be used, and either the electrical signal ES2P or the electrical signal ES2N is used. It may be used. For example, when the electrical signal ES1P and the electrical signal ES2P are added, the amplitude of the detection signal DS2b becomes 0 when the average phase error δ2 is 1.5π. As described above, the dependency of the amplitude of the detection signal DS2b on the average phase error δ2 is shifted by π. However, if this is noted, the average phase error δ2 is from the control voltage VT2 corresponding to 0.5π and 1.25π. A control voltage VT2 corresponding to an average phase error δ2 of 0 or π can be obtained. As a result, the phase shift amount α2 given to the optical signal PS2 propagating through the optical path 221 is controlled to coincide with α1 + 0.5π or α1 + 1.5π, which is the target value.

  Further, the period of the low frequency signal LFS1 is a change in the symbol period in the optical signal PS and the phase shift amount α1 in the optical circuit 21 so that the detection signal DS1 has a waveform as shown in FIGS. Must be longer than the response time. When the phase shift amount α1 is changed using the thermo-optic effect as in the first embodiment, the response time of the change is generally about 1 ms. Similarly, the period of the low-frequency signal LFS2 must be longer than the symbol period in the optical signal PS and the response time of the change in the phase shift amount α2 in the optical circuit 22.

  Further, the method of giving the phase shift to the optical signal is not limited to the method using the thermo-optic effect, but may be a method using the electro-optic effect or a method of adjusting the optical path length mechanically.

<A-3. Effect>
According to the first embodiment of the present invention, in the optical receiver, the delay interferometer 2 including the optical circuits 21 and 22 that are the first and second optical circuits belonging to the first and second series, and the optical circuit 21. , 22 are respectively received by the first and second twin photodiodes 31 and 32, and the differentially modulated optical signals PS1 and PS2 are separately used by using the first and second series. The phase of the optical signal PS1 in the optical circuit 21 is controlled using the demodulator 1 that outputs the electrical signals ES1 and ES2 that are the first and second demodulated signals that are deviated from each other by a predetermined phase and the electrical signal ES1. The optical circuit 22 using the control unit 50 that is the first control unit, the adder 75 that is the calculation unit that calculates the electric signals ES1 and ES2 and outputs the calculation signal, and the electric signal ES2 and the calculation signal. Signal PS Since the phase difference between the first sequence and the second sequence can be determined by providing the control unit 70 that is the second control unit that controls the phase state of the first phase, the configuration for phase difference control is simplified and normal Communication can be performed at high speed.

  In addition, according to the first embodiment of the present invention, in the optical receiver, the control unit 70 as the second control unit is configured so that the electrical signal ES1 and the electrical signal ES2 are shifted by 0 or π phase. By controlling the phase state of the optical signal PS2 at, it can be controlled to converge to a normal operating point.

  Further, according to the first embodiment of the present invention, in the optical receiver, the adder 75 as the calculation unit adds the electric signal ES1 and the electric signal ES2 and outputs the calculation signal, thereby calculating the calculation signal. Can be used to determine the normal operating point and control the phase difference.

  Further, according to the first embodiment of the present invention, in the optical receiver, the control unit 50 that is the first control unit detects the electrical signal ES1 and outputs the detection signal DS that is the first detection signal. An envelope detector 51 which is one detector, an average value calculating unit 531 which is a first average value calculating unit which calculates an average value of the detection signal DS and outputs it as an average signal A1 which is a first average value; Based on the signal A 1, a determination unit 532 that is a first determination unit that determines a control output to the optical circuit 21 is provided, thereby determining a control voltage to the heaters 212 and 213, and an optical signal PS 1 in the optical circuit 21. The phase state can be controlled.

  Further, according to the first embodiment of the present invention, in the optical receiver, the control unit 70 as the second control unit detects the electrical signal ES2 and outputs the detection signal DS2a as the second detection signal. An envelope detector 71a which is a two-detector, an average value calculating unit 731a which is a second average value calculating unit which calculates an average value of the detection signal DS2a and outputs it as an average signal A2a which is a second average value; Based on the signal A2a, a determination unit 732 that is a second determination unit that determines the control output to the optical circuit 22 is provided, so that the control voltage to the heaters 222 and 223 is determined and the optical signal PS2 in the optical circuit 22 is determined. The phase state can be controlled.

  Also, according to the first embodiment of the present invention, in the optical receiver, the control unit 70 detects an arithmetic signal and outputs an envelope signal that is a third detector that outputs a detection signal DS2b that is a third detection signal. A detector 71b is further provided, and the determination unit 732 determines a normal operating point using the detection signal DS2b by determining a control output to the optical circuit 22 based also on the detection signal DS2b, and determines a phase difference. Can be controlled.

  Also, according to the first embodiment of the present invention, in the optical receiver, the control unit 70 is the detection signal DS2b or the third average value before controlling the phase state of the optical signal PS2 in the optical circuit 22. By further including a sweep unit 733 that sweeps the average signal A2b, the voltage is swept by the amount by which the average phase error δ2 changes by 2π or more, and the control voltage VT2 that minimizes the amplitude of the detection signal DS2b can be obtained.

<B. Second Embodiment>
<B-1. Configuration>
FIG. 15 is a diagram showing a configuration of an optical receiving apparatus according to Embodiment 2 of the present invention. In the second embodiment, a method for determining control voltages VT1 and VT2 by a method different from that of the first embodiment will be described.

  Hereinafter, the optical receiving apparatus according to the second embodiment will be described with a focus on the control unit 700 that is different from the optical receiving apparatus according to the first embodiment. Since the components other than the control unit 700 are the same as those in the first embodiment, detailed description thereof is omitted.

  The control unit 700 generates an envelope detector 71a that detects an electrical signal ES2N that is a demodulated signal, an envelope detector 71b that detects a signal (calculation signal) added by the adder 75, and a low-frequency signal LFS2. From the signal generator 72, the operation amount determination unit 800 to which the detection signals DS2a and DS2b output from the envelope detectors 71a and 71b and the low frequency signal LFS2 are input, the low frequency signal LFS2, and the operation amount determination unit 73 Are added, and an adder 74 that adds the output of the adder 76 and the control voltage VT2 from the operation amount determination unit 800 is provided.

  The operation amount determination unit 800 includes a multiplier 730a that multiplies the low-frequency signal LFS2 and the detection signal DS2a, a multiplier 730b that multiplies the low-frequency signal LFS2 and the detection signal DS2b (third detection signal), and a multiplier 730a. An average value calculation unit 731a that calculates the average value of the output signals from, and an average value calculation unit 731b that is a third average value calculation unit that calculates the average value of the output signal (third detection signal) from the multiplier 730b; The control voltage VT2 and its initial value are calculated using the average signal A2a, which is the second average value output from the average value calculation unit 731a, and the average signal A2b, which is the third average value output from the average value calculation unit 731b. The determination unit 732 is a second determination unit that determines the sweep voltage, and the sweep unit 733 to which the sweep voltage is input. The control voltage VT2 is an operation amount for the heaters 222 and 223 of the demodulator 1, that is, a voltage supplied to the heaters 222 and 223. The average value calculator 731a calculates the average value of the output signal of the multiplier 730a, and thereby performs synchronous detection.

  The envelope detectors 71a and 71b are configured by, for example, a half-wave rectifier circuit or a full-wave rectifier circuit using a Schottky barrier diode.

  The average value calculation units 731a and 731b are configured by a low-pass filter, an arithmetic circuit that performs integration calculation or Fourier transform calculation executed in the cycle of the low-frequency signal LFS2, and calculates a DC component.

  The sweep is not limited to monotonous change. For example, synchronous detection is performed by giving an initial value every 0.25π, and the average value signal is controlled to be zero. In order to converge to a convergence point close to the initial value, the average phase error δ2 can be found to be 0.5π, 1.25π, 1.5π, 1.75π by changing the average phase error δ2 by 2π or more. Moreover, it is not restricted to this.

<B-2. Operation>
Next, the operation of the control unit 700 will be described. In order to reduce the bit error of the transmission data reproduced in the data reproducing unit 60, it is necessary to control the phase shift amounts α1 and α2 to be the target values.

  In the optical receiving apparatus according to the second embodiment, the target value of Δ1 is 0, 0.5π, π, or 1.5π, and the desired value of Δ2 is either Δ1 or Δ1 + π. Features. First, according to the method described in the first embodiment, the control unit 50 is operated to converge the average phase error δ1 to any desired 0, 0.5π, π, or 1.5π. Here, a case where the average phase error δ1 converges to 0 will be described.

  Thereafter, control in the control unit 700 is started. First, the adder 75 outputs the result of adding the electrical signal ES1P output from the demodulator 1 and the electrical signal ES2N.

  FIG. 16 shows the relationship between the average value signal by the average value calculation unit 731b and the average phase error δ2. From FIG. 16, the determining unit 732 causes the sweep unit 733 to sweep the voltage in a direction in which the average phase error δ2 increases by 2π or more, so that the average phase error δ2 is 0.5π, 1.25π, 1.5π. It can be seen that the average value signal becomes 0 at 1.75π. When the average phase error δ2 is 0.5π and 1.5π, the average value increases as the average phase error δ2 increases. When the average phase error δ2 is 1.25π and 1.75π, The average value decreases. Next, using this characteristic, a control voltage VT2 corresponding to the average phase error δ2 of 0.5π, 1.25π, 1.5π, and 1.75π is obtained. From these results, a control voltage VT2 corresponding to an average phase error δ2 of 0 or π is obtained. Finally, the control voltage VT2 corresponding to 0 or π average phase error δ2 is set to its initial value.

  Thereafter, the voltage sweep by the sweep unit 733 is stopped, and the same control as in the first embodiment is executed. That is, the initial value of the control voltage VT2 and the low frequency signal LFS2 are added by the adder 74, and the control voltage VT2 is applied to the heater 222.

  The envelope detector 71a detects the envelope of the electric signal ES2N, and multiplies the low frequency signal LFS2 by the multiplier 730a. The average value of the multiplication results is calculated by the average value calculation unit 731a, and the control voltage VT2 for the heaters 222 and 223 is determined.

  The manipulated variable determination unit 800 is controlled so that the average phase error δ2 converges to a desired value by continuing to execute the above processing for each cycle of the low frequency signal LFS2.

<B-3. Effect>
According to the second embodiment of the present invention, in the optical receiver, the control unit 700 that is the second control unit calculates the average value of the detection signal DS2b that is the third detection signal, and is the third average value. An average value calculation unit 731b that is a third average value calculation unit that outputs the average signal A2b is further included, and the determination unit 732 that is the second determination unit outputs a control output to the optical circuit 22 based also on the average signal A2b. By determining, it is possible to determine a normal operating point using the average signal A2b and to control the phase difference.

<C. Embodiment 3>
<C-1. Configuration>
FIG. 17 is a diagram showing the configuration of the optical receiving apparatus according to Embodiment 3 of the present invention. In the third embodiment, a method for determining control voltages VT1 and VT2 by a method different from that of the first embodiment will be described.

  Hereinafter, the optical receiver according to the third embodiment will be described focusing on the subtractor 77 that is different from the optical receiver according to the first embodiment. The subtractor 77 subtracts the electrical signals ES2P and ES1P which are demodulated signals. Since the components other than the subtractor 77 are the same as those in the first embodiment, detailed description thereof is omitted.

<C-2. Operation>
Next, the operation of the control unit 70 will be described. In order to reduce the bit error of the transmission data reproduced in the data reproducing unit 60, it is necessary to control the phase shift amounts α1 and α2 to be the target values.

  In the optical receiving apparatus according to the third embodiment, the desired value of Δ1 is 0, 0.5π, π, or 1.5π, and the desired value of Δ2 is either Δ1 or Δ1 + π. It is characterized by. First, according to the method described in the first embodiment, the control unit 50 is operated to converge the average phase error δ1 to any desired 0, 0.5π, π, or 1.5π. Here, a case where the average phase error δ1 converges to 0 will be described.

  Thereafter, control in the control unit 70 is started. First, since the subtractor 77 outputs a result obtained by subtracting the electrical signal ES1P and the electrical signal ES2P output from the demodulator 1, the output signal of the subtractor 77 conforms to Expression (15) in the first embodiment.

  Thereafter, in the same manner as in the first embodiment, the sweep unit 733 sweeps the voltage by an amount by which the average phase error δ2 changes by 2π or more, whereby the control voltage VT2 corresponding to the average phase error δ2 of 0.5π and 1.25π. Ask for. Next, a control voltage VT2 corresponding to an average phase error δ2 of 0 or π is obtained. Finally, the control voltage VT2 corresponding to 0 or π average phase error δ2 is set to its initial value.

  Thereafter, similarly to the first embodiment, the voltage sweep by the sweep unit 733 is stopped, and the average phase error δ2 is controlled to converge to a desired value by PID control. As described above, the phase shift amount α2 given to the optical signal PS2 propagating through the optical path 221 is controlled so as to coincide with α1 + 0.5π or α1 + 1.5π which is a target value.

  In addition, although the example which subtracts the electric signal ES1P and the electric signal ES2P was shown, either the electric signal ES1P or the electric signal ES1N may be used, and either the electric signal ES2P or the electric signal ES2N may be used. For example, when the electric signal ES1N and the electric signal ES2P are subtracted, the amplitude of the detection signal DS2 becomes 0 when the average phase error δ2 is 1.5π. Thus, the dependency of the amplitude of the detection signal DS2 on the average phase error δ2 is shifted by π. However, if this is noted, the average phase error δ2 is from the control voltage VT2 corresponding to 0.5π and 1.25π. A control voltage VT2 corresponding to an average phase error δ2 of 0 or π can be obtained. As described above, the phase shift amount α2 given to the optical signal PS2 propagating through the optical path 221 is controlled to coincide with the target value.

<C-3. Effect>
According to the third embodiment of the present invention, in the optical receiver, the subtractor 77 that is a calculation unit subtracts the electric signal ES1P and the electric signal ES2P and outputs the calculation signal, thereby using the calculation signal. Thus, the normal operating point can be determined and the phase difference can be controlled.

<D. Embodiment 4>
<D-1. Configuration>
FIG. 18 is a diagram showing a configuration of an optical receiving apparatus according to Embodiment 4 of the present invention. In the fourth embodiment, a method for determining control voltages VT1 and VT2 by a method different from that of the first embodiment will be described.

  Hereinafter, the optical receiving apparatus according to the fourth embodiment will be described with a focus on the multiplier 78 that is different from the optical receiving apparatus according to the first embodiment. The multiplier 78 multiplies the electric signals ES2N and ES1P which are demodulated signals. Since the components other than the multiplier 78 are the same as those in the first embodiment, detailed description thereof is omitted.

<D-2. Operation>
Next, the operation of the control unit 70 will be described. In order to reduce the bit error of the transmission data reproduced in the data reproducing unit 60, it is necessary to control the phase shift amounts α1 and α2 to be the target values.

  In the optical receiving apparatus according to the fourth embodiment, the desired value of Δ1 is 0, 0.5π, π, or 1.5π, and the desired value of Δ2 is either Δ1 or Δ1 + π. It is characterized by. First, according to the method described in the first embodiment, the control unit 50 is operated to converge the average phase error δ1 to any desired 0, 0.5π, π, or 1.5π. Here, a case where the average phase error δ1 converges to 0 will be described.

Thereafter, control in the control unit 70 is started. The output signal of the multiplier 78 is
−A ² (t) cos (Δφi + 0.25π + β · sin (ωt)) × cos (Δφi−0.25π + β · sin (ωt) + δ2) (16)
It is expressed. FIG. 19 shows the relationship between the amplitude of the detection signal DS2b output from the envelope detector 71b and the average phase error δ2. When δ2 = 1.5π, the amplitude of the detection signal DS2b has a minimum value, and when δ2 = 0.25π and 0.75π, the amplitude of the detection signal DS2b has a maximum value. Therefore, the determination unit 732 causes the sweep unit 733 to sweep the voltage by an amount that the average phase error δ2 changes by 2π or more, whereby the control voltage VT2 corresponding to the average phase error δ2 of 0.25π, 0.75π, and 1.5π. Ask for. Next, from these results, a control voltage VT2 corresponding to an average phase error δ2 of 0 or π is obtained. Finally, the control voltage VT2 for the heater 222 corresponding to the average phase error δ2 of 0 or π is set to its initial value.

  Thereafter, similarly to the first embodiment, the voltage sweeping by the sweep unit 733 is stopped, the initial value of the control voltage VT2 and the low frequency signal LFS2 are added by the adder 74 and applied to the heaters 222 and 223. The envelope detector 71a detects the envelope of the electric signal ES2N, and multiplies the low frequency signal LFS2 by the multiplier 730a. The average value of the multiplication results is calculated by the average value calculation unit 731a, and the control voltage VT2 for the heaters 222 and 223 is determined. The manipulated variable determining unit 73 is controlled so that the average phase error δ2 converges to a desired value by continuing to execute the above processing for each cycle of the low frequency signal LFS2.

  Here, since the initial value of the control voltage VT2 is set to an operation amount corresponding to an average phase error δ2 of 0 or π, the average phase error δ2 does not converge to 0.5π or 1.5π. As described above, the phase shift amount α2 given to the optical signal PS2 propagating through the optical path 221 is controlled so as to coincide with α1 + 0.5π or α1 + 1.5π which is a target value.

  In addition, although the example which multiplies the electric signal ES1P and the electric signal ES2N was shown, either the electric signal ES1P or the electric signal ES1N may be used, and either the electric signal ES2P or the electric signal ES2N may be used.

<D-3. Effect>
According to the fourth embodiment of the present invention, in the optical receiver, the multiplier 78 that is a calculation unit multiplies the electric signal ES1 and the electric signal ES2 and outputs the calculation signal, thereby using the calculation signal. Thus, the normal operating point can be determined and the phase difference can be controlled.

<E. Embodiment 5>
<E-1. Configuration>
FIG. 20 is a diagram showing the configuration of the optical receiving apparatus according to the fifth embodiment of the present invention. In the fifth embodiment, a method of determining control voltages VT1 and VT2 by a method different from that of the first embodiment will be described.

  Hereinafter, the optical receiver according to the fifth embodiment will be described with a focus on the multiplier 78 and the control unit 703 that are different from the optical receiver according to the first embodiment. Since the components other than the multiplier 78 and the control unit 703 are the same as those in the first embodiment, detailed description thereof is omitted.

  The control unit 703 that is the second control unit includes an operation amount determination unit 803 to which a signal (multiplication signal) multiplied by the multiplier 78 that is a multiplication unit is input. The operation amount determination unit 803 is an average value calculation unit 731b that is a third average value calculation unit that calculates an average value (third average value) of the multiplication signals, and a third average value that is output from the average value calculation unit 731b. A determination unit 732 that is a second determination unit that determines the control voltage VT2, its initial value, and sweep voltage using a certain average signal A2b. The control voltage VT2 is an operation amount for the heaters 222 and 223 of the demodulator 1, that is, a voltage supplied to the heaters 222 and 223.

<E-2. Operation>
Next, the operation of the control unit 703 will be described. In order to reduce the bit error of the transmission data reproduced in the data reproducing unit 60, it is necessary to control the phase shift amounts α1 and α2 to be the target values.

  The invention in Embodiment 5 is characterized in that the desired value of Δ1 is any one of 0, 0.5π, π, and 1.5π, and the desired value of Δ2 is any of Δ1 and Δ1 + π. And

First, according to the method described in the first embodiment, the control unit 50 is operated to converge the average phase error δ1 to any desired 0, 0.5π, π, or 1.5π. Here, a case where the average phase error δ1 converges to 0 will be described. Thereafter, control in the control unit 703 is started. The output signal of the multiplier 78 is
-A ² (t) cos (Δφi + 0.25π + β · sin (ωt)) × cos (Δφi−0.25π + δ2)
= −A ² (t) / 2 {cos (Δφi + δ1 + δ2) + sin (δ2)} (17)
It is expressed. FIG. 21 shows the relationship between the average value output by the average value calculator 731b and the average phase error δ2. When δ2 = 0 or π, the average value is 0. Therefore, the operation amount determination unit 73 is controlled so that the average phase error δ2 converges to a desired value by continuously determining the control voltage VT2 for the heater 222 so that the average value becomes zero.

  For example, if the average value is positive by PID control, the average phase error δ2 is decreased, and if the average value is negative, if the average phase error δ2 is increased, the average phase error δ2 converges to zero. As described above, the phase shift amount α2 given to the optical signal PS2 propagating through the optical path 221 is controlled so as to coincide with α1 + 0.5π or α1 + 1.5π which is a target value.

  In addition, although the example which multiplies the electric signal ES1P and the electric signal ES2N was shown, either the electric signal ES1P or the electric signal ES1N may be used, and either the electric signal ES2P or the electric signal ES2N may be used.

<E-3. Effect>
According to the fifth embodiment of the present invention, in the optical receiver, the delay interferometer 2 including the first and second optical circuits 21 and 22 belonging to the first and second series, and the first and second optical circuits. Twin photodiodes 31 and 32 that are first and second twin photodiodes respectively receiving the output lights 21 and 22, and differentially modulated optical signals PS1 and PS2 using the first and second series. The phase state of the optical signal PS1 in the optical circuit 21 is determined by using the demodulator 1 that outputs the electrical signals ES1 and ES2 that are first and second demodulated signals that are separately demodulated and shifted from each other by a predetermined phase, and the electrical signal ES1. The phase state of the optical signal PS2 in the optical circuit 22 is controlled using the control unit 50, which is the first control unit to be controlled, the multiplier 78 that multiplies the electrical signals ES1 and ES2, and outputs the multiplication signal, and the multiplication signal. Second control unit By providing a certain control unit 703, the phase difference between the first sequence and the second sequence can be discriminated, so that the configuration for phase difference control can be simplified and normal communication can be performed at high speed. .

  Further, according to the fifth embodiment of the present invention, in the optical receiver, the control unit 703 calculates the average value of the multiplication signals and outputs the third average value that is output as the average signal A2b that is the third average value. The average value calculation unit 731b that is a unit and the determination unit 732 that is a second determination unit that determines the control output to the optical circuit 22 based on the average signal A2b, thereby using the average signal A2b. The operating point can be determined and the phase difference can be controlled.

<Modification 1>
FIG. 22 is a diagram showing the configuration of the optical receiving apparatus according to the first modification of the present invention. Hereinafter, description will be made centering on demodulator 1000 that is different from the optical receiver according to the first embodiment. Since the components other than the demodulator 1000 are the same as those in the first embodiment, detailed description thereof is omitted.

  In the first embodiment, the light receiving element 4 includes two photodiodes 4a and 4b connected in series, and the two optical signals output from the optical circuit 21 are irradiated to the photodiodes 4a and 4b, respectively. Is done. From the light receiving element 4, the difference between the current generated by the photodiode 4a and the current generated by the photodiode 4b is output as a differential current signal. Similarly, the light receiving element 5 includes two photodiodes 5a and 5b connected in series, and the two optical signals output from the optical circuit 22 are irradiated to the photodiodes 5a and 5b, respectively. From the light receiving element 5, the difference between the current generated by the photodiode 5a and the current generated by the photodiode 5b is output as a differential current signal.

  In the demodulator 1000 according to the first modification, the light receiving element 4 includes two photodiodes 4a and 4b, and the photodiodes 4a and 4b are irradiated with two optical signals output from the optical circuit 21, respectively. Are not connected in series. Similarly, the light receiving element 5 includes two photodiodes 5a and 5b. The photodiodes 5a and 5b are irradiated with two optical signals output from the optical circuit 22, but are connected in series. Absent.

  In the first embodiment, the transimpedance amplifier 6 converts the differential current signal output from the light receiving element 4 into a voltage signal, and outputs the voltage signal as an electric signal ES1P and an electric signal ES1N. Similarly, the transimpedance amplifier 7 converts the differential current signal output from the light receiving element 5 into a voltage signal, and outputs the voltage signal as an electric signal ES2P and an electric signal ES2N.

  In the demodulator 1000 according to the first modification, the transimpedance amplifier 6 differentially detects the current signals output from the two photodiodes 4a and 4b, converts them into voltage signals, and converts the voltage signals into the electric signal ES1P and the electric signal. Output as signal ES1N. Similarly, the transimpedance amplifier 7 differentially detects current signals output from the two photodiodes 5a and 5b, converts them into voltage signals, and outputs the voltage signals as the electric signal ES1P and the electric signal ES1N.

  Thus, even if differential detection is performed by the transimpedance amplifiers 6 and 7, the same result can be obtained.

  The above configuration is applicable not only in the first embodiment but also in the second to fifth embodiments.

<Modification 2>
FIG. 23 is a diagram showing a configuration of an optical receiving apparatus according to the second modification of the present invention. Hereinafter, description will be made centering on demodulator 1001, which is different from the optical receiver according to Embodiment 1. Since the components other than the demodulator 1001 are the same as those in the first embodiment, detailed description thereof is omitted.

  In the first embodiment, envelope detectors 51, 71a, 71b are included in control units 50, 70, respectively, and adder 75 is included in control unit 70. However, the demodulator 1001 in the second modification includes envelope detectors 51, 71a, 71b, and an adder 75. Even with such a configuration, an equivalent result can be obtained.

  The above configuration is applicable not only in the first embodiment but also in the second to fifth embodiments.

  1,1000,1001 demodulator, 2 delay interferometer, 3 photoelectric conversion device, 4,5 light receiving element, 4a, 4b, 5a, 5b photodiode, 6,7 transimpedance amplifier, 20 splitter, 21, 22 optical circuit, 31, 32 twin photodiodes, 50, 70, 700, 703 control unit, 51, 71a, 71b envelope detector, 52, 72 signal generator, 53, 73, 800, 803 manipulated variable determination unit, 54, 74, 75,76 adder, 60 data reproducing unit, 61,62 CDR circuit, 63 reproducing unit, 77 subtractor, 78,530,730a, 730b multiplier, 100 graph, 101,102,103,104 white circle, 210,211 220, 221 optical path, 212, 213, 222, 223 heater, 230 silicon substrate, 231 Underclad layer, 232 Overclad layer, 531, 731a, 731b Average value calculation unit, 532, 732 determination unit, 733 sweep unit, 734 detection monitor unit.

Claims (10)

A delay interferometer that includes first and second optical circuits belonging to the first and second series, and first and second twin photodiodes that respectively receive the output light of the first and second optical circuits. A demodulator that separately demodulates a dynamically modulated optical signal using the first and second sequences and outputs first and second demodulated signals that are shifted from each other by a predetermined phase;
A first control unit that controls a phase state of the optical signal in the first optical circuit using the first demodulated signal;
A computing unit that computes the first and second demodulated signals and outputs a computed signal;
A second control unit that controls a phase state of the optical signal in the second optical circuit using the second demodulated signal and the arithmetic signal;
Optical receiver. The second control unit controls a phase state of the optical signal in the second optical circuit so that the first demodulated signal and the second demodulated signal are shifted in phase by 0 or π.
The optical receiver according to claim 1. The arithmetic unit adds, subtracts, or multiplies the first demodulated signal and the second demodulated signal, and outputs the arithmetic signal.
The optical receiver according to claim 1. The first controller is
A first detector for detecting the first demodulated signal and outputting a first detected signal;
A first average value calculating unit that calculates an average value of the first detection signal and outputs the average value as a first average value;
A first determining unit that determines a control output to the first optical circuit based on the first average value;
The optical receiver according to claim 1. The second controller is
A second detector for detecting the second demodulated signal and outputting a second detected signal;
A second average value calculating unit that calculates an average value of the second detection signal and outputs the second average value;
A second determining unit that determines a control output to the second optical circuit based on the second average value;
The optical receiver according to claim 1. The second controller is
A third detector for detecting the operation signal and outputting a third detection signal;
The second determination unit determines a control output to the second optical circuit based on the third detection signal;
The optical receiver according to claim 5. The second controller is
A third average value calculating unit that calculates an average value of the third detection signal and outputs the average value as a third average value;
The second determination unit determines a control output to the second optical circuit based on the third average value;
The optical receiver according to claim 6. The second controller is
A sweep unit that sweeps the third detection signal or the third average value before controlling a phase state of the optical signal in the second optical circuit;
The optical receiver according to claim 7. A delay interferometer that includes first and second optical circuits belonging to the first and second series, and first and second twin photodiodes that respectively receive the output light of the first and second optical circuits. A demodulator that separately demodulates a dynamically modulated optical signal using the first and second sequences and outputs first and second demodulated signals that are shifted from each other by a predetermined phase;
A first control unit that controls a phase state of the optical signal in the first optical circuit using the first demodulated signal;
A multiplier for multiplying the first and second demodulated signals and outputting a multiplied signal;
A second control unit that controls the phase state of the optical signal in the second optical circuit using the multiplication signal;
Optical receiver. The second controller is
A third average value calculating unit that calculates an average value of the multiplication signals and outputs the average value as a third average value {average signal A2b};
A second determination unit that determines a control output to the second optical circuit based on the third average value;
The optical receiver according to claim 9.

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