JP2011061384A - Quadrature modulator and transmitter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To improve the average efficiency by limiting power consumption during high output. <P>SOLUTION: A quadrature modulator is equipped with: an oscillation means 104 for generating a local signal having a predetermined frequency; a phase shifting means 105 for generating a first local signal and a second local signal having a phase difference of 90 degrees from the local signal; a phase rotation means 103 for obtaining a second in-phase signal and a second quadrature signal by rotating the phases of a first in-phase signal and a first quadrature signal by 45+90×n degrees (n is an arbitrary integer); and a multiplication means 106 for obtaining a first output signal by multiplying the first local signal by the second in-phase signal, and obtaining a second output signal by multiplying the second local signal by the second quadrature signal. <P>COPYRIGHT: (C)2011,JPO&INPIT

Description

  The present invention relates to a quadrature modulator used in a wireless transmission device and a transmission device.

  One important performance index of a transmitter is power efficiency given by the ratio of output power to consumed power. As the power efficiency is higher, the power consumption can be suppressed, and as a result, the power consumption of the transmission unit can be reduced. In general, the ratio of the power consumption of the power amplifier to the power consumption of the transmitter is large, and the power amplifier has a great influence on the power efficiency. Therefore, as a method for increasing the power efficiency, there is a method using a power mixer that obtains a large output power directly from a modulator (see Non-Patent Document 1, for example).

S. Kousai et al, “An Octave-Range Watt-Level Fully Integrated CMOS Switching Power Mixer Array for Linearization and Back-Off Efficiency Improvement”, ISSCC2009, 22.2.

  However, since the polar modulation method used in Non-Patent Document 1 has a wider signal band for amplitude information and phase information than the signal band for I and Q signals used for the orthogonal modulation method, the signal processing unit is widened. There is a need to. For this reason, it is conceivable to configure the modulator using an orthogonal modulation method that can be used in a narrow signal band. However, in the quadrature modulation method, efficiency is most degraded at the maximum output with respect to a signal system having a square signal point arrangement such as 16QAM or 64QAM. When the efficiency varies depending on the signal point, if the efficiency is low at the time of high output, the wasteful power increases in proportion to the output, so the average efficiency decreases.

  The present invention has been made to solve the above-described problems, and an object of the present invention is to provide a quadrature modulator and a transmission device that can suppress power consumption at high output and improve average efficiency. And

  In order to solve the above-described problem, an orthogonal modulator according to the present invention includes an oscillating unit that generates a local signal having a predetermined frequency, and a first local signal and a second local signal that are 90 degrees out of phase from the local signal. The phase-shifting means for generating the signal and the phase of the first in-phase signal and the first quadrature signal are rotated by 45 degrees + 90 × n degrees (n is an arbitrary integer) to obtain the second in-phase signal and the second quadrature signal. A first rotation signal is multiplied by the first local signal and the second in-phase signal to obtain a first output signal, and the second local signal and the second quadrature signal are multiplied to obtain a second output signal. Obtaining multiplication means.

  Further, the transmission apparatus according to the present invention includes an oscillating unit that generates a local signal having a predetermined frequency, and a phase shifting unit that generates a first local signal and a second local signal that are 90 degrees out of phase from the local signal. And phase rotation means for rotating the phase of the first in-phase signal and the first quadrature signal by 45 degrees + 90 × n degrees (n is an arbitrary integer) to obtain the second in-phase signal and the second quadrature signal, Multiplying means for multiplying the first local signal and the second in-phase signal to obtain a first output signal and multiplying the second local signal and the second quadrature signal to obtain a second output signal; It is characterized by comprising combining means for combining the first output signal and the second output signal to obtain a third output signal, and an antenna for transmitting the third output signal.

  According to the quadrature modulator and the transmission apparatus of the present invention, it is possible to suppress power consumption at high output and improve average efficiency.

1 is a block diagram showing a quadrature modulator according to a first embodiment. The figure which shows the vector locus | trajectory of the baseband signal of 16QAM. The figure which shows the vector locus | trajectory of the baseband signal of 16QAM rotated by 45 degree | times clockwise. The figure which shows the structural example of a signal phase rotation part. The figure which shows an example in which a signal phase rotation part is comprised by the digital domain. The figure which shows an example of a structure of the power mixer which concerns on 1st Embodiment. The figure which shows another example of a structure of the power mixer which concerns on 1st Embodiment. The block diagram which shows the quadrature modulator which concerns on 2nd Embodiment. The figure which shows the output possible range and signal point distribution range of the orthogonal modulator which concern on 1st Embodiment. The figure which shows the output possible range of the quadrature modulator which concerns on 2nd Embodiment. The figure which shows an example of a structure of a controller and a power mixer. The figure which shows an example of a structure of a controller and a power mixer when the LSB side of a signal is made into an analog signal. The figure which shows an example of a structure of the mixer unit which concerns on 2nd Embodiment. The block diagram which shows the transmitter which concerns on 3rd Embodiment. The block diagram which shows another example of the transmitter which concerns on 3rd Embodiment.

(First embodiment)
Hereinafter, a quadrature modulator and a transmission apparatus according to embodiments of the present invention will be described in detail with reference to the drawings. In the following embodiments, the same reference numerals are assigned to the same numbered parts, and repeated description is omitted.
The configurations of the quadrature modulator and the transmission apparatus according to the present embodiment will be described in detail with reference to FIG.
The quadrature modulator 100 according to the present embodiment includes input units 101 and 102, a signal phase rotation unit 103, a local oscillation unit 104, a phase shift unit 105, power mixers (also referred to as multiplication units) 106 and 107, and a synthesis unit 108. .

  Input units 101 and 102 receive a signal corresponding to a baseband signal to be transmitted from a baseband unit (not shown). In the example of this embodiment, an I signal (also referred to as an in-phase signal) is input to the input unit 101, and a Q signal (also referred to as a quadrature signal) is input to the input unit 102.

  The signal phase rotation unit 103 receives the I signal and the Q signal from the input unit 101 and the input unit 102, respectively, and 45 degrees + 90 × n degrees (n is an arbitrary integer) on the IQ plane with respect to the I signal and the Q signal. ) Phase rotation. The operation of the signal phase rotation unit 103 will be described later with reference to FIGS. 2, 3, and 4.

  The local oscillation unit 104 generates a local signal corresponding to a carrier frequency of an RF (Radio Frequency) signal.

  The phase shift unit 105 receives the local signal from the local oscillation unit 104 and generates two orthogonal first and second local signals whose phases are different by 90 degrees.

  The power mixer 106 receives the I signal from the signal phase rotation unit 103 and the first local signal from the phase shift unit 105, and generates a first output signal obtained by multiplying the I signal and the first local signal.

  The power mixer 107 receives the Q signal from the signal phase rotation unit 103 and the second local signal from the phase shift unit 105, and generates a second output signal obtained by multiplying the Q signal and the second local signal. The power mixer 106 may multiply the I signal and the second local signal, and the power mixer 107 may multiply the Q signal and the first local signal.

  The synthesizing unit 108 receives the first and second output signals from the power mixers 106 and 107, and outputs a transmission signal obtained by adding the two output signals to the outside.

  Here, an example of the vector locus of the I signal and the Q signal input to the input units 101 and 102 will be described in detail with reference to FIG. FIG. 2 shows a 16QAM vector locus on the IQ plane. The IQ plane is a plane coordinate having an I signal on the horizontal axis and a Q signal on the vertical axis.

  In a signal system such as 16QAM or 64QAM, the vector trajectory of the I signal and the Q signal input to the input units 101 and 102 is a range surrounded by a square line symmetrical with respect to the I axis and the Q axis on the IQ plane. Distributed inside. The amplitudes of the first and second output signals output from the power mixers 106 and 107 are proportional to the distance from the origin, and are maximized at the signal points 201, 202, 203, and 204 at each vertex of the square shown in FIG. .

Assume that the power consumption of the power mixers 106 and 107 is proportional to the absolute value of the output signal. When the signal point is not rotated by the signal phase rotation unit 103, the power mixers 106 and 107 have the size of “a” at the signal points 201, 202, 203, and 204 at each vertex of the square on the IQ plane. 1. Since the second output signal is output, the power consumption can be expressed as 2 × a × P _unit . Here, P _unit indicates the power required to output a signal point that is a unit distance away from the origin on the I axis or the Q axis.

  When polar modulation is used as the modulation method, the power consumption is mainly determined by the distance from the origin on the IQ plane to the output signal point, so the power consumption at the maximum output is determined by the distance to each vertex. On the other hand, when quadrature modulation is used, it is determined by the sum of the amplitude of the I signal and the amplitude of the Q signal at the signal point. Since either the Q component or the I component is 0 at the points on the I axis and the Q axis, the sum of the amplitude of the I signal and the amplitude of the Q signal is the same as the distance from the origin to the signal point. Therefore, the power consumption is the same as that of polar modulation. However, at other signal points, the sum of the amplitude of the I signal and the amplitude of the Q signal is greater than the distance from the origin to the signal point, so that power consumption is greater than with polar modulation. The ratio of the difference between the distance from the origin and the I component + Q component is the largest on the diagonal of the square in which the signal points are distributed, which is √2. That is, the power consumption when the same signal point on the diagonal line is output becomes √2 times larger in the quadrature modulation than in the polar modulation, and the efficiency is degraded accordingly.

Next, a vector locus when the signal point is rotated by the signal phase rotating unit 103 will be described in detail with reference to FIG.
The example of FIG. 3 is a signal point arrangement when the signal point is rotated in the positive direction (clockwise) by 45 degrees. The I signal and Q signal after rotation of the signal point are referred to as I ′ signal and Q ′ signal, respectively. Here, the signal points 201, 202, 203, and 204, which are the maximum output points shown in FIG. 2, correspond to the signal points 301, 302, 303, and 304 on the I ′ axis and the Q ′ axis, respectively. . In order to rotate the phase by 45 degrees in the positive direction, for example, the following calculation may be performed.

In the example of FIG. 3, the signal point is rotated 45 degrees in the positive direction. However, the present invention is not limited to this, and any signal point obtained by rotating the signal point by 45 degrees + 90 × n degrees may be used. Therefore, since the signal point that becomes the maximum output when the signal point is rotated exists on the I ′ axis and the Q ′ axis, the distance from the origin is √2 × a, and the power required for the maximum output Becomes √2 × a × P _unit . This is because the power consumption is 1 / √2 compared to the case where the signal point is not rotated, so that the power efficiency at the maximum output is improved by 1.5 dB.

Here, consider a case where the maximum power consumption is the same. In the example of FIG. 2, assuming that the maximum output amplitude of “a” in the I-axis direction and “a” in the Q-axis direction is output and the power consumption at that time is 2 × a × P _unit , the example of FIG. In this case, the maximum power that can be output with the power consumption of 2 × a × P _unit is a point separated from the origin by a distance of 2 × a in either of the directions on the I ′ axis and the Q ′ axis. At this time, the maximum output amplitude is √2 times larger than √2 × a. Since the output power is proportional to the square of the output amplitude, the maximum output power at the same power consumption is 3 dB higher than the signal point shown in FIG. 2 for the signal point shown in FIG.

A configuration example of the signal phase rotation unit 103 will be described in detail with reference to FIG.
The signal phase rotation unit 103 shown in FIG. 4 is a case where the phase is rotated in the positive direction by 45 degrees as shown in Expression (1). The signal phase rotation unit 103 receives the I signal and the Q signal from the input units 101 and 102 respectively, and the adders 401 and 402 add the respective signals. In the example of FIG. 4, the adder 401 adds an I signal and an inverted Q signal, that is, a “−Q” signal, to generate an I ′ signal. Similarly, the adder 402 adds the I signal and the Q signal to generate a Q ′ signal. As a result, an I ′ signal and a Q ′ signal, which are signals obtained by rotating the phase in the positive direction by 45 degrees, can be obtained. Then, the signal phase rotation unit 103 sends the I ′ signal and the Q ′ signal to the power mixers 106 and 107, respectively.
For example, when the phase is to be rotated 135 degrees in the positive direction, the “−I” signal and the “−Q” signal are input to the adder 401, and the I signal and the “−Q” signal are input to the adder 402. Input and add each. In this way, if the addition and subtraction are interchanged and positive / negative inversion is used in the same configuration, the signal phase rotation unit 103 of 45 degrees + 90 × n degrees can be configured with respect to an arbitrary n. Furthermore, since the amplitudes of the I ′ signal and the Q ′ signal are increased by √2 times compared to the I signal and the Q signal, it is necessary to make the amplitudes of the I signal, the Q signal, the I ′ signal, and the Q ′ signal uniform. May include a multiplier that multiplies 1 / √2 to either of the inputs and outputs.

Next, a configuration example of the power mixer used in the power mixers 106 and 107 will be described in detail with reference to FIG. FIG. 5 shows an example using a double balance mixer type configuration. The power mixer also serves as a drive amplifier or a power amplifier.
At this time, the combining unit 108 may be realized by sharing the loads of a plurality of power mixers.
Another example of the configuration of the power mixer will be described in detail with reference to FIG.
In FIG. 7, the power mixer includes a plurality of power mixer units 601, and each power mixer unit 601 uses, for example, a double balance mixer having the same configuration as in FIG. 5. Input signals are independently input to the power mixer units 601, and the power mixer units 601 are controlled by the respective input signals. Note that the output power may be controlled by switching the power mixer unit 601 on or off as a digital signal. Also, some input signals may be analog signals corresponding to the MSB (Most Significant Bit) side of the digital signal, and other input signals may be corresponding to the LSB (Least Significant Bit) side of the digital signal. By doing in this way, the number of power mixer units can be increased / decreased according to the request | requirement required for digital control.

  According to the first embodiment described above, the signal point on the IQ plane is rotated by 45 degrees + 90 × n degrees, thereby suppressing the power consumption at the maximum output and improving the overall average efficiency. The power efficiency equivalent to that of the polar modulation method can be obtained while the signal processing unit is configured with a signal band narrower than that of the polar modulation method.

(Modification of the first embodiment)
A configuration example in which the signal phase rotation unit 103 is realized in the digital domain will be described in detail with reference to FIG. A digital filter 701 and DACs (Digital-analog converters) 702 and 703 are included in the subsequent stage of the signal phase rotation unit 103 of the quadrature modulator 700 according to this modification.

  Further, in the example of FIG. 7, the signal phase rotation unit 103 is arranged before the digital filter. However, when the digital filter 701 is arranged, the signal phase rotation unit 103 is placed either before or after the digital filter 701. You may arrange | position in the position of.

  According to the modification of the first embodiment described above, since the phase rotation process is performed in the digital domain, the phases of the I signal and the Q signal can be accurately rotated.

(Second Embodiment)
In the first embodiment, the power mixer is arranged independently for each of the I signal and the Q signal. However, in this embodiment, it is first to use a common power mixer for the I signal and the Q signal. Different from the first embodiment.

  The quadrature modulator 800 according to the present embodiment includes an input unit 101 and an input unit 102, a signal phase rotation unit 103, a local oscillation unit 104, a phase shift unit 105, a controller 801, and a power mixer 802.

  Since the input unit 101, the input unit 102, the signal phase rotation unit 103, the local oscillation unit 104, and the phase shift unit 105 perform the same operations as those in the first embodiment, a detailed description thereof is omitted here.

  The controller 801 receives the I ′ signal and the Q ′ signal from the signal phase rotation unit 103 and outputs a local selection signal and a phase selection signal. The local selection signal is a signal corresponding to any of the I ′ component, the Q ′ component, and the no signal. The no signal may be either “0” transmitted as a signal or a case where no signal is transmitted. In the following, it is assumed that no signal transmits “0” as a signal. Detailed operation of the controller 801 will be described later with reference to FIGS. 11 and 12.

  The power mixer 802 receives the local selection signal and the phase selection signal from the controller 801, and generates an output signal according to the input signal. Detailed operation of the power mixer 802 will be described later with reference to FIGS. 11, 12, and 13.

Here, the relationship between the signal output possible range and the signal point distribution range will be described in detail with reference to FIG. 9 and FIG.
In the first embodiment, assuming that the maximum outputs from the power mixer 106 and the power mixer 107 are points separated from the origin by “a”, the signal output possible range 901 is indicated by a solid line as shown in FIG. This is a range surrounded by a square composed of straight lines parallel to the I ′ axis and the Q ′ axis. However, the signal point distribution range 902 is a range rotated by 45 degrees from the square of the output possible range 901 as indicated by a dotted line. For this reason, since the output possible range 901 and the signal point distribution range 902 are different from each other, an extra device area that does not actually output a signal is provided.
On the other hand, the output possible range in this embodiment is shown in FIG. In this embodiment, the sum of the absolute value of the I ′ signal and the absolute value of the Q ′ signal, that is, the sum of the amplitudes is set so as not to exceed a predetermined maximum value. When the sum of absolute values is “a”, the output possible range 1001 can be matched with the signal point distribution range 902 shown in FIG.

  For example, as a specific example, assuming that the device area necessary for configuring a device that outputs a point separated by “a” from the origin is “A”, in the first embodiment, the I′-axis direction and the Q′-axis Since it is necessary to output only “a” in each direction, a total device area of “2A” is required. On the other hand, in this embodiment, since the sum of the absolute value of the I ′ signal and the absolute value of the Q ′ signal is “a”, the required device area is “A”. That is, if a power mixer for multiplying the I ′ signal and the first local signal and a power mixer for multiplying the Q ′ signal and the second local signal are prepared separately, two power mixers are necessary. 2A "device area is required. However, in this embodiment, focusing on the fact that the sum of the absolute value of the I ′ signal and the absolute value of the Q ′ signal is “a”, one power mixer is multiplied by the I ′ signal and the first local signal. And the device area can be set to “A” by sharing the multiplication of the Q ′ signal and the second local signal.

  Actually, the output of the power mixer 802 needs to be controlled by the controller 801 in accordance with the I ′ signal and the Q ′ signal, but the controller 801 can be realized with a sufficiently small area as compared with the power mixer 802. For this reason, the entire device can be configured with a small device area, and the size of a necessary device can be reduced, so that the influence of parasitic capacitance and the like can be reduced.

  A configuration example of the controller 801 and the power mixer 802 in this embodiment is shown in FIG. The power mixer 802 includes a (a is an arbitrary natural number) mixer units 1101. The mixer unit 1101 according to this embodiment includes an LO selection unit 1102 and a power mixer unit 601.

The power mixer unit 601 has the same configuration as that of the first embodiment.
The LO selection unit 1102 receives the local selection signal and the phase selection signal from the controller 801, and selects and outputs a signal corresponding to the local selection signal.

Here, detailed operations of the controller 801 and the power mixer 802 will be described in detail with reference to FIG. Here, it is assumed that a bit string of a digital signal is input as the I ′ signal and the Q ′ signal.
The controller 801 sends a local selection signal and a phase selection signal to the a number of mixer units 1101 to the LO selection unit 1102. One of “1”, “0”, and “−1” is selected as the phase selection signal.

  The LO selection unit 1102 receives two local signals that are 90 degrees out of phase from the phase shift unit 105 and a local selection signal from the controller 801, and corresponds to the I ′ signal or the Q ′ signal according to the local selection signal. Select and output a local signal. When the local selection signal is “0”, that is, when there is no signal, “0” is output as the local signal.

  The power mixer unit 601 receives the phase selection signal from the controller 801 and the local signal selected from the LO selection unit 1102 by the local selection signal, and performs multiplication. When “1” is input as the phase selection signal, a signal in phase with the selected local signal is output, and when “−1” is input, a signal having a phase opposite to that of the selected local signal is output. Is output. When “0” is input as the phase selection signal, the output signal is also “0”. As a method for setting the output from the mixer unit 1101 to “0”, whether the local selection signal from the controller 801 is no signal or the phase selection signal is “0” is selected. Alternatively, a method in which both of them are selected is conceivable, but either method may be used.

As a specific example, FIG. 11 shows a case where N bit signals whose I ′ component is + N are input as I ′ signals and M bit signals whose Q ′ component is −M are input as Q ′ signals. Yes.
Here, since the sum of the absolute value of the I ′ signal and the absolute value of the Q ′ signal does not always exceed “a” within the possible range of the signal, the controller 801 has a total of a mixer units. A local selection signal with “I ′” selected as the I ′ signal is output to the N mixer units 1101 in 1101, and “1” is output as the phase selection signal because the I ′ component is “+”. To do. Similarly, a local selection signal with “Q ′” selected as the Q ′ signal is output to the M mixer units 1101, and since the Q ′ component is “−”, “−1” is selected as the phase. Output as a signal. For the remaining (a−N−M) mixer units 1101, the local selection signal is set to no signal, and “0” is output as the phase selection signal. Finally, by adding all the outputs from the a number of mixer units 1101, the power mixer 802 is desired that the sum of the absolute value of the I ′ signal and the absolute value of the Q ′ signal does not always exceed “a”. Output signal can be obtained.

In the example of FIG. 11, the phase selection signal has a value of “−1”, “0”, or “1”. However, as in the case of the first embodiment, some input signals are An analog signal corresponding to the LSB side of the digital signal may be used.
FIG. 12 shows a case where the LSB side is an analog signal. The controller 801 shown in FIG. 11 receives the MSB side of the digital signal indicating the I ′ signal and the Q ′ signal, and inputs the LSB side of the digital signal indicating the I ′ signal and the Q ′ signal to the DAC. Get. By doing so, the number of mixer units 1101 in the power mixer 802 can be reduced, and the device area of the power mixer 802 can be reduced.

A configuration example of the mixer unit in the second embodiment will be described in detail with reference to FIG.
As shown in FIG. 13, a LO mixer 1102 and a power mixer as shown in FIG. 6 are combined to form one mixer unit 1101, and a power mixer 802 can be configured by using a plurality of them.

  According to the second embodiment described above, by matching the signal point distribution range and the output possible range, the entire device area can be reduced, and power consumption is reduced while reducing the influence of parasitic capacitance and the like. Can improve the average efficiency.

(Third embodiment)
FIG. 14 shows a transmission apparatus 1400 according to the third embodiment. Transmitting apparatus 1400 includes quadrature modulator 100 and antenna 1401.
The antenna 1401 receives an output signal from the combining unit 108 related to the quadrature modulator 100 and radiates the output signal to space.

  Further, when it is necessary to supply an output signal larger than the sum of the absolute values of the output signals from the power mixer 106 and the power mixer 107 to the antenna 1401, as in the transmission device 1500 shown in FIG. (Power Amplifier) 1501 may be disposed in front of the antenna 1401.

  According to the third embodiment described above, it is possible to configure a transmission device that improves average power efficiency.

  Note that the present invention is not limited to the above-described embodiment as it is, and can be embodied by modifying the constituent elements without departing from the scope of the invention in the implementation stage. In addition, various inventions can be formed by appropriately combining a plurality of constituent elements disclosed in the embodiment. For example, some components may be deleted from all the components shown in the embodiment. Furthermore, constituent elements over different embodiments may be appropriately combined.

100, 500, 800 ... quadrature modulator, 101, 102 ... input unit, 103 ... signal phase rotation unit, 104 ... local oscillation unit, 105 ... phase shift unit, 106, 107, 802 ... Power mixer, 108 ... Synthesis unit, 201, 202, 203, 204, 301, 302, 303, 304 ... Signal point, 401,402 ... Adder, 601 ... Power mixer Unit, 701 ... Digital filter, 702,703 ... DAC, 801 ... Controller, 901,1001 ... Output possible range, 902 ... Signal point distribution range, 1101 ... Mixer unit, 1102 ... LO selection unit, 1400, 1500 ... transmitting device, 1401 ... antenna, 1501 ... PA.

Claims (4)

Oscillating means for generating a local signal having a predetermined frequency;
Phase shifting means for generating a first local signal and a second local signal having phases different from each other by 90 degrees from the local signal;
Phase rotation means for rotating the phases of the first in-phase signal and the first quadrature signal by 45 degrees + 90 × n degrees (n is an arbitrary integer) to obtain the second in-phase signal and the second quadrature signal;
Multiplying means for multiplying the first local signal and the second in-phase signal to obtain a first output signal and multiplying the second local signal and the second quadrature signal to obtain a second output signal; A quadrature modulator comprising:   The first in-phase signal and the first quadrature signal are digital signals, and the phase rotation means adds or subtracts the first in-phase signal and the first quadrature signal to thereby add the second in-phase signal. The quadrature modulator according to claim 1, wherein the signal and the second quadrature signal are obtained. The second in-phase signal, the second quadrature signal, and the no signal from a plurality of bits so that the sum of the absolute value of the second in-phase signal and the absolute value of the second quadrature signal does not exceed a threshold value If the local selection signal for selecting any one of the signal and the in-phase component of the second in-phase signal or the quadrature component of the second quadrature signal are positive, a signal in phase with the local signal is output and the in-phase signal is output. If the component or the quadrature component is negative, it further comprises control means for outputting a phase selection signal for outputting a signal having a phase opposite to that of the local signal,
The quadrature modulator according to claim 2, wherein the multiplication unit multiplies the phase selection signal and the local signal for each bit. Oscillating means for generating a local signal having a predetermined frequency;
Phase shifting means for generating a first local signal and a second local signal having phases different from each other by 90 degrees from the local signal;
Phase rotation means for rotating the phases of the first in-phase signal and the first quadrature signal by 45 degrees + 90 × n degrees (n is an arbitrary integer) to obtain the second in-phase signal and the second quadrature signal;
Multiplying means for multiplying the first local signal and the second in-phase signal to obtain a first output signal and multiplying the second local signal and the second quadrature signal to obtain a second output signal;
Combining means for combining the first output signal and the second output signal to obtain a third output signal;
An antenna for transmitting the third output signal;

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