JP2010178580A - Motor inverter system - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To reduce iron loss by decreasing magnetic flux ripple by PWM control of a drive current. <P>SOLUTION: A coil iA and a coil iB are wound around a teeth i (i=1 -9) so that the number of turns of the coil A is equal to that of the coil B. One end of the coil B is connected to a -V3 phase of a 9-phase inverter 10. The other end of the coil B is connected to one end of the coil 2A. The other end of the coil 2A is connected to a neutral point 11. Hereinafter, the same connection is performed and one end of the coil 9B is connected to a -V2 phase of the 9-phase inverter 10. The other end of the coil 9B is connected to one end of the coil 1A. Phases of a carrier of U1, V1, W1 phase, a carrier of U2, V2, W2 phase, and a carrier of U3, V3, W3 phase of a PWM control circuit for driving the 9-phase inverter 10 are shifted by 120°, respectively. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a motor / inverter system that drives a motor from an inverter that is PWM-modulated at a carrier frequency.

  In an AC servo motor that supplies PWM controlled current to each phase of U, V, and W, a plurality of coils are provided in each phase in order to reduce the ripple width of the magnetic flux waveform and reduce iron loss. There is known a technique for driving a coil with a PWM wave having a phase shift (for example, Patent Document 1). According to this technology, when two systems of coils are provided for each phase, the current ripple peaks of one system and the other are driven by driving the coils of each system with PWM waves that are 180 ° out of phase with each other. Since the current ripple valleys of the system overlap with each other in time, the width of the resultant magnetic flux ripple is reduced to reduce the iron loss.

JP-A-6-197593

  However, in the conventional example, two three-phase inverters are required to drive the three-phase motor, so the number of inverters is twice as many as that of a normal three-phase motor. In addition, since the current flowing through the switching element of each inverter and the voltage applied to the switching element are not different from those of a normal three-phase motor inverter, there is a problem that the inverter loss is doubled.

  In order to solve the above problems, the present invention supplies m-phase current to a motor having m teeth (m is an integer of 3 or more) each having two coils and m half bridges. A motor / inverter system comprising: a coil wound around one tooth and a coil wound around another tooth connected in series, and two coils connected in series The induced voltage phases generated in the inverters are different from each other by 360 / m [°], and the currents flowing through the two coils wound around the teeth are different from each other in the phase of the PWM carrier wave in the inverter.

  According to the present invention, since the currents of the two coils wound around one tooth have different carrier phases when generated by PWM in the inverter, the ripples of magnetic flux generated by the two coil currents cancel each other. There is an effect that iron loss can be reduced without increasing the number of inverters.

BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a circuit diagram illustrating a connection between a winding and an inverter of a 9-phase, 9-slot, 10-pole motor that is Embodiment 1 of the motor / inverter system according to the present invention. It is a figure which shows the phase of the induced voltage in Example 1, and the electric current phase sent through a coil. It is a figure which shows the structure which wound one coil around 1 teeth of the comparative example. It is a figure which shows the time change of the coil electric current and magnetic flux in a comparative example. 3 is a diagram illustrating a configuration of two coils per tooth in Example 1. FIG. It is a figure which shows the time change of the synthetic | combination magnetic flux when the electric current in which a 40 degree phase differs is sent through the two coils per tooth | gear in Example 1. FIG. It is a figure explaining how to shift the PWM carrier phase in the first embodiment. (A) Reduction amount of iron loss in Example 1, (b) It is a figure which shows the decreasing rate of the iron loss in Example 1. FIG. (A) The amount of reduction | decrease of the total loss in Example 1, (b) It is a figure which shows the reduction rate of the total loss in Example 1. FIG. FIG. 6 is a main part connection diagram illustrating a modification of the first embodiment.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

[Circuit connection and current supply]
FIG. 1 is a circuit diagram illustrating the connection between a winding of a 9-phase 9-slot 10-pole motor, which is Embodiment 1 of the motor / inverter system according to the present invention, and an inverter. In this example, the value of m indicating the number of phases is 9. The motor shown in FIG. 1 includes nine teeth from a top to a teeth 1, teeth 2,. Two coils, i.e., coil iA and coil iB, are wound around each tooth i (i = 1 to 9). The induced voltage phase U2 generated in the tooth 2 is delayed by 40 [°] with respect to the induced voltage phase U1 generated in the tooth 1, and the induced voltage phase U3 generated in the tooth 3 is delayed by 80 [°]. Is delayed.

  The nine-phase inverter 10 is composed of -V3, -W1, -W2, -W3, -U1, -U2, -U3, -V1, and -V2 from DC supplied from the DC power supply E and the smoothing capacitor C. It has nine half bridges that convert to phase alternating current. A PWM control circuit (not shown) controls each half bridge of the inverter 10 and supplies a nine-phase drive current to the motor.

  One end of the coil 1B wound around the tooth 1 is connected to the -V3 phase of the 9-phase inverter 10. The other end of the coil 1B is connected to one end of a coil 2A wound around the tooth 2. The other end of the coil 2 </ b> A is connected to the neutral point 11. Therefore, the coil 1B and the coil 2A are connected in series.

  One end of the coil 2B wound around the tooth 2 is connected to the -W1 phase of the 9-phase inverter 10. The other end of the coil 2 </ b> B is connected to one end of a coil 3 </ b> A wound around the tooth 3. The other end of the coil 3 </ b> A is connected to the neutral point 11. Therefore, the coil 2B and the coil 3A are connected in series.

  Thereafter, the same connection is performed, and one end of the coil 9B wound around the tooth 9 is connected to the -V2 phase of the 9-phase inverter 10. The other end of the coil 9 </ b> B is connected to one end of a coil 1 </ b> A wound around the tooth 1. The other end of the coil 1 </ b> A is connected to the neutral point 11. Therefore, the coil 9B and the coil 1A are connected in series. Thus, one of the coils connected in series is connected to one at the neutral point 11 and the other is connected to the half bridge of the 9-phase inverter.

  With the above connection, when viewed with respect to each tooth i, the two coils iA and iB wound around the tooth i are supplied from the 9-phase inverter 10 with currents shifted from each other by 360 / m = 40 [°] phase. Will be.

  The phase of the current flowing from the half bridge to the coil is the same as the induced voltage generated in the coil, and when the advance angle control is performed, the phase is advanced based on the induced voltage phase. By performing the connection and energization as described above, a coil magnetic flux advanced by 90 [°] from the magnetic magnetic flux generated in each of the teeth 1 to 9 can be generated.

[Geometric Teeth Arrangement]
Next, with reference to Table 1, the geometric tooth arrangement will be described.

  Assuming that the tooth 1 is the reference position, the tooth 6 is located at a position geometrically shifted by 40 [°] from the tooth 1. There is a tooth 2 at a position displaced by 80 [°], and a tooth 7 is disposed at a position displaced by 120 [°]. Similarly, the arrangement shown in Table 1 is made.

[Coil series connection]
Two teeth are wound around each tooth, one being coil A and the other being coil B. The same number written in the column of the coil A and the number written in the column of the coil B, for example, the coil A of the tooth 1 and the coil B of the tooth 2 are connected in series.

[Current phase]
FIG. 2 is a diagram showing the phase of the induced voltage and the phase of the current flowing through the coil. In the figure, a solid arrow indicates an induced voltage phase, and a broken arrow indicates a current phase. A coil through which currents -V2 and -V3 flow is wound around the tooth where the induced voltage U1 is generated. A coil through which currents -V3 and -W1 flow is wound around the tooth where the induced voltage U2 is generated. Similarly, a coil through which a current of a phase indicated by two dashed arrows sandwiching a solid arrow is wound around each tooth where the induced voltage phase indicated by the solid line in FIG. 2 is generated.

[Current and magnetic flux of motor of comparative example]
Next, regarding the current flowing through the coil, the present embodiment is compared with a comparative example in which one coil is wound around one tooth having a normal configuration. The magnetic flux generated by the coil is proportional to the product of the current flowing through the coil and the number of turns of the coil. FIG. 3 shows a comparative example in which one coil of N turns is wound around one tooth and current I flows. FIG. 4 is a graph drawn assuming that a magnetic flux having an amplitude NI is generated when the current amplitude is I and the number of coil turns is N.

ここで、
Ａ：コイルＡに流れる電流振幅×巻数
Ｂ：コイルＢに流れる電流振幅×巻数
Ｘ：コイルＡに流れる電流に対する合成磁束の位相遅れ
φ：コイルＡに流れる電流とコイルＢに流れる電流の位相差
である。 [Current and magnetic flux of Example 1]
FIG. 5 shows a state where the coil A and the coil B are wound around one tooth of the first embodiment. The number of turns of each of the coils A and B is 0.5N, and the total number of the two coils is N. FIG. 6 shows current amplitude × number of turns necessary to generate a magnetic flux with amplitude NI when two coils are wound around one tooth 0.5N times and currents of 40 [°] phase are passed. It is the graph which showed. In order to generate a magnetic flux having the same amplitude NI as that of the comparative example of FIG. 3, the value of the current amplitude of each coil A and B × the number of turns needs to be about 0.532 NI. It can be seen that the current needs to flow 1.064I. This current value is obtained by the following equations (1) and (2).

here,
A: current amplitude flowing in coil A × number of turns B: current amplitude flowing in coil B × number of turns X: phase lag of combined magnetic flux with respect to current flowing in coil A φ: phase difference between current flowing in coil A and current flowing in coil B is there.

In the case of 9 phases as in this example,
φ = 360/9 = 40 [°]
X = φ / 2 = 20 [°]
And
A = B≈0.532
Is required.

  Thus, since the number of turns of the two coils wound around the individual teeth is made equal, if the phase current amplitude flowing in the two coils is made the same, the amplitude of the ripple current becomes the same, effectively canceling the magnetic flux ripple. There is an effect that can be.

  FIG. 7 shows an example of a method for shifting the carrier phase in the present embodiment. When one of the divisors of m (number of phases) and 1 other than m is K, the m voltage command vectors are added. In this example, K voltage command vectors that become 0 are grouped into one group, the phase of the carrier wave compared with the voltage command of the same group is the same, and the phase of the PWM carrier wave is different for each group. In FIG. 7, K = 3.

  In FIG. 7, the carrier phases of the U1, V1, and W1 phases are the same, the carrier phases of the U2, V2, and W2 phases are the same, and the carrier phases of the U3, V3, and W3 phases are the same. Then, the carrier phase of the U2 phase (V2 phase, W2 phase) is delayed by 120 [°] with respect to the carrier phase of the U1 phase (V1 phase, W1 phase), and the carrier phase of the U2 phase (V2 phase, W3 phase) is obtained. On the other hand, the carrier phase of the U3 phase (V3 phase, W3 phase) is further delayed by 120 [°].

  By shifting the carrier phase in this way, the current flowing through the two coils wound around each tooth is generated by PWM having different carrier phases, and the phase of the current ripple is shifted by PWM control. . Therefore, the magnetic flux ripples generated in the teeth are canceled out to reduce the iron loss. FIGS. 8 and 9 show the effect of reducing the iron loss on the rotation speed-torque map.

  FIG. 8A is a graph showing the iron loss reduction amount of Example 1 in [W] with respect to the comparative example. The size of the circle in the graph is proportional to the iron loss reduction. It can be seen that the iron loss reduction amount is 1046.3 [W] under the conditions of 4500 [rpm] and 200 [A] in the high rotation and high load region, and the absolute value of the iron loss reduction amount is large. FIG.8 (b) is the graph which showed the reduction rate of the iron loss of Example 1 with% with respect to the comparative example. The size of the circle in the graph is proportional to the iron loss reduction rate. The iron loss reduction rate is 21.3 [%] under the conditions of 4500 [rpm] and 200 [A] in the high rotation and high load region, and the iron loss is performed under the conditions of 500 [rpm] and 50 [A] in the low rotation and low load region. The reduction rate is 44.3 [%], and the iron loss reduction rate is 44.5 [%] under the conditions of 500 [rpm] and 350 [A] in the low rotation and high load region. The rate is high.

  As described above, according to the present embodiment, magnetic flux ripples generated by two coils wound around one tooth cancel each other, and iron loss can be reduced. However, compared with the comparative example, it is necessary to flow the current about 1.06 times, so that the inverter loss and the motor copper loss slightly increase. FIG. 9 shows the result of calculating the total loss of the inverter loss, the copper loss, and the iron loss and comparing it with the comparative example.

  FIG. 9A is a graph showing the total loss reduction amount of Example 1 as [W] with respect to the comparative example. The size of the circle in the graph is proportional to the amount of change in total loss. A hatched circle indicates a reduction in total loss, and a white circle indicates an increase in total loss. Compared with the comparative example, the present embodiment increases the total loss in the low rotation and high load region, but reduces the loss in the low rotation and low load region considered to be frequently used. FIG. 9B shows the reduction rate of the total loss in%. Loss in the low load region is reduced. The size of the circle in the graph is proportional to the rate of change in total loss. A hatched circle indicates a reduction in total loss, and a white circle indicates an increase in total loss. Compared with the comparative example, the present embodiment is considered to be frequently used although the total loss increases by 6.9 [%] under the conditions of 500 [rpm] in the low rotation and high load region and 350 [A] current. The total loss is reduced by 18.9 [%] under the conditions of 500 [rpm] and current 350 [A] in the low rotation and low load region.

  As described above, in the low-rotation and high-load region, in this embodiment, the increase amount of the inverter loss and the copper loss tends to be larger than the iron loss reduction amount. However, compared with the comparative example, in this embodiment, the iron loss reduction amount is larger than the increase in the inverter loss and the copper loss in the region other than the low rotation and high load region, and the total loss can be reduced. Therefore, when applied to a motor for driving an automobile that is frequently used in a low load region, the efficiency can be improved comprehensively.

  According to the present embodiment described above, the currents of the two coils wound around one tooth have different carrier phases when generated by PWM in the inverter, and therefore the phase of the ripple current flowing through the two coils is shifted. In addition, the ripples of the magnetic flux generated by the two coil currents cancel each other, and the iron loss can be reduced.

  In addition, according to this embodiment, one coil wound around one tooth and one coil wound around another tooth are connected in series, and an induced voltage generated in two coils connected in series. There is an effect that the torque can be efficiently generated by connecting the phases so that the phases are different from each other by 360 / m [°] and flowing the current so as to be in phase with the induced voltage phase connected in series.

  In addition, according to the present embodiment, by connecting two coils in series, it is not necessary to double the number of switching elements as in the conventional example, and the cost can be reduced and the size can be reduced by reducing the number of elements. It is. Furthermore, the inverter loss does not increase by a factor of two.

  Further, according to the present embodiment, since the number of turns of the two coils wound around the individual teeth is made equal, if the phase current amplitude flowing in the two coils is made the same, the amplitude of the ripple current becomes the same, and the magnetic flux There is an effect that ripple can be effectively canceled.

[Modification of Example 1]
As a modification of the present embodiment, FIG. 10 shows a motor / inverter system that can prevent the increase in total loss by switching the wiring connections in a low rotation and high load region. In FIG. 10, the coil iA and the coil iB are wound around the teeth i (i = 1 to 9) as in the first embodiment of FIG. 1. In the modification of FIG. 10, for the coil of the tooth 1, a switch 21 </ b> A that opens and closes between the other end of the coil 1 </ b> B and one end of the coil 2 </ b> A, and a switch that opens and closes between one end of the coil 1 </ b> A and the other end of the coil 1 </ b> B. 21B is added. The switches 22A to 29A and the switches 22B to 29B are similarly added to the coils of the teeth 2 to 9 (however, the illustration in FIG. 10 is up to a part of the coils of the tooth 5). Other configurations are the same as those of the embodiment of FIG.

  In FIG. 10, when the switch 21A is short-circuited and the switch 21B is opened, the connection of the present embodiment in which the coil 1B and the coil 2A are connected in series is obtained. When the switch 21A is opened and the switch 21B is short-circuited, the two coils 1A and 1B wound around the same tooth 1 are connected in series, and the same connection as that obtained when one coil is wound around one normal tooth is obtained. . Therefore, in the low rotation high load region, the efficiency reduction in the low rotation high load region can be prevented by opening the switch 21A and short-circuiting the switch 21B. The same applies to the switches 22A to 29A and 22B to 29B for the teeth 2 to 9.

  According to this modification described above, there is an effect that it is possible to avoid an increase in total loss in the low rotation and high load region, which is a drawback of the first embodiment.

  Next, as Example 2, a case where the present invention is applied to a 9-phase 18-slot 20-pole motor will be described. Corresponding to 18 slots, in this embodiment, teeth 1 to 18 are provided. Each tooth i (i = 1 to 18) is wound with two coils, i.e., a coil iA and a coil iB. The 9-phase inverter that drives these coils has the same configuration as the 9-phase inverter 10 shown in FIG.

  Here, coils wound around two teeth having the same induced voltage phase are connected in parallel. That is, since the induced voltage phases of the tooth 1 and the tooth 10 are equal, the coil 1A of the tooth 1 and the coil 10A of the tooth 10 are connected in parallel, and the coil 1B of the tooth 1 and the coil 10B of the tooth 10 are connected in parallel. Similarly, since the induced voltage phases of the tooth 2 and the tooth 11 are equal, the coil 2A of the tooth 2 and the coil 11A of the tooth 11 are connected in parallel, and the coil 2B of the tooth 2 and the coil 11B of the tooth 11 are connected in parallel. . Similarly, coils having the same induced voltage phase are connected in parallel. Since the induced voltage phases of the tooth 9 and the tooth 18 are equal, the coil 9A of the tooth 9 and the coil 18A of the tooth 18 are connected in parallel, and the coil 9 of the tooth 9 and the coil 18B of the tooth 18 are connected in parallel.

  If each of these two coils connected in parallel is regarded as one coil and the code of the younger one of the original two coils is changed, coil A wound on one tooth and coil B wound on another tooth are connected in series. The connection is the same as in the first embodiment, and the connection with the nine-phase inverter 10 is the same as in FIG.

  In this way, it is possible to lower the induced voltage by supplying coils by connecting coils of a pair of teeth having the same induced voltage phase in parallel, compared to supplying current in series. There is an effect that the control range can be expanded.

  Next, a third embodiment of the motor / inverter system according to the present invention will be described. In the present embodiment, for example, when the present invention is applied to a 5-phase 15-slot 16-pole motor, coils wound around a plurality of teeth having a difference in induced voltage phase are driven in the same current phase. This is an example.

Table 2 shows a connection method when the present invention is applied to a motor of 5 phases, 15 slots and 16 poles.

  In a present Example, it has two coils each wound around 15 teeth of the teeth 1-15. Since this is driven in five phases, it is necessary to arrange three sets of coils having similar induced voltage phases in parallel. When the induced voltage phase generated in the tooth 1 is used as a reference, the induced voltage generated in the tooth 1 by the tooth 2 in which an induced voltage delayed by 24 [°] and the tooth 15 in which an induced voltage advanced by 24 [°] is generated The phase is closest. These teeth 1, 2, and 15 will be referred to as teeth group 0.

[Teeth group]
The above description is generalized as follows. Let m be the number of phases and n be an integer that satisfies 0 ≦ n ≦ m−1. With respect to the teeth where the induced voltage phase [°] satisfies 360 × (n−1) / (2 m) ≦ 360 × n / m ≦ 360 × n / (2 m), the induced voltage satisfying the above inequality is obtained when n = 0. A plurality of teeth generated are teeth group 0, and when n = 1, a plurality of teeth that generate an induced voltage satisfying the above inequality is referred to as a teeth group 1, and hereinafter the teeth group m-1 is similarly named.

[Parallel connection A]
One of the two coils wound around each tooth of the tooth group 0 is connected in parallel with one of the two coils wound around the other teeth of the tooth group 0. This parallel connection coil is referred to as a coil A-0.

[Parallel connection B]
The other coil of the two coils wound around each tooth of the tooth group 0 is connected in parallel with the other coil of the two coils wound around the other teeth of the tooth group 0. This parallel connection coil is referred to as a coil B-0.

[Parallel connection A (generalized)]
Similarly, one of the two coils wound around each tooth of the tooth group n is connected in parallel with one of the two coils wound around the other teeth of the tooth group n. The This parallel connection coil is called parallel connection coil An.

[Parallel connection B (generalized)]
The other coil of the two coils wound around each tooth of the tooth group n is connected in parallel with the other coil of the two coils wound around the other teeth of the tooth group n. . This parallel connection coil is referred to as coil parallel connection Bn.

[Direct connection]
Next, as shown in Table 3, the parallel connection coil A-0 and the parallel connection coil B-1 are connected in series. This is named a series connection coil 0.

[Series connection (generalized)]
Similarly, the parallel connection coil An is connected in series with the parallel connection coil B- (n + 1). This is named a series connection coil n. This is named a series connection coil n. The parallel connection coil A- (m-1) is connected in series with the parallel connection coil B-0. This is named a series connection coil m-1.

[How to flow current]
One end of the series connection coil 0 is connected to an inverter having an m-phase half bridge, and the other end is connected to a neutral point. Then, a current having a current phase of −180 / m [°] flows from the inverter to the series connection coil 0.

[How to flow current (generalization)]
Similarly, one end of the series connection coil n is connected to an inverter having an m-phase half bridge, and the other end is connected to a neutral point. Then, a current having a current phase of 360 n / m-180 / m [°] is supplied from the inverter to the coil n connected in series.

  In FIG. 7 of the first embodiment, assuming that K = 3, K voltage command vectors that are added to 0 are set as one group, the phase of the carrier wave to be compared with the voltage command of the same group is the same, and the PWM for each group An example with different carrier wave phases is shown.

  In the fourth embodiment, how to shift the carrier phase when K is made as small as possible will be described.

表４は、８相８スロット１０極モータの誘起電圧位相とキャリア位相のずらし方を示している。キャリア位相の欄に書かれているアルファベットが同じものは、キャリア位相が同一であることを示している。キャリア位相のずらし方（１）もキャリア位相のずらし方（２）も電流ベクトルが０になる電流についてはキャリア位相が同じである。

Table 4 shows how to shift the induced voltage phase and the carrier phase of the 8-phase 8-slot 10-pole motor. Those having the same alphabet written in the column of carrier phase indicate that the carrier phase is the same. Both the carrier phase shifting method (1) and the carrier phase shifting method (2) have the same carrier phase for the current having a current vector of zero.

  However, since the carrier phase shift method (2) has the same carrier phase for two currents, there are four carriers whose phases can be changed. On the other hand, since the carrier phase shift method (1) has the same carrier phase for the four currents, there are two carriers whose phases can be changed.

  Accordingly, the carrier phase (2) has a greater degree of freedom with respect to how the carrier phase is shifted, and the effect of reducing iron loss can be increased. Thus, the greater the number of carriers whose phase can be changed with respect to the current, the greater the degree of freedom in how to shift the carrier phase. In other words, the smaller the number of currents relative to the number of carriers whose phase can be changed, the greater the degree of freedom in how to shift the carrier phase. Therefore, if the smallest value K among the divisors of the number of phases m is the number of currents per carrier whose phase can be changed, and if the carriers are the same with respect to currents having different phases of 360 / K [°], the carrier phase The degree of freedom of shifting can be maximized.

  As the angle for shifting the carrier phase, by providing a map of the carrier phase shift angle on the rotation speed / torque curve, the optimum carrier phase shift angle can always be selected according to the operating state of the motor.

  According to the present embodiment described above, by setting K to the smallest value other than 1 among the divisors of m, the number of groups that should have the same carrier phase becomes the largest, and the carrier phase among many groups. Therefore, there is an effect that the degree of freedom in shifting the carrier phase is increased and the magnetic flux ripple can be further reduced.

1-9 Teeth 1A-9A, 1B-9B Coil 10 Inverter 11 Neutral point

Claims (5)

When an integer of 3 or more is m,
A motor with m teeth, each with two coils,
an inverter for supplying m-phase current from m half bridges to the motor,
In the motor, one coil wound around one tooth and one coil wound around another tooth are connected in series, and induced voltage phases generated in two coils connected in series are 360 / m [ °] different motors,
A motor / inverter system, wherein the currents flowing through the two coils wound around each tooth have different phases of the PWM carrier wave in the inverter.   The motor / inverter system according to claim 1, wherein the number of turns of two coils wound around each tooth of the motor is equal. Assuming that one of the divisors of m other than 1 and m is K, among the m voltage command vectors, K voltage command vectors that add up to 0 are grouped together,
3. The motor / inverter system according to claim 1, wherein the phase of the carrier wave to be compared with the voltage command of the same group is the same, and the phase of the PWM carrier wave for each group is different.   4. The motor / inverter system according to claim 3, wherein K is the smallest value other than 1 among divisors of m.   A current of phase a-180 / m [°] is supplied to one coil of two coils wound around a tooth where an induced voltage of phase a [°] is generated, and phase a + 180 / m [is supplied to the other coil. The motor / inverter system according to claim 2, wherein a current of [°] is passed.

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