JP2010154320A - Radio communication apparatus and radio communication method - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To reduce operational complexity in reception processing for obtaining an information signal from a reception signal. <P>SOLUTION: A radio communication apparatus is provided with: an acquisition section 302 for acquiring an equalization coefficient of (g) bits ((g) is a natural number) for equalizing a channel gain; a processing section 301 for receiving a reception signal of (y) bits ((y) is a natural number) obtained by adding a perturbation signal, that is constituted of an integer multiple of a constant τ, is added to the information signal, performing quantization so that quantization widths q<SB>y</SB>and q<SB>g</SB>of the reception signal and the equalization coefficient satisfy 2<SP>s</SP>q<SB>y</SB>q<SB>g</SB>=τ, to perform multiplication processing relating to low order (s) bits ((s) is a natural number of less than (g+y)) of the reception signal and the equalization coefficient; a determination section 303 for determining a modulation point from a result of the multiplication processing; and a demodulation section 304 for performing demodulation processing on the basis of the determined modulation point. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a wireless communication apparatus and a wireless communication method.

  As a transmission method capable of speeding up existing wireless communication, a transmission method called MIMO (Multiple Input Multiple Output) that performs communication using a plurality of transmission antennas and a plurality of reception antennas has been proposed. The MIMO transmission method is a technique in which streams independent from a plurality of transmission antennas are multiplexed and transmitted at the same frequency, and an interference stream is separated by spatial filtering or maximum likelihood determination in a receiving apparatus (user). In the MIMO transmission method, spatial division multiplexing (SDMA) can be performed by beam forming, and communication can be performed simultaneously with a plurality of users.

  However, when there are a plurality of receiving apparatuses in the vicinity, it is difficult to separate the receiving apparatuses by beam forming in the transmitting apparatus (base station), and the frequency utilization efficiency is lowered. As a method for solving this problem, nonlinear precoding is known. Nonlinear precoding includes THP (Tomlinson-Harashima Precoding) and VP (Vector Perturbation).

  In the THP transmission processing, precoding processing is performed on a transmission signal addressed to a receiving device that receives interference between receiving devices, and interference components are thinned out in advance. However, since the signal after the precoding process has a component due to the propagation path, the transmission power may exceed the specified value.

  Therefore, a modulo arithmetic process is performed on the signal after the precoding process. The modulo operation processing is to perform a remainder operation of a predetermined value for the real part and the imaginary part of the input signal. This is equivalent to adding a perturbation signal having a value that is an integral multiple of a specified value to the real part and imaginary part of the input signal. A signal whose transmission power is suppressed within a specified value by modulo calculation is transmitted by beam forming.

  On the other hand, in the THP reception process, a modulo arithmetic process is performed again on a received signal in which the perturbation signal is added to a desired signal (information signal including data), and the perturbation signal component is removed to obtain an information signal. Is taken out.

  However, the received signal before the modulo arithmetic processing is caused by a perturbation signal component, and the signal point amplitude increases. Therefore, the reception processing circuit requires a large bit width, and the reception processing arithmetic amount increases.

  As a technique for solving such a problem, in the transmission process, a signal called a shaping signal is added to the transmission signal before the precoding process, and the real part component and the imaginary part component of the perturbation signal to be added in the modulo operation are defined. A technique for preventing the value from being exceeded is known (see, for example, Patent Document 1). With this method, the amount of computation for reception processing is reduced.

However, in such a conventional method, the calculation amount of transmission processing increases in order to appropriately set the shaping signal. An increase in the amount of computation in the transmission processing may increase processing delay and power consumption, and may cause a reduction in communication capacity.
US Pat. No. 5,854,812

  An object of the present invention is to provide a wireless communication apparatus and a wireless communication method capable of reducing the amount of calculation of reception processing for obtaining an information signal from a received signal.

  A wireless communication device according to an aspect of the present invention includes an acquisition unit that acquires an equalization coefficient of g bits (g is a natural number) that equalizes a channel gain, and y bits (y is a natural number) obtained by adding a perturbation signal to the information signal. ), A processing unit that performs a multiplication process related to lower s bits (s is a natural number less than g + y) of the received signal and the equalization coefficient, and a modulation point is determined from the result of the multiplication process A determination unit; and a demodulation unit that performs demodulation processing based on the determined modulation point.

  A wireless communication device according to an aspect of the present invention includes an acquisition unit that acquires propagation path information with N (N is a natural number) other wireless communication devices, and a kth other wireless communication device (k is 1 ≦ k ≦ N). A modulation unit that performs quadrature amplitude modulation on the k-th transmission data destined to an integer satisfying the above-mentioned integer), and receives the k-th transmission data subjected to the quadrature amplitude modulation as an s-bit digital value. Is an integer satisfying 1 ≦ m <k) and a lower ε bit (ε is an integer satisfying s + 1 ≦ ε <s + λ) of λ bits (λ is a natural number) based on the propagation path information. A processing unit that performs a subtraction process for subtracting a predetermined s bit of the multiplication processing result from the k-th transmission data by an s bit without a carry, and the propagation path information is added to the transmission data subjected to the subtraction process. Based beamforming weight A beam forming unit for multiplying, in which and a transmission unit which transmits the transmission data to which the beamforming weights are multiplied to the other wireless communication device.

  According to the present invention, it is possible to reduce the amount of calculation of reception processing for obtaining an information signal from a reception signal.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings.

  (First Embodiment) FIG. 1 shows a configuration of a multi-user environment MIMO communication system including a radio communication apparatus according to a first embodiment of the present invention. The multi-user environment MIMO communication system includes a transmission device 100 and K reception devices 110 (K is an integer of 1 or more).

The transmission apparatus 100 includes a digital signal processing unit 101, k (k is an integer satisfying 1 ≦ k ≦ K) analog circuit units 102, upper information signal processing units 103, and N _t (N _t is an integer of 1 or more). The antenna 104 is provided.

  The receiving apparatus 110 includes a digital signal processing unit 111, one or more analog circuit units 112, an upper information signal processing unit 113, and one or more antennas 114. The number of antennas of each receiving device 110 may be different.

  FIG. 2 shows a schematic configuration of the digital signal processing unit 101 of the transmission apparatus 100. The digital signal processing unit 101 includes a channel information acquisition unit 201, a modulation unit 202, a non-linear precoding unit 203, and a beam forming unit 204.

ここで、[]^Ｔは転置行列を表す。 A transmission process in the transmission apparatus (base station) 100 will be described. The modulation unit 202 is notified of N _s information signals (N _s is a natural number equal to or less than k) from the higher-order information signal processing unit 113, performs quadrature amplitude modulation (QAM), and is expressed by Equation 1 below. Such an N _s- dimensional transmission signal vector s is generated and notified to the non-linear precoding unit 203.

Here, [] ^T represents a transposed matrix.

The N _s information signals may be subjected to any randomization, error correction code encoding, and bit interleaving as long as the receiving apparatus 110 can perform derandomization, error correction code decoding, and bit deinterleaving. Further, it is assumed that a signal directed to a certain receiving device 110 is subjected to the same modulation every time and the modulation method is known to the receiving device 110.

In the following, for simplicity of explanation, it is assumed that the number of receiving apparatuses 110 to be spatially multiplexed is the same as the number of antennas 104 of the transmitting apparatus 100 and one stream is transmitted to each of the receiving apparatuses 110 to be spatially multiplexed. That is, K = N _t = N _s . Although the description will be made assuming that the number of the antennas 114 of all the receiving apparatuses 110 is one, the number of the antennas 114 is not limited to one.

The channel information acquisition unit 201 notifies the K × N _t channel matrix H acquired in advance to the nonlinear precoding unit 203 and the beam forming unit 204. The propagation path matrix H may indicate an impulse response of the propagation path between the transmission / reception apparatus antennas, or may quantize the impulse response, and may be any that represents the state of the propagation path.

  As a method for obtaining the propagation path matrix H, an analog value may be fed back, an index of a sample having the highest correlation with the quantized one may be fed back, and the transmission apparatus 100 is known to be transmitted from the reception apparatus 110. The signal may be received and estimated, and is not limited to the above method. In the example described in the present embodiment, single carrier transmission is handled. However, the present invention can be similarly applied to multicarrier transmission represented by OFDM (Orthogonal Frequency Division Multiplexing).

ただし、[]^Ｈはエルミート行列を表す。また、数式２のＱはユニタリ行列、Ｒは上三角行列である。Ｒは次式で表わされる。

When transmission is performed according to ZF (Zero-Forcing) standard THP (Tomlinson-Harashima Precoding), weights used in the nonlinear precoding unit 203 and the beamforming unit 204 are determined using QR decomposition of the channel matrix H. . Specifically, by performing QR decomposition on the Hermitian matrix of the propagation path matrix H, the following Equation 2 is obtained.

However, [] ^H represents a Hermitian matrix. In Equation 2, Q is a unitary matrix and R is an upper triangular matrix. R is represented by the following formula.

ただし、ｎはＫ次元雑音ベクトルであり、[]^＊は複素共役を表す。数式４において、Ｒ^Ｈ中のｒ^＊ _ｌｋ（ｌ＜ｋ）はｌ番目の受信装置１１０からｋ番目の受信装置１１０への干渉成分であり、後に順序付けられている受信装置１１０ほど多くの干渉を受けることを示している。 The beamforming unit 204 sets the unitary matrix Q as the transmission beamforming weight W _THP . At this time, the K-dimensional received signal vector y in the 1st to Kth receiving apparatuses 110 is expressed by the following Equation 4.

Here, n is a K-dimensional noise vector, and [] ^* represents a complex conjugate. In Equation 4, r ^* _lk (l <k) in R ^H is an interference component from the l-th receiving device 110 to the k-th receiving device 110, and _causes more interference as the receiving device 110 later ordered. Indicates that you will receive.

Therefore, the nonlinear precoding unit 203 in THP uses the Hermitian matrix ^RH of the upper triangular matrix R obtained by Equation 2 as a precoding matrix in order to remove the interference component, and performs precoding processing represented by Equation 5 below. To obtain a K-dimensional transmission signal vector s ′.

  As a result, the receiving apparatus 110 can receive a signal in which the interference component is canceled. However, since the transmission signal vector s ′ in Expression 5 has a component due to the propagation path, there is a possibility that the transmission power exceeds the specified value. Therefore, the non-linear precoding unit 203 when THP is used keeps the transmission power within a specified value by performing modulo operation on each component of the transmission signal vector s ′ in addition to the precoding of Expression 5.

κ_Ｍは送信電力の平均値を正規化するための係数であり、例えばＱＰＳＫ（Ｍ＝４）、１６ＱＡＭ（Ｍ＝１６）、６４ＱＡＭ（Ｍ＝６４）それぞれに対して、

となる。κ_Ｍはさらに、モジュロ演算により信号点が前記領域内に一様分布することを考慮し、送信電力の平均値を正規化するための値としてもよく、使用する全送信アンテナでの総送信電力を既定値とするための値として設定してもよい。 Modulo calculation means that the modulation multi-value number is M, the reference value is τ _M , and the signal point is within the range of ± τ _M square, so that the real part and the imaginary part are components of integer multiples of 2τ _M respectively. To add a perturbation signal. Here, τ _M can be expressed as, for example, Equation 6 below.

against kappa _M is a coefficient for normalizing the mean value of the transmission power, for example, QPSK (M = 4), 16QAM (M = 16), 64QAM (M = 64) , respectively,

It becomes. kappa _M further considering that the signal point by modulo uniformly distributed in the region, the total transmit power of the entire transmission antenna is good, use a mean value of the transmission power as a value for normalizing May be set as a value for setting as a default value.

ただし、

であり、

は負方向の整数に丸めこむ処理を表す。このように、非線形プリコーディング部２０３によりモジュロ演算を施されたＫ次元送信信号ベクトルｓ"は次式で表わされる。

The value of the tau _M may be greater than _M to (√M-1) / κ be the values of all of the modulation point M pieces falls within the region, not limited to the above values. Results in the case of performing a modulo operation to a reference value 2.tau _M against complex signal u is expressed by the following equation.

However,

And

Represents the process of rounding to a negative integer. As described above, the K-dimensional transmission signal vector s ″ subjected to the modulo operation by the nonlinear precoding unit 203 is expressed by the following equation.

FIG. 3 shows the configuration of the nonlinear precoding unit 203 when K = 3. A signal 12 (corresponding to the signal s _k ′) after sequentially performing interference suppression on the input signal 11 (corresponding to the signal s _k (k = 1, 2, 3)) is a modulo arithmetic unit 401. Shows a state in which a modulo operation is performed and a signal 13 (corresponding to the signal s _k ") is output.

FIG. 4 shows the movement of the transmission signal point on the IQ plane by the above-described processing of the nonlinear precoding unit 203 when QPSK is used. A region indicated by a dot in the center of FIG. 4 is a region of ± τ _M square including _M modulation points of the original constellation.

The signal 11 (signal s _k ) input to the non-linear precoding unit 203 is subjected to pre-coding expressed by Equation 5, and moves to a point indicated by a signal 12 (signal s _k ′) in the figure. For modulo arithmetic to keep the signal point of the signal 12 to ± tau _M square region indicated by dots, -1 times the real axis direction 2.tau _M, -2-fold 2.tau _M imaginary axis added respectively (i.e. In Equation 12, p _{Re, k} = -1, p _{Im, k} = -2), and signal 13 (signal s _k ").

ただし、diag(g)は対角行列を表わす。数式１３で示される受信信号から所望の信号を得る受信処理に関しては後述する。 By replacing s in Equation 4 with s "represented by Equation 11, the received signal y at the receiving apparatus 110 in THP can be expressed by the following equation. In the following, noise n is ignored.

However, diag (g) represents a diagonal matrix. A reception process for obtaining a desired signal from the reception signal represented by Expression 13 will be described later.

  In the present embodiment, the ZF normative THP has been described for the sake of simplicity, but it may be an MMSE (Minimum Mean Square Error) norm or other norms. The transmission signal vector s ″ represented by Equation 11 processed by the nonlinear precoding unit 203 is notified to the beamforming unit 204.

ここでγは総送信電力を規格化するための係数であり、次式で表わされる。

ただし、‖‖はユークリッドノルムを表す。１〜Ｋ番目の受信装置におけるＫ次元受信信号ベクトルｙおよび平均受信ＳＮＲ（Signal-to-Noise Ratio：信号対雑音電力比）は次式で表わされる。

Next, the processing of the non-linear precoding unit 203 and the beam forming unit 204 when transmitting by VP (Vector Perturbation) based on ZF will be described. The ZF-standard transmission beamforming weight W _ZF is a generalized inverse matrix H ^H (HH ^H ) ^{−1 of the} propagation path matrix H. At this time, the K-dimensional signal vector x after transmission beamforming weight multiplication is expressed by the following equation.

Here, γ is a coefficient for normalizing the total transmission power, and is expressed by the following equation.

However, ‖‖ represents the Euclidean norm. The K-dimensional received signal vector y and the average received SNR (Signal-to-Noise Ratio) in the 1st to Kth receivers are expressed by the following equations.

ただし、ｐ_ＶＰは摂動信号を成分とするＫ次元摂動ベクトルであり、次式で表わせる。

ただし、ｐ′_Re,k、ｐ′_Im,k（１≦ｋ≦Ｋ）は整数である。ここでも、説明の簡単のためにＺＦ規範のＶＰについて説明を行ったが、ＭＭＳＥ規範やその他の規範であってもよい。 From Equation 16, it can be seen that the reception quality deteriorates when the value of γ is large. Conversely, if the value of γ can be reduced, the average received SNR can be increased and the communication quality can be improved. Therefore, a perturbation signal having a component that is an integral multiple of the reference value 2τ _M is added to the real part and imaginary part of each component of the transmission signal vector s so that the value of γ is minimized in the VP nonlinear precoding process. . Here, τ _M is a value determined by a modulation method or the like, similar to that described in the THP transmission process. The transmission signal vector after this non-linear precoding process can be expressed by Equation 17 below.

Here, p _VP is a K-dimensional perturbation vector whose component is a perturbation signal, and can be expressed by the following equation.

However, p ′ _{Re, k} and p ′ _{Im, k} (1 ≦ k ≦ K) are integers. Here, the ZF norm VP has been described for the sake of simplicity, but an MMSE norm or other norms may be used.

とする。図５に示す例の場合、実軸方向に２τ_Ｍの−１倍を、虚軸方向に２τ_Ｍの−２倍をそれぞれ加えた（すなわち、数式１８においてｐ′_Re,k＝−１、ｐ′_Im,k＝−２とした）信号点（信号１５）に相当する。 FIG. 5 shows an example of processing performed by the nonlinear precoding unit 203 in VP using QPSK. As shown in FIG. 5, and those arranged infinitely 2.tau _M intervals replication of ± tau _M square area in the original constellation on the IQ plane called extended constellation 601. In VP nonlinear precoding section 203, the input signal 11 (signal s _k) and the symbol shown in FIG. 5, selects the signal points satisfying Equation 18 from the signal points arranged at 2.tau _M intervals, which signal

And In the example shown in FIG. 5, a -1-fold 2.tau _M in real axis direction, the imaginary axis plus respective -2 times 2.tau _M (i.e., p _'Re in Equation 18, k = -1, p _'Im, and a k = -2) corresponds to the signal point (signal 15).

はビームフォーミング部２０４へ通知される。 Transmission signal vector expressed by Equation 17 processed by the nonlinear precoding unit 203

Is notified to the beam forming unit 204.

  The beam forming unit 204 multiplies the notified transmission signal vector by the transmission beam forming weight, and notifies the analog circuit 102 of the signal vector obtained thereby.

  The analog circuit 102 performs processing for converting the notified digital signal into an analog signal, and transmits the signal by notifying the antenna 104 of the output. The analog circuit 102 at the time of transmission has a general configuration including a filter, a digital analog (D / A) converter, a frequency modulator, and a power amplifier, and detailed description thereof is omitted.

  Next, reception processing in the reception device 110 will be described. FIG. 6 shows a schematic configuration of the digital signal processing unit 111 of the receiving apparatus 110. The digital signal processing unit 111 includes a nonlinear reception processing unit 301, a channel equalization coefficient information acquisition unit 302, a determination unit 303, and a demodulation unit 304.

If the receiving device 110 the antenna 114 receives a reception signal shown in Equation 13, the processing is performed in the analog circuit 112, it is notified to the digital signal processor 111 as a received signal y _k of the digital. The analog circuit 112 has a general configuration including a low noise amplifier, an analog / digital (A / D) converter, and a filter, and detailed description thereof is omitted.

However, in the quantization process in the analog / digital (A / D) converter, when the quantization width is equal, the number of bits to be encoded is b, and the quantization width is q, [−2 ^b−1 q, It is assumed that the range of 2 ^b-1 q) can be expressed.

  Further, when the number of antennas is plural, any receiving method such as maximum ratio combining reception may be used. Hereinafter, the noise is ignored, and the reception process of the kth (1 ≦ k ≦ K) -th receiving device 110 will be described.

ここで、数式２１中のｇ_ｋはチャネル利得であり、上述したＺＦ規範のＴＨＰとＶＰではそれぞれｒ^＊ _ｋｋと１／√γとに置き換えて考えることができる。また、数式２１中の２τ_Ｍ（ｐ_Re,k＋ｊｐ_Im,k）はＴＨＰ及びＶＰの送信処理において付加された摂動信号を示す。チャネル利得ｇ_ｋの値は送信装置１００より送信された既知信号を受信装置１１０が受信することによりチャネル等化係数情報取得部３０２において推定してもよく、送信装置１００より通知されてもよく、いかなる方法であってもよい。 The received signal y _k notified to the digital signal processing unit 111 of the k-th receiving device can be expressed by the following Equation 21.

Here, g _{k in} Equation 21 is a channel gain, and can be considered by replacing with R ^* _kk and 1 / √γ in the ZF norm THP and VP described above, respectively. Further, 2τ _M (p _{Re, k} + jp _{Im, k} ) in Equation 21 represents a perturbation signal added in the THP and VP transmission processing. The value of the channel gain g _k may be estimated by the channel equalization coefficient information acquisition unit 302 when the reception device 110 receives a known signal transmitted from the transmission device 100, or may be notified from the transmission device 100. Any method may be used.

FIG. 7 shows a configuration of the nonlinear reception processing unit 301 ′ in the reception method according to the comparative example. The non-linear reception processing unit 301 ′ in the reception method according to the comparative example includes a channel equalization unit 701 and a modulo arithmetic unit 702. Channel equalizer 701 performs channel equalization processing on the received signal y _k. Specifically, using the channel gain g _k (signal 22) between the antenna 104 and the antenna 114 acquired by the channel equalization coefficient information acquisition unit 302, the received signal y _k (signal 21) expressed by Equation 21 is used. ) Is normalized. If the normalized signal is z _k (signal 23), z _k is expressed by the following Equation 22.

From Equation 22, it can be seen that the signal z _k includes a desired signal (information signal) s _k and a perturbation signal 2τ _M (p _{Re, k} + jp _{Im, k} ) added in the transmission process of the transmission apparatus 100. Therefore, the signal z _k is input to the modulo arithmetic unit 703. As described in the THP transmission process, the modulo arithmetic unit 703 performs a process of storing the input signal in the ± τ _M square area. As a result, the perturbation signal component of the second term on the right side of Equation 22 is removed, and the desired signal s _k (signal 24) can be extracted.

FIG. 8 shows movement of signal points on the IQ plane by the processing of the nonlinear reception processing unit 301 ′. Received signal _{y k} (signal 21) is moved to a point of the signal _{z k} (signal 23) by the channel equalization, 1x 2.tau _M to the real axis direction by a modulo operation, double 2.tau _M imaginary axis direction Each is added to show signal s _k (signal 24).

Here, it can be seen that the signal point amplitude of the signal z _k (signal 23) increases as the value of the channel gain g _k decreases. In communication systems using nonlinear precoding typified by THP and VP, the bit width necessary to represent the signal point of this signal z _k (signal 23) increases, and the amount of computation increases accordingly. Is a problem. In this embodiment, such a problem is solved.

Hereinafter, the signals y _k , 1 / g _k , z _k , and s _k will be expressed as signals 21, 25, 23, and 24, and the real part and the imaginary part will be considered in binary expression. FIG. 9 shows an example of quantization of the signal 24 when QPSK is used as the modulation method. A state in which both the real part and the imaginary part are expressed by 2 bits of code is shown. If the number of bits representing the signal 24 is b _s and the quantization width is q _s , these satisfy the following relationship.

ここで、ｄ_ｙ、ｄ_ｇは０以上の整数であり、信号２１、２５を２進数表現するために準備されたビット数ｂ_ｙ、ｂ_ｇのうち、ｑ_ｓよりも小さな範囲を表現するためのビット数をそれぞれ示す。 Further, if the number of bits representing each of the signals 21 and 25 is b _y and b _g (b _y ≧ b _s , b _g ≧ b _s ) and the quantization widths are q _y and q _g , the signals 24, 21, 25 The quantization relationship is designed so as to satisfy the relationship of the following equation, as indicated by 24-1, 21-1, and 25-1 in FIG.

Here, d _y and d _g are integers of 0 or more, and in order to express a range smaller than q _s among the bit numbers b _y and b _g prepared for binary representation of the signals 21 and 25. Indicates the number of bits.

  In the following, when explanation is given in binary representation, only one of the real part and the imaginary part will be explained, but the same applies to the other part.

FIG. 11 shows the relationship of bits when channel equalization is performed by multiplication of the signal 21 and the signal 25 in binary representation. The bit sequence of y _k and 1 / g _k shown in FIG. 11 represents the low-order bits from the right, and when the relationship of Expression 24 is satisfied, the width that can be expressed when the bits are expressed in decimal numbers is q from the right. _It is divided into three in the meaning of less than _s , q _s to τ _M , τ _M or more, and the pattern is changed.

Due to the relationship between the bits before and after the multiplication, the signal 23 can be divided similarly, and the number of bits is also shown as A, B, and C as shown in FIG. The number of bits shown as A, B, and C in the number of bits b _z (= b _y + b _g ) for expressing the signal 23 satisfies the relationship of Equation 25 below.

これより、信号２３は次式で表せる。

Therefore, if the quantization width of the signal 23 is q _z , 2 ^{dy + dg} q _z = q _s is obtained, and by substituting this into the equation 23, the following relationship is obtained.

Thus, the signal 23 can be expressed by the following equation.

Equation 27 expresses the lower (b _s + d _y + d _g ) bits and the upper (b _y + b _g −b _s −) among b _z (= b _y + b _g ) bits representing the signal 23 (signal z _k ). d _y −d _g ) bits represent the signal 24 (signal s _k ) and the perturbation signal 2τ _M (p _{Re, k} + jp _{Im, k} ) independently of each other.

Therefore, the nonlinear reception processing unit 301 in the present embodiment includes a digital signal lower-order multiplier 1201 as shown in FIG. 12, and the digital signal lower-order multiplier 1201 is connected to the signal 21 as shown in FIG. The multiplication of the signal 25 is performed only for the lower (b _s + d _y + d _g ) bits, and only the upper b _s bits of the (b _s + d _y + d _g ) bits are output. The digital signal lower-order multiplier 1201 does not process or output the higher-order (b _y + b _g −b _s −d _y −d _g ) bits. In FIG. 11, the processing portion that can be reduced as compared with the conventional multiplier is indicated by hatching, and it can be seen that the amount of calculation is reduced accordingly.

Although the most significant bit indicating the sign, the most significant bit of the signal 23 to indicate ^{-2 bz-1} _{q z,} which is also included in the second term second equation right side of Equation 27, the signal 24 (signal s Note that it is independent of _k ).

For example, in the example shown in FIG. 11, when multiplying the 4-bit signal 21-2 (signal y _k ) and the 4-bit signal 25-2 (signal 1 / g _k ), the multiplication result is only the lower 3 bits. It only has to be obtained. Accordingly, the digital signal lower multiplier 1201 performs a multiplication process for obtaining only the lower 3 bits without obtaining all 8 bits of the multiplication result. Then, only the upper 2 bits representing the information signal (s _k ) among the 3 bits of the processing result are output.

  As described above, the digital signal lower-order multiplier 1201 can remove the perturbation signal component added by the nonlinear precoding unit 203 of the transmission apparatus 100 and extract a desired signal (information signal) 24.

FIG. 14 shows how signal points are moved by the processing of the nonlinear reception processing unit 301. The received signal y _k (signal 21) moves directly to the desired signal (information signal) s _k (signal 24). Since the signal amplitude point does not increase as shown in FIG. 8 and the bit width does not increase, the amount of calculation can be reduced.

The relationship between the quantization widths of the signals 21 and 22 is not limited to Expression 24, and it is only necessary to satisfy the following expression.

Further, when not the signal 22 (signal g _k ) but the signal 25 (signal 1 / g _k ) is notified from the transmission apparatus 100 as the channel gain, all b _g bits may be notified, and the digital signal lower part multiplication is performed. 2 ^{dg + bs} q _g expressed only by the lower-order d _g + b _s bits related to the processing in the device 1201 may be notified. The signal s _k (signal 24) extracted by the above-described processing in the nonlinear reception processing unit 301 is notified to the determination unit 303.

Determination unit 303, to the signal s _k notified from the non-linear receive processing unit 301 determines one of the points of the signal points of the modulation multi-level number M of modulation scheme applied (determined). The determination method at this time includes a method using LLR (Log-Likelihood Ratio), and any method may be used. The determination result is notified to the demodulation unit 304.

  The demodulation unit 304 performs demodulation processing based on the determination result notified from the determination unit 303 and notifies the higher information signal processing unit 113 as a digital data sequence.

  The upper information signal processing unit 113 extracts information by performing bit deinterleaving, error correction code decoding, derandomization, and the like on the notified signal as necessary.

  Thus, according to the present embodiment, when performing multi-user MIMO communication using nonlinear precoding, in the reception process, the configuration of the nonlinear reception processing unit 301 ′ according to the comparative example illustrated in FIG. As can be seen from the comparison with the configuration of the nonlinear reception processing unit 301 in this embodiment, the reception processing in this embodiment is simplified, and the amount of reception processing calculation can be reduced.

The channel equalization coefficient information acquisition unit 302 may output the channel equalization coefficient 1 / g _k (signal 25) instead of the channel gain g _k (signal 22) to the nonlinear reception processing unit 301. .

  (Second Embodiment) The configuration of a radio communication apparatus according to the second embodiment of the present invention is the same as that shown in FIG. 1, and the transmission process is also the same as that of the first embodiment. However, in the present embodiment, as shown in FIG. 15, the digital signal processing unit 111 of the receiving apparatus 110 further includes an inter-modulation-mode signal amplitude normalization unit 1501 and multipliers 1502 and 1503. Different from form.

  The receiving apparatus 110 according to the present embodiment changes the quantization width design of the nonlinear reception processing unit 301 of the receiving apparatus 110 even in a communication system to which adaptive modulation using an appropriate modulation scheme is applied according to the state of the propagation path. Without being limited, the nonlinear reception processing described in the first embodiment can be applied. Hereinafter, the reception processing related to the inter-modulation-system signal amplitude normalization unit 1501 will be described.

In a wireless communication system that employs adaptive modulation, the modulation scheme is not fixed, and is appropriately selected by the transmission apparatus 100 according to the state of the propagation path. A received signal y _k in the k th receiving apparatus 110 is expressed by Equation 21.

However, when adaptive modulation is applied, τ _{M in} Equation 21 is different for each modulation method selected by the transmission apparatus 100 and is not a constant value. In addition, it is not easy to adaptively change the quantization width of the receiving device 110. For this reason, when the modulation method changes due to adaptive modulation, the desired signal (information signal) and the perturbation signal are completely separated from each other with a certain bit as a boundary as shown in Equation 27 shown in the first embodiment. I can't.

  Therefore, inter-modulation-system signal amplitude normalization section 1501 shown in FIG. 15 acquires information on the modulation system applied to the received signal, and outputs the held normalization coefficient to multiplier 1502 in accordance with the modulation system. . Here, as a method for receiving apparatus 110 to acquire information on the modulation scheme, notification from transmitting apparatus 100 may be made in advance, or receiving apparatus 110 may estimate.

The received signal y _k (signal 21) notified from the analog circuit 112 to the digital signal processing unit 111 is multiplied by the normalization coefficient output from the inter-modulation scheme signal amplitude normalization unit 1501 by the multiplier 1502, and is transmitted between the modulation schemes. Is normalized with different signal point amplitude criteria.

ここで、αは定数であり、ｓ_Ｍ，ｋ＝αｓ_ｋ／τ_Ｍである。また、上記数式２９で示される信号２６に対し、チャネル利得ｇ_ｋ（信号２２）を用いてチャネル等化された信号ｚ_k′は以下の数式３０で表わされる。

ここで数式２３におけるτ_Ｍをαに、ｓ_ｋをｓ_Ｍ、ｋに置き換えた次式の関係を満たすように受信装置１１０のｓ_Ｍ、ｋに対する量子化幅ｑ_ｓＭを設計する。

これは、変調方式に依存しない区間［−α、α）を、ｂ_ｓＭビットで表現できることを意味する。また、信号２６とチャネル利得ｇ_ｋの逆数１／ｇ_ｋである信号２５それぞれの量子化幅ｑ_ｙ′、ｑ_ｇに関しても数式２４と同様に次式で示す関係を満たすものとする。

ただし、ｄ_ｙ′、ｄ_ｇは０以上の整数であり、信号２６と信号２５を２進数表現するために準備されたビット数ｂ_ｙ′、ｂ_ｇのうち、ｑ_ｓＭよりも小さな範囲を表現するビット数をそれぞれ示す。以上の関係より、上記第１の実施形態における数式２７と同様に次式の関係が成り立つ。

Specifically, the signal 21 is multiplied by a normalization coefficient α / τ _M as shown in the following Expression 29. Assuming that the normalized signal 26 is y _k ′, the signal 26 is expressed as follows.

Here, α is a constant, and s _{M, k} = αs _k / τ _M. Further, a signal z _k ′ that is channel-equalized using the channel gain g _k (signal 22) with respect to the signal 26 expressed by the above-described expression 29 is expressed by the following expression 30.

Here, the tau _M in equation 23 alpha, a _{s k s M, s M} of the receiving device 110 so as to satisfy the following equation relationships replaced with _k, designing the quantization width _{q sM} for _k.

This means that an interval [-α, α) that does not depend on the modulation scheme can be expressed by b _sM bits. Further, the quantization widths q _{y ′} and q _g of the signal 26 and the signal 25 which is the reciprocal 1 / g _k of the channel gain g _k also satisfy the relationship expressed by the following equation similarly to the equation 24.

However, d _{y ′} and d _g are integers of 0 or more, and express a range smaller than q _sM among the bit numbers b _{y ′} and b _g prepared for binary representation of the signal 26 and the signal 25. Indicates the number of bits to be used. From the above relationship, the following relationship is established in the same manner as Equation 27 in the first embodiment.

Among the b _{z ′} (= b _{y ′} + b _g ) bits representing z _k ′, the lower (b _sM + d _{y ′} + d _g ) bits and the upper (b _{y ′} + b _g −b _sM −d _{y ′} − d _g ) bits represent the desired signal s _{M, k} (= αs _k / τ _M ) and the perturbation signal 2α (p _{Re, k} + jp _{Im, k} ), each independently normalized. It was done. Therefore, in the nonlinear reception processing unit 301, when the signal 26 and the signal 22 are multiplied, only the low-order (b _sM + d _{y ′} + d _g ) bits are processed, and the (b _sM + d _{y ′} + d _g ) bits are included. Only the upper _bsM bits are output, and this output signal is referred to as a signal 27. The non-linear reception processing unit 301 does not process or output the upper (b _{y ′} + b _g −b _sM −d _{y ′} −d _g ) bits.

As a result, the perturbation signal component added in the nonlinear precoding unit 203 of the transmission apparatus 100 is removed using the digital signal lower-order multiplier 1201 described in the first embodiment, and the signal 27 (normalized) The desired signal s _{M, k} ) can be extracted.

The signal 27 extracted by the above-described processing in the nonlinear reception processing unit 301 is multiplied by the inverse number τ _M / α of the previous normalization coefficient output from the inter-modulation-system signal amplitude normalization unit 1501 in the multiplier 1503. . As a result, a signal 23 (desired signal s _k ) as shown in the following equation is obtained.

  The desired signal 23 obtained in this way is notified to the determination unit 303. Since the process after the determination part 303 is the same as that of the said 1st Embodiment, description is abbreviate | omitted.

The relationship between the quantization widths of the signals 25 and 26 is not limited to Expression 32, and it is only necessary to satisfy the following expression.

  As described above, according to the present embodiment, the digital signal lower-order multiplier 1201 and the determination unit in the nonlinear reception processing unit 301 described in the first embodiment are also applied to the wireless communication system to which adaptive modulation is applied. The present invention can be applied without changing the design of 303, and the same effect as in the first embodiment can be obtained.

  (Third Embodiment) The configuration of a wireless communication apparatus according to the third embodiment of the present invention is the same as that of the first embodiment shown in FIG. The difference between the present embodiment and the first embodiment is that the present embodiment is applied to a communication system to which a modulation mapping method called flipped constellation is applied. Hereinafter, transmission / reception processing in the communication system to which the flipped constellation is applied will be described.

  In the first embodiment, in the case of VP, the nonlinear precoding unit 203 of the transmission apparatus 100 performs the nonlinear precoding process represented by Expression 17 in order to maximize the average received SNR. In this process, for example, when QPSK (M = 4) is used as the modulation method, the VP nonlinear precoding process selects an optimum point from the signal point candidates corresponding to the extended constellation shown in FIG. It is also shown that it is equivalent to doing.

Here, when replicating a region of ± τ _M square of the original constellation, the one that is inverted so as to be sequentially inverted in the real axis direction and the imaginary axis direction at 2τ _M as shown in FIG. Called constellation.

  In the VP nonlinear precoding process using the flipped constellation, the optimum signal point that maximizes the received SNR is selected from the signal point candidates indicated by the same symbol among the signal points shown in FIG.

  In the following, processing of signals received by the receiving device when flipped constellation is used in the VP transmission processing of the transmitting device will be described in detail. In the following, for the sake of simplicity, the case where adaptive modulation is not applied will be described. However, even when adaptive modulation is applied, processing for normalizing the reference value of the signal amplitude between the modulation schemes as in the second embodiment. It is feasible by performing.

In the non-linear precoding process using the flipped constellation in the VP, if the transmission signal is the signal 14-1 (signal s _k ⁽⁰⁾ ) in FIG. 16, the signal point candidates indicated by circles are selected. One optimal point that satisfies Equation 18 is selected. This means that for any one of four signal points of the signal in FIG. 16 14-1 to 14-4 (signal ^{_{s k (0) ~s k (}} 3)), the real and imaginary parts in is equivalent to adding an optimal perturbation signal with integer multiples of the components of 4.tau _M. For this reason, the transmission signal after the nonlinear precoding processing in which the flipped constellation is applied to the transmission signal s _k ⁽⁰⁾ addressed to the kth receiving apparatus 110 can be expressed by the following equation.

ここで、信号ｙ_k ^F、１／ｇ_k、ｓ_k ^Fそれぞれを表現するビット数ｂ_ｙ、ｂ_ｇ、ｂ_ｓ（ｂ_ｙ≧ｂ_ｓ、ｂ_ｇ≧ｂ_ｓ）および量子化幅ｑ_ｙ、ｑ_ｇ、ｑ_ｓが、数式２４と次式の関係を満たすものとする。

When the transmission signal represented by Expression 36 is transmitted by the VP processing, the reception signal y _k ^F in the k-th receiving apparatus 110 can be expressed by the following expression.

Here, the signal ^{_{y k F, 1 / g k}} , s k F number of bits to represent each _{b y, b g, b s} (b y ≧ b s, b g ≧ b s) and quantization width _q y, It is assumed that q _g and q _s satisfy the relationship of Expression 24 and the following expression.

となり、以下の数式４０に示す関係が得られる。

As a result, the relationship of 2 ^{dy + dg} q _z = q _s regarding the number of bits b _z (= b _y + b _g ) and the quantization width q _z representing the signal z _k ^F as a result of channel equalization processing of y _k ^F is obtained. Satisfied,

Thus, the relationship shown in the following formula 40 is obtained.

Equation 40 expresses the lower (b _s + d _y + d _g ) bits and the upper (b _y + b _g −b _s −d _y −d _g ) among b _z (= b _y + b _g ) bits representing z _k. ) Bits independently represent the desired signal s _k ^F and the perturbation signal 4τ _M (p ^F _{Re, k} + jp ^F _{Im, k} ) in the flipped constellation.

Therefore, the digital signal lower-order multiplier 1201 included in the non-linear reception processing unit 301, as shown in FIG. 13, applies 1 / g _k to the signal 26 that is y _k ^F , as shown in FIG. Channel equalization is performed by performing multiplication of a certain signal 22. In this multiplication processing, only multiplication processing relating to the lower (b _s + d _y + d _g ) bits of the multiplication result is performed, and (b _s + d _y + d _g). ) Only the upper b _s +1 bits of the bits are output. Top _{(b y + b g -b s} -d y -d g) multiplication process and output for bit is not performed.

As a result, the digital signal lower-order multiplier 1201 can remove the perturbation signal component added by the nonlinear precoding unit 203 of the transmission apparatus 100 and extract the signal s _k ^F as the signal 27. The signal 27 extracted by the above-described processing in the nonlinear reception processing unit 301 is notified to the determination unit 303.

The relationship between the quantization widths q _y and q _g is not limited to Expression 24, but only needs to satisfy the following expression.

The determination unit 303 refers to the flipped constellation as shown in FIG. 16 for the signal 27 (signal s _k ^F ) notified from the nonlinear reception processing unit 301, and determines the modulation multilevel number M of the original constellation. One of the signal points is determined. The determination method at this time includes a method using LLR (Log-Likelihood Ratio), and any method may be used. The determination result is notified to the demodulation unit 304.

  The processing after the demodulator 304 is the same as that described in the first embodiment.

  As described above, even in the VP reception processing using the flipped constellation, the digital signal lower-order multiplier 1201 can be applied by optimally performing the quantization design in accordance with the perturbation signal. The same effects as those of the first and second embodiments can be obtained.

  (Fourth Embodiment) The configuration of a wireless communication apparatus according to the fourth embodiment of the present invention is the same as that in FIG. 1, and the reception process is the same as in any of the first to third embodiments. In the present embodiment, regarding THP transmission processing, the non-linear precoding unit 203 in the digital signal processing unit 101 is configured as shown in FIG. 18 using the digital signal lower-order multiplier 1801 shown in FIG. This is different from the first to third embodiments.

  The digital signal lower part multiplier 1801 outputs only the number of bits representing the transmission signal input to the nonlinear precoding unit 203 and processes only the lower bits related to this output. Hereinafter, transmission processing using THP according to the present embodiment will be described.

In the transmission process using THP, as described in the first embodiment, the nonlinear precoding unit 203 performs the nonlinear precoding process represented by Expression 11. At this time, to the transmission signal s _k of the k-th receiver 110 addressed, l (1 ≦ l ≦ k -1) th interference component xi] _lk from the transmission signal of the receiving device 110 destined, on shown in Equation 3 triangle Using the Hermitian matrix component of the matrix R, it is expressed by the following equation.

Xi] _lk shown in Equation 42, λ _lk, s _l "the number of bits representing the respective real and imaginary parts _{b ξ, b λ, b s} , a quantization width q _ξ, q λ, and _{q s} In the following description, only one of the real part and the imaginary part will be described when explaining in binary notation, but the same can be applied to the other part. The number of bits and the quantization width are shown, but the same applies to the transmission processing described here, and the value of the reference value τ _M is set to the real and imaginary axes on the IQ plane with respect to the modulation multilevel number M. Is set so that all M modulation points can be included in the region of ± τ _M square, that is, the following relationship is established.

ただし、ｄ_λは整数であり、ｄ_λが０より大きい場合は、λ_lkを２進数表現するために準備されたビット数ｂ_λのうち、ｑ_ｓよりも小さな領域を表現するためのビット数を示す。また、ｄ_λが０より小さい場合は、ｑ_λよりもｑ_ｓの方がｄ_λビット分小さな領域を表現できることを示す。このとき、２進数の乗算におけるビットの関係より、ｑ_ξとｑ_ｓについて次式の関係が成り立つ。

Further, q _λ and q _s satisfy the relationship of the following equation.

However, d _λ is an integer, and when d _λ is larger than 0, out of the number of bits b _λ prepared for expressing λ _lk in binary, the number of bits for expressing an area smaller than q _s Indicates. Further, when d _λ is smaller than 0, it indicates that q _s can represent a region smaller than q _λ by d _λ bits. At this time, the relationship of the following equation holds for q _ξ and q _{s based} on the bit relationship in binary multiplication.

これはξ_lkを表現するｂ_ξ（＝ｂ_λ＋ｂ_ｓ）ビットのうち、（ｄ_λ＋ｂ_ｓ＋１）ビット目よりも上位のビットが２τ_Ｍの整数倍の分解能で領域を表現していることを意味する。ＴＨＰを用いた場合の非線形プリコーディング部２０３の処理では、上記第１の実施形態で示したように、モジュロ演算を行う。このモジュロ演算は、上述の量子化設計を行っているため、（ｄ_λ＋ｂ_ｓ＋１）ビット目よりも上位のビットを削除することで容易に実現できる。そこで、図１７に示すデジタル信号下位部乗算器１８０１では、信号１３（信号ｓ_l"）と信号１４（信号λ_lk）を入力とし、ξ_lkを表現するｂ_ξ（＝ｂ_λ＋ｂ_ｓ）ビットのうち、（ｄ_λ＋ｂ_ｓ＋１）ビット目よりも上位のビットに関しては処理および出力を行わずに、下位の（ｂ_λ＋ｂ_ｓ）ビットにのみ関係する処理を行う。そして、得られた（ｂ_λ＋ｂ_ｓ）ビットのうち、上位のｂ_ｓビットだけを出力する。このｂ_ｓビットで出力される信号１５を信号ε_lkとする。 Therefore, the relationship of the following equation is obtained from Equation 44 and Equation 46.

This _{b ξ (= b λ + b} s) of the bits representing the xi] _lk, it expresses the region upper bits is an integer multiple of the resolution of 2.tau _M than _(d λ + _b s +1) th bit Means. In the processing of the non-linear precoding unit 203 when THP is used, modulo arithmetic is performed as shown in the first embodiment. This modulo operation can be easily realized by deleting the higher-order bits than the (d _λ + b _s +1) -th bit because the above-described quantization design is performed. Therefore, the digital signal lower part multiplier 1801 shown in FIG. 17, the signal 13 (signal s _l ") and an input signal 14 (signal lambda _lk), b representing the _{ξ lk ξ (= b λ +} b s) bit Among them, the processing related to only the lower (b _λ + b _s ) bits is performed without performing the processing and output for the upper bits than the (d _λ + b _s +1) -th bit, and obtained ( _b λ + b _s) of the bits, and outputs only the _{b s} significant bits. the signal 15 output by the _{b s} bits as signals epsilon _lk.

In the nonlinear precoding unit 203 illustrated in FIG. 18, the signal 14 (14-1 to 14-3) and the signal 15 (15-1 to 15-3) obtained from the signal 13 are sequentially applied to the input signal 11. 3 shows a state of the subtraction process, and the THP nonlinear precoding process represented by Expression 11 equivalent to the nonlinear precoding unit shown in FIG. 3 can be performed. However, the subtraction unit 16-1 to 16-3, and outputs only the lower of _{b s} bits of the result of the subtraction between the _{b s} bits. That is, the subtraction units 16-1 to 16-3 have a circuit configuration that does not support carry to the b _s bit.

  The transmission signal vector s "thus obtained is further subjected to signal amplitude correction for adjusting the total transmission power, and then notified to the beam forming unit 204. The processing after the beam forming unit 204 is described above. Since it is the same as that of embodiment of this, description is abbreviate | omitted.

  As described above, according to the present embodiment, by performing optimal quantization design, it is possible to realize nonlinear precoding processing with only a limited number of bits in THP transmission processing. Compared with the configuration of the THP nonlinear precoding unit according to the first embodiment shown in FIG. 3, the configuration of the nonlinear precoding unit according to the present embodiment shown in FIG. 19 is different from the multiplier in the modulo operation unit (401). Since the number of bits of the multiplier can be designed to be small, the circuit scale and the calculation amount can be reduced.

  In the above embodiment, description has been made using THP and VP as examples of nonlinear precoding, but nonlinear precoding other than THP and VP may be used.

  At least a part of the wireless communication device described in the above-described embodiment may be configured by hardware or software. When configured by software, a program for realizing at least a part of the functions of the wireless communication apparatus may be stored in a recording medium such as a flexible disk or a CD-ROM, and read and executed by a computer. The recording medium is not limited to a removable medium such as a magnetic disk or an optical disk, but may be a fixed recording medium such as a hard disk device or a memory.

  Further, a program for realizing at least a part of the functions of the wireless communication device may be distributed via a communication line (including wireless communication) such as the Internet. Further, the program may be distributed in a state where the program is encrypted, modulated or compressed, and stored in a recording medium via a wired line such as the Internet or a wireless line.

  Note that the present invention is not limited to the above-described embodiment as it is, and can be embodied by modifying the constituent elements without departing from the scope of the invention in the implementation stage. In addition, various inventions can be formed by appropriately combining a plurality of components disclosed in the embodiment. For example, some components may be deleted from all the components shown in the embodiment. Furthermore, constituent elements over different embodiments may be appropriately combined.

It is a block block diagram of the radio | wireless communication apparatus which concerns on the 1st thru | or 4th embodiment of this invention. It is a block block diagram of the digital signal processing part contained in the transmitter which concerns on the 1st thru | or 4th embodiment. It is a block block diagram of the nonlinear precoding part which concerns on the 1st Embodiment. It is a figure which shows an example of the movement of the signal point at the time of the transmission process of THP. It is a figure which shows an example of the movement of the signal point at the time of the transmission process of VP. It is a block block diagram of the digital signal processing part contained in the receiver which concerns on the said 1st Embodiment. It is a block block diagram of the nonlinear reception process part by a comparative example. It is a figure which shows an example of the movement of the signal point at the time of the reception process by a comparative example. It is a figure which shows an example of the quantization of a transmission signal. It is a figure which shows the relationship of the quantization of the input-output signal of a nonlinear reception process part. It is a figure which shows an example of the multiplication process of a digital signal. It is a block block diagram of the nonlinear reception process part which concerns on the 1st Embodiment. It is a figure which shows the relationship of the input / output bit number of the digital signal low-order part multiplier which concerns on the same 1st Embodiment. It is a figure which shows an example of the movement of the signal point at the time of the reception process which concerns on the 1st Embodiment. It is a block block diagram of the digital signal processing part contained in the receiver which concerns on the 2nd Embodiment of this invention. It is a figure explaining the flipped constellation which concerns on the 3rd Embodiment of this invention. It is a figure which shows the relationship of the input / output bit number of the digital signal low-order part multiplier contained in the nonlinear precoding part which concerns on the 4th Embodiment of this invention. It is a block block diagram of the nonlinear precoding part which concerns on the same 4th Embodiment.

Explanation of symbols

100 Transmitting apparatus 101, 111 Digital signal processing unit 102, 112 Analog circuit 103, 113 Upper information signal processing unit 104, 114 Antenna 110 Receiving apparatus

Claims (9)

An acquisition unit for acquiring an equalization coefficient of g bits (g is a natural number) for equalizing a channel gain;
A received signal of y bits (y is a natural number) with a perturbation signal added to the information signal is received, and a multiplication process related to the lower s bits (s is a natural number less than g + y) of the received signal and the equalization coefficient is performed. A processing unit;
A determination unit for determining a modulation point from the result of the multiplication process;
A demodulator that performs demodulation processing based on the determined modulation point;
A wireless communication device comprising:   2. The wireless communication according to claim 1, wherein the processing unit notifies the determination unit of high-order sd bits (d is an integer of 0 or more and less than s) among the s-bit multiplication processing results. apparatus. The perturbation signal is a signal in which a real part and an imaginary part are represented by an integer multiple of a predetermined constant τ (τ is a real number).
3. The quantization width representing the received signal is q _y , and the quantization width representing the equalization coefficient is q _g , wherein 2 ^s q _y q _g = τ is satisfied. Wireless communication device. The acquisition unit notifies the processing unit of the lower s bits of the g-bit equalization coefficient,
The radio processing apparatus according to claim 3, wherein the processing unit performs the multiplication process using the received signal and the lower s bits of the equalization coefficient. A normalization unit that obtains information on a modulation scheme applied to the received signal and obtains a normalization coefficient based on the modulation scheme;
A first multiplier for multiplying the received signal by the normalization coefficient and notifying the processing unit;
A second multiplier that multiplies the result of the multiplication process by the processing unit with the inverse of the normalization coefficient and notifies the determination unit;
The wireless communication apparatus according to claim 1, further comprising: An acquisition unit for acquiring propagation path information with N (N is a natural number) other wireless communication devices;
A modulator that performs quadrature amplitude modulation on the kth transmission data destined for the kth other wireless communication device (k is an integer satisfying 1 ≦ k ≦ N);
The quadrature amplitude-modulated k-th transmission data is received as an s-bit digital value, and m-th transmission data (m is an integer satisfying 1 ≦ m <k) and λ bits (λ is based on the propagation path information) (Multiple natural number) interference component and low-order ε bits (ε is an integer satisfying s + 1 ≦ ε <s + λ), and a predetermined s bits of the multiplication result are carried from the k-th transmission data. A processing unit for performing a subtraction process for subtracting with no s bits;
A beam forming unit that multiplies the transmission data subjected to the subtraction processing by a beam forming weight based on the propagation path information;
A transmission unit that transmits the transmission data multiplied by the beamforming weight to the other wireless communication device;
A wireless communication device comprising:   The wireless communication apparatus according to claim 6, wherein the predetermined s bits are upper s bits. Obtain an equalization coefficient of g bits (g is a natural number) for equalizing the channel gain,
Receiving a reception signal of y bits (y is a natural number) in which a perturbation signal is added to an information signal;
A multiplication process related to the lower s bits (s is a natural number less than g + y) of the received signal and the equalization coefficient;
A modulation point is determined from the result of the multiplication process,
A wireless communication method for performing demodulation processing based on the determined modulation point. Obtain propagation path information with N (N is a natural number) other wireless communication devices,
K-th transmission data destined for the k-th other wireless communication device (k is an integer satisfying 1 ≦ k ≦ N) is subjected to quadrature amplitude modulation,
Receiving the kth transmission data subjected to the quadrature amplitude modulation as an s-bit digital value;
Lower ε bits (ε is an integer satisfying s + 1 ≦ ε <s + λ) between m-th transmission data (m is an integer satisfying 1 ≦ m <k) and an interference component of λ bits (λ is a natural number) based on the propagation path information )
Performing a subtraction process of subtracting a predetermined s bit of the multiplication process result from the k-th transmission data by an s bit without a carry,
Multiplying the subtracted transmission data by a beamforming weight based on the propagation path information,
A wireless communication method for transmitting the transmission data multiplied by the beamforming weight to the other wireless communication device.

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