JP5909104B2 - Radio transmitting apparatus, radio receiving apparatus, radio communication system, and precoding method - Google Patents

Description

  The present invention relates to a technique for performing multi-user MIMO (Multiple Input Multiple Output) transmission to a plurality of radio receiving apparatuses.

  In the wireless communication system, it is always desired to improve the transmission speed in order to provide various broadband information services. Although the transmission speed can be improved by expanding the communication bandwidth, since the usable frequency band is limited, it is essential to improve the frequency utilization efficiency. Multiple input multiple output (MIMO) technology, which performs wireless transmission using multiple transmission / reception antennas, is attracting attention as a technology that can significantly improve frequency utilization efficiency, and has been put into practical use in cellular systems and wireless LAN systems. . The amount of improvement in frequency utilization efficiency by the MIMO technology is proportional to the number of transmission / reception antennas. However, the number of receiving antennas that can be arranged in the terminal device is limited. Therefore, multi-user MIMO (Multi User-MIMO (MU-MIMO)) that spatially multiplexes transmission signals from the base station device to each terminal device is regarded as a virtual large-scale antenna array. It is effective for improving the utilization efficiency.

  In MU-MIMO, transmission signals destined for each terminal apparatus are received by the terminal apparatus as inter-user-interference (IUI), so it is necessary to suppress IUI. For example, in long term evolution (LTE) adopted as one of the 3.9th generation mobile radio communication systems, a linear filter calculated based on propagation path information notified from each terminal device is used for the base station device. Linear precoding that suppresses the IUI by multiplying in advance is employed.

  Further, as a method for realizing MU-MIMO that can further improve the frequency utilization efficiency, MU-MIMO technology using nonlinear precoding in which nonlinear processing is performed on the base station apparatus side is attracting attention. If the terminal device is capable of modulo operation, it can add a perturbation vector whose element is a complex number (perturbation term) obtained by multiplying an arbitrary Gaussian integer by a constant real number to the transmitted signal. It becomes. Therefore, if the perturbation vector is appropriately set according to the propagation path state between the base station apparatus and the plurality of terminal apparatuses, it is possible to significantly reduce the required transmission power as compared with linear precoding. Non-patent document 1 describes Vector perturbation (VP) as a method that can realize optimal transmission characteristics as nonlinear precoding. While VP can achieve excellent transmission characteristics, the amount of computation increases exponentially in proportion to the number of spatially multiplexed terminals. On the other hand, in Tomlinson Harashima precoding (THP) described in Non-Patent Document 2, although the calculation amount is extremely small, the transmission characteristics are inferior to VP.

  By the way, in mobile radio communication, since the propagation path (channel) between the base station apparatus and the terminal apparatus has time and frequency selectivity, the signal transmitted from the base station apparatus has a phase and amplitude. It is received by the terminal device in a fluctuating state. In order to demodulate a desired signal from a received signal affected by channel fluctuation, channel equalization processing (also called synchronous detection or the like) that removes the influence of channel fluctuation is required. In order to perform channel equalization, propagation path estimation (channel estimation) is required to grasp the state of the propagation path (also referred to as propagation path information or channel state information (CSI)) by the terminal device. As a well-known channel estimation method, pilot channel estimation in which the terminal apparatus performs propagation path estimation by transmitting a known signal (pilot signal) from the base station apparatus to the terminal apparatus in advance is well known. It is actually used in generation mobile radio communication systems, wireless LAN systems, and the like.

  In MU-MIMO, transmission signals are precoded, so in order for the terminal device to demodulate a desired signal, in addition to propagation path information, what kind of precoding is performed on the transmission signal is grasped. There is a need. Precoding similar to the data signal is applied to some pilot signals (also referred to as unique reference signals or demodulation reference signals), and the result of precoding is reflected by performing the same pilot channel estimation as in the conventional terminal device. It is possible to estimate the propagation path information. Since the unique reference signal is for estimating information unique to each terminal device, the unique reference signals need to be orthogonal to each other. Since the linear MU-MIMO precoding is to spatially orthogonalize transmission data addressed to each terminal apparatus, by performing precoding similar to the data signal on the unique reference signal, the unique reference signal is obtained with the same radio resource. It is possible to send. Since the unique reference signal is a redundant signal, many radio resources are allocated for insertion of the unique reference signal, which increases overhead and decreases frequency utilization efficiency. According to this method, time division multiplexing is performed. As a result, overhead is reduced and high frequency utilization efficiency can be achieved compared to ensuring orthogonality of the unique reference signal.

  On the other hand, performing precoding similar to a data signal on a unique reference signal in nonlinear MU-MIMO means that a perturbation vector must also be assigned to the unique reference signal. In the terminal device, in order to remove the perturbation vector, it is necessary to perform a modulo operation on the received signal, but the modulo operation is performed after the channel equalization process. This means that a perturbation vector cannot be simply added to a specific reference signal for estimating propagation path information necessary for channel equalization processing.

JP 2009-182894 A JP 2010-114605 A

Yamagishi et al., “THP MU-MIMO using nonlinear multiplexing DMRS”, IEICE 2011 Society Conference, B-5-37, September 2011. BM Hochwald, et. Al., “A vector-perturbation technique for near-capacity multiantenna multiuser communication-Part II: Perturbation,” IEEE Trans. Commun., Vol. 53, No. 3, pp.537-544, March 2005 . M. Joham, et. Al., “MMSE approaches to multiuser spatio-temporal Tomlinson- Harashima precoding”, Proc. 5th Int. ITG Conf. On Source and Channel Coding, Erlangen, Germany, Jan. 2004.

  Patent Document 1 proposes a method of transmitting a unique reference signal by using another radio resource such as time without performing spatial multiplexing on a specific reference signal even in nonlinear MU-MIMO. However, many radio resources are used for transmission of the unique reference signal, resulting in a decrease in frequency utilization efficiency.

  In Patent Document 2, non-linear spatial multiplexing of a unique reference signal and a data signal is realized by separately reporting control information associated with a perturbation vector added to the unique reference signal. In this case, since it is necessary to notify the control information separately, the overhead is increased.

  In Non-Patent Document 1, normally, a part of the unique reference signal that is periodically transmitted is transmitted by radio resources other than space as in Patent Document 1, and the propagation path estimation result is used to A method for removing a perturbation vector given to a specific reference signal has been proposed. According to this method, spatial multiplexing of some unique reference signals can be realized, so that high frequency utilization efficiency can be realized. However, in an environment where the time and frequency selectivity of the propagation path is severe, propagation of other unique reference signals is possible. In the channel estimation result, a unique reference signal that is not correctly subjected to the modulo calculation is generated, so that the accuracy of the channel estimation is remarkably impaired and the transmission characteristics are greatly deteriorated.

  As described above, it is very difficult to realize appropriate nonlinear precoding for the unique reference signal. An increase in overhead due to the insertion of the unique reference signal causes a significant decrease in the frequency utilization efficiency of the nonlinear MU-MIMO.

  The present invention has been made in view of such circumstances, and a wireless transmission device and a wireless reception device capable of realizing transmission of a specific reference signal that significantly suppresses a specific reference signal insertion loss of nonlinear MU-MIMO. An object of the present invention is to provide a wireless communication system and a precoding method.

  (1) In order to achieve the above object, the present invention takes the following measures. That is, a wireless transmission device of the present invention is a wireless transmission device that includes a plurality of antennas and performs multi-user MIMO (Multiple Input Multiple Output) transmission to a plurality of wireless reception devices, each of the plurality of wireless reception devices. And a DMRS (Demodulation reference) in which each element is spatially multiplexed with the same radio resource, with the element being a unique reference signal transmitted to the data and a data signal destined for at least one radio receiver other than the destination of the unique reference signal. signal) vector is pre-encoded, and the pre-coded DMRS vector is spatially multiplexed and transmitted to each radio receiving apparatus.

  As described above, the unique reference signal transmitted to each of the plurality of wireless reception devices and the data signal addressed to at least one wireless reception device other than the destination of the unique reference signal are used as elements, and each of the elements has the same wireless signal. Since precoding is performed on DMRS vectors that are spatially multiplexed with resources, it is possible to improve the frequency utilization efficiency of nonlinear multiuser MIMO.

  (2) In addition, the wireless transmission device of the present invention uses a data signal addressed to the plurality of wireless reception devices as an element, and the transmission data vector in which each element is spatially multiplexed with the same wireless resource, and the DMRS vector. Thus, nonlinear precoding is performed on data signals included in the transmission data vector and the DMRS vector in different radio resources.

  As described above, in the data resources addressed to the plurality of radio reception devices as elements, the transmission data vector in which each element is spatially multiplexed with the same radio resource and the DMRS vector in different radio resources, Since non-linear precoding is performed on the transmission data vector and the data signal included in the DMRS vector, even in the non-linear multi-user MIMO, an eigen reference signal insertion loss equivalent to that of the conventional linear multi-user MIMO can be realized. Transmission is possible. As a result, it is possible to improve the frequency utilization efficiency of nonlinear multiuser MIMO.

  (3) Further, the radio transmission apparatus of the present invention is characterized in that linear precoding is performed on the unique reference signal included in the DMRS vector.

  As described above, since the linear precoding is performed on the unique reference signal included in the DMRS vector, the transmission characteristic of the unique reference signal can be improved.

  (4) Further, the radio transmission apparatus of the present invention multiplies each of the transmission data vector and the DMRS vector by a power normalization term that makes a total sum of transmission powers in at least one radio resource constant. Features.

  Thus, since each of the transmission data vector and the DMRS vector is multiplied by a power normalization term that makes the total sum of transmission power in at least one radio resource constant, transmission power control in nonlinear multiuser MIMO is performed. It becomes possible to carry out appropriately.

  (5) In the wireless transmission device of the present invention, the ratio between the power normalization term multiplied by the transmission data vector and the power normalization term multiplied by the DMRS vector is associated with the phase of the specific reference signal. It is characterized by being.

  Thus, since the ratio of the power normalization term multiplied by the transmission data vector and the power normalization term multiplied by the DMRS vector and the phase of the specific reference signal are related, Difference information relating to the power normalization term can be given to the phase.

  (6) Also, in the precoding, the radio transmission apparatus of the present invention performs ordering on the transmission data vector and the DMRS vector, ordering applied to the transmission data vector, and the DMRS vector. It is characterized by common ordering.

  Thus, in the precoding, the transmission data vector and the DMRS vector are each ordered, and the ordering applied to the transmission data vector and the ordering applied to the DMRS vector are common. It is possible to realize spatial multiplexing of the unique reference signal and the data signal without adding a perturbation term to the reference signal. In addition, it is possible to reduce an error between the power normalization term estimated by the unique reference signal and the power normalization term applied to the data signal portion.

  (7) In the wireless transmission device of the present invention, the transmission data vector includes a first data signal and a second data signal, and the wireless resource to which the first data signal is transmitted is The unique reference signal is associated with a radio resource to be transmitted.

  As described above, the transmission data vector includes the first data signal and the second data signal, and the radio resource to which the first data signal is transmitted is the radio resource to which the unique reference signal is transmitted. Therefore, even for some data signals, it is possible to improve the coding gain by channel coding without adding the perturbation term.

  (8) Further, the radio reception apparatus of the present invention includes at least one antenna, and receives a radio signal transmitted by a multi-user MIMO (Multiple Input Multiple Output) from a radio transmission apparatus having a plurality of antennas. A transmission data vector having elements of data signals addressed to a plurality of radio reception devices, a unique reference signal transmitted to each of the plurality of radio reception devices, and the unique reference signal. DMRS (Demodulation Reference Signal) vectors whose elements are data signals addressed to at least one radio receiving apparatus other than the destination are pre-coded, spatially multiplexed, and transmitted by different radio resources. Based on the unique reference signal addressed to the device, the propagation path state is estimated to generate propagation path state information, Based on the serial channel state information, wherein demodulating the data signal addressed to the device itself included in the DMRS vector and the transmission data vector.

  Thus, the wireless signal from the wireless transmission device includes a transmission data vector whose elements are data signals addressed to a plurality of wireless reception devices, a unique reference signal transmitted to each of the plurality of wireless reception devices, and the Since DMRS (Demodulation reference signal) vectors whose elements are data signals addressed to at least one radio receiving apparatus other than the destination of the unique reference signal are precoded in different radio resources, even in nonlinear multi-user MIMO, It is possible to transmit a unique reference signal that can realize a unique reference signal insertion loss equivalent to that of conventional linear multiuser MIMO. As a result, it is possible to improve the frequency utilization efficiency of nonlinear multiuser MIMO.

  (9) Further, the radio reception apparatus of the present invention provides a ratio between a power normalization term multiplied by the transmission data vector and a power normalization term multiplied by the DMRS vector, and a unique reference signal addressed to the own device. The propagation path state information is corrected on the basis of the phase of the unique reference signal addressed to the own apparatus.

  Thus, the ratio between the power normalization term multiplied by the transmission data vector and the power normalization term multiplied by the DMRS vector is associated with the phase of the unique reference signal addressed to the own device. Thus, it is possible to obtain difference information related to the power normalization term from the phase of the unique reference signal and obtain appropriate propagation path state information.

  (10) In addition, the radio reception apparatus of the present invention may add the transmission data vector and the DMRS vector to the transmission data vector and the DMRS vector based on the position of the radio resource to which the DMRS vector including the unique reference signal addressed to the own apparatus is transmitted. The method is characterized in that the method for demodulating the data signal addressed to its own device is switched.

  As described above, based on the position of the radio resource to which the DMRS vector including the unique reference signal addressed to the own device is transmitted, the transmission data vector and the data signal addressed to the own device included in the DMRS vector are changed. Since the demodulation method is switched, higher error correction capability can be realized when channel data is decoded as a nonlinear data signal when linear data signals and nonlinear data signals are mixed and transmitted.

  (11) A wireless communication system according to the present invention includes the wireless transmission device according to any one of (2) to (7), and the wireless reception device according to any one of (8) to (10). It is characterized by comprising.

  With this configuration, it is possible to improve the frequency utilization efficiency of nonlinear multiuser MIMO.

  (12) The precoding method of the present invention is a precoding method applied to a radio transmission apparatus that includes a plurality of antennas and performs multi-user MIMO (Multiple Input Multiple Output) transmission to a plurality of radio reception apparatuses. A data signal addressed to the plurality of radio receiving devices, and a transmission data vector in which each element is spatially multiplexed with the same radio resource, and a unique transmitted to each of the plurality of radio receiving devices. With respect to a DMRS (Demodulation Reference Signal) vector in which at least one data signal addressed to the wireless reception device other than the destination of the reference signal and the specific reference signal is an element, and each element is spatially multiplexed with the same radio resource In different radio resources, they are included in the transmission data vector and the DMRS vector. And performing a non-linear precoding data signals it is.

  With this configuration, it is possible to improve the frequency utilization efficiency of nonlinear multiuser MIMO.

  According to the present invention, even in non-linear MU-MIMO, the efficiency of frequency utilization of non-linear MU-MIMO can be greatly improved because the insertion loss of inherent reference signal can be made equal to that of conventional linear MU-MIMO.

It is a figure which shows the outline of the radio | wireless communications system which concerns on the 1st Embodiment of this invention. It is a block diagram which shows the structure of the base station apparatus 1 which concerns on the 1st Embodiment of this invention. It is a block diagram which shows the apparatus structure of the antenna part 109 which concerns on the 1st Embodiment of this invention. It is a block diagram which shows the apparatus structure of the precoding part 107 which concerns on the 1st Embodiment of this invention. An example of a frame format when DMRS spatial multiplexing is not performed is shown. 2 shows an example of a frame format in the first embodiment of the present invention. 7 is a flowchart for explaining signal processing for a DMRS vector in a perturbation vector search unit 303 and a transmission signal generation unit 305, including a case of correcting a power normalization term, in the first embodiment of the present invention. In the 1st Embodiment of this invention, it is the figure which showed an example of the resource allocation in the case of U = 4 and _Nt = 4. It is a block diagram which shows the structure of the terminal device 3 which concerns on the 1st Embodiment of this invention. It is a flowchart explaining the signal processing performed with respect to DMRS in the propagation path estimation part 411 which concerns on the 1st Embodiment of this invention. It is the figure which showed an example of the bit error rate (BER) characteristic in the 1st Embodiment of this invention. It is a figure which shows an example of the frame format of DMRS which concerns on the 2nd Embodiment of this invention. It is a flowchart of the signal processing of the precoding part 107 with respect to the DMRS vector and transmission data vector which concern on the 2nd Embodiment of this invention. It is the figure which showed an example of the resource allocation which concerns on the 2nd Embodiment of this invention. In the 2nd Embodiment of this invention, it is a figure which shows an example of the frame format in the case of changing an ordering method with a DMRS vector and a transmission data vector. In the 2nd Embodiment of this invention, it is the figure which showed an example of the resource allocation in the case of changing an ordering method with a DMRS vector and a transmission data vector. It is a flowchart explaining the signal processing of the precoding part 107 which concerns on the 3rd Embodiment of this invention. It is a flowchart explaining the signal processing of the precoding part 107 which concerns on the 4th Embodiment of this invention. It is a figure which shows an example of the transmission frame format when not adding a perturbation term to a part of transmission data in the 4th Embodiment of this invention.

  Hereinafter, an embodiment in a case where a wireless communication system of the present invention is applied will be described with reference to the drawings. In addition, the matter demonstrated in this embodiment is an aspect for understanding invention, and the content of invention is not interpreted limited to embodiment.

In the following, ^AT is a transposed matrix of matrix A, A ^H is an adjoint (Hermitian transpose) matrix of matrix A, A ^-1 is an inverse matrix of matrix A, A ⁺ is a pseudo (or general) inverse matrix of matrix A, diag (A) is a diagonal matrix obtained by extracting only the diagonal components of the matrix A. floor (c) is a floor that returns the largest Gaussian integer whose real part and imaginary part do not exceed the values of the real part and imaginary part of the complex number c, respectively. Function, E [x] is the ensemble average of the random variable x, abs (c) is a function that returns the amplitude of the complex number c, angle (c) is a function that returns the argument of the complex number c, || a || Norm and x% y represent the remainder when integer x is divided by integer y. [A; B] represents a matrix obtained by combining two matrices A and B in the row direction, and [A, B] represents a matrix obtained by combining the matrices A and B in the column direction.

[1. First Embodiment]
FIG. 1 is a diagram showing an outline of a radio communication system according to the first embodiment of the present invention. In the first embodiment, has a transmission antenna of the N _t present, (also referred to as a wireless transmission apparatus) nonlinear precoding capable base station apparatus with respect to 1, the terminal device having one receive antenna (radio (Also referred to as a receiving device) 3 (terminal devices 3-1, 3-2,..., 3-u,. For MIMO transmission, N _t = U, but N _t and U may be different, but N _t > U is desirable. As a transmission method, orthogonal frequency division multiplexing (OFDM) having _Nc subcarriers (subcarriers) is assumed. The base station apparatus 1 acquires propagation path information to each terminal apparatus 3 from the control information notified from each terminal apparatus 3, and performs precoding for each subcarrier on transmission data based on the propagation path information. And Note that the number of reception antennas of the terminal device 3 is not limited to one. In the present embodiment, the number of data streams (also referred to as the rank number) transmitted to each terminal device 3 is 1. However, the present embodiment includes a case where the rank number is larger than 2.

First, propagation path state information between the base station apparatus 1 and the terminal apparatus 3 is defined. In the present embodiment, a quasi-static frequency selective fading channel is assumed. When the complex channel gain of the k-th subcarrier between the n-th transmission antenna (n = 1 to N _t ) and the u-th terminal apparatus 3-u (u = 1 to U) is h _{u, n} (k), A propagation path matrix H (k) is defined as shown in Equation (1).

Note that h _u (k) = [h _{u, 1} (k),..., H _{u, Nt} (k)] is a propagation path vector composed of complex channel gains observed by the u th terminal apparatus 3-u. Represents.

[1.1 Base station apparatus 1]
FIG. 2 is a block diagram showing a configuration of the base station apparatus 1 according to the first embodiment of the present invention. As illustrated in FIG. 2, the base station apparatus 1 includes a channel coding unit 101, a data modulation unit 103, a mapping unit 105, a precoding unit 107, an antenna unit 109, a control information acquisition unit 111, a CSI. The acquisition unit 113 is included. Precoding section 107 is the number of subcarriers _{N c,} the antenna unit 109 is present respectively by the number of transmit antennas _{N t.} The transmission data sequence addressed to each terminal apparatus 3 is subjected to channel coding in channel coding section 101 and then subjected to digital data modulation such as QPSK and 16QAM in data modulation section 103. An output from the data modulation unit 103 is input to the mapping unit 105.

  The mapping unit 105 performs mapping (also referred to as scheduling or resource allocation) in which each data is allocated to a specified radio resource (also referred to as resource element or simply resource). Here, the radio resource mainly refers to frequency, time, code, and space. The radio resource to be used is determined based on the reception quality observed by the terminal device 3, the orthogonality of the propagation path between the spatially multiplexed terminals, and the like. In the present embodiment, it is assumed that the radio resources to be used are determined in advance and can be grasped by both the base station device 1 and each terminal device 3. In addition, in mapping section 105, a known reference signal sequence for performing channel estimation in each terminal apparatus 3 is also multiplexed.

  The reference signals addressed to the respective terminal devices 3 are multiplexed so as to be orthogonal to each other so that they can be separated in the received terminal device 3. Also, the reference signal is multiplexed with two reference signals, Cell-specific reference signal (CRS), which is a reference signal for channel estimation, and Demodulation reference signal (DMRS), which is a specific reference signal for demodulation. However, another reference signal may be further multiplexed. The CRS is for estimating the propagation path matrix represented by the equation (1), and the DMRS is for estimating the propagation path information reflecting the result of precoding described later. In the present invention, the CRS is assumed to be completely orthogonal to the data signal and DMRS, and the CRS transmitted from each transmission antenna is also arranged in the radio resource so as to be orthogonal to each other. . The data signal and DMRS multiplexing method will be described in detail after describing signal processing in the precoding unit 107 described later.

  The output of mapping section 105 is input to corresponding subcarrier precoding section 107. The signal processing in the precoding unit 107 will be described later. Hereinafter, the signal processing for the output of the precoding unit 107 will be described first. The output of precoding section 107 for each subcarrier is input to antenna section 109 of the corresponding transmission antenna.

FIG. 3 is a block diagram showing a device configuration of the antenna unit 109 according to the first embodiment of the present invention. As shown in FIG. 3, the antenna unit 109 includes an IFFT unit 201, a GI insertion unit 203, a wireless transmission unit 205, a wireless reception unit 207, and an antenna 209. Each antenna unit 109, the output of the corresponding precoding section 107 is input to the IFFT unit 201, an inverse fast Fourier transform of _{N c} points (IFFT), or inverse discrete Fourier transform (IDFT) is applied, _{N c} sub An OFDM signal having a carrier is generated and output from the IFFT unit 201. Here, the number of subcarriers and the number of points of IFFT are described as being the same, but when a guard band is set in the frequency domain, the number of points is larger than the number of subcarriers. The output of the IFFT unit 201 is input to the GI insertion unit 203, and after being given a guard interval, is input to the wireless transmission unit 205. In the wireless transmission unit 205, the baseband transmission signal is converted into a radio frequency (RF) transmission signal. The output signal of the wireless transmission unit 205 is transmitted from the antenna 209.

  The radio reception unit 207 receives information associated with the propagation path state information estimated by the terminal device 3 and outputs the information to the control information acquisition unit 111.

[1.2 Precoding unit 107]
The signal processing performed in the precoding unit 107 will be described. Hereinafter, the k-th subcarrier precoding unit 107 will be described, and first, a case where a data signal component of the mapping unit 105 output is input will be described.

FIG. 4 is a block diagram showing a device configuration of the precoding unit 107 according to the first embodiment of the present invention. As shown in FIG. 4, the precoding unit 107 includes a linear filter generation unit 301, a perturbation vector search unit 303, and a transmission signal generation unit 305. The precoding unit 107 includes the k-th subcarrier component {d _u (k); u = 1 to U} of the output of the mapping unit 105 including the transmission data addressed to each terminal device 3, and the output of the CSI acquisition unit 113. The propagation path matrix H (k) of the kth subcarrier is input. H (k) is estimated by the terminal device 3 based on the above-described CRS and notified to the base station device 1. In the following description, H (k) is ideally acquired by the CSI acquisition unit 113, and the index k is omitted for simplicity.

  First, the linear filter generation unit 301 generates a linear filter W for IUI suppression. In the following, a method based on THP will be described.

In THP, first, LQ decomposition is applied to the acquired propagation path matrix H to decompose into a lower triangular matrix L and a unitary matrix Q such that H = LQ (note that QR decomposition is applied to H ^H. It is also possible to obtain the same calculation result). At this time, the linear filter W can be expressed by W = Q ^H {diag (L)} ⁻¹ . The THP using the linear filter calculated in this way is based on the Zero-forcing (ZF) standard. THP based on the minimum mean square error (Minimum MSE (MMSE)) standard that minimizes the mean square error (MSE) between the transmitted signal and the received signal is possible instead of the ZF standard. , [H; γI] is regarded as a channel matrix, and LQ decomposition may be performed. Here, γ is a parameter for controlling the residual interference. Normally, the reciprocal of the square root of the transmission signal power to noise power ratio per terminal apparatus is input, but the value differs depending on the type of CSI acquired by the precoding unit 107. For example, when the received power is also included in the CSI, γ is a standard deviation of the thermal noise applied by the terminal device 3. The generated linear filter W is input to the perturbation vector search unit 303 and the transmission signal generation unit 305.

The signal processing in the perturbation vector search unit 303 will be described. Transmission data vector s = [s ₁ ,..., S _Nt ] ^T = Wd obtained by multiplying transmission data vector d = [d ₁ ,..., D _U ] ^T by linear filter W from base station apparatus 1 In the case of transmission, a received signal vector r = [r ₁ ,..., R _U ] ^T composed of received signals observed by each terminal device 3 is given by Expression (2).

Here, η = [η ₁ ,..., Η _Nr ] ^T is a noise vector constituted by noise applied to the received signal in each terminal device 3. L _{i, j} represents the i-th row and j-th column component of the matrix L {diag (L)} ⁻¹ . From equation (2), it can be seen that the first terminal device 3-1 can achieve transmission without IUI, while the second terminal device 3-2 and subsequent devices are affected by IUI.

Therefore, in addition to the multiplication of the linear filter, signal processing for suppressing the IUI is performed in the precoding unit 107 of the base station apparatus 1. Specifically, the transmission signal of the u terminal device 3-u addressed, the d _u rather, the process of the transmission code x _u given by Equation (3) is carried out.

If a transmission code vector x = [x ₁ ,..., X _U ] ^T composed of the transmission code x _u given by Expression (3) is transmitted instead of d, transmission without IUI can be realized. However, the power of x _u is dependent on the channel matrix H, the state of the H may become a much greater power than d _u. Therefore, in THP, modulo arithmetic is sequentially performed on {x _u ; u = 2 to U}. A modulo operation modulo _δ (x _u ) that returns an output such that the real part and the imaginary part of the input complex number x _u fall within −δ to δ, respectively, is given by Expression (4).

Here, z _{t, u} = −floor (x _u / 2δ + (1 + j) / 2) is called a perturbation term. The perturbation term is a complex number (also called a Gaussian integer) whose real part and imaginary part are integers. By performing the modulo operation, the power of x _u does not depend on the propagation path matrix H, and δ is given as a parameter. The optimum value of δ is calculated according to the data modulation method, and is 2 ^{1/2 for} QPSK and 4/10 ^{1/2 for} 16 QAM, but the base station apparatus 1 and terminal apparatus 3 If it is shared between the two, it is not necessary to set this value. Using a perturbation vector z _t = [z _{t, 1} ,..., Z _{t, U} ] ^T composed of perturbation terms, the transmission code vector is given by x = (HW) ⁻¹ (d + 2δz _t ). The signal processing performed by the perturbation vector search unit 303 is to calculate the transmission code vector x. The transmission code vector x output from the perturbation vector search unit 303 is input to the transmission signal generation unit 305.

Transmission signal generation section 305 calculates transmission signal vector s = (2P) ^1/2 βWx and outputs it as an output of precoding section 107. Here, P represents the average transmission power per terminal device. β is a power normalization term, and is given by Equation (5) when E [| d _u | ² ] = 1.

ここで、Ｃ_ｖは送信符号ベクトルｘの共分散行列であり、式（６）で与えられる。

Here, _Cv is a covariance matrix of the transmission code vector x and is given by Equation (6).

  The above is the description regarding the signal processing when only the data signal is input to the precoding unit 107. When CRS is input, the signal processing described so far is not basically performed, and only power adjustment is performed.

A case where DMRS is input will be described. The DMRS is a signal for estimating propagation path information necessary for the terminal device 3 to demodulate a desired signal by reception signal processing. Although details will be described later, when nonlinear precoding by THP is performed, the terminal device 3 needs to grasp (2P) ^1/2 β.

Here, similarly to linear MU-MIMO, it is assumed that pilot signal vector p = [p ₁ ,..., P _U ] ^T composed of known signals is subjected to precoding similar to a data signal. At this time, the received signal is given by equation (7).

From the equation (7), for the first terminal device 3-1, the received signal-to-noise power ratio (Signal-to-Noise power Ratio (SNR)) can be sufficiently large, dividing the p ₁ from the received signal It can be seen that desired information can be acquired. On the other hand, in the second terminal device 3-2, even by dividing the received signal by p ₂ is a known signal, for perturbation term z _{t, 2} is unknown, it is impossible to obtain the desired information . The same applies to the third terminal device 3-3 and later. Thus, for DMRS, methods that do not perform spatial multiplexing have been studied.

FIG. 5 shows an example of a frame format when DMRS spatial multiplexing is not performed. In FIG. 5, the number of spatially multiplexed terminals is U = 4. p _u is DMRS transmitted to the u th terminal apparatus 3-u. D _u (i) represents the i-th transmission symbol transmitted to the u-th terminal apparatus 3-u. {E [| p _u | ² ] = 1, u = 1 to U} and {E [| d _u (i) | ² ] = 1, u = 1 to U}. The power may be different, or may be different for each terminal device 3. In the following, the position of the spatial resource may be referred to as a number called an antenna port number.

  DMRS is transmitted to each terminal device 3 at different times, and no data is transmitted to other terminal devices 3 while DMRS is transmitted to a certain terminal device 3. In this case, there is no IUI to be suppressed in the received DMRS signal. Therefore, since it is not necessary to add the perturbation term by the modulo calculation, each terminal apparatus 3 can obtain desired information as long as appropriate transmission power control is performed. However, in this case, since many radio resources are consumed for DMRS transmission, the frequency utilization efficiency is greatly reduced. Taking FIG. 5 as an example, the transmission rate is halved due to DMRS insertion. In this embodiment, the DMRS and a part of the data signal are spatially multiplexed to improve the frequency use efficiency.

FIG. 6 shows an example of a frame format in the first embodiment of the present invention. In FIG. 6, the number of spatially multiplexed terminals is U = 4, but there is no limit on the number of multiplexed terminals. As shown in FIG. 6, in this embodiment, a data signal addressed to another terminal device 3 is transmitted together with a DMRS addressed to a certain terminal device 3. In this case, the transmission rate is 7/8 due to DMRS insertion, which is approximately 1.75 times that of the conventional method. From time t = 1 to time t = 4, transmission data vectors d _p (1), d _p (2), d _p (3), and d _p (4) in which DMRS and data signals are mixed are consecutive. It is assumed that after time t = 5, a transmission data vector d (t) composed entirely of data signals is transmitted. It is assumed that d _p (1), d _p (2), d _p (3), and d _p (4) are given by equation (8).

Hereinafter, a transmission data vector in which a DMRS and a data signal are mixed is referred to as a DMRS vector, and a vector composed only of a data signal is referred to as a transmission data vector. First, a case where d _p (1) is input to the precoding unit 107 will be described. When signal processing similar to precoding for the data signal already described is performed, the received signal vector is given by equation (9).

The first terminal device 3-1 can acquire desired information by dividing p ₁ from the received signal. As for the devices after the second terminal device 3-2, the signal cannot be demodulated at this stage, so that the received signal is stored.

Next, a case where d _p (2) is input to the precoding unit 107 will be described. When signal processing similar to precoding for the data signal already described is performed, the received signal vector is given by equation (10). In order to simplify the explanation, it is assumed that the time selectivity of the propagation path is sufficiently weak.

The first terminal device 3-1 can demodulate d ₁ (1) because information necessary for signal demodulation is obtained by signal processing on r (1). On the other hand, since the perturbation term is given to the known signal for the second terminal device 3-2, desired information cannot be acquired.

Therefore, in the perturbation vector search unit 303 of the first embodiment, when a DMRS vector is input, a perturbation term is not added to the DMRS. That is, when d _p (2) is input, the output x _p (2) = [x _{p, 1} (2), x _{p, 2} (2), x _{p, 3} (2) of the perturbation vector search unit 303 is input. , X _{p, 4} (2)] ^T is given by equation (11).

  A transmission signal is generated by inputting the transmission code vector given by Expression (11) to the transmission signal generation unit 305. However, unlike the equation (6), the covariance matrix of the transmission code vector is given by the equation (12), so the power normalization term β also has a slightly different value.

Hereinafter, the power normalization term when d _p (t) is transmitted is represented by β ^(t) . Note that β ⁽¹⁾ = β.

When the transmission signal vector obtained in this way is transmitted from the base station apparatus 1, the received signal is given by the equation (13). Therefore, the second terminal apparatus 3-2 determines (2P) from the received signal. ^1/2 β ⁽²⁾ can be grasped.

By using this information, the data signal d ₂ (1) can be demodulated from r (1) once stored. Also, the first terminal device 3-1, from information obtained by r _(1), it becomes possible to demodulate _d 1 (1). Hereinafter, for d _p (3) and d _p (4), the perturbation vector search unit 303 and the transmission signal generation unit 305 may perform signal processing similar to that of d _p (2). However, depending on the state of the propagation path, β ^{(2 )} to β ⁽⁴⁾ may be significantly different from β. When the difference is so large that it cannot be ignored (for example, when the difference is larger than a predetermined threshold Γ), the difference may be notified by the method described below.

As method 1, there is a transmission power control method. This is a power normalization term for all resources (including transmission data vectors ⁾ among β ^{(1) to} β ^{( 4)} . By controlling in this way, an error depending on the error of the power normalization term does not occur, but the received power is slightly reduced. As method 2, there is a phase difference notification method. This gives difference information regarding the power normalization term to the phase of the DMRS addressed to each terminal device 3.

FIG. 7 is a flowchart for explaining signal processing for the DMRS vector in the perturbation vector search unit 303 and the transmission signal generation unit 305 including the case of correcting the power normalization term in the first embodiment of the present invention. ^First , β ^{(1) to} β ⁽⁴⁾ are calculated based on the input linear filter W (step S101). Note that β = β ⁽¹⁾ . Subsequently, it is determined whether or not an error between β ⁽²⁾ to β ⁽⁴⁾ and β exceeds a threshold Γ (step S102). It should be noted that an appropriate value of Γ may be grasped in advance by computer simulation or the like. When the error between β ⁽²⁾ to β ⁽⁴⁾ and β is small (step S102: YES), no correction is performed on the power normalization term, and as described above, d _p (1) to d against _p (4), THP is applied to calculate the transmitted code vector _{x p (1) ~x p (} 4) ( step S103). Then, a transmission signal vector is calculated based on the calculated transmission code vector and the linear filter W (step S104).

On the other hand, when the error between β ⁽²⁾ to β ⁽⁴⁾ and β is large (step S102: NO), signal processing for correcting the error of the power normalization term is performed. When correcting by the method 1 (step S105: N = 1), a search is made for a minimum value among β ^{(1) to} β ^{( 4)} , and this is set as β ^min (step S106). Then, after signal processing similar to that in step S103 is performed (step S107), a transmission signal vector is generated. In this case, β ^min is always used as the power normalization term (step S108). When Method 1 is used, β ^min is used as the power normalization term for the transmission data vector.

When correcting by the method 2 (step S105: N = 2), α ⁽¹⁾ = β ⁽¹⁾ / β ⁽⁴⁾ and {α ^(t) = β ^(t) / β ⁽¹⁾ ; t = 2 to 2 4} is calculated (step S109). Subsequently, based on the calculated α ^(t) , the DMRS addressed to each terminal device 3 is set to p _t exp (jα ^(t) ) (step S110). Thereafter, the same signal processing as in steps S103 and S104 is performed (steps S111 and S112). Although details will be described later, when all DMRS vectors are corrected by method 2, signal demodulation may not be performed correctly. For this reason, at least one DMRS vector needs to be corrected by method 1 or transmitted using orthogonal resources as in the prior art.

  Whether method 1 or method 2 is used may be determined in advance according to the desired reception quality or the like, or may be controlled to be adaptively changed according to the instantaneous reception quality. good. When the change is adaptively performed, control information associated with the correction method used is notified to each terminal device 3.

In the above description, the DMRS vectors d _p (1), d _p (2), d _p (3), and d _p (4) are assumed to be time-division multiplexed. It may be multiplexed by division multiplexing or code division multiplexing, or may be multiplexed in two dimensions of time and frequency. According to the multiplexing method of d _p (1), d _p (2), d _p (3), and d _p (4), mapping section 105 multiplexes the data signal, DMRS, and CRS.

FIG. 8 is a diagram illustrating an example of resource allocation in the case of U = 4 and N _t = 4 in the first embodiment of the present invention. #N represents a CRS transmitted from the nth transmission antenna, and a signal is transmitted only from the nth transmission antenna in the corresponding radio resource. Data signals or control signals are multiplexed in the blank portion, and signals addressed to a plurality of terminal apparatuses 3 are spatially multiplexed and transmitted. In addition, it is assumed that the base station apparatus 1 notifies in advance which radio resource DMRS is transmitted.

  In the description so far, the DMRS vector includes one DMRS, but the number of DMRSs included is not limited to one. When spatially multiplexing a plurality of DMRSs, control may be performed so as not to add perturbation terms for any DMRS, but perturbation terms may be added for some DMRSs. However, for example, at time t = 1, the DMRS addressed to the first terminal device 3-1 and the second terminal device 3-2 is spatially multiplexed, and a perturbation term is added to the DMRS addressed to the second terminal device 3-2. In this case, at a different time (for example, time t = 2), it is necessary to transmit a DMRS that has not been added with a perturbation term to the second terminal apparatus 3-2. By using the information estimated by the DMRS to which the perturbation term is not added, the DMRS to which the perturbation term is added can also be used for propagation path estimation.

[1.3 Terminal device 3]
FIG. 9 is a block diagram showing a configuration of the terminal device 3 according to the first embodiment of the present invention. As illustrated in FIG. 9, the terminal device 3 includes an antenna 401, a radio reception unit 403, a GI removal unit 405, an FFT unit 407, a reference signal separation unit 409, a propagation path estimation unit 411, and a feedback information generation unit. 413, a wireless transmission unit 415, a propagation path compensation unit 417, a demapping unit 419, a data demodulation unit 421, and a channel decoding unit 423.

In the terminal device 3, a signal received by the antenna 401 is input to the wireless reception unit 403 and converted into a baseband signal. The signal converted into the baseband is input to the GI removal unit 405, and after the guard interval is removed, is input to the FFT unit 407. In the FFT unit 407, N _c point fast Fourier transform (FFT) or discrete Fourier transform (DFT) is applied to the input signal to convert it into N _c subcarrier components. The output of the FFT unit 407 is input to the reference signal separation unit 409. The reference signal separation unit 409 separates the input signal into a data signal component, a CRS component, and a DMRS component. The data signal component is output toward the propagation path compensation unit 417, and the CRS and DMRS are output toward the propagation path estimation unit 411. The signal processing described below is basically performed for each subcarrier.

  The propagation path estimation unit 411 performs propagation path estimation based on the input known reference signals CRS and DMRS. First, channel estimation using CRS will be described. Since the CRS is transmitted without applying precoding, the component corresponding to each terminal device 3 (for example, the u-th terminal device 3) in the channel matrix H (k) represented by the equation (1). If -u, it is possible to estimate the u-th component of H (k). Usually, since CRS is periodically multiplexed with respect to radio resources, it is not possible to directly estimate the propagation path information of all subcarriers, but CRS at time intervals and frequency intervals that satisfy the sampling theorem. By transmitting, it is possible to estimate the propagation path information of all subcarriers by appropriate interpolation. A specific propagation path estimation method is not particularly limited. For example, it is conceivable to use two-dimensional MMSE propagation path estimation.

  The propagation path information estimated by the CRS is input to the feedback information generation unit 413. The feedback information generation unit 413 generates information to be fed back to the base station apparatus 1 according to the propagation path information format fed back by each terminal apparatus 3. In the present invention, the propagation path information format is not limited to anything. For example, a method is conceivable in which estimated channel information is quantized with a finite number of bits and the quantized information is fed back. Further, feedback may be performed based on a code book that has been agreed with the base station apparatus 1 in advance. Information generated by feedback information generation section 413 is input to radio transmission section 415 and notified to base station apparatus 1. Hereinafter, this propagation path information is also referred to as first propagation path information, and control information associated with the first propagation path information is also referred to as first control information.

  Next, propagation path estimation using DMRS will be described. In the following, as described in the base station apparatus 1, when the number of spatially multiplexed terminals is U, it is assumed that the DMRS vector is transmitted from time t = 1 to time t = U, and the u-th terminal apparatus 3-u It is assumed that the addressed DMRS is transmitted at time t = u. In this case, the signal input to the propagation path estimation unit 411 is only the signal transmitted at time t = u, and the signals transmitted at other times are output to the propagation path compensation unit 417. .

  The signal processing for DMRS differs depending on the correction processing performed for the power normalization term performed in precoding section 107. Which correction process is being performed can be grasped from the control information notified from the base station apparatus 1. Control may be performed so as to perform one predetermined correction process.

FIG. 10 is a flowchart illustrating signal processing performed on DMRS in propagation path estimation section 411 according to the first embodiment of the present invention. First, the u-th terminal apparatus 3-u divides the known signal p _u from the signal transmitted at time t = u, and is required for demodulation of the data signal (2P) ^1/2 β ^(u). Is calculated (step S201). When the power normalization term is not corrected or is corrected by the method 1 (step S202: NO), the information calculated and acquired here is output to the propagation path compensation unit 417. . In this case, in addition to the demodulation of the data signal included in the transmission data vector, the value estimated here is also used for the demodulation of the data signal addressed to the terminal device 3 included in the DMRS vector.

  A case where method 2 is performed (step S202: YES) will be described. In this case, in the u-th terminal device 3-u, the reception signal of the signal transmitted at time t = u is given by Expression (14).

Here, exp (jφ) represents a phase shift caused by a frequency offset between the oscillator of the base station apparatus 1 and the oscillator of the terminal apparatus 3 (usually, this phase shift is estimated by being included in the propagation path information). Therefore, the description is omitted in other received signal notations in the specification). In the method 2, the terminal device 3 needs to know the value of exp (jφ). exp (jφ) can be grasped by transmitting some DMRS vectors without correcting the power normalization term, or by transmitting DMRS using orthogonal resources, as in the conventional method. For example, when the power normalization term is not corrected, the received signal given by the equation (14) is r _u (u) = (2P) ^1/2 β ^(u) exp (jφ) p _u + η _u ( Since it is given by u), the value of φ can be estimated by calculating angle (r _u (u)). When there is an offset in frequency between the oscillators, exp (jφ) fluctuates over time. Therefore, it is desirable that a DMRS that does not perform correction be transmitted using radio resources in the vicinity of the DMRS that performs correction. In the following, exp (jφ) is assumed to be estimable, and the description is omitted.

Subsequently, the propagation path estimation unit 411 calculates abs (r _u (u) / p _u ) and angle (r _u (u) / p _u ) (step S203). For the terminal device 3 (that is, u> 1) in which DMRS is transmitted from antenna port 2 and after (step S204: Yes), if the DMRS transmission power is sufficiently higher than the noise power, abs (r _u (u) / P _u ) ≈ (2P) ^1/2 β ^(u) , and angle (r _u (u) / p _u ) ≈α ^(u) . Since α ^(u) = β ⁽¹⁾ / β ^(u) , by multiplying abs (r _u (u) / p _u ) by angle (r _u (u) / p _u ), (2P) ^1/2 β ⁽¹⁾ (= (2P) ^1/2 β) can be estimated (step S205). This value is output to the propagation path compensator 417 for demodulation of transmission data included in the transmission data vector.

Next, a method for estimating data signal demodulation information included in the DMRS vector will be described. In the u-th terminal device 3-u, β ⁽¹⁾ and β ^(u) are obtained by the above method, but in order to demodulate the data signal, all information of β ^{(2) to} β ⁽⁴⁾ is obtained. Need to know. In the first embodiment, β ^{(2) to} β ⁽⁴⁾ are estimated by linear interpolation using β ⁽¹⁾ and β ^(u) (step S206). The interpolation method is not limited to linear interpolation.

For the first terminal device 3-1 to which DMRS is transmitted via the antenna port 1 (step S204: No), abs (r ₁ (1) / p ₁ ) ≈ (2P) ^1/2 β ⁽¹⁾ is transmitted. It outputs as information for data vector demodulation (step S207). After that, abs (r ₁ (1) / p ₁ ) is multiplied by angle (r ₁ (1) / p ₁ ⁾ to estimate (2P) ^1/2 β ⁽⁴⁾ . Information for demodulating the data signal included in the DMRS vector is calculated by the same signal processing as that performed by the u terminal apparatus 3-u (step S208).

  In the description so far, it is assumed that the DMRS addressed to each terminal apparatus 3 is transmitted only by one radio resource. However, as shown in FIG. 8, the DMRS is periodically transmitted by a plurality of radio resources. You may send it. In this case, it is possible to improve propagation path estimation accuracy by performing appropriate interpolation or the like using a plurality of propagation path estimation results. When the power normalization term is corrected by the method 2, some DMRS vectors need to be transmitted without being corrected by the method 2. In this case, the DMRS vector that has been corrected by the method 2 is used. It is desirable that both DMRS vectors not subjected to correction exist in the same OFDM symbol.

Returning to FIG. 9, out of the output of the reference signal separation unit 409, the propagation path information (that is, (2P) ^1/2 β) obtained by DMRS out of the output of the propagation path estimation unit 411 is Input to the propagation path compensation unit 417. The channel compensation unit 417 performs channel equalization processing on the data signal component. The received signal in the data signal portion of the u-th terminal device 3-u is given by equation (15).

In addition, description is abbreviate | omitted about the index showing time and a frequency. Introduction The channel compensation section 417, is divided by the information that has been estimated _{r u} by DMRS ^{(2P) 1/2} β. That is, r _u / (2P) ^1/2 β = (d _u + 2δz _{t, u} ) + η _u / (2P) ^1/2 β. If the received SNR is sufficiently high and η _u / (2P) ^1/2 β≈0, then r _u / (2P) ^1/2 β = d _u + 2δz _{t, u} , so r _u / (2P ) against ^1/2 beta, by applying the same modulo operation with formula (4), it is possible to demodulate the _{d u.}

  The output of the propagation path compensation unit 417 is then input to the demapping unit 419, and each terminal apparatus 3 extracts transmission data addressed to itself from radio resources used for transmission of transmission data addressed to itself. The output of demapping section 419 is then input to data demodulation section 421 and channel decoding section 423, where data demodulation and channel decoding are performed. Note that the output of the reference signal separation unit 409 is input to the demapping unit 419 first, only the radio resource component corresponding to the own device is input to the propagation path compensation unit 417, and the output of the propagation path compensation unit 417 is input to the data demodulation unit. It is good also as a structure which inputs into 421. FIG.

FIG. 11 is a diagram showing an example of bit error rate (BER) characteristics in the first embodiment of the present invention. Modulation scheme is QPSK, the number of transmitting antennas N _t multiplexed number of terminals U are both set to 4. Error correction is not performed, and it is assumed that there is no time variation of the propagation path. The frame format is such that the same frame as in FIG. 6 is repeatedly transmitted, but the number of transmission data vectors is 12. In addition, linear interpolation using two DMRSs is assumed. Note that feedback of propagation path information from the terminal device 3 to the base station device 1 is ideal. For comparison, a case where ideal channel estimation is assumed and a case where DMRS is transmitted using orthogonal resources (that is, a case where transmission is performed in the frame format of FIG. 5), which is a conventional method, are shown together. The total transmission power at each time is assumed to be constant, and the average value of the error rates of all terminal apparatuses 3 for only the transmission data vector portion is shown. For reference, the correction by the method 2 and the DMRS transmission power being quadrupled are also shown.

  As can be seen from FIG. 11, when the DMRS and the data signal are spatially multiplexed, the required SNR is increased by about 3 dB as compared with the conventional method even if correction is performed by the method 2. This is because the transmission power of DMRS is ¼ compared with the conventional method. If the transmission power of the DMRS is the same, the BER characteristics are substantially the same between the method of the present embodiment corrected by the method 2 and the method of transmitting the DMRS using orthogonal resources. When DMRS is transmitted using orthogonal resources, pilot signal insertion loss (number of data signals / (total number of resources)) is 0.75, whereas when DMRS and data signals are spatially multiplexed, Since the insertion loss is 0.9375, the transmission efficiency is about 1.25 times.

By the way, the signal candidate point of r _u / (2P) ^1/2 β is one of signal points in which the signal candidate point of the original modulation signal is periodically repeated in the signal point space. In the modulo calculation, a signal point closest to r _u / (2P) ^1/2 β is detected. The log-likelihood ratio can be calculated based on the distance (likelihood) between a periodically repeated signal point and r _u / (2P) ^1/2 β without performing a modulo operation. When data demodulation or channel decoding is performed based on the log likelihood ratio, the propagation channel compensator 417 does not have to perform a modulo operation.

  In this embodiment, it is assumed that OFDM signal transmission is performed and precoding is performed for each subcarrier. However, there is no limitation on the transmission scheme (or access scheme) and the precoding application unit. For example, the present embodiment is also applicable when precoding is performed for each resource block in which a plurality of subcarriers are grouped. Similarly, a single carrier-based access scheme (for example, single carrier frequency division multiple access (SC- (FDMA) method).

  As described above, in the downlink MU-MIMO transmission using nonlinear precoding, it is possible to suppress an increase in overhead related to DMRS insertion.

[2. Second Embodiment]
The first embodiment is directed to a method of suppressing an increase in overhead related to DMRS insertion by spatially multiplexing DMRS and a data signal. However, the method of the first embodiment can be realized with a device configuration almost the same as that of the conventional THP, but the channel estimation accuracy by DMRS is slightly lowered. The second embodiment is directed to a method for improving propagation path estimation accuracy by DMRS.

[2.1 Base station apparatus 1]
The configuration of the base station apparatus 1 according to the second embodiment is the same as that in FIG. However, since the signal processing in the mapping unit 105 and the precoding unit 107 is different from that of the first embodiment, the signal processing of the mapping unit 105 and the precoding unit 107 will be described below.

[2.2 Precoding unit 107 and mapping unit 105]
First, signal processing in the precoding unit 107 will be described. In the first embodiment, the perturbation vector search unit 303 does not add a perturbation term to the DMRS, thereby enabling spatial multiplexing of the DMRS and the data signal. By the way, in the non-linear precoding by THP, the perturbation term is not added to the transmission code addressed to the terminal device 3 where the transmission code is generated first. Therefore, in the second embodiment, addition of perturbation terms to the DMRS is avoided by appropriately setting the ordering for changing the order in which the transmission codes are generated.

  FIG. 12 is a diagram illustrating an example of a DMRS frame format according to the second embodiment of the present invention. As in FIG. 6, it is assumed that U = 4, a signal in which DMRS and data are mixed is transmitted from time t = 1 to time 4, and a signal in which only the data signal is spatially multiplexed is transmitted at subsequent times. Shall. The ordering position is represented by an antenna port number.

FIG. 13 is a flowchart of signal processing of the precoding unit 107 for DMRS vectors and transmission data vectors according to the second embodiment of the present invention. First, as shown in FIG. 12, the DMRSs are arranged so that the DMRSs addressed to the terminal apparatuses 3 are ordered in order to the antenna port 1, that is, the position where transmission encoding is performed first (step S301). By ordering in this way, it is not necessary to add a perturbation vector to the DMRS addressed to each terminal device 3. The ordering may be performed in the precoding unit 107, but may be performed in the mapping unit 105 by appropriately setting an antenna port for transmitting DMRS and data signal in advance. At this time, the DMRS vectors d ′ _p (1), d ′ _p (2), d ′ _p (3), and d ′ _p (4) in the second embodiment are d _p used in the first embodiment. Using (1), d _p (2), d _p (3), and d _p (4), it is given by equation (16).

Here, Π _u represents a permutation matrix for exchanging the rows of the matrix, and the permutation matrix appearing in Equation (16) is given by Equation (17), respectively.

Subsequently, the linear filter W is generated in the linear filter generation unit 301. In the second embodiment, a matrix _{ｕ u} H obtained by multiplying the permutation matrix _{ｕ u} by the propagation channel matrix H is regarded as a propagation channel matrix. The linear filter W is calculated as in the first embodiment (step S302). For example, for d ′ _p (1), a linear filter W may be generated by regarding Π ₁ H as a channel matrix. The following shall represent this linear filter W _1.

Subsequently, in the perturbation vector search unit 303, a transmission code vector is generated, and d ′ _p (1) is regarded as a transmission symbol vector, and the transmission code vector is generated in the same manner as the data signal in the first embodiment. (Step S303). For example, the transmission code vector generated from d ′ _p (1) is given by x _p (1) = ((Π ₁ H) W ₁ ) ⁻¹ (Π ₁ d _p (1) + 2δz _t (1)). It will be. Thereafter, transmission code vectors may be generated similarly for d ′ _p (2), d ′ _p (3), and d ′ _p (4).

  Subsequently, the transmission signal generation unit 305 calculates a power normalization term β (step S304). When power normalization is performed in the same manner as in the first embodiment, the power normalization term given by Equation (5) has a different value for each transmission signal vector. This is because the values of the linear filters used for the transmission signal vectors are different (the covariance matrix of the transmission code vector x does not change). Therefore, in the second embodiment, power normalization is performed so that the total transmission power of a plurality of transmission signal vectors is constant. For example, when power normalization is performed with transmission signal vectors transmitted at time t = 1 to 4, the power normalization term β is given by Equation (18).

Then, the transmission signal vector _{s p (t) = βW t} x p (t) is generated in the transmission signal generating unit 305 (step S305). This is the signal processing in the precoding unit 107 for the DMRS vector.

Subsequently, after time t = 5, that is, precoding is performed on the transmission data vector. In the second embodiment, as in the data signal, as shown in FIG. 12, the same ordering as that in the DMRS part is performed on four consecutive transmission data vectors. That is, when the transmission data vector at time t is d (t), ordering is performed so that Π _{((t−1)% 4 + 1)} d (t) is the transmission data vector (step S306). Then, using the linear filter W _{((t−1)% 4 + 1)} , the precoding as performed in the data signal portion in the first embodiment is performed (step S307). Subsequently, in the transmission signal generation unit 305, a transmission signal vector is applied, and the power normalization term generated in step S304 is used (step S308).

  By doing so, it is possible to make the power normalization term common to the DMRS portion and the data signal portion. That is, in the second embodiment, perturbation to DMRS is achieved by sharing a resource block composed of a plurality of DMRS vectors and an ordering applied to a resource block composed of a plurality of transmission data vectors. Spatial multiplexing of the data signal and DMRS is realized without adding terms. In the second embodiment, the correction for the power normalization term as in the first embodiment is not necessary.

  The mapping unit 105 may multiplex the data signal, DMRS, and CRS by appropriately ordering the DMRS and data signal as described above.

FIG. 14 is a diagram showing an example of resource allocation according to the second embodiment of the present invention. Basically, it is the same as that shown in FIG. 8, but the resource block composed of four data signal portions surrounded by a square with a thick solid line is a resource composed of a plurality of DMRS vectors as described above. The same ordering as the ordering performed on the block is performed, and the power normalization is performed with the portion surrounded by a square as one unit. Here, the same ordering means that the ordering is performed by using Π _{1 to} Π ₄ once in the resource block. For example, ordering may be performed using Π _{1 for} the resource indicated by i, Π _{2 for} the resource indicated by ii, Π _{3 for} the resource indicated by iii, and _{４ 4} for the resource indicated by iv. The same applies when DMRS vectors and data vectors are mixed in the resource block. For example, in the resource block in which d ′ _p (1) and d ′ _p (2) are arranged, the remaining 2 One of the data vectors may be performed ordering using [pi ₃ and [pi _4. As for the portion surrounded by a dotted line, it is sufficient to form four data signal groups together with the portions surrounded by other dotted lines to perform ordering and power normalization.

  The configuration of the resource block shown in FIG. 14 is merely an example, and the resource block may be configured by a combination of resources different from that in FIG.

  In addition, it is also possible to give a degree of freedom to the ordering for the data signal portion by changing the number of transmission signal vectors for power normalization. For example, unlike the equation (18), if the power normalization is performed including the data signal part, the ordering for the data signal part may be freely performed.

  FIG. 15 is a diagram illustrating an example of a frame format when the ordering method is changed between the DMRS vector and the transmission data vector in the second embodiment of the present invention. In this case, the signal processing in the linear filter generation unit 301 and the perturbation vector search unit 303 is the same as the method described so far except for the ordering method, but the power normalization performed in the transmission signal generation unit 305 is as follows. The total transmission power of the eight transmission signal vectors including the DMRS vector and the transmission data vector is constant.

  FIG. 16 is a diagram illustrating an example of resource allocation in the case where the ordering method is changed between the DMRS vector and the transmission data vector in the second embodiment of the present invention. The same ordering and power normalization are performed for each of the eight resources surrounded by a square. As before, the same ordering means that the eight permutation matrices for realizing the frame format of FIG. 15 are each used once for ordering. As for the part surrounded by the dotted line, a resource block composed of eight resources may be formed together with the part enclosed by the other dotted lines to perform ordering and power normalization. The grouping method is not limited to this, and the number of one resource block may be arbitrarily determined.

[2.3 Terminal device 3]
The configuration of the terminal device 3 is the same as that in FIG. 9, and the signal processing performed in each device is also performed in the propagation path estimation unit 411, except that the correction for the power normalization term β is not performed. Since it becomes the same as a form, description is abbreviate | omitted.

  As described in the first embodiment, when DMRS is transmitted periodically, propagation path estimation accuracy can be improved by performing appropriate interpolation. In some cases, it is necessary to know in which resource block unitization is performed. The unit of the resource block for which power normalization is performed may be determined in advance between the base station apparatus 1 and the terminal apparatus 3, and is adaptively changed according to the state of the propagation path and the reception quality. The base station apparatus 1 may notify control information associated with a resource block unit.

  In the second embodiment, by sharing an ordering method and a power normalization term for each resource block composed of a plurality of resources, spatial multiplexing of the DMRS and the data signal can be performed without adding a perturbation term to the DMRS. The target was to achieve this. According to the method of the second embodiment, unlike the first embodiment, the error between the power normalization term estimated by the DMRS and the power normalization term applied to the data signal portion is reduced. Can do.

[3. Third Embodiment]
In the first and second embodiments, the case where THP is used as a precoding technique has been targeted. By the way, the transmission signal vector s in the first embodiment can be transformed as shown in Expression (19).

In Expression (19), it can be seen that if the perturbation vector z _t is removed, a transmission signal vector of MU-MIMO using linear precoding based on the well-known ZF criterion is obtained. In other words, non-linear precoding is merely adding a perturbation term to a linear MU-MIMO transmission symbol based on linear precoding, and the transmission quality varies greatly depending on the added perturbation term. To do. The THP targeted in the first and second embodiments searches for a perturbation term that can reduce the transmission power of the transmission signal vector given by Equation (19). Other methods for searching for such perturbation terms include THP using vectoring and lattice reduction (LR), Vector perturbation (VP), and the like. It can be expressed in the form of equation (19). The third embodiment is directed to downlink MU-MIMO transmission using nonlinear precoding other than THP.

[3.1 Base station apparatus 1]
Since the configuration of the base station apparatus 1 is the same as that of the first and second embodiments, and only the signal processing in the precoding unit 107 is different, the precoding unit according to the third embodiment will be described below. The signal processing at 107 will be described.

[3.2 Precoding unit 107]
The device configuration of the precoding unit 107 is the same as that in FIG. 4, but the signal processing in each component device is different. In the following description, the frame format shown in FIG. 6 is assumed as the frame format of the transmission signal, but the frame format is not limited to this.

FIG. 17 is a flowchart illustrating signal processing of the precoding unit 107 according to the third embodiment of the present invention. In the precoding unit 107, first, the linear filter generation unit 301 calculates the linear filter W (step S401). As the linear filter W, a ZF filter represented by W = H ⁻¹ (or H ⁺ ), an MMSE filter represented by W = H ^H (HH ^H + γI) ⁻¹ , or the like can be considered.

  Subsequently, the perturbation vector search unit 303 searches for a perturbation term. The perturbation term search method is determined in accordance with the desired transmission quality and the amount of computation that can be realized by the arithmetic unit included in the base station apparatus 1. For example, when a VP that can achieve the highest reception quality is used, the perturbation term can be obtained by solving the minimization problem expressed by Equation (20).

Here, Z ^U _G represents a set of U-dimensional Gaussian integer vectors. The solution of equation (20), that is, the perturbation vector search method is not limited to a specific method, and is realized by a method such as Sphere encoding (SE) or QRM-VP based on M algorithm and QR decomposition. Also good. Further, instead of VP, the perturbation term may be searched by lattice base reduction. Further, the perturbation term may be searched for by LRA-THP (LR aided-THP), which is a THP advancement technique targeted in the first and second embodiments, or by ordering THP using ordering. The ordering referred to here includes ordering such as the BLAST method for the purpose of improving the reception quality in addition to what is applied to the DMRS vector in the second embodiment.

  When the DMRS vector is input to the perturbation vector search unit 303, the perturbation vector search unit 303 searches for the perturbation term by solving the minimization problem expressed by the equation (21) (step S402).

  That is, although the perturbation term is searched as in the conventional VP, the perturbation term is not added to the DMRS included in the DMRS vector. The same applies when searching for the perturbation term by a method other than VP. When the number of spatially multiplexed terminals is large, high-precision nonlinear precoding such as VP or ordering THP can obtain a large spatial diversity effect. Therefore, there is no significant difference in transmission power between the case where the perturbation term is searched based on Equation (20) and the case where the perturbation term is searched based on Equation (21). Even if the addition is not performed, the reception quality is not greatly reduced.

After perturbation vector _z t (t) is probed, the transmission signal generating unit 305, transmission signal to the DMRS Vector _{s p (t) = W (} d p (t) + 2δz t (t)) is generated once (Step S403).

Subsequently, the perturbation vector search unit 303 searches for the perturbation term {z _t (t); t> 4} for the transmission data vector {d (t); t> 4} (step S404). Unlike the DMRS vector, there is no need to give a restriction such as not adding a perturbation term to some signals. Thereafter, the transmission signal generation unit 305 once generates a transmission signal vector s (t) = W (d (t) + 2δz _t (t)) for the transmission data vector (step S405).

  In precoding section 107, power normalization is finally performed on the generated transmission signal vector (step S406). Unlike the THP targeted in the first or second embodiment, when nonlinear precoding such as VP is performed, the power of each transmission signal vector is not only transmitted to the state of the propagation path but also to each terminal device 3 Since it varies depending on the transmission symbol, when power normalization is performed for each resource, information necessary for demodulating the received signal cannot be estimated only by periodically transmitting DMRS.

  Therefore, in the third embodiment, power normalization is performed for each resource block composed of a plurality of resources. For example, what is necessary is just to obtain | require power normalization term (beta) like Formula (22).

Here, N _d is the number of resources included in the resource block that is performing power normalization at the same time, and may be an appropriate value as appropriate. In order to improve the accuracy of estimation of the power normalization term by DMRS, It is desirable that a resource block to be converted includes a DMRS vector. Note that increasing the number of resources included in the resource block means that the number of transmission signal vectors with constant average received power increases. Therefore, depending on the coding rate of channel coding applied in the channel coding unit 101, power normalization with a very large resource block may not be preferable. Therefore, control may be performed so as to change the size of the resource block in accordance with the coding rate. For example, when the coding rate is high, it is conceivable to increase the number of resources for power normalization, and when the coding rate is low, control may be performed to reduce the number of resources.

  The above is the signal processing in the precoding unit 107 according to the third embodiment. Since the signal processing of other parts is the same as in the first and second embodiments, the description thereof is omitted.

[3.3 Terminal device 3]
The configuration of the terminal device 3 is the same as that in FIG. 9, and the signal processing performed in each device is also performed in the propagation path estimation unit 411, except that the correction for the power normalization term β is not performed. Since it becomes the same as a form, description is abbreviate | omitted.

  In the third embodiment, nonlinear precoding other than THP is targeted. According to the third embodiment, MU-MIMO based on more accurate nonlinear precoding can be realized.

[4. Fourth Embodiment]
The third embodiment is directed to a method for realizing spatial multiplexing of DMRS and a data signal for any nonlinear precoding. In the third embodiment, only the DMRS included in the DMRS vector does not perform perturbation term addition. The fourth embodiment is directed to a method for improving transmission characteristics by not adding a perturbation term for a partial data signal included in a transmission data vector. Hereinafter, a data signal without adding a perturbation term is called a linear data signal (first data signal), and a data signal with adding a perturbation term is called a non-linear data signal (second data signal). Taking the first and second embodiments as an example, the signal transmitted through the antenna port 1 is a linear data signal, and the signals transmitted through other antenna ports are nonlinear data signals.

[4.1 Base station apparatus 1]
The configuration of the base station apparatus 1 is the same as that of the third embodiment, and only the signal processing in the precoding unit 107 is different. Therefore, in the following, the signal in the precoding unit 107 according to the fourth embodiment is described. Processing will be described.

[4.2 Precoding unit 107]
The device configuration of the precoding unit 107 is the same as that in FIG. 4, but the signal processing in each component device is different. In the following description, the frame format shown in FIG. 6 is assumed as the frame format of the transmission signal, but the frame format is not limited to this.

  FIG. 18 is a flowchart illustrating signal processing of the precoding unit 107 according to the fourth embodiment of the present invention. The signal processing for the DMRS vector, that is, steps S501 to S503 are the same as steps S401 to S403 in FIG. Subsequently, the perturbation term to be added to the transmission data vector is searched by the perturbation vector search unit 303. In the fourth embodiment, unlike the third embodiment, the transmission data vector is also a DMRS vector. Further, the perturbation term is searched for without adding the perturbation term to the partial data signal (step S504). The determination method of the terminal device 3 that does not add the perturbation term to the transmission data may be arbitrarily determined.

  FIG. 19 is a diagram illustrating an example of a transmission frame format when a perturbation term is not added to part of transmission data in the fourth embodiment of the present invention. Here, the data signal shown in black is a linear data signal, and the others are non-linear data signals.

  The mapping information indicating how the linear data signal and the nonlinear data signal are multiplexed needs to be understood by the terminal device 3 as well. As a method of notifying the mapping information, it may be notified separately by control information, and if a periodic multiplexing method as shown in FIG. 19 is used, linear data is transmitted between the base station apparatus 1 and the terminal apparatus 3. An interval at which signals are multiplexed may be determined in advance. Note that the multiplexing interval of the linear data signal is associated with the antenna port to which the DMRS is transmitted, and the antenna port to which the DMRS is transmitted is notified to each terminal device 3, thereby notifying the multiplexing position of the linear data signal. You may control to do.

  The nonlinear data multiplexing method is not limited to a specific method. As shown in FIG. 19, the signals may be periodically multiplexed. In an extreme case, a signal addressed to a specific terminal device 3 is a linear data signal, and other signals addressed to the terminal device 3 are nonlinear data. You may control so that it may become a signal. However, it is desirable to multiplex based on specific regularity so that the position of the nonlinear data signal can be estimated from the antenna port through which DMRS is transmitted.

  Returning to FIG. 18, after the search for the perturbation term in the perturbation vector search unit 303 is completed, the transmission signal generation unit 305 generates a transmission signal vector for the transmission data vector (step S505). Thereafter, power normalization is performed on the generated transmission signal vector to calculate a transmission signal vector to be output from the precoding unit 107 (step S506). As a method of power normalization, normalization may be performed for each resource block composed of a plurality of resources, as in the third embodiment, but in the fourth embodiment, a DMRS vector portion and a transmission The ratio of the signal to which the perturbation term is added is the same in the data vector portion. Therefore, as in the first and second embodiments, power normalization may be performed for each resource.

[4.3 Terminal device 3]
The configuration of the terminal device 3 is the same as in FIG. 9 and the signal processing performed in each device is substantially the same as in the third embodiment, but the signal processing in the propagation path compensation unit 417 and the channel decoding unit 423 is different.

  In the channel compensation unit 417, the channel equalization based on the channel estimation value estimated by the DMRS is performed on the data signal as in the third embodiment. In the embodiment, the modulo calculation performed after that is not performed by the propagation path compensation unit 417.

  Next, signal processing in the channel decoding unit 423 will be described. In the following, for channel codes applied to data signals, decoding that requires a log likelihood ratio (LLR) of the received signal for decoding (for example, Log-MAP decoding, etc.) ) Shall apply. Hereinafter, an LLR calculation method in the channel decoding unit 423 in the fourth embodiment will be described.

  As described in the description of the precoding unit 107, in the fourth embodiment, a linear data signal and a nonlinear data signal are mixedly received in the data signal received by each terminal device 3. Each terminal device 3 receives a linear data signal and nonlinear data based on control information separately notified from the base station device 1 or control information for notifying an antenna port to which DMRS is transmitted. It is assumed that it is possible to grasp which of the signals. Channel decoding section 423 changes the LLR calculation method according to the type of data signal being received.

Now, it is assumed that precoding of the base station apparatus 1 and channel equalization of the terminal apparatus 3 are ideally performed, and the received signals of the linear data signal and the nonlinear data signal are represented by r _L and r _N , respectively. Each is given by equation (23). For simplicity, all indexes indicating resources and the like are omitted.

  Here, n ′ represents equivalent noise after channel equalization. For simplicity, the transmission symbol d is assumed to be a BPSK modulated signal.

First, an LLR calculation method when a linear data signal is received will be described. The LLR has a conditional probability p (r _L | d = + 1) of the received signal r _L when the transmission symbol is d = + 1, that is, “1” is transmitted as a transmission bit, and the transmission symbol is d = −1. It is calculated by taking the logarithm of the ratio with the conditional probability p (r _L | d = −1) of the reception signal r _L when “0” is transmitted as the transmission bit.

On the other hand, in the LLR calculation method when a non-linear data signal is received, the condition of the received signal r _N when the transmission symbol is d = + 1, that is, “1” is transmitted as the transmission bit, as with the linear data signal. It is necessary to obtain a conditional probability in consideration of the perturbation term added to the data signal. That is, when not only d = + 1 but also the conditional probabilities in the case of d = + 1 ± 2δ, + 1 ± 4δ,..., And all of them are added, the transmission bit is “1” The conditional probability of However, since it is difficult to consider all the perturbation terms, only the point closest to the received signal among d = + 1 ± 2δ, + 1 ± 4δ,. The conditional probability when the transmission bit is 0 is similarly obtained, and the LLR of the nonlinear data signal can be calculated by taking the logarithm of the ratio of the two probabilities.

  Usually, when the received power is the same, the LLR of the linear data signal and the LLR of the non-linear data signal can take values with higher accuracy. Therefore, as in the fourth embodiment, a linear data signal and a non-linear data signal are mixed and transmitted, so that a high error correction capability can be obtained as compared with the case where channel decoding is performed as a non-linear data signal. Is possible.

  In the fourth embodiment, as in the case where DMRS and data signals are spatially multiplexed, a method in which perturbation terms are not added is also targeted for some data signals included in the transmission data vector. According to this method, it is possible to improve the coding gain by channel coding.

[5. Common to all embodiments]
The embodiment of the present invention has been described in detail with reference to the drawings. However, the specific configuration is not limited to this embodiment, and the design and the like within the scope of the present invention are also within the scope of the claims. include.

  The program that operates in the mobile station apparatus and the base station apparatus 1 related to the present invention is a program (a program that causes a computer to function) that controls the CPU and the like so as to realize the functions of the above-described embodiments related to the present invention. Information handled by these devices is temporarily stored in the RAM at the time of processing, then stored in various ROMs and HDDs, read out by the CPU, and corrected and written as necessary. As a recording medium for storing the program, a semiconductor medium (for example, ROM, nonvolatile memory card, etc.), an optical recording medium (for example, DVD, MO, MD, CD, BD, etc.), a magnetic recording medium (for example, magnetic tape, Any of a flexible disk etc. may be sufficient. In addition, by executing the loaded program, not only the functions of the above-described embodiment are realized, but also based on the instructions of the program, the processing is performed in cooperation with the operating system or other application programs. The functions of the invention may be realized.

  In the case of distribution in the market, the program can be stored and distributed in a portable recording medium, or transferred to a server computer connected via a network such as the Internet. In this case, the storage device of the server computer is also included in the present invention. Moreover, you may implement | achieve part or all of the mobile station apparatus and base station apparatus 1 in embodiment mentioned above as LSI which is typically an integrated circuit. Each functional block of the mobile station apparatus and the base station apparatus 1 may be individually made into a processor, or a part or all of them may be integrated into a processor. Further, the method of circuit integration is not limited to LSI, and may be realized by a dedicated circuit or a general-purpose processor. In addition, when an integrated circuit technology that replaces LSI appears due to progress in semiconductor technology, an integrated circuit based on the technology can also be used.

DESCRIPTION OF SYMBOLS 1 Base station apparatus 3, 3-1, 3-2, 3-3, 3-4, 3-u, 3-U terminal apparatus 101 Channel encoding part 103 Data modulation part 105 Mapping part 107 Precoding part 109 Antenna part 111 control information acquisition unit 113 CSI acquisition unit 201 IFFT unit 203 GI insertion unit 205 wireless transmission unit 207 wireless reception unit 209 antenna 301 linear filter generation unit 303 perturbation vector search unit 305 transmission signal generation unit 401 antenna 403 wireless reception unit 405 GI removal Unit 407 FFT unit 409 reference signal separation unit 411 propagation channel estimation unit 413 feedback information generation unit 415 wireless transmission unit 417 propagation channel compensation unit 419 demapping unit 421 data demodulation unit 423 channel decoding unit

Claims (9)

Multi-user MIMO (Multiple MIMO) for multiple wireless receivers with multiple antennas
input multiple output) a wireless transmission device that performs transmission,
A unique reference signal transmitted to each of the plurality of wireless reception devices and a data signal addressed to at least one wireless reception device other than the destination of the unique reference signal are used as elements, and each element is a space with the same radio resource. Precoding is performed on a DMRS (Demodulation Reference Signal) vector to be multiplexed,
Linear precoding is performed on the unique reference signal included in the DMRS vector,
A radio transmitting apparatus, wherein the pre-coded DMRS vector is spatially multiplexed and transmitted to each radio receiving apparatus.   A transmission data vector in which the data signals addressed to the plurality of radio reception devices are elements, and each element is spatially multiplexed with the same radio resource, and the transmission data vector and the DMRS vector in different radio resources. The radio transmission apparatus according to claim 1, wherein nonlinear precoding is performed on a data signal included in the DMRS vector. 3. The radio according to claim 1, wherein each of the transmission data vector and the DMRS vector is multiplied by a power normalization term that makes a sum of transmission powers in at least one of the radio resources constant. Transmitter device. The ratio of a power normalization term to be multiplied to the transmission data vector and a power normalization term to be multiplied to the DMRS vector, and a phase of the unique reference signal are associated with each other . Wireless transmission device. In the precoding, the transmission data vector and the DMRS vector are each ordered.
The radio transmission apparatus according to claim 2, wherein the ordering applied to the transmission data vector and the ordering applied to the DMRS vector are common . The transmission data vector includes a first data signal and a second data signal,
The radio transmission apparatus according to claim 2 , wherein the radio resource to which the first data signal is transmitted is associated with the radio resource to which the unique reference signal is transmitted . A radio reception device that includes at least one antenna and receives a radio signal transmitted by a multi-user MIMO (Multiple input multiple output) from a radio transmission device having a plurality of antennas,
A transmission data vector whose elements are data signals addressed to a plurality of radio reception devices from the radio transmission device, and a unique reference signal transmitted to each of the plurality of radio reception devices and at least other than the destination of the unique reference signal DMRS (Demodulation Reference Signal) vectors whose elements are data signals addressed to one radio reception apparatus receive radio signals that are precoded and spatially multiplexed in different radio resources,
Based on the unique reference signal addressed to its own device, the propagation path state is estimated to generate propagation path state information,
A ratio between a power normalization term multiplied by the transmission data vector and a power normalization term multiplied by the DMRS vector, and a phase of the unique reference signal addressed to the own device are associated;
Based on the phase of the unique reference signal addressed to the device itself, correct the propagation path state information,
A radio receiving apparatus that demodulates a data signal addressed to the own apparatus included in the transmission data vector and the DMRS vector based on the propagation path state information. A method of demodulating a data signal addressed to the own device included in the transmission data vector and the DMRS vector based on a position of a radio resource in which a DMRS vector including a unique reference signal addressed to the own device is transmitted. The radio reception apparatus according to claim 7, wherein the radio reception apparatus is switched . Multi-user MIMO (Multiple MIMO) for multiple wireless receivers with multiple antennas
input multiple output) a precoding method applied to a wireless transmission device that performs transmission,
A unique reference signal transmitted to each of the plurality of wireless reception devices and a data signal addressed to at least one wireless reception device other than the destination of the unique reference signal are used as elements, and each element is a space with the same radio resource. Precoding is performed on a DMRS (Demodulation Reference Signal) vector to be multiplexed,
A precoding method comprising performing linear precoding on a specific reference signal included in the DMRS vector.

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