JP2009527968A - In-phase and quadrature path imbalance compensation - Google Patents

Abstract

The direct down-conversion receiver (19) has an imbalance compensation parameter determination stage (20) and a phase and amplitude imbalance compensation stage (21). The decision stage (20) is connected to receive the I and Q signals I ′ (t), Q ′ (t) output from the I and Q paths (5, 6) of the receiver (19). The determination stage (20) determines imbalance compensation parameters g _I cos (φ ₁ ), g _I sin (φ ₁ ), g _Q cos (φ ₃ ), and g _Q sin (φ ₃ ) from the training sequence of the received signal. To do. Thereafter, the decision stage calculates the average value of each of the determined imbalance compensation parameters g _I cos (φ ₁ ), g _I sin (φ ₁ ), g _Q cos (φ ₃ ), and g _Q sin (φ ₃ ). with calculating, from these average values are calculated further imbalance compensation parameter _{g I g Q sin (φ 3} -φ 1). The phase and amplitude imbalance compensation stage (21) includes the determined imbalance compensation parameters g _I cos (φ ₁ ), g _I sin (φ ₁ ), g _Q cos (φ ₃ ), g _Q sin (φ ₃ ). And the calculated further imbalance compensation parameter g _I g _Q sin (φ ₃ −φ ₁ ) and the phase between the I and Q paths (5, 6) of the receiver during reception of the payload of the received signal and Compensate for amplitude imbalance.

Description

  The present invention relates to in-phase (I) and quadrature (Q) path imbalance compensation. In particular, the present invention relates to, but is not limited to, a receiver such as a direct down-conversion radio receiver that incorporates I / Q path imbalance compensation.

  Many receivers use quadrature modulation where the received signal needs to be processed over both an in-phase (I) path and a quadrature (Q) path. For example, referring to FIG. 1, a conventional direct down-conversion receiver 1 has an antenna 2 for receiving a signal. The antenna 2 is connected so as to output the received signal to a radio frequency (RF) filter 3 that is a band pass filter for filtering the received signal. The RF filter 3 is connected to output a filtered signal to a low noise amplifier (LNA) 4 for amplifying the filtered signal. This amplification helps to reduce noise in subsequent processing of the received signal. The LNA 4 is connected to output an amplified signal to both the in-phase (I) path 5 and the quadrature (Q) path 6 of the receiver 1. The I path 5 includes an I mixer 7, an analog low-pass filter 8, an automatic gain controller (AGC) 9, an analog-digital converter (ADC) 10, and a digital low-pass filter 11 connected in series in this order. I have. Q path 6 is identical to the I path except that it has Q mixer 12 instead of I mixer 7. Therefore, the Q path 6 includes a Q mixer 12, an analog low-pass filter 13, an AGC 14, an ADC 15, and a digital low-pass filter 16. The signals output by the I and Q paths 5 and 6 are sent to a demodulation and decoding stage 17 to demodulate and decode the signals to produce a data output 18.

  Each of the I mixer 7 and the Q mixer 12 mixes the amplified signal output by the LNA 4 with the local oscillation. If the receiver 1 is a direct down-conversion receiver, the local oscillation signal must have approximately the same frequency as the carrier frequency of the desired signal modulated with the signal received by the antenna 2. Thereby, the mixers 7 and 12 can down-convert a desired signal to the baseband. Ideally, the local oscillator signals used by the I and Q mixers are phase shifted by 90 ° from each other so that the desired signal is properly demodulated from the signals on the I path 5 and Q path 6. it can. However, this is difficult to implement in practice, and in many cases the phase difference between the local oscillator signals used by the I and Q mixers 7 and 12 varies. This variation is known as phase imbalance. Similarly, the gains over I path 5 and Q path 6 may not be the same. Therefore, the signals output by the I and Q paths 5 and 6 may have an unintended amplitude difference. This is known as amplitude imbalance. Phase and amplitude imbalance can cause errors in demodulation. This can cause aliasing of image interference to the desired signal band.

  Many methods for correcting phase and amplitude imbalance have been proposed, most of which involve calibrating the receiver using a calibration signal. In general, the calibration signal has a known frequency, phase and amplitude. As the calibration signal is passed through the receiver, the signal output by the I and Q paths can be analyzed to determine the phase and amplitude imbalance between them. There are many problems associated with such methods. For example, the calibration signal is most often generated locally by the receiver to be calibrated so that its frequency, phase and amplitude can be carefully controlled. However, this tends to require a dedicated circuit for generating the calibration signal, which is undesirable because it is expensive and complex. Some methods circumvent this by receiving an external calibration signal via an antenna for the receiver to normally receive the signal. This avoids the additional cost and complexity of generating the calibration signal locally at the receiver, but is not practical because it occupies space in the frequency spectrum that can be used for communication or the like. All these methods also prevent the receiver from receiving other signals during calibration. Therefore, communication via the receiver or reception of the broadcast signal at the receiver may be interrupted during calibration. This is clearly undesirable. Predict when a signal will be received, even for receivers operating in time division multiple access (TDMA) mode, time division duplex (TDD) mode, etc. Is adversely affected by this problem because it can be difficult and therefore can be difficult to allocate the appropriate time for calibration. Thus, conventional methods for correcting phase and amplitude imbalances are far from ideal.

  The present invention seeks to overcome the aforementioned problems.

According to the first aspect of the present invention,
A receiver for receiving a signal including a training sequence and a payload,
An in-phase path and a quadrature path for converting the received signal into an in-phase signal and a quadrature signal for output to the demodulator;
Processing means for determining an imbalance compensation parameter from in-phase and quadrature signals representing a training sequence;
An imbalance compensation circuit for compensating for a phase and amplitude imbalance between the in-phase and quadrature signals representing the payload using the determined imbalance compensation parameters;
A receiver characterized by comprising:

According to a second aspect of the invention,
A method for receiving a signal including a training sequence and a payload, comprising:
Converting received signals into in-phase and quadrature signals for output to a demodulator;
Determining an imbalance compensation parameter from the in-phase and quadrature signals representing the training sequence;
Compensating the phase and amplitude imbalance between the in-phase and quadrature signals representing the payload using the determined imbalance compensation parameters;
Is provided.

  That is, the phase and amplitude imbalance compensation during reception of the signal payload can be based on the signal training sequence. Training sequences are an inherent part of many current communication system signals. As such, the present invention has the significant advantage of being able to compensate for phase and amplitude imbalances without the use of dedicated calibration signals. The calibration signal need not be generated locally at the receiver. Similarly, there is no need to ensure communication capacity for receiving dedicated calibration signals.

  Thus, the training sequence may also be used for purposes other than determining imbalance compensation parameters. This may include further signal processing. That is, the processing means may use the training sequence for further signal processing. Similarly, the method may include using a training sequence for further signal processing. In one example, further signal processing may include training the equalizer. In other examples, further signal processing may include any one of synchronization, channel quality assessment, signal level adjustment, and the like.

  It will be appreciated that the training sequence may take various forms. Fundamentally, the training sequence is usually part of a known received signal. For example, a signal that is expected to be received during a training sequence may be known. The training sequence may include a preamble, a pilot sequence, an access code, a synchronization word, a synchronization sequence, and the like. In any case, the training sequence is usually adapted to a specific purpose in addition to amplitude and phase imbalance compensation. For example, the training sequence may meet certain conventional communication standard requirements. In particular, the training sequence may be suitable for training the equalizer or may be suitable for enabling synchronization. In general, this means that the training sequence may comprise a sequence with a significant peak in its autocorrelation function.

  The training sequence and payload may be included in the data packet. Thus, compensation of the signal representing the payload of the data packet may be performed using an imbalance compensation parameter determined from the signal representing the training sequence of the data packet. That is, imbalance compensation can be achieved from a signal representing a single data packet. Indeed, imbalance compensation can be performed on a packet-by-packet basis as needed.

The compensation parameter is preferably determined in the time domain rather than in the frequency domain. This is because the computation in the time domain is quick and easy, thereby avoiding the use of Fourier transforms. For example, the receiver preferably comprises a selector for selecting a part of the training sequence in-phase and quadrature signals at two different times. Thus, the imbalance compensation parameter can be determined from the in-phase and quadrature signal portions selected at two different times. These times are indicated as times t ₁ and t ₂ in the following description of the preferred embodiment of the present invention. In general, some of the in-phase and quadrature signals are selected from some of the training sequences that are also used for further processing as described above. That is, a specific part of the training sequence that is used only for amplitude and phase imbalance compensation is not necessary.

It is also beneficial to determine the imbalance compensation parameter based on a signal representing a training sequence that is expected to be received. These are indicated by baseband signals A (t ₁ ), A (t ₂ ), B (t ₁ ), B (t ₂ ) in the following description of the preferred embodiment of the present invention.

An imbalance compensation parameter may be determined based on the in-phase signal, and other imbalance compensation parameters may be determined based on the quadrature signal. Indeed, it is preferred that a pair of imbalance compensation parameters is determined from the in-phase signal and another pair of imbalance compensation parameters is determined from the quadrature signal. In the following description of the preferred embodiment of the present invention, these are imbalance compensation parameters g _I cos (φ ₁ ), g _I sin (φ ₁ ) and imbalance compensation parameters g _Q cos (φ ₃ ), g _Q It is indicated by each of sin (φ ₃ ). Further, it is preferable that a further imbalance compensation parameter is determined from an average value of these imbalance compensation parameters. This is, in the following description of preferred embodiments of the present invention is illustrated by further imbalance compensation parameter _{g I g Q sin (φ 3} -φ 1).

  Phase and amplitude imbalance compensation can use these parameters. More specifically, the compensation can include multiplying the in-phase signal and the quadrature signal by the determined imbalance compensation parameter. The compensation preferably includes multiplying the in-phase signal by an imbalance compensation parameter determined from the quadrature signal, and multiplying the quadrature signal by an imbalance compensation parameter determined from the in-phase signal. The compensation also preferably includes multiplying the in-phase signal and the quadrature signal by the reciprocal of the determined further imbalance compensation parameter. This is illustrated by equations (6) and (7) in the following description of the preferred embodiment of the present invention.

  The use of the terms “processing means”, “circuit”, “selector” and the like is intended to be general rather than specific. The present invention may be implemented using such separate components. However, the present invention may be similarly implemented using individual processors such as a digital signal processor (DSP) or a central processing unit (CPU). Similarly, the present invention may be implemented using one or more wiring circuits, such as application specific integrated circuits (ASICs), or by embedded software. Indeed, it will also be understood that the present invention may be implemented using computer program code. Thus, according to a further aspect of the present invention there is provided computer software or computer program code adapted to perform the method described above when processed by the processing means. Computer software or computer program code can be carried by a computer-readable medium. The medium may be a physical storage medium such as a read only memory (ROM) chip. Alternatively, the medium may be a disk such as a digital versatile disk (DVD-ROM) or a compact disk (CD-ROM). The medium can also be a signal such as an electronic signal, an optical signal, or a radio signal to a satellite via a wiring. The invention also extends to a processor executing software or code, such as a computer configured to perform the method described above.

  Now, preferred embodiments of the present invention will be described with reference to the accompanying drawings.

  Referring to FIG. 2, the direct down-conversion receiver 19 according to the first preferred embodiment of the present invention has many components similar to those of the prior art receiver 1 described above with reference to FIG. Therefore, the same reference numerals are used for similar components. However, between the I and Q paths 5, 6 and the demodulation and decoding stage 17, the receiver 19 according to the first preferred embodiment of the present invention has an unbalance compensation parameter determination stage 20 and a phase and amplitude imbalance. And a compensation stage 21.

The imbalance compensation parameter determination stage 20 is connected to the I and Q paths 5 and 5 of the receiver 19 after filtering by the analog low-pass filters 8 and 13, conversion from analog to digital by the ADCs 10 and 15, and filtering by the digital low-pass filters 11 and 16. 6 is connected to receive the I and Q signals I ′ (t) and Q ′ (t) output from 6. The decision stage 20 is shown in more detail in FIG. As can be seen from FIG. 3, the decision stage comprises a selector 22 for selecting a part of the I and Q signals I ′ (t), Q ′ (t) output and received by the I and Q paths 5 and 6. And processing selected portions of the signals I ′ (t) and Q ′ (t) to obtain imbalance compensation parameters g _I cos (φ ₁ ), g _I sin (φ ₁ ), g _Q cos (φ ₃ ). , and a _g Q sin _{(φ 3)} and _{g I g Q sin (φ 3} -φ 1) processor 23 for calculating to determine.

ここで、Ａ（ｔ）及びＢ（ｔ）はキャリア信号ｃｏｓ（ｗ_ｃｔ＋φ_１）及びｓｉｎ（ｗ_ｃｔ＋φ_１）へ変調されたベースバンド信号、φ_１はキャリア信号の位相、キャリア周波数ｆ_ｃ＝ｗ_ｃ／２πである。レシーバ１９のＩ及びＱミキサ７，１２は、受信信号Ｓ（ｔ）と、９０°だけ互いに対して位相シフトされた二つの局部発振信号とを混合させて、信号Ｓ（ｔ）をＩ及びＱ経路５，６上のベースバンド信号へと変換する。混合後のＩ経路５上及びＱ経路６上の信号はそれぞれ以下のように表すことができる。

及び

ここで、θは、所望の９０°位相差からの任意の変化を表しており、ｇ_Ｉ，ｇ_Ｑは、Ｉ経路５及びＱ経路６における全体の利得係数である。理想的には、位相変化θはゼロであり、即ち、θ＝０であり、また、利得係数ｇ_Ｉ，ｇ_Ｑは等しく、即ち、ｇ_Ｉ＝ｇ_Ｑである。しかしながら、回路欠陥や温度変化等に起因して、位相変化θはゼロではなく、即ち、θ≠０であり、また、利得係数ｇ_Ｉ，ｇ_Ｑは等しくなく、即ち、ｇ_Ｉ≠ｇ_Ｑである。周波数約２ｗ_ｃを有する混合後のＩ経路５上及びＱ経路６上の信号Ｉ（ｔ），Ｑ（ｔ）の成分は、ローパスフィルタ８，１１，１３，１６によって比較的容易に除去される。従って、Ｉ及びＱ経路５，６から出力される信号Ｉ’（ｔ），Ｑ’（ｔ）はそれぞれ、以下のように更に簡単に表すことができる。

及び

ここで、φ_３＝φ_１−（θ−π／２）＝φ_１−θ＋π／２である。そのため、これらの方程式（４）及び（５）は、セレクタ２２によって選択されるＩ及びＱ信号Ｉ’（ｔ），Ｑ’（ｔ）の部分を表すために考慮され得る。 A signal received by the antenna 2 can be expressed as follows.

Here, A (t) and B (t) are carrier signals cos (w _c t + φ ₁ ) and baseband signals modulated into sin (w _c t + φ ₁ ), φ ₁ is the phase of the carrier signal, and carrier frequency f _c = W _c / 2π. The I and Q mixers 7 and 12 of the receiver 19 mix the received signal S (t) with two local oscillation signals that are phase shifted relative to each other by 90 ° to generate the signal S (t) as I and Q. Convert to baseband signals on paths 5 and 6. The signals on the I path 5 and the Q path 6 after mixing can be expressed as follows.

as well as

Here, θ represents an arbitrary change from the desired 90 ° phase difference, and g _I and g _Q are overall gain coefficients in the I path 5 and the Q path 6. Ideally, the phase change θ is zero, ie, θ = 0, and the gain factors g _I , g _Q are equal, ie, g _I = g _Q. However, due to circuit defects and temperature changes, the phase change θ is not zero, that is, θ ≠ 0, and the gain coefficients g _I and g _Q are not equal, that is, g _I ≠ g _Q. is there. The components of the signals I (t) and Q (t) on the mixed I path 5 and Q path 6 having a frequency of about 2 w _c are relatively easily removed by the low-pass filters 8, 11, 13, and 16. . Therefore, the signals I ′ (t) and Q ′ (t) output from the I and Q paths 5 and 6 can be expressed more simply as follows.

as well as

Here, φ ₃ = φ ₁ − (θ−π / 2) = φ ₁ −θ + π / 2. Therefore, these equations (4) and (5) can be considered to represent the parts of the I and Q signals I ′ (t), Q ′ (t) selected by the selector 22.

及び

ここで、φ_３≠φ_１±ｎπ，ｎ＝０，１，２．．．．．．である。 Equations (4) and (5) can be rewritten to give:

as well as

Here, φ ₃ ≠ φ ₁ ± nπ, n = 0, 1, 2,. . . . . . It is.

During transmission of the training sequence, φ ₁ , φ ₃ , g _I and g _Q are unknown, but the baseband signals A (t) and B (t) used to modulate the carrier signal are known. That is, the signal expected to be received during the training sequence is known. Therefore, the selector 22 receives a portion I ′ (t) of the signals I ′ (t) and Q ′ (t) output from the I and Q paths 5 and 6 at two different times t ₁ and t ₂ during reception in the training sequence. t ₁ ), I ′ (t ₂ ), Q ′ (t ₁ ), Q ′ (t ₂ ) are selected. In this embodiment, the training sequence is received at the beginning of a data packet that also includes a payload and is used to train an equalizer (not shown) in the receiver 19.

More specifically, the imbalance compensation parameters g _I cos (φ ₁ ), g _I sin (φ ₁ ) are expressed as (A (t ₁ ) B (t ₂ ) using the following equations (8) and (9): _{≠ A (t 2) B (} t 1) on the condition that), the signal I output from the I path 5 'signal portions being selected by (t) from the selector _{22 I' (t 1),} I '(t 2 ) And the known baseband signals A (t ₁ ), A (t ₂ ) B (t ₁ ), B (t ₂ ) of the training sequence at two different times t ₁ and t ₂ .

Similarly, the compensation parameters g _Q cos (φ ₃ ), g _Q sin (φ ₃ ) are calculated using the following equations (10) and (11) (A (t ₁ ) B (t ₂ ) ≠ A ( t ₂ ) under the condition of B (t ₁ ), signal portions Q ′ (t ₁ ), Q ′ (t ₂ ) selected by the selector 22 from the signal Q ′ (t) output from the Q path 6, It is determined from the known baseband signals A (t ₁ ), A (t ₂ ) B (t ₁ ), B (t ₂ ) of the training sequence at two different times t ₁ and t ₂ .

In this embodiment, the imbalance compensation parameters g _I cos (φ ₁ ), g _I determined using the equations (8), (9), (10), (11) during the entire training sequence. sin (φ ₁ ), g _Q cos (φ ₃ ), g _Q sin (φ ₃ ) are averaged over a predetermined time, and the parameters g _I cos (φ ₁ ), g _I sin (φ ₁ ), g _Q An average value of cos (φ ₃ ) and g _Q sin (φ ₃ ) can be given.

Therefore, the determined imbalance compensation parameters g _I cos (φ ₁ ), g _I sin (φ ₁ ), g _Q cos (φ ₃ ), g _Q sin (φ ₃ ) and further calculated imbalance compensation parameters substituting g _I g _Q sin a (phi ₃ -.phi ₁₎ in equation (6) and (7), used to modulate the carrier signal _{cos (w c t + φ 1} ) and _{sin (w c t + φ 1} ) It can be seen that baseband signals A (t) and B (t) can be obtained. In practice, this allows the baseband signals A (t), B (t) used to modulate the carrier signals cos (w _c t + φ ₁ ) and sin (w _c t + φ ₁ ) to be known (eg, , During the training sequence) the imbalance compensation parameters g _I cos (φ ₁ ), g _I determined and calculated from the signals I ′ (t), Q ′ (t) output by the I and Q paths 5, 6 _{sin (φ 1), g Q} cos (φ 3), g Q sin (φ 3), using the _{g I g Q sin (φ 3} -φ 1), at other times, for example, a carrier signal cos ( I and W when the baseband signals A (t), B (t) used to modulate w _c t + φ ₁ ) and sin (w _c t + φ ₁ ) are unknown (eg, in the payload portion of the received signal) Output by Q path 5 and 6 Signal I that can compensate for the phase and amplitude imbalance in the '(t), Q' (t).

Therefore, the phase and amplitude imbalance compensation stage 21 shown in more detail in FIG. 4 includes imbalance compensation parameters g _I cos (φ ₁ ), g _I sin (φ ₁ ), g _Q cos (φ ₃ ), g _Q sin. (phi ₃₎ and _g I _g Q using sin (φ ₃ -φ _1), phase and amplitude in equation (6) and (7) using the signal I '(t), Q' (t) Compensate for imbalances. More specifically, the phase and amplitude imbalance compensation stage 21 includes a pair of I signal multipliers 24 and 25 to which an I signal I ′ (t) is input and a Q signal Q ′ (t ) Are input to a pair of Q signal multipliers 26 and 27. One multiplier 24 of the pair of I signal multipliers 24 and 25 has a first imbalance compensation parameter g _Q cos (determined from the I signal I ′ (t) and the Q signal Q ′ (t). Multiply by φ ₃ ). The other multiplier 25 of the pair of I signal multipliers 24 and 25 includes a second imbalance compensation parameter g _Q sin () determined from the I signal I ′ (t) and the Q signal Q ′ (t). Multiply by φ ₃ ). Similarly, one of the pair of Q signal multipliers 26 and 27 has a first imbalance compensation parameter g determined from the Q signal Q ′ (t) and the I signal I ′ (t). Multiply by _I cos (φ ₁ ). Similarly to the above, the other multiplier 27 of the pair of Q signal multipliers 26 and 27 has a second imbalance determined from the Q signal Q ′ (t) and the I signal I ′ (t). The compensation parameter g _I sin (φ ₁ ) is multiplied.

The phase and amplitude imbalance compensation stage 21 has a pair of adders 28 and 29 and a further pair of multipliers 30 and 31. The first adder 28 of the pair of adders 28 and 29 receives the signals output by the first I signal multiplier 24 and the first Q signal multiplier 26. The second adder of the pair of adders 28 and 29 receives the signals output by the second I signal multiplier 25 and the second Q signal multiplier 27. The adders 28 and 29 add the signals received by the adders 28 and 29 to each other, and output the resultant added signals to the corresponding multipliers 30 and 31 of the pair of multipliers 30 and 31. . Multipliers 30 and 31, by multiplying the signals they receive, and 2 times the reciprocal of the calculated further imbalance compensation parameter _{g I g Q sin (φ 3} -φ 1), are compensated Q signal and I signal are output respectively. These compensated Q and I signals are ideally the respective basebands used to modulate the radio frequency (RF) carrier cos (w _c t + φ ₁ ) and sin (w _c t + φ ₁ ). Should be equal to the signals B (t) and A (t).

Referring to FIG. 5, the heterodyne receiver 32 according to the second preferred embodiment of the present invention is the direct down-conversion receiver 19 of the first preferred embodiment of the present invention described above with reference to FIGS. Have many components that are similar to each other, and therefore the same reference numerals are used for similar components. However, the receiver 32 according to the second preferred embodiment of the present invention has a conversion stage 33 between the LNA 4 and the I and Q paths 5, 6, which is amplified and filtered. Mixer 34 for mixing the received signal and the local oscillation signal of frequency f ₀ , an intermediate frequency filter 35 for filtering the signal after mixing, and a signal for output to the I and Q mixers 7, 12 AGC 36 for adjusting the gain. In this embodiment, the local oscillation signal used by the I and Q mixers 7 and 12 is an intermediate frequency (IF). The combination of mixing with the local oscillation signal at frequency f _{0 and} the local oscillation signal at intermediate frequency (IF) converts the received signal to baseband. Therefore, importantly, the desired signal is in the baseband in the I and Q paths 5,6.

  In this specification and in the claims, the term “a” or “an” preceding an element does not exclude the presence of a plurality of such elements. Also, the term “comprising” does not exclude the presence of elements or steps other than those listed. The inclusion of reference signs in parentheses in the claims is intended to aid understanding and is not intended to be limiting.

It is the schematic of the direct down conversion receiver which concerns on a prior art. 1 is a schematic diagram of a direct down-conversion receiver according to a first preferred embodiment of the present invention. FIG. 3 is a schematic diagram of an imbalance compensation parameter determination stage of the direct down-conversion receiver shown in FIG. 2. FIG. 3 is a schematic diagram of a phase and amplitude compensation stage of the direct down-conversion receiver shown in FIG. 2. It is the schematic of the heterodyne receiver which concerns on the 2nd preferable embodiment of this invention.

Claims (27)

A receiver for receiving a signal including a training sequence and a payload,
An in-phase path and a quadrature path for converting the received signal into an in-phase signal and a quadrature signal for output to the demodulator;
Processing means for determining an imbalance compensation parameter from in-phase and quadrature signals representing a training sequence;
An imbalance compensation circuit for compensating for a phase and amplitude imbalance between the in-phase and quadrature signals representing the payload using the determined imbalance compensation parameters;
A receiver comprising: The signal comprises a data packet including a training sequence and a payload;
The imbalance compensation circuit compensates in-phase and quadrature signals representing the payload of the data packet using imbalance compensation parameters determined from the in-phase and quadrature signals representing the training sequence of the data packet;
The receiver according to claim 1.   The receiver according to claim 1 or 2, wherein the training sequence is also used for further signal processing.   A selector for selecting a part of the in-phase signal and the quadrature signal at two different times, the processing means comprising an imbalance compensation parameter from the part of the in-phase signal and the quadrature signal selected at two different times The receiver according to claim 1, wherein the receiver is determined.   5. A receiver as claimed in any preceding claim, wherein the processing means determines an imbalance compensation parameter based on a signal representative of a training sequence expected to be received.   The selector selects a part of the in-phase signal and the quadrature signal, from which the processing means determines an imbalance compensation parameter when the signals expected to be received are different; The receiver according to claim 4.   The said processing means determines an imbalance compensation parameter based on an in-phase signal, and determines an imbalance compensation parameter based on a quadrature signal. Receiver.   The processing means determines a pair of imbalance compensation parameters based on the in-phase signal and determines a pair of imbalance compensation parameters based on the quadrature signal. The receiver according to item.   9. The receiver according to claim 7, wherein the processing means calculates a further imbalance compensation parameter from the determined imbalance compensation parameter.   The processing means calculates an average of each imbalance compensation parameter based on the in-phase signal and the quadrature signal, and calculates a further imbalance compensation parameter based on the calculated average imbalance compensation parameter. 10. A receiver according to claim 9, characterized in that   The imbalance compensation circuit compensates for a phase and amplitude imbalance between the in-phase signal and the quadrature signal by multiplying the in-phase signal and the quadrature signal by the determined imbalance compensation parameter. The receiver according to any one of claims 1 to 10.   The imbalance compensation circuit multiplies the in-phase signal and the imbalance compensation parameter determined from the quadrature signal, and multiplies the imbalance compensation parameter determined from the quadrature signal and the in-phase signal. The receiver according to claim 7, wherein the receiver compensates for a phase and amplitude imbalance between the phase signal and the quadrature signal.   The imbalance compensation circuit compensates for phase and amplitude imbalance between the in-phase signal and the quadrature signal by multiplying the in-phase signal and the quadrature signal by the reciprocal of the calculated further imbalance compensation parameter. The receiver according to claim 9 or 10, wherein: A method for receiving a signal including a training sequence and a payload, comprising:
Converting received signals into in-phase and quadrature signals for output to a demodulator;
Determining an imbalance compensation parameter from the in-phase and quadrature signals representing the training sequence;
Compensating the phase and amplitude imbalance between the in-phase and quadrature signals representing the payload using the determined imbalance compensation parameters;
A method comprising the steps of: The signal comprises a data packet including a training sequence and a payload;
The compensation includes compensating in-phase and quadrature signals representing the payload of the data packet using an imbalance compensation parameter determined from the in-phase and quadrature signals representing the training sequence of the data packet.
15. The method of claim 14, wherein:   16. A method according to claim 14 or 15, further comprising using the training sequence for further signal processing.   Selecting a portion of the in-phase signal and the quadrature signal at two different times, wherein the imbalance compensation parameter is determined from the portion of the in-phase signal and the quadrature signal selected at two different times. 17. A method according to any one of claims 14 to 16, characterized in that   18. A method according to any one of claims 14 to 17, wherein the determination of an imbalance compensation parameter is based on a signal representative of a training sequence expected to be received.   The method of claim 17, wherein the in-phase and quadrature signals that determine the imbalance compensation parameter are selected when the signals expected to be received are different.   20. A method according to any one of claims 14 to 19, comprising determining an imbalance compensation parameter based on the in-phase signal and determining an imbalance compensation parameter based on the quadrature signal.   20. The method according to claim 14, further comprising: determining a pair of imbalance compensation parameters based on the in-phase signal and determining a pair of imbalance compensation parameters based on the quadrature signal. The method described.   22. A method according to claim 20 or 21, comprising calculating further imbalance compensation parameters from the determined imbalance compensation parameters.   Calculating an average of each imbalance compensation parameter based on the in-phase signal and the quadrature signal, and calculating a further imbalance compensation parameter based on the calculated average imbalance compensation parameter; The method of claim 22.   24. A method according to any one of claims 14 to 23, wherein the compensation comprises multiplying in-phase and quadrature signals by a determined imbalance compensation parameter.   The compensation includes multiplying the in-phase signal and the imbalance compensation parameter based on the quadrature signal and multiplying the quadrature signal and the imbalance compensation parameter based on the in-phase signal. The method according to any one of 20 to 23.   24. A method according to claim 22 or 23, wherein the compensation comprises multiplying the in-phase signal and the quadrature signal by a reciprocal of the calculated further imbalance compensation parameter.   Computer program code configured to perform the method of any one of claims 13 to 24 when processed by a computer processing means.

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