JP2009303435A - Motor controller - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To easily set the phase of a vibration suppression signal used for suppressing vibrations due to the fluctuation of a load torque to the optimum value. <P>SOLUTION: An axis perpendicular to the direction of the magnetic flux of a magnet is a q-axis and an estimated axis for control corresponding to the q-axis is a δ-axis. From a speed deviation based on a rotational speed command value (ω<SP>*</SP>) and an estimated rotational speed (ω<SB>e</SB>), a δ-axis current command value (iδ<SP>*</SP>) is generated and a vibration suppression signal (iδ<SB>C</SB>) which changes in conformity with a change in load torque is also generated. A phase adjustment section (31) adjusts the phase of the vibration suppression signal based on the estimated rotational speed (ω<SB>e</SB>) and a δ-axis voltage command value (vδ<SP>*</SP>) for specifying an application voltage to the motor. The vibration suppression signal after phase adjustment is superimposed on the δ-axis current command value (iδ<SP>*</SP>), and the δ-axis voltage command value (vδ<SP>*</SP>) is generated based on a signal obtained by this superimposing operation. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a motor control device that controls a motor, and more particularly to a motor control device that performs vector control on a motor that drives a load whose load torque fluctuates periodically. The present invention also relates to a motor drive system and a compressor using the motor control device.

  Motor loads often involve periodic load fluctuations. A compressor used in an air conditioner or the like is a typical example including such a load. In a hermetic compressor used for an air conditioner, it is known that a change in refrigerant gas pressure between intake, compression, and discharge strokes affects load torque. It is also known that the load torque due to the refrigerant gas pressure fluctuates in synchronization with the rotation of the compressor (rotation of the motor), and accordingly, the rotation speed of the compressor fluctuates periodically. Such periodic fluctuations in the rotational speed of the compressor cause vibration in the compressor itself and cause noise.

  Various methods have been proposed to solve the vibration caused by such load torque fluctuations. In a conventional method, the rotational speed of the motor is estimated by calculation based on the voltage information and current information of the motor, and the additional torque current command value (described later) having the same period as the rotational period of the estimated rotational speed is calculated using the estimated rotational speed. By generating a vibration suppression signal, speed fluctuation is suppressed (see, for example, Patent Document 1 below).

A block diagram of a conventional motor drive system corresponding to this conventional method is shown in FIG. In this motor drive system, a speed deviation (ω ^* −ω _e ) between the rotational speed command value ω ^* that specifies the rotational speed of the motor and the rotational speed ω _e estimated by the speed estimator is obtained. The δ-axis current command value iδ ^* that the q-axis current should follow is created by proportional-integral control so that the deviation converges to zero. The q-axis current is a current related to torque, and the δ-axis is a control estimation axis that should follow the q-axis. On the other hand, in the motor drive system of FIG. 15, a vibration suppression signal for suppressing vibration due to load torque fluctuation is generated from the speed deviation (ω ^* −ω _e ), and this vibration suppression signal is used as a δ-axis current command. The supply current to the motor is determined according to the value added to the value iδ ^* . Multiplied by the torque constant K _T to the supply current to the motor corresponds to the generated torque Tm of the motor, multiplied by the 1 / Js on the difference between the generated torque Tm and the load torque Tdis is equivalent to the actual rotational speed ω To do. Here, J is inertia and s is a Laplace operator.

JP 2000-41400 A

  In the motor drive system shown in FIG. 15, in order to suppress vibrations satisfactorily, a vibration suppression signal is generated so that the generated torque is strengthened when the load torque is relatively strong and the generated torque is weakened when the load torque is relatively weak. There is a need to. Therefore, the degree of suppression of vibration due to load torque fluctuation strongly depends on the phase of the vibration suppression signal (corresponding to the above-described additional torque current command value).

  In the conventional method, the phase suitable for vibration suppression is adjusted through experiments. The phase suitable for vibration suppression depends on the motor characteristics and also depends on the motor rotation speed and control constants used for vector control (proportional constant and integral constant in proportional integral control, etc.). Then, it is necessary to obtain a phase suitable for vibration suppression by experiment. As described above, the work of examining the phase suitable for vibration suppression by experiment is extremely complicated.

  Accordingly, an object of the present invention is to provide a motor control device, a motor drive system, and a compressor that contribute to facilitating setting of a control parameter (corresponding to the phase of a vibration suppression signal) suitable for vibration suppression.

  A motor control device according to the present invention is a motor control device that performs vector control on a motor that drives a load whose load torque fluctuates periodically, by estimating the rotation speed of the motor. When the axis orthogonal to the direction of the magnetic flux generated by the permanent magnet provided in the child is the q axis and the estimated control axis corresponding to the q axis is the δ axis, the motor control device estimates the estimated rotational speed. Speed control means for generating a δ-axis current command signal based on the speed signal, and a δ-axis current correction signal that fluctuates in response to fluctuations in the load torque, and the δ-axis using the δ-axis current correction signal Correction means for correcting the current command signal, and the δ-axis component of the applied voltage to the motor should follow so that the δ-axis component of the current supplied to the motor follows the corrected δ-axis current command signal. Generate shaft voltage command signal And a current control means for, said correcting means, on the basis of the δ-axis voltage command signal, and adjusts the phase of the δ-axis current correction signal.

  If the δ-axis current command signal is corrected using the δ-axis current correction signal that changes in response to the change in the load torque, it is possible to suppress vibrations resulting from the load torque change. However, when the rotational speed is estimated, the phase of the δ-axis current correction signal can deviate from the optimum one by the estimation error.

  On the other hand, the δ-axis voltage command signal has a signal component that fluctuates in the same cycle as the fluctuation cycle in the actual rotation speed of the motor, and the phase of the signal component is the phase of the fluctuation component in the actual rotation speed of the motor. And generally agree. Therefore, based on the δ-axis voltage command signal, the above estimation error can be compensated. Considering this, the motor control device is formed as described above. As a result, it is possible to easily set the phase of the δ-axis current correction signal suitable for vibration suppression, and optimization of vibration suppression can be achieved with this setting.

  Specifically, for example, the correction unit adjusts the phase of the δ-axis current correction signal based on the δ-axis voltage command signal and the estimated speed signal.

  More specifically, for example, the correction unit adjusts the phase of the δ-axis current correction signal based on the phase information of the δ-axis voltage command signal and the phase information of the estimated speed signal.

  More specifically, for example, the correction unit receives a control signal that fluctuates in response to fluctuations in the load torque, and generates the δ-axis current correction signal by emphasizing a periodic fluctuation component of the control signal. And a phase of the δ-axis current correction signal is adjusted based on a difference between the phase of the δ-axis voltage command signal and the phase of the estimated speed signal.

  A motor drive system according to the present invention includes a motor, an inverter that drives the motor, and a motor control device according to any one of the above that performs vector control on the motor via the inverter. It is characterized by.

  The compressor which concerns on this invention uses the rotational force of the motor with which the said motor drive system was equipped as a drive source, It is characterized by the above-mentioned.

  ADVANTAGE OF THE INVENTION According to this invention, it becomes possible to provide the motor control apparatus, motor drive system, and compressor which contribute to the easy setting of the parameter for control suitable for vibration suppression (corresponding to the phase of a vibration suppression signal).

  The significance or effect of the present invention will become more apparent from the following description of embodiments. However, the following embodiment is merely one embodiment of the present invention, and the meaning of the term of the present invention or each constituent element is not limited to that described in the following embodiment. .

  Hereinafter, embodiments of the present invention will be specifically described with reference to the drawings. In each of the drawings to be referred to, the same part is denoted by the same reference numeral, and redundant description regarding the same part is omitted in principle. The first to fourth embodiments will be described later. First, matters that are common to each embodiment or items that are referred to in each embodiment will be described.

  FIG. 1 is a schematic block diagram of a motor drive system according to an embodiment of the present invention. The motor drive system of FIG. 1 includes a motor 1, a PWM (Pulse Width Modulation) inverter 2, and a motor control device 3.

  The motor 1 is a three-phase permanent magnet synchronous motor, and includes a rotor (not shown) provided with permanent magnets and a stator (not shown) provided with armature windings for three phases.

A PWM inverter (hereinafter simply referred to as an inverter) 2 supplies a three-phase AC voltage to the motor 1 in accordance with the rotor position of the motor 1. The three-phase AC voltage applied to the motor 1 by the inverter 2 is a U-phase voltage v _u representing the voltage applied to the U-phase armature winding, and a V-phase voltage representing the voltage applied to the V-phase armature winding. v _v and a W-phase voltage v _w representing a voltage applied to the W-phase armature winding. Is a combined voltage of the U-phase voltage v _u, V-phase voltage v _v and the W-phase voltage v _w, to the motor 1, the entire applied voltage is called a motor voltage (motor terminal voltage), representing it by the symbol V _a .

By application of the motor voltage V _a, U-phase component of the current supplied from the inverter 2 to the motor 1, V-phase component and a W-phase component, i.e. U-phase, the current flowing in the armature winding of the V-phase and W-phase, These are called U-phase current i _u , V-phase current i _v and W-phase current i _w , respectively. The total supply current to the motor 1, which is a combined current of the U-phase current i _u , the V-phase current i _v and the W-phase current i _w , is called a motor current (armature current), and is represented by the symbol I _a . .

The motor control device 3 provides the inverter circuit 2 with a PWM signal for realizing desired vector control, based on the detected value of the motor current _{Ia and} the like.

  FIGS. 2A and 2B are analysis model diagrams of the motor 1. In the following description, the armature winding refers to that provided in the motor 1. FIG. 2A shows U-phase, V-phase, and W-phase armature winding fixed axes. 1 a is a permanent magnet provided on the rotor of the motor 1. In a rotating coordinate system that rotates at the same speed as the rotational speed of the magnetic flux produced by the permanent magnet 1a, the axis along the direction of the magnetic flux produced by the permanent magnet 1a is defined as the d axis, and the control rotational axis corresponding to the d axis is the γ axis. And The direction of the d axis matches the direction of the magnetic flux produced by the permanent magnet 1a. Further, as shown in FIG. 2B, an axis whose phase is advanced by 90 degrees in electrical angle from the d axis is a q axis, and an axis whose phase is advanced by 90 degrees in electrical angle from the γ axis is a δ axis. 2A and 2B, the counterclockwise direction corresponds to the phase advance direction. The d axis and the q axis are collectively referred to as a dq axis, and a coordinate system having the d axis and the q axis as coordinate axes is referred to as a dq coordinate system. The γ-axis and the δ-axis are collectively called a γδ axis, and a coordinate system having the γ-axis and the δ-axis as coordinate axes is called a γδ coordinate system.

The dq axis and the dq coordinate system are rotating, and the rotation speed is represented by ω. The γδ axis and the γδ coordinate system are also rotating, and the rotation speed is represented by ω _e . In addition, on the dq axis, the angle (phase) of the d axis viewed from the U-phase armature winding fixed axis is represented by θ. Similarly, in the γδ axis, the angle (phase) of the γ axis viewed from the U-phase armature winding fixed axis is represented by θ _e . The angles represented by θ and θ _e are angles in electrical angles, and they are generally also called rotor positions or magnetic pole positions. The rotational speed represented by ω and ω _e is an angular velocity in electrical angle. An axial error Δθ between the d-axis and the γ-axis is expressed by Δθ = θ−θ _e .

Hereinafter, θ or θ _e is referred to as a rotor position, and ω or ω _e is referred to as a rotation speed. When the rotor position and the rotational speed are derived by estimation, the γ axis and δ axis can be called control estimation axes. In addition, θ _e and ω _e are also referred to as an estimated rotor position and an estimated rotational speed, respectively, and ω is also referred to as an actual rotational speed. ω is an actual rotational speed of the rotor of the motor 1, and ω _e is an estimated rotational speed of the rotor of the motor 1.

The motor control device 3 performs vector control so that θ and θ _e match. When θ and θ _e coincide with each other, the d axis and the q axis coincide with the γ axis and the δ axis, respectively.

Symbols related to the control of the motor drive system are defined as follows.
D-axis component of the motor voltage V _a, q-axis component, gamma-axis component, a δ-axis component, respectively, d-axis voltage, q-axis voltage, the symbols together referred to as gamma-axis voltage and the δ-axis voltage v _d, v _q, v? And vδ.
The d-axis component, q-axis component, γ-axis component, and δ-axis component of the motor current I _a are referred to as d-axis current, q-axis current, γ-axis current, and δ-axis current, respectively, and symbols i _d , i _q , iγ. And iδ.
Φ _a represents an armature interlinkage magnetic flux by the permanent magnet 1a.
L _d and L _q represent the d-axis inductance (d-axis component of the inductance of the armature winding) and the q-axis inductance (q-axis component of the inductance of the armature winding), respectively.
R _a represents a resistance value per phase of the armature winding.
Φ _a , L _d , L _q and R _a are determined in advance according to the characteristics of the motor 1 and are used as parameters for controlling the motor drive system.

The target values of the γ-axis voltage vγ and the δ-axis voltage v _δ that should be followed by the γ-axis voltage v _γ and the δ-axis voltage v _δ are represented by a γ-axis voltage command value v _γ ^* and a δ-axis voltage command value vδ ^* , respectively. .
The target values of the γ-axis current iγ and the δ-axis current i _δ that should be followed by the γ-axis current i _γ and the δ-axis current i _δ are represented by a γ-axis current command value i _γ ^* and a δ-axis current command value i δ ^* , respectively. .
Estimated rotation speed omega _e to be followed, representing the rotational speed command value a target value of the estimated rotation speed ω _e ω ^*.

Note that v _γ can also be used as a symbol representing the value of the γ-axis voltage. The same applies to symbols representing state quantities other than vγ (including state quantities relating to voltage and current). In addition, in this specification, for simplicity of description, a state quantity or the like corresponding to the symbol may be expressed by using only a symbol (such as iγ). That is, in this specification, for example, “iγ” and “γ-axis current iγ” or “γ-axis current value iγ” indicate the same thing.

In the motor control device 3, the γ-axis voltage value vγ and the δ-axis voltage value vδ follow the γ-axis voltage command value vγ ^* and the δ-axis voltage command value vδ ^* , respectively, and the γ-axis current value iγ and δ-axis. Vector control is performed so that the current value iδ follows the γ-axis current command value iγ ^* and the δ-axis current command value iδ ^* , respectively.

  The motor 1 rotationally drives a load whose load torque fluctuates periodically. This load is, for example, a load provided in a compressor (see FIG. 14), a washing machine or a dryer (not shown). The compressor is used for an air conditioner or the like. In the compressor, a change in the refrigerant gas pressure during a process including suction, compression, and discharge, which is executed periodically, acts on the load torque, and thereby the load torque varies periodically.

  Specifically, the washing machine or the dryer is, for example, a drum-type washing machine or a drum-type dryer. In a drum-type washing machine, a drum for storing laundry rotates about an axis that is not parallel to the vertical line, lifts the laundry, and performs washing by dropping. When this drum is used as the load of the motor 1, the load torque becomes relatively large when the laundry is lifted, and the load torque becomes relatively small when the laundry is not lifted. The same applies to the drum dryer.

  The principle of vector control according to the present embodiment, which is useful when driving such a load, will be described. FIG. 3 is a block diagram showing only the torque-related part of the motor drive system of FIG. It can be considered that the motor drive system according to the present embodiment includes each part referred to by reference numerals 101 to 110. In the motor drive system corresponding to FIG. 3, sensorless control (sensorless vector control) is performed without using a rotor sensor (not shown) that measures the rotor position.

  First, the function of each part referred to by reference numerals 101 to 108 will be described. Assuming that a load whose load torque fluctuates periodically is rotated, the load torque is represented by Tdis.

The subtractor 101 obtains a speed deviation (ω ^* −ω _e ) by subtracting the estimated rotational speed ω _e calculated by the speed estimation unit 108 from the rotational speed command value ω ^* given from the outside. The PI control unit 102 uses proportional integral control to calculate the δ-axis current command value iδ ^* so that the speed deviation (ω ^* −ω _e ) from the subtracter 101 converges to zero. The adder 103, [delta] by adding the output value i? _C 'from the phase adjusting section 110 to the axis current value i? ^*, Corrects the [delta] -axis current value i? ^* From the PI controller 102.

The corrected δ-axis current command value, that is, (iδ ^* + iδ _C ′) is converted into a δ-axis current iδ by the current control unit 104. Multiplier 105 multiplies δ-axis current iδ obtained by this conversion by torque constant K _{T. ^{Thus, (iδ * + iδ C '}} ) to the based δ-axis current i? Is converted into the motor torque Tm. The current control unit 104 generates a δ-axis voltage value v? ^* Based on the corrected δ-axis current value _{^{(iδ * + iδ C ')}} . The subtractor 106 subtracts the load torque Tdis having a periodic variation from the motor torque Tm, and the multiplier 107 multiplies the subtraction result (Tm−Tdis) by 1 / Js. In this “1 / Js”, J is an inertia and s is a Laplace operator. (Tm−Tdis) / Js obtained through the subtractor 106 and the multiplier 107 represents the actual rotational speed ω of the motor. The speed estimation unit 108 obtains a rotational speed ω _e that represents an estimated value of the rotational speed ω.

The functions of the vibration suppression signal generation unit 109 and the phase adjustment unit 110 which are characteristic parts will be described with reference to FIGS. Now, it is assumed that the rotation speed command value ω ^* is maintained at a constant value. Further, it is assumed that vector control is performed so that i _d = 0. In the graphs of FIGS. 4 and 5, the horizontal axis corresponds to time, and the vertical axis corresponds to the signal value. 4 and 5, a straight line 301 is a signal waveform having the rotation speed command value ω ^* as a signal value, and a curve 302 is a signal waveform representing the estimated rotation speed ω _e . A curve 303 is a waveform of the vibration suppression signal generated by the vibration suppression signal generation unit 109 of FIG. The signal value of the vibration suppression signal generated by the vibration suppression signal generator 109 is expressed by i? _C.

When the load torque varies periodically, the rotational speed of the motor also varies periodically around ω ^* due to the delay of the control system. Variations in the rotational speed of the motor cause vibrations of equipment (such as a compressor) incorporating the motor drive system, and the vibrations cause noise. The vibration suppression signal generation unit 109 in FIG. 3 generates a vibration suppression signal for suppressing this vibration based on the speed deviation (ω ^* −ω _e ). Specifically, (ω ^* -ω _e) vary with the same period as the fluctuation period of and (ω ^* -ω _e) an electrical angle of 90 degrees only vibration suppression advanced phase than the phase of the signal represented Generate a signal. This vibration suppression signal is generated as a correction signal to the δ-axis current command value iδ ^{* related} to the torque.

  When the rotation speed of the motor is accelerating and the magnitude of the speed change is maximum (at a time corresponding to the point 304 in FIG. 4), vibration suppression is performed so that the generated torque of the motor 1 is suppressed to the greatest extent. When the signal is generated and the rotational speed of the motor is being reduced and the magnitude of the speed change is maximized (at the time corresponding to the point 305 in FIG. 4), the torque generated by the motor 1 increases the most. Thus, a vibration suppression signal is generated. As a result, ideally, the rotational speed of the motor is stabilized and vibrations of the motor and the entire system are minimized.

However, when sensorless control is performed, the estimated rotational speed ω _e is calculated based on the motor current information and the like, so that the phase of the estimated rotational speed ω _e is shifted from the phase of the actual rotational speed ω as shown in FIG. there is a possibility. In FIG. 5, a curve 312 is a signal waveform representing the actual rotational speed ω. If such a shift exists, the phase of the vibration suppression signal generated based on the estimated rotation speed ω _e is also shifted from the optimum phase for vibration suppression.

The difference between the phase of the signal representing the estimated rotational speed ω _e (hereinafter also referred to as the estimated speed signal) and the phase of the signal representing the actual rotational speed ω (hereinafter also referred to as the actual speed signal) is represented by This is represented by a phase difference Δε. The estimated speed signal and the actual speed signal are signals that fluctuate in the same period as the fluctuation period of the load torque. If the phase difference Δε is zero and the amplitudes of the estimated speed signal and the actual speed signal are equal, The estimated speed signal matches the actual speed signal.

  The phase difference Δε is not constant because it depends on the rotational speed of the motor 1 and control constants used for vector control. The control constant is, for example, a proportional constant and an integral constant used for proportional-integral control of the PI control unit 102. As can be seen from the simulation results of FIG. 6, the phase difference Δε tends to increase as the rotational speed of the motor 1 increases. A curve 320 in FIG. 6 shows the dependency of the phase difference Δε on the rotational speed of the motor 1. In FIG. 6, the horizontal axis represents the rotational speed of the motor 1, and the vertical axis represents the phase difference Δε. The unit of the numerical value of the rotational speed indicated on the horizontal axis of FIG. 6 is rps (revolutions per second), and the unit of the numerical value of the phase difference Δε indicated on the vertical axis of FIG. degree).

In sensorless control, since the rotational speed of the motor is not detected by actual measurement, the phase difference Δε cannot be detected based on the actual measurement result. On the other hand, as can be seen from the simulation results of FIG. 7, the q-axis voltage v _q varies in the same cycle as the actual speed signal and the estimated speed signal, and the phase of the signal representing the q-axis voltage v _q is the actual rotation. The phase of the signal representing the speed ω is substantially the same. In FIG. 7, a straight line 351 is a waveform of a signal having the rotational speed command value ω ^* as a signal value, and curves 352 to 354 are a waveform of a signal representing the estimated rotational speed ω _e and an actual rotational speed ω, respectively. signal waveform representing, and a waveform of a signal representing the q-axis voltage v _q.

In consideration of such characteristics of the q-axis voltage, in the present embodiment, information on the δ-axis voltage corresponding to the q-axis voltage is referred to as information for estimating the fluctuation of the rotation speed, and the phase difference is determined using the information. Perform phase adjustment adapted to Δε. The phase adjustment function of FIG. 3 is responsible for this phase adjustment function. The phase adjustment unit 110 adjusts the phase of the vibration suppression signal generated by the signal suppression signal generation unit 109 based on the δ-axis voltage command value vδ ^* from the current control unit 104, and the vibration suppression signal after the phase adjustment Is output to the adder 103. The signal value of the vibration suppression signal after the phase adjustment is represented by iδ _C ′. More specifically, assuming that q-axis voltage v _q coincides with the δ-axis voltage value v? ^*, The q-axis voltage v _q is in consideration of the characteristics that varies in synchronism with the actual rotation speed omega, to estimate the phase difference Δε based on vδ ^* and ω _e. Then, the phase of the vibration suppression signal (iδ _C ) is changed based on the phase difference Δε so that the phase of the vibration suppression signal represented by iδ _C ′ advances by 90 degrees in electrical angle with reference to the phase of the actual speed signal. Thus, the vibration suppression signal (iδ _C ′) after the phase adjustment is obtained. The adder 103 adds the output value iδ _C ′ of the phase adjustment unit 110 to the δ-axis current command value iδ ^* , so that the δ-axis current command value to be given to the current control unit 104 is optimal for vibration suppression. It is corrected to a thing.

The reason why the actual rotational speed ω and the q-axis voltage v _q have substantially the same phase characteristics can be understood from the vector diagram relating to the motor voltage in FIG. FIG. 8 is a vector diagram when it is assumed that i _d = 0. The q-axis voltage v _q is represented by “v _q = R _a · i _q + ω · Φ _a ” and is a function of ω, and therefore v _q also fluctuates in synchronization with fluctuation of ω.

  In the conventional method corresponding to FIG. 15, it is necessary to obtain the optimum phase of the vibration suppression signal by experiment. However, as in this embodiment, the phase of the vibration suppression signal is adjusted using information on the q-axis voltage (δ-axis voltage). By providing the motor control device with the function to perform this, it is possible to easily find the optimal phase of the vibration suppression signal (the optimal phase for vibration suppression).

  The degree of vibration suppression depends not only on the phase of the vibration suppression signal but also on the amplitude of the vibration suppression signal. In the conventional method corresponding to FIG. 15, it is necessary to treat the phase and amplitude of the vibration suppression signal as parameters to be adjusted and to experimentally search for the optimum values of the two parameters that depend on each other. Searching for optimum values of a plurality of parameters that depend on each other is complicated and sometimes fails to search for optimum values. Further, if the phase of the vibration suppression signal is not optimal, there is a high possibility that a vibration suppression signal having a larger amplitude is required as compared with the case where it is optimal. An increase in amplitude in the vibration suppression signal causes an increase in power consumption.

  On the other hand, according to the present embodiment, since the optimum phase of the vibration suppression signal is automatically found, it is possible to search for an amplitude suitable for vibration suppression while the phase of the vibration suppression signal is fixed to the optimal phase. . For this reason, it is possible to easily and reliably find the optimum value of the amplitude of the vibration suppression signal. As a result, it becomes possible to drive the motor under conditions optimal for vibration suppression and power consumption suppression.

In the above description, it is assumed that the d-axis current i _d is zero. However, even if i _d ≠ 0, the magnitude of i _d is usually sufficiently larger than the magnitude of i _q. Since it is small, the phase adjustment by the phase adjustment unit 110 functions effectively.

  Below, the 1st-4th Example is demonstrated as an Example explaining the structure of the motor drive system which concerns on this embodiment, or the matter which concerns it.

<< First Example >>
First, a first embodiment according to the present invention will be described. FIG. 9 is a detailed block diagram of the motor drive system according to the first embodiment. The motor drive system of FIG. 9 includes the motor 1 and the inverter 2 shown in FIG. 1, a motor control device 3 a that functions as the motor control device 3 of FIG. 1, and a phase current sensor 11. The motor control device 3a includes each part referred to by reference numerals 12 to 20 and 30 to 32. It can also be considered that the phase current sensor 11 is included in the motor control device 3a. Each part in the motor control device 3a can freely use each value generated in the motor control device 3a.

Different parts constituting the motor drive system of this example and other examples described later, themselves calculated at a predetermined update period (or detection) to the command value to be output (i? ^*, I? ^*, V? ^*, V? ^* , V _u ^* , v _v ^* and v _w ^* ), state quantities (including i _u , i _v , iγ, iδ, θ _e and ω _e ) or δ-axis current correction values iδ _C and iδ _C ′ , And perform necessary calculations using the latest values.

The estimated rotational speed ω _e changes with time, and the waveform of the signal (estimated speed signal) obtained by sequentially plotting the value of ω _e with time as the horizontal axis is, for example, the curve 302 in FIG. 4 or the curve 352 in FIG. It becomes a waveform like this. Similarly, the waveform of the signal (δ-axis current correction signal) obtained by sequentially plotting the δ-axis current correction value iδ _C or iδ _C ′ with time as the horizontal axis becomes a waveform such as a curve 303 in FIG. The waveform of the signal (δ-axis voltage command signal) obtained by sequentially plotting the δ-axis voltage correction value vδ ^* with the time as the horizontal axis is, for example, a waveform as shown by a curve 354 in FIG. Similar to ω _e , iδ _C , iδ _C ′ and vδ ^* , other command values and state quantities also form signal waveforms that change with time.

Phase current sensor 11 detects the U-phase current value i _u and the V-phase electric-current value i _v is a fixed axis component of the motor current I _a supplied from the inverter 2 to the motor 1. The W-phase current value i _w is calculated from the relational expression “i _w = −i _u −i _v ”. Coordinate converter 12 performs coordinate transformation of the U-phase current value i _u and the V-phase electric-current value i _v into current values on the γδ-axis based on the estimated rotor position theta _e, gamma-axis current value iγ and δ-axis The current value iδ is calculated. In the first embodiment, the estimated rotor position θ _e and the estimated rotation speed ω _e are calculated by the position / speed estimator 20.

The subtractor 19 refers to the estimated rotational speed ω _e and the rotational speed command value ω ^* from a rotational speed command value generation unit (not shown) provided outside the motor control device 3a, and the speed deviation between the two. (Ω ^* −ω _e ) is calculated.

The speed controller 17 calculates and outputs the δ-axis current command value iδ ^* so that the speed deviation (ω ^* −ω _e ) converges to zero by using proportional integral control or the like. The adder 32 adds the δ-axis current correction value iδ _C ′ from the phase adjustment unit 31 to iδ ^* from the speed control unit 17, and outputs the addition value (iδ ^* + iδ _C ′) to the subtractor 13. Originally, the output value i δ ^* of the speed control unit 17 is a target value of i δ, but it is corrected by the resonance filter 30, the phase adjustment unit 31, and the adder 32, and actually the corrected value (i δ ^* + iδ _C ') becomes the target value of iδ. The operations of the resonance filter 30 and the phase adjustment unit 31 will be described later.

The magnetic flux controller 16 determines the γ-axis current command value iγ ^* and outputs it to the subtractor 14. iγ ^* can take various values depending on the type of vector control executed in the motor drive system and the rotational speed of the motor. In this embodiment, in order to estimate the dq axis, iγ ^* = 0 is set in the case of performing control to make the d-axis current zero. When performing maximum torque control and flux-weakening control, iγ ^* is a negative value corresponding to the estimated rotational speed ω _e . The operation of the phase adjustment unit 31 does not depend on the value of iγ ^* , but when iγ ^* = 0 or iγ ^* << iδ ^* , the phase adjustment is performed particularly accurately.

The subtractor 14 subtracts the γ-axis current value iγ output from the coordinate converter 12 from the γ-axis current command value iγ ^* output from the magnetic flux controller 16 to calculate a current error (iγ ^* −iγ). The subtracter 13 'subtracts the from output from the coordinate converter 12 [delta] -axis current value i?, The current error (i? ^* + I? _C is the value (iδ ^* + iδ _C)' output from the adder 32 - i?) calculate.

Current controller 15, so that the current error (i? ^* - i?) And _{^{(iδ * + iδ C '-iδ}} ) converges to zero both perform current feedback control using a proportional integral control. At this time, by using the decoupling control for eliminating interference between the γ axis and the δ-axis, (i? ^* - i?) And _{^{(iδ * + iδ C '-iδ}} ) converge to zero both γ An axial voltage command value vγ ^* and a δ-axis voltage command value vδ ^* are calculated. In calculating vγ ^* and vδ ^* , ω _e , iγ and iδ can also be referred to.

The coordinate converter 18 performs coordinate conversion of vγ ^* and vδ ^* given from the current control unit 15 on a three-phase fixed coordinate axis based on the estimated rotor position θ _e output from the position / speed estimator 20. Thus, the three-phase voltage command value is calculated and output. Three-phase voltage command value, U-phase, V-phase and W-phase voltage value _{v u,} v v and v U-phase to specify the _w, V-phase and W-phase voltage command value v _u ^*, v _v ^* and v _w ^* Formed from.

In the inverter 2, the actual U-phase, V-phase and W-phase voltage values v _u , v _v and v _w are the same as the U-phase, V-phase and W-phase voltage command values v _u ^* , v _v ^* and v _w ^* , respectively. Thus, the motor 1 is driven by supplying the motor current I _a according to the three-phase voltage command value to the motor 1.

Position and speed estimator 20 can perform such a proportional integral control by using all or part of v? ^* And v? ^* Of from iγ and iδ and the current control unit 15 from the coordinate converter 12, the estimated rotation The child position θ _e and the estimated rotational speed ω _e are obtained. θ _e and ω _e are obtained so that the axial error Δθ (= θ−θ _e ) between the d axis and the γ axis converges to zero. As a method for estimating the rotor position and the rotational speed (that is, a method for deriving θ _e and ω _e ), any estimation method including a known method can be used.

FIG. 10 shows an example of an internal block diagram of the position / speed estimator 20. The position / velocity estimator 20 in FIG. 10 includes each part referred to by reference numerals 21 to 23. The axis error estimator 21 calculates an axis error Δθ based on iγ, iδ, vγ ^*, and vδ ^* . For example, the axis error Δθ is calculated using the following equation (1) also disclosed in Japanese Patent No. 311878. The proportional-integral calculator 22 performs proportional-integral control to calculate the estimated rotational speed ω _e so that the axis error Δθ converges to zero. The integrator 23 calculates the estimated rotor position θ _e by integrating the estimated rotation speed ω _e . The calculated θ _e and ω _e are given to each part in the motor control device 3a that requires the values.

As described above, in the present embodiment, it is assumed that the motor 1 rotationally drives a load whose load torque varies periodically. In this case, iδ ^* may deviate from the ideal value due to fluctuations in the load torque, but the resonance filter 30, the phase adjustment unit 31, and the adder 32 work in a direction to suppress this deviation.

Although related to the vibration suppression principle described above with reference to FIG. 3 and the like, the shift suppression principle will be described with reference to FIG. In FIG. 11, the horizontal axis represents the phase of current in electrical angle, and the vertical axis represents the current value. During the steady rotation of the motor 1, the current phase changes with time, so the horizontal axis in FIG. 11 also corresponds to time. In FIG. 11, a curve 401 represents a current converted value (current component) of the load torque, a curve 402 represents a δ-axis current command value iδ ^* calculated by the speed control unit 17, and a curve 403 is represented by the curve 401. An error waveform between the current converted value of the load torque and the δ-axis current command value iδ ^* represented by the curve 402 is represented. A curve 404 represents a waveform obtained by inverting the polarity of the error waveform represented by the curve 403 and also represents a waveform of the δ-axis current correction value iδ _C ′ to be output from the phase adjustment unit 31.

In order to cause the motor 1 to generate a torque that matches the load torque, a δ-axis current iδ corresponding to the current converted value of the load torque may be passed. When it is considered that there is no δ-axis current correction value, if the current converted value of the load torque and iδ ^* are completely matched, the speed fluctuation due to the load torque fluctuation is reduced and the vibration is reduced.

However, in reality, since a delay of the control system occurs, the δ-axis current command value iδ ^* is delayed from a value that should be truly calculated (current converted value of load torque). In order to eliminate this delay, it is conceivable to superimpose the δ-axis current correction value iδ _C obtained from the fluctuation component of the speed deviation (ω ^* −ω _e ) on iδ ^* . However, since there is a phase difference Δε between the actual speed signal (ω) and the estimated speed signal (ω _e ) (see FIG. 5), it is also necessary to compensate for this. Therefore, in this embodiment, after obtaining the δ-axis current correction value iδ _C from the fluctuation component of the speed deviation (ω ^* −ω _e ), the phase of the signal represented by the δ-axis current correction value iδ _C is changed to the phase difference Δε. By adjusting the δ-axis current correction value i δ _C ′ after the phase adjustment to i δ ^* , (i δ ^* + i δ _C ′) is matched with the current converted value of the load torque.

The operations of the resonance filter 30 and the phase adjustment unit 31 responsible for this function will be described. The resonant filter 30 receives the subtraction result (ω ^* −ω _e ) of the subtractor 19 as an input signal, and extracts a periodic fluctuation component derived from the load torque fluctuation from the input signal. Then, it outputs the extracted variation component as i? _C. The transfer function H _A (s) of the resonance filter 30 is expressed by the following equation (2) (or equation (3) described later).

Here, b _o is a gain coefficient, b ₁ is a phase adjustment amount, ζ is an attenuation coefficient, and ω _r is a natural angular frequency. S is a Laplace operator. Ideally, the resonance filter 30 extracts and outputs only the frequency component of ω _r of the input signal. The resonance filter 30 amplifies (emphasizes) the component of the natural angular frequency ω _{r in} the input signal to a degree corresponding to the gain coefficient b _o, and outputs frequency components other than ω _r as much as possible to the output signal. Do not include. Further, by passing through the resonant filter 30, the phase of the input signal to the resonant filter 30 advances by 90 degrees in electrical angle.

The natural angular frequency ω _r is set to be equal (or as equal as possible) to the angular frequency of the periodic load torque fluctuation of the load of the motor 1. The value of the natural angular frequency ω _r changes according to ω ^* or ω _e . This is because the load torque fluctuation cycle changes depending on the rotation speed of the motor 1. How to set ω _r according to ω ^* or ω _e is defined in advance at the time of designing the motor drive system. For example, table data for determining ω _r from ω ^* or ω _e is given to the motor control device 3a in advance.

  The attenuation coefficient ζ is a value that determines the degree of resonance (resonance characteristics) of the resonance filter 30 and can be set to an arbitrary value of 0 ≦ ζ <1. For example, ζ = 0.01 or ζ = 0.1. The attenuation coefficient ζ can be determined in advance at the time of designing the motor drive system.

The phase adjustment amount b ₁ is a value for adjusting the phase of the δ-axis current correction value iδ _C. The value of the phase adjustment amount b ₁ changes according to ω ^* or ω _e . How b ₁ is set according to ω ^* or ω _e is defined in advance at the time of designing the motor drive system. For example, table data for determining b ₁ from ω ^* or ω _e is given in advance to the motor control device 3a. Incidentally, when there is no need to adjust the phase of i? _C by the resonance filter 30, b ₁ = 0, it is.

Gain of the resonance filter 30, i.e., the degree of emphasis with respect to the signal component of natural angular frequency omega _r of the input signal of the resonance filter 30 is determined by the gain coefficient b _o. When b ₁ = 0, the gain of the resonance filter 30 can be changed by changing _bo in the above equation (2). For b ₁ ≠ 0, the above equation (2) is deformed as the following equation (3), it is possible to change the gain of the resonance filter 30 by changing the b _o in formula (3) .

The pre-phase adjustment δ-axis current correction value iδ _C output from the resonance filter 30 is given to the phase adjustment unit 31. Based on the estimated rotational speed ω _e from the position / speed estimator 20 and the δ-axis voltage command value vδ ^* from the current control unit 15, the phase adjustment unit 31 suppresses vibrations represented by the δ-axis current correction value iδ _C. The phase of the signal (δ-axis current correction signal) is adjusted, and the vibration suppression signal after the phase adjustment is output to the adder 32. The signal value of the vibration suppression signal after the phase adjustment is represented by iδ _C ′.

More specifically, for example, the phase adjustment unit 31 extracts the signal component of the natural angular frequency ω _r from the signal (δ-axis voltage command signal) represented by the δ-axis voltage command value vδ ^* , and the estimated rotational speed ω. The signal component of the natural angular frequency ω _r is extracted from the signal represented by _e (estimated velocity signal), and the phase difference Δε is estimated by comparing the extracted signal components. Then, based on the phase difference Δε, the vibration suppression signal (iδ _C) that is the input signal is set so that the phase of the vibration suppression signal represented by i δ _C ′ advances by 90 degrees in electrical angle with reference to the phase of the actual speed signal. ) And the vibration suppression signal after the phase adjustment is output. The signal value of the vibration suppression signal after the phase adjustment (that is, iδ _C ′) is given from the phase adjustment unit 31 to the adder 32.

<< Second Example >>
Next, a second embodiment according to the present invention will be described. FIG. 12 is a detailed block diagram of the motor drive system according to the second embodiment. The motor drive system of FIG. 12 includes the motor 1 and the inverter 2 shown in FIG. 1, a motor control device 3 b that functions as the motor control device 3 of FIG. 1, and a phase current sensor 11. The motor control device 3b includes each part referred to by reference numerals 12 to 20 and 30 to 32. It can also be considered that the phase current sensor 11 is included in the motor control device 3b. Each part in the motor control device 3b can freely use each value generated in the motor control device 3b.

In the motor control device 3a of FIG. 9, the input signal of the resonance filter 30 is a speed deviation (ω ^* −ω _e ). However, in the motor control device 3b of FIG. It is not the deviation (ω ^* −ω _e ) but the δ-axis current command value iδ ^* from the speed control unit 17. Therefore, the resonance filter 30 in the second embodiment extracts the periodic fluctuation component derived from the load torque fluctuation from the signal represented by the δ-axis current command value iδ ^* (δ-axis current command signal), and extracts it. and it outputs the variation component to the phase adjusting unit 31 as i? _C. Except for this difference regarding the input signal of the resonance filter 30, the motor control devices 3a and 3b are the same.

When the speed deviation (ω ^* −ω _e ) varies due to the load torque variation, the δ-axis current command value iδ ^* also varies in synchronization with the speed deviation (ω ^* −ω _e ). For this reason, even if it comprises like the motor control apparatus 3b, the effect similar to 1st Example is acquired.

<< Third Example >>
Next, a third embodiment according to the present invention will be described. FIG. 13 is a detailed block diagram of the motor drive system according to the third embodiment. The motor drive system of FIG. 13 includes the motor 1 and the inverter 2 shown in FIG. 1, a motor control device 3 c that functions as the motor control device 3 of FIG. 1, and a phase current sensor 11. The motor control device 3c includes each part referred to by reference numerals 12 to 20, 31 and 32, and a resonance filter 30c. It can also be considered that the phase current sensor 11 is included in the motor control device 3c. Each part in the motor control device 3c can freely use each value generated in the motor control device 3c.

In the motor control device 3a of FIG. 9, the resonance filter 30 calculates the δ-axis current correction value iδ _C from the speed deviation (ω ^* −ω _e ), but in the motor control device 3c of FIG. 13, the resonance filter 30c. Calculates the δ-axis current correction value iδ _C from the axis error Δθ. The motor control devices 3a and 3c are the same except for the difference regarding the resonance filter. Hereinafter, description of the same part between the two will be omitted, and the input / output of the resonance filter 30c will be described.

The resonance filter 30c receives the axial error Δθ estimated by the position / speed estimator 20 as an input signal, and extracts a periodic fluctuation component derived from the load torque fluctuation from the axial error Δθ. Then, it outputs the variation component thus extracted to the phase adjustment unit 31 as the δ-axis current correction value i? _C. The transfer function H _B (s) of the resonance filter 30c is expressed by the following equation (4).

Here, b _o is the gain coefficient, b ₁ is the phase adjustment amount, the attenuation coefficient zeta, the omega _r a natural angular frequency, they are the same as those described in the first embodiment. S is a Laplace operator. The resonance filter 30c amplifies in degree according to the frequency components of omega _r of the input signal to the gain coefficient b _o (emphatically) output, ideally, only the frequency components of omega _r of the input signal Extract and output.

Since the shaft error Δθ fluctuates in synchronization with the periodic load torque variation, iδ _C obtained by multiplying the shaft error Δθ by the transfer function H _B (s) has a periodicity of the shaft error Δθ. The fluctuation component will be emphasized. For this reason, even if it comprises like the motor control apparatus 3c, the effect similar to 1st Example is acquired.

Incidentally, in the motor control device 3c in FIG. 13 is the input to the resonance filter 30c has an axis error [Delta] [theta], the input to the resonance filter 30c, the variation of the axis error [Delta] [theta] (the axis at the natural angular frequency omega _r Any value may be used as long as the value fluctuates in synchronization with the fluctuation of the error Δθ. This is because the value that varies in synchronization with the variation of the shaft error Δθ includes a periodic variation component synchronized with the periodic load torque variation, similarly to the shaft error Δθ.

  For example, a signal that is proportional (or substantially proportional) to the axial error Δθ can be used as an input signal to the resonance filter 30c.

  For example, the torque fluctuation component ΔTm estimated using the axis error Δθ may be used as an input signal to the resonance filter 30c. Non-Patent Document 1 “Yasuo Notohara and 4 others,“ Reduction Control Method of Periodic Torque Disturbance for Compressor ”, 2004 Annual Meeting of the Institute of Electrical Engineers of Japan , September 14, 2004, 1-57 (I-337 to I-340) ”, for example, is a value approximately calculated by the following equation (5), and the axial error Δθ In Equation (5), P represents the number of poles of the motor 1, J represents inertia, and the torque fluctuation component ΔTm causes a speed fluctuation, and finally Specifically, it appears as a fluctuation of the axis error Δθ.

<< 4th Example >>
FIG. 14 shows a compressor 500 as a device to which any of the motor drive systems described above in this embodiment is applied. FIG. 14 is an external view of the compressor 500. Any of the motor drive systems described above in the present embodiment is provided in the compressor 500. The compressor 500 compresses refrigerant gas (not shown) using the rotational force of the motor 1 as a drive source.

  For example, the compressor 500 is a scroll compressor. When the compressor 500 is a scroll compressor, a pair of spiral bodies (not shown) having the same shape are provided in the compressor 500, and one spiral pair is fixed. The other spiral body is circularly moved by the rotational force of the motor 1 to change the volume in the compression chamber and compress the gas (refrigerant gas or the like) in the compression chamber. Before and after the gas compression step, gas is sucked and discharged. In this case, the load directly driven by the rotational force of the motor 1 is the above-described spiral body, but the scroll compressor itself can also be regarded as the load of the motor 1.

  Of course, the compressor 500 may be a compressor other than the scroll compressor, or may be a reciprocating compressor, a rotary compressor, or the like. When the compressor 500 is a reciprocating compressor, the compressor 500 includes a cylinder that forms a compression chamber with the piston, and changes the volume in the cylinder by reciprocating the piston in the cylinder by the rotational force of the motor 1. To compress the gas (refrigerant gas, etc.) in the cylinder. In this case, the load directly driven by the rotational force of the motor 1 is the above-described piston, but the reciprocating compressor itself can also be regarded as the load of the motor 1.

<< Deformation, etc. >>
The matters described in one embodiment can be applied to other embodiments as long as there is no contradiction. At the time of this application, it is interpreted that there is no code difference (a difference between 3, 3a, 3b, and 3c, etc.) as appropriate for a part having the same name. As modifications or annotations of the above-described embodiment, notes 1 to 4 are described below. The contents described in each comment can be arbitrarily combined as long as there is no contradiction.

[Note 1]
Various command values above (i? ^*, I? ^*, V? ^* And v? ^*, Etc.) and state quantities including (i?, I? Etc.), derived technique of all the values to be derived is arbitrary. That is, for example, they may be derived by calculation within the motor control device (3, 3a, 3b or 3c), or may be derived from preset table data.

[Note 2]
In each of the embodiments described above, the U-phase current value i _u and the V-phase current value _iv are directly detected using the phase current sensor 11, but based on the DC current on the power source side of the inverter 2, May be detected.

[Note 3]
A part or all of the functions of the motor control device (3, 3a, 3b or 3c) is realized by using software (program) incorporated in a general-purpose microcomputer, for example. When a motor control device is realized using software, a block diagram showing a configuration of each part of the motor control device represents a functional block diagram. Of course, it is also possible to form the motor control device not by software (program) but only by hardware or by a combination of software and hardware.

[Note 4]
The following points should be noted in this specification and the drawings. Γ and δ expressed as so-called subscripts in the description of the formulas in the brackets (formula (1), etc.) expressed as the above number or in the drawing are subscripts outside the brackets. Can be written as non-standard characters. The difference between the γ and δ subscripts and the standard characters should be ignored.

1 is a schematic block diagram of a motor drive system according to an embodiment of the present invention. It is an analysis model figure of the motor of FIG. 1 is a block diagram of a motor drive system according to an embodiment of the present invention. It is a figure which shows the signal waveform corresponding to the block diagram of FIG. It is a figure which shows the signal waveform corresponding to the block diagram of FIG. FIG. 4 is a diagram illustrating the motor speed dependency of the phase difference (Δε) between the actual speed signal and the estimated speed signal, according to the block diagram of FIG. 3. It is a figure which shows the signal waveform corresponding to the block diagram of FIG. It is a vector diagram regarding a motor voltage when d-axis current is zero. 1 is a detailed block diagram of a motor drive system according to a first embodiment of the present invention. FIG. 10 is an internal block diagram of the position / velocity estimator of FIG. 9. According to the first embodiment of the present invention, the load torque current conversion value, the δ-axis current command value, the error between the load torque current conversion value and the δ-axis current command value, and the δ-axis current correction value after phase adjustment It is a figure which shows the waveform of It is a detailed block diagram of the motor drive system which concerns on 2nd Example of this invention. It is a detailed block diagram of the motor drive system which concerns on 3rd Example of this invention. It is an external view of the compressor which concerns on 4th Example of this invention. It is a block diagram of the conventional motor drive system.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Motor 2 Inverter 3, 3a, 3b, 3c Motor control apparatus 15 Current control part 17 Speed control part 20 Position / speed estimator 30, 30c Resonance type filter 32 Phase adjustment part

Claims (6)

In a motor control device that executes vector control for a motor that drives a load whose load torque fluctuates periodically through estimation of the rotational speed of the motor,
When the axis orthogonal to the direction of the magnetic flux created by the permanent magnet provided in the rotor of the motor is q-axis, and the estimated control axis corresponding to the q-axis is δ-axis,
The motor control device
Speed control means for generating a δ-axis current command signal based on an estimated speed signal representing an estimated rotational speed;
A correction unit that generates a δ-axis current correction signal that fluctuates in response to a change in the load torque, and that corrects the δ-axis current command signal using the δ-axis current correction signal;
Current control means for generating a δ-axis voltage command signal that the δ-axis component of the applied voltage to the motor should follow so that the δ-axis component of the current supplied to the motor follows the corrected δ-axis current command signal And comprising
The motor controller according to claim 1, wherein the correction means adjusts the phase of the δ-axis current correction signal based on the δ-axis voltage command signal. 2. The motor control device according to claim 1, wherein the correction unit adjusts a phase of the δ-axis current correction signal based on the δ-axis voltage command signal and the estimated speed signal. 3. The motor control according to claim 2, wherein the correction unit adjusts the phase of the δ-axis current correction signal based on phase information of the δ-axis voltage command signal and phase information of the estimated speed signal. apparatus. The correction means includes a resonance type filter that receives a control signal that fluctuates in response to fluctuations in the load torque and generates the δ-axis current correction signal by emphasizing a periodic fluctuation component of the control signal, 4. The phase of the δ-axis current correction signal is adjusted based on the difference between the phase of the δ-axis voltage command signal and the phase of the estimated speed signal. 5. Motor control device. A motor,
An inverter for driving the motor;
A motor control system according to any one of claims 1 to 4, wherein vector control for the motor is executed via the inverter. 6. A compressor using a rotational force of a motor provided in the motor drive system according to claim 5 as a drive source.

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