JP2009261103A - Motor controller - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To enable combined use of sensorless drive system and flux-weakening control without impeding smooth drive of a motor. <P>SOLUTION: A current command value generating section 11 generates a current command value I<SB>dq</SB><SP>*</SP>(d-axis current command value and q-axis current command value). A position estimator 20 estimates the induced voltage of a motor 3, and determines the estimated rotational position θ^ of the rotor in the motor 3 based on the induced voltage thus estimated. The estimated rotational position θ^ is used in coordinate converters 14 and 18 which perform coordinate transformation between a rotary coordinate system and a fixed coordinate system. In order to increase torque output from the motor 3 in a high-speed rotation range, the current command value generation section 11 performs flux-weakening control using a d-axis current command value of a negative significant value. The d-axis current command value is limited so that an induced voltage required for maintaining precision at the position estimator 20 for estimating rotational position of the rotor can be ensured. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a motor control device for sensorless driving of a brushless motor. The brushless motor is used, for example, as a source for generating a steering assist force in an electric power steering apparatus.

  A motor control device for driving and controlling a brushless motor is generally configured to control the supply of motor current in accordance with the output of a position sensor for detecting the rotational position (electrical angle) of the rotor. However, the environmental resistance of the position sensor becomes a problem, and the expensive position sensor and the wiring associated therewith hinder the cost reduction and the size reduction. Therefore, a sensorless driving method for driving a brushless motor without using a position sensor has been proposed. The sensorless driving method is a method for estimating the phase of the magnetic pole (electrical angle of the rotor) by estimating the induced voltage accompanying the rotation of the rotor (see Patent Document 1).

  On the other hand, the motor control device performs a sine wave drive that applies a voltage that changes in a sinusoidal manner to each phase of the stator based on the rotational position of the rotor. For example, in an electric power steering device, the motor control device sets a command value of a two-phase current in dq coordinates, that is, a d-axis current command value and a q-axis current command value, based on a steering torque detected by a torque sensor. To do. Further, the motor control device detects the d-axis current and the q-axis current actually flowing in the motor, obtains the deviation of the d-axis current and the q-axis current with respect to each command value, and the d-axis voltage corresponding to the deviation. The command value and the q-axis voltage command value are calculated. The motor control device converts the d-axis voltage command value and the q-axis voltage command value into three-phase (U-phase, V-phase, W-phase) voltage values, and applies the voltages of these values to each phase of the motor. (See Patent Document 2).

In the low-medium speed rotation range, the d-axis current command value is set to zero, while the q-axis current command value is set to a value corresponding to the steering torque, whereby the necessary torque can be generated from the motor. However, in the high-speed rotation range, the output (torque) is insufficient due to the counter electromotive force of the motor. Therefore, in order to increase the output of the motor, the d-axis current command value is set to a significant value other than zero, and flux-weakening control is performed so that current flows in the direction of weakening the field.
JP 2007-37299 A WO2006 / 109809

However, if the flux-weakening control is performed, the induced voltage becomes small, so that a change in the induced voltage cannot be detected accurately. For this reason, if the flux-weakening control is used in combination with the sensorless driving method, the accuracy of the rotor position estimation using the induced voltage is deteriorated. As a result, the motor may not be able to be driven smoothly.
Accordingly, an object of the present invention is to suppress a decrease in the estimated position accuracy of the rotor caused by the flux weakening control, thereby enabling the combined use of the sensorless driving method and the flux weakening control without hindering the smooth drive of the motor. A motor control device is provided.

  The invention described in claim 1 for achieving the above object is a motor control device for controlling a motor (3) including a rotor (50) and a stator (51-53) facing the rotor. The motor induced voltage detection means (9, 35) for detecting the motor induced voltage generated by the rotation of the rotor, and the rotational position of the rotor based on the motor induced voltage detected by the motor induced voltage detection means. A position estimating means (20, 37) to be obtained, a weak magnetic flux control means (11, S7) for performing a weak magnetic flux control for increasing the output of the motor by weakening the field of the rotor, and the weak magnetic flux control means. It is a motor control apparatus including a limiting means (11, S8, S9) for limiting the magnetic flux weakening control. The alphanumeric characters in parentheses indicate corresponding components in the embodiments described later. The same applies hereinafter.

  According to this configuration, by limiting the weak magnetic flux control, the motor induced voltage can be maintained at a large value even when the weak magnetic flux control is performed. Thereby, it can suppress or prevent that the precision of rotor rotation position estimation using a motor induced voltage deteriorates. As a result, both the sensorless driving method and the flux-weakening control can be achieved. Therefore, even when the flux-weakening control is performed, the rotor rotational position can be accurately estimated and the motor can be driven smoothly.

According to a second aspect of the present invention, the motor control device controls the d-axis current and the q-axis current on the dq coordinate, and the limiting means calculates the d-axis current command value I _d ^* by the following formula ( The motor control device according to claim 1, wherein the motor control device is limited to a range satisfying A).
I _d ^* ≧ (K−ωφ) ÷ (ωL _d ) (A)
Where K is a predetermined voltage threshold, ω is the rotational angular velocity of the rotor, φ is √ (3/2) of the maximum number of stator winding linkage magnetic fluxes due to the field, and L _d is the d-axis inductance.

According to this configuration, the motor of the size required for the rotor rotational position estimation calculation is obtained by adding the restriction according to the formula (A) to the d-axis current command value I _d ^* for the flux weakening control. An induced voltage is secured. As a result, even when the flux-weakening control is performed, the rotor rotational position can be accurately estimated reliably, so that the motor can be driven more smoothly.
The motor voltage equation in the dq coordinate is given by the following equations (B) and (C). Of these, if the value of (ωL _d I _d ^* + ωφ), which is the sum of the second term and the third term of the formula (C), is small, it is difficult to detect the induced voltage change, and the motor can be driven smoothly. It becomes difficult.

ただし、Ｖ_dはｄ軸電圧、
Ｖ_qはｑ軸電圧、
Ｒはステータ巻線抵抗、
ωはロータの回転角速度、
Ｌ_dはｄ軸インダクタンス
Ｌ_qはｑ軸インダクタンス
Ｉ_d ^*はｄ軸電流指令値、
Ｉ_q ^*はｑ軸電流指令値、
φは界磁によるステータ巻線鎖交磁束数最大値の√（３／２）である。

Where V _d is the d-axis voltage,
V _q is the q-axis voltage,
R is the stator winding resistance,
ω is the rotational angular speed of the rotor,
L _d is d-axis inductance
L _q is q-axis inductance
I _d ^* is the d-axis current command value,
I _q ^* is the q-axis current command value,
φ is √ (3/2) of the maximum number of stator winding interlinkage magnetic fluxes due to the field.

On the other hand, in the flux weakening control, the value of (ωL _d I _d ^* + ωφ) is reduced by increasing the d-axis current command value I _d ^* (<0) to a significant value, thereby increasing the torque output.
Therefore, if the voltage threshold value K is determined so that the rotor rotational position can be estimated with necessary accuracy from the induced voltage change and the following equation (D) is satisfied, even when the flux-weakening control is performed, the motor induced voltage can be reduced. There is no problem in the estimation of the rotor rotation position based on it. When the equation (D) is solved for the d-axis current command value I _d ^* , the equation (A) is obtained.

ωL _d I _d ^* + ωφ ≧ K (D)

Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings.
FIG. 1 is a block diagram for explaining an electrical configuration of an electric power steering apparatus to which a motor control apparatus according to an embodiment of the present invention is applied. The electric power steering apparatus includes a torque sensor 1 that detects a steering torque applied to a steering wheel of a vehicle, a motor 3 (electric motor) that applies a steering assist force to a steering mechanism 2 of the vehicle, and drive control of the motor 3. And a motor control device 5. The motor control device 5 drives the motor 3 according to the steering torque detected by the torque sensor 1, thereby realizing appropriate steering assistance according to the steering situation.

  In this embodiment, the motor 3 is a three-phase brushless motor, and as shown schematically in FIG. 2, the rotor 50 as a field and the U-phase, V-phase and W-phase arranged in the stator facing the field 50 are provided. Phase stator windings 51, 52, 53. The motor 3 may be of an inner rotor type in which a stator is disposed facing the outside of the rotor, or may be of an outer rotor type in which a stator is disposed facing the inside of a cylindrical rotor.

The motor control device 5 detects a current that flows through the microcomputer 7, a drive circuit (inverter circuit) 8 that is controlled by the microcomputer 7 and supplies power to the motor 3, and a stator winding of each phase of the motor 3. And a current sensor 9.
The microcomputer 7 includes a CPU and a memory (such as a ROM and a RAM), and functions as a plurality of function processing units by executing a predetermined program. The plurality of function processing units include a current command value generation unit 11, a PI (proportional integration) control unit 12, an instruction voltage generation unit 13, a γδ / αβ coordinate conversion unit 14, and an αβ / UVW coordinate conversion unit 15. A PWM control unit 16, a UVW / αβ coordinate conversion unit 17, an αβ / γδ coordinate conversion unit 18, a deviation calculation unit 19, a position estimation unit 20, and an angular velocity calculation unit 21.

The current command value generation unit 11 generates a command value I _d ^* of a d-axis current component along the rotor magnetic pole direction of the motor 3 and a command value I _q ^* of a q-axis current component orthogonal to the d axis. Hereinafter, these are collectively referred to as “current command value I _dq ^* ”. However, the dq coordinate plane is a plane along the rotation direction of the rotor 50, and the d axis and the q axis define a two-phase rotation coordinate system (dq) that rotates together with the rotor 50 (see FIG. 2).

When the current command value I ^* representing the amplitude of the current (sine wave current) to be applied to the U phase, V phase and W phase of the motor 3 is used, the d axis current command value I _d ^* and the q axis current command value I _q ^*. Is expressed by the following equations (1) and (2).
I _d ^* = 0 (1)
I _q ^* = − (3/2) ^1/2・ I ^* (2)
Therefore, the current command value generation unit 11 generates the d-axis current command value I _d ^* = 0, while generating the q-axis current command value I _q ^* corresponding to the steering torque detected by the torque sensor 1. To work. More specifically, the q-axis current command value I _q ^* may be generated using a map (table) that stores the q-axis current command value I _q ^* corresponding to the steering torque. Since the torque generated by the motor 3 corresponds to the motor current, the current command value I _dq ^* can be rephrased as a “torque command value” for commanding the torque to be generated from the motor 3.

On the other hand, since the torque output of the motor 3 is insufficient in a high speed rotation range (for example, a rotation speed range exceeding 800 rpm), the current command value generation unit 11 sets the d-axis current command value I _d ^* ≠ 0, So-called weakening magnetic flux control is performed to increase the output of the motor 3.
Although details of the operation of the current command value generation unit 11 will be described later, in this embodiment, the current command value generation unit 11 takes into account the motor rotation angular velocity ω calculated by the angular velocity calculation unit 21 and takes into account the current command value I _dq. ^{* Is} set.

The current sensor 9 detects the U-phase current I _U , the V-phase current I _V and the W-phase current I _W of the motor 3 (hereinafter referred to as “three-phase detection current I _UVW ” when collectively referred to). The detected value is sampled at a predetermined sampling frequency f _s , taken into the microcomputer 7, and given to the UVW / αβ coordinate converter 17.
The UVW / αβ coordinate converter 17 converts the three-phase detection current I _UVW into the currents I _α and I _β on the two-phase fixed coordinate system (α-β) (hereinafter referred to as “two-phase detection current I _The coordinates are converted to “ _αβ ”. The two-phase fixed coordinate system (α-β) is a fixed coordinate system in which the rotation center of the rotor 50 is the origin and the α axis and the β axis orthogonal to the rotation axis of the rotor 50 are defined (see FIG. 2). ). The coordinate-converted two-phase detection current I _αβ is given to the αβ / γδ coordinate conversion unit 18.

The αβ / γδ coordinate conversion unit 18 converts the two-phase detection current I _αβ into a two-phase rotation coordinate system (hereinafter referred to as “estimated rotation position θ ^”) estimated by the position estimation unit 20. Coordinates are converted into currents I _γ and I _δ on γ−δ) (hereinafter, collectively referred to as “two-phase detection current I _γδ ”). The two-phase rotational coordinate system (γ−δ) is a rotational coordinate system defined by a γ axis along the rotor magnetic pole direction and a δ axis orthogonal to the γ axis when the rotor 50 is at the estimated rotational position θ ^. (See FIG. 2). When the estimated rotational position θ ^ has no error and coincides with the actual rotor rotational position, the two-phase rotational coordinate system (dq) and the two-phase rotational coordinate system (γ-δ) coincide.

The two-phase detection current I _γδ is supplied to the deviation calculation unit 19. The deviation calculator 19 calculates a deviation δ _Id of the γ-axis current I _{γ with} respect to the d-axis current command value I _d ^* and a deviation δ _Iq of the δ-axis current I _{δ with} respect to the q-axis current command value I _q ^* . These current deviations δ _Id and δ _Iq are given to the PI control unit 12 and subjected to PI calculation processing. Then, in accordance with these calculation results, the command voltage generation unit 13 causes the γ-axis command voltage V _γ ^* and the δ-axis command voltage V _δ ^* (hereinafter referred to as “two-phase command voltage V _γδ ^* ” when collectively referred to). .) Is generated and provided to the γδ / αβ coordinate converter 14.

The γδ / αβ coordinate converter 14 converts the γ-axis command voltage V _γ ^* and the δ-axis command voltage V _δ ^* into α-axis command voltages V _α ^* and β that are command voltages of the two-phase fixed coordinate system (α-β). Coordinates are converted to an axis command voltage V _β ^* (hereinafter referred to as “two-phase command voltage V _αβ ^* ” when these are collectively referred to). The two-phase instruction voltage V _αβ ^* is given to the αβ / UVW coordinate conversion unit 15.
The αβ / UVW coordinate converter 15 converts the α-axis command voltage V _α ^* and the β-axis command voltage V _β ^* into the command voltages of the three-phase fixed coordinate system, that is, the command voltages V _U ^{* of the} U phase, the V phase, and the W phase ^. , V _V ^* , V _W ^* (hereinafter collectively referred to as “three-phase indicating voltage V _UVW ^* ”).

The PWM control unit 16 generates a drive signal having a duty ratio controlled according to the three-phase instruction voltages V _U ^* , V _V ^* , and V _W ^* and supplies the drive signal to the drive circuit 8. As a result, a voltage is applied to each phase of the motor 3 with a duty ratio corresponding to the instruction voltages V _U ^* , V _V ^* , and V _W ^* of the corresponding phase.
With such a configuration, when a steering torque is applied to a steering wheel (not shown) as an operation member coupled to the steering mechanism 2, this is detected by the torque sensor 1. Then, the current command value generation unit 11 generates a current command value I _dq ^* corresponding to the detected steering torque. A deviation between the current command value I _dq ^* and the two-phase detection current I _γδ is obtained by the deviation calculating unit 19, and PI calculation is performed by the PI control unit 12 so as to lead this deviation to zero. A two-phase command voltage V _γδ ^* corresponding to this calculation result is generated by the command voltage generation unit 13, and this is converted into a three-phase command voltage V _UVW ^* via the coordinate conversion units 14 and 15. The drive circuit 8 operates at a duty ratio corresponding to the three-phase instruction voltage V _UVW ^* by the action of the PWM control unit 16 to drive the motor 3 and assist torque corresponding to the current command value I _dq ^*. Is given to the steering mechanism 2. Thus, steering assistance can be performed according to the steering torque. The three-phase detection current I _UVW detected by the current sensor 9 passes through the coordinate conversion units 17 and 18 and is expressed in two-phase rotational coordinate system (γ−δ) so as to correspond to the current command value I _dq ^*. After being converted to the phase detection current I _γδ , it is given to the deviation calculating unit 19.

In order to perform coordinate conversion between the rotating coordinate system and the fixed coordinate system, a phase angle (electrical angle) θ representing the rotational position of the rotor 50 is required. An estimated rotational position θ ^ representing this phase angle is generated by the position estimation unit 20 and is provided to the γδ / αβ coordinate conversion unit 14 and the αβ / γδ coordinate conversion unit 18.
The position estimation unit 20 performs two-phase detection current I _αβ given from the UVW / αβ coordinate conversion unit 17 and two-phase generated by the γδ / αβ coordinate conversion unit 14 for each sampling period Ts (= 1 / f _s ). Based on the command voltage V _αβ ^* , the rotational position of the rotor 50 is estimated, and an estimated rotational position θ ^ is generated.

The angular velocity calculation unit 21 calculates the rotational angular velocity ω of the rotor 50 by performing a time differential calculation on the estimated rotation position θ ^ obtained by the position estimation unit 20. This rotational angular velocity ω is used for generation of the current command value I _dq ^* by the current command value generation unit 11.
FIG. 3 is a block diagram for explaining the configuration of the position estimation unit 20. The position estimation unit 20 includes a signal processing unit 31 and a rotor position estimation unit 32. The signal processing unit 31 includes a voltage filter 33 configured by a low-pass filter that removes a high-frequency component of the two-phase indication voltage V _αβ ^* , and a low-pass filter that removes a high-frequency component of the two-phase detection current I _αβ. Current filter 34. The rotor position estimation unit 32 is supplied with the two-phase command voltage V _αβ ^* and the two-phase detection current I _αβ after signal processing (filtering) by the signal processing unit 31. The rotor position estimation unit 32 removes a high-frequency component from the disturbance observer 35 that estimates the induced voltage of the motor 3 as a disturbance and the estimated induced voltage output by the disturbance observer 35 based on a motor model that is a mathematical model of the motor 3. Based on an estimated value filter 36 composed of a low-pass filter and an estimated induced voltage (value after filtering) output from the estimated value filter 36, an estimated position generation unit that generates an estimated rotational position θ ^ of the rotor 50 37. Then, the two-phase command voltage V _αβ ^* filtered by the voltage filter 33 and the two-phase detection current I _αβ filtered by the current filter 34 are input to the disturbance observer 35 of the rotor position estimation unit 32, respectively. It has become.

FIG. 4 is a block diagram for explaining an example of the disturbance observer 35 and a configuration related thereto. A motor model that is a mathematical model of the motor 3 can be expressed as, for example, (R + pL) ⁻¹ . Here, R is an armature winding resistance, L is an αβ axis inductance, and p is a differential operator. It can be considered that the two-phase command voltage V _αβ ^* and the induced voltage E _αβ (α-axis induced voltage E _α and β-axis induced voltage E _β ) are applied to the motor 3.

The disturbance observer 35 receives a two-phase detection current I _αβ as an input and estimates an inverse motor model (inverse model of the motor model) 38 for estimating a motor voltage, and a motor voltage estimated by the inverse motor model 38 and a two-phase indication voltage V _αβ. And a voltage deviation calculation unit 39 for obtaining a deviation from ^* . The voltage deviation calculating unit 39 obtains a disturbance with respect to the two-phase indicating voltage V _αβ ^* . As is apparent from FIG. 4, this disturbance is an estimated value E ^ _αβ (α-axis induced voltage corresponding to the induced voltage E _αβ. The estimated value E ^ _α and the β-axis induced voltage estimated value E ^ _β (hereinafter collectively referred to as “estimated induced voltage E ^ _αβ ”) The reverse motor model 38 is represented by R + pL, for example.

The estimated value filter 36 can be constituted by a low-pass filter represented by a / (s + a), for example. a is a design parameter, and the cut-off frequency ω _c of the estimated value filter 36 is determined by the design parameter a.
The induced voltage E _αβ can be expressed by the following equation (3). However, K _E is an induced voltage constant, θ is a rotor rotational position, and ω is a rotor rotational speed.

したがって、推定誘起電圧Ｅ^_αβが求まれば、次の(4)に従って、推定回転位置θ^が求まる。この演算が、推定位置生成部３７によって行われるようになっている。

Therefore, when the estimated induced voltage E ^ _αβ is obtained, the estimated rotational position θ ^ is obtained according to the following (4). This calculation is performed by the estimated position generation unit 37.

図５は、電流指令値生成部１１の動作を説明するための図であり、モータ回転角速度ωに応じたｑ軸電流指令値Ｉ_q ^*の上限値Ｉ_{q_uplim}および下限値Ｉ_{q_downlim}が示されている。これらの上限値Ｉ_{q_uplim}および下限値Ｉ_{q_downlim}の間でｑ軸電流指令値Ｉ_q ^*が設定されることによって、電動モータ３のＵ相、Ｖ相およびＷ相に正弦波電圧を印加する正弦波駆動が可能になる。

Figure 5 is a diagram for explaining the operation of the current command value generating unit 11, and q-axis current command value corresponding to the motor rotational angular velocity omega I _q ^* upper limit value I _{Q_uplim} and the lower limit I _{Q_downlim} is shown Yes. By setting the q-axis current command value I _q ^* between the upper limit value I _{q_uplim} and the lower limit value I _{q_downlim} , a sine wave that applies a sine wave voltage to the U phase, V phase, and W phase of the electric motor 3 Drive becomes possible.

When the flux-weakening control is not performed, that is, when the d-axis current command value I _d ^* = 0, the q-axis current command value I _q ^* is limited according to the characteristics L1 ⁺ and L1 ⁻ indicated by broken lines. Characteristic L1 ⁺ indicates an upper limit value when positive q-axis current command value I _q ^* is set in order to generate torque in one direction (for example, clockwise direction), and characteristic L1 ⁻ indicates the other direction (for example, left direction). The lower limit value when the negative q-axis current command value I _q ^* is set in order to generate torque in the rotation direction). In this case, the upper limit value and the lower limit value of the q-axis current command value I _q ^* are respectively in a low and medium speed rotation range where the absolute value of the motor rotation angular speed ω is equal to or less than a predetermined first threshold value ω1 (for example, ω1 = 800 rpm) In the high speed region where the absolute value of the motor rotational angular velocity ω exceeds the first threshold ω1, the upper limit value of the q-axis current command value I _q ^* is constant values α and −α (where α is a positive constant). The lower limit of the q-axis current command value I _q ^* increases linearly as the absolute value of the motor rotational angular velocity ω increases (the absolute value decreases). ) Characteristic. That is, the absolute values of the upper limit value and the lower limit value of the q-axis current command value I _q ^* are held at a constant value in the medium / low speed rotation range that is equal to or less than the first threshold value ω1, while In FIG. 4, the characteristic decreases linearly as the motor rotational angular velocity ω increases.

On the other hand, when the flux-weakening control is performed, the q-axis current command value I _q ^* is limited according to the characteristics L2 ⁺ and L2 ⁻ indicated by the solid lines. Characteristic L2 ⁺ indicates the upper limit value when setting a positive q-axis current command value I _q ^* in order to generate torque in one direction (e.g., clockwise), characteristic L2 ^- the other direction (e.g., left The lower limit value when the negative q-axis current command value I _q ^* is set in order to generate torque in the rotation direction). In this case, in the rotational speed region where the absolute value of the motor rotational angular speed ω is equal to or smaller than the second threshold ω2 (> ω1, for example, ω2 = 900 rpm) larger than the first threshold ω1, the q-axis current command value I _q ^* The upper limit value and the lower limit value are constant values α and −α, respectively. In the rotational speed range where the absolute value of the motor rotational angular speed ω exceeds the second threshold ω2, the upper limit value of the q-axis current command value I _q ^* decreases nonlinearly as the motor rotational angular speed ω increases. The lower limit value of the q-axis current command value I _q ^* is a characteristic that increases nonlinearly (absolute value decreases) as the absolute value of the motor rotational angular velocity ω increases. In other words, the absolute values of the upper limit value and the lower limit value of the q-axis current command value I _q ^* are kept constant in the rotation speed range below the second threshold value ω2, while in the rotation speed range exceeding the second threshold value ω2. The characteristic is non-linearly decreasing as the motor rotational angular velocity ω increases.

The q-axis current corresponds to the output torque of the electric motor 3. Therefore, when the flux-weakening control is not performed, as understood from the characteristics L1 ⁺ and L1 ⁻ , the absolute value of the motor rotational angular velocity ω increases in a high-speed rotational range where the absolute value of the motor rotational angular velocity ω exceeds the first threshold value ω1. Accordingly, the output torque decreases, and it becomes impossible to generate torque at a certain rotational speed ω10 (for example, ω10 = 1500 rpm). Therefore, in the high-speed rotation region exceeding the first threshold value ω1, the d-axis current command value I _d ^* is set to a significant value other than zero and the magnetic flux weakening control is performed. The output of the electric motor 3 can be increased.

The upper limit value I _{q_uplim} of the q-axis current command value I _q ^* when the flux-weakening control is performed is given by the following formula (5), and the lower limit value I _{q_downlim} is given by the following formula (6). However, the upper limit value I _{q_uplim} is applied to the positive q-axis current command value I _q ^* , and the lower limit value I _{q_downlim} is applied to the negative q-axis current command value I _q ^* .

Ｒはモータの固定子巻線抵抗
φは界磁によるステータ鎖交磁束から計算されるｄｑ座標上の磁束
（すなわち、界磁のＵ相、Ｖ相、Ｗ相ステータ巻線鎖交磁束数最大値の√（３／２））
ωはモータの電気角における回転角速度
Ｌ_dはｄ軸インダクタンス
Ｌ_qはｑ軸インダクタンス
Ｖ_limは正弦波駆動が可能なｄｑ座標での制限電圧
αは正の定数
である。

R is the stator winding resistance of the motor φ is the magnetic flux on the dq coordinate calculated from the stator interlinkage magnetic flux by the field (ie, the maximum number of field U-phase, V-phase, and W-phase stator interlinkage magnetic fluxes √ (3/2))
ω is the rotational angular velocity at the electrical angle of the motor L _d is the d-axis inductance L _q is the q-axis inductance V _lim is the limit voltage α in the dq coordinate that allows sinusoidal driving, and α is a positive constant.

Further, the d-axis current command value I _d ^* for the flux weakening control is given by the following formula (7).

である。

It is.

However, for the d-axis current command value I _d ^* determined by the equation (7), the current command value generation unit 11 further determines whether or not the d-axis current upper limit value I _{d_limit} shown by the following equation (8) is exceeded. To do. When the d-axis current command value I _d ^* obtained according to the equation (7) exceeds the d-axis current upper limit value I _{d_limit} , the current command value generation unit 11 sets I _d ^* = I _{d_limit} as The d-axis current command value I _d ^* is limited to the d-axis current upper limit value I _{d_limit} . That is, the d-axis current command value I _d ^* is limited to the d-axis current upper limit value I _{d_limit} or less.

I _{d_limit} = (K− _ωφ ) / (ωL _d ) (8)
FIG. 6 is a flowchart for explaining the operation of the current command value generation unit 11 when the flux-weakening control is performed. Current command value generation section 11, characteristic L2 ⁺ in FIG. 5, L2 ^- accordingly sets the upper limit value I _{Q_uplim} and lower limit value I _{Q_downlim} corresponding to the motor rotational angular velocity omega (step S1). Furthermore, the current command value generation unit 11 sets the q-axis current command value I _q ^* according to the steering torque or the like (step S2).

Next, the current command value generation unit 11 compares the q-axis current command value I _q ^* with the upper limit value I _{q_uplim} (step S3). If the q-axis current command value I _q ^* is larger than the upper limit value I _{q_uplim} (step S3: YES), the q-axis current command value I _q ^* is limited. That is, the upper limit value I _{q_uplim} is substituted as the q-axis current command value I _q ^* for use in subsequent control (step S4).

On the other hand, if the q-axis current command value I _q ^* is equal to or lower than the upper limit value I _{q_uplim} (step S3: NO), then the current command value generation unit 11 then determines the q-axis current command value I _q ^* and the lower limit value I _{q_downlim.} Are compared (step S5). If the q-axis current command value I _q ^* is less than the lower limit value I _{q_downlim} (step S5: YES), the q-axis current command value I _q ^* is limited. That is, the lower limit I _{q_downlim} is substituted as the q-axis current command value I _q ^* for use in subsequent control (step S6).

If the q-axis current command value I _q ^* set in step S2 is not _more than the upper limit value I _{q_uplim} (step S3: NO) and not _less than the lower limit value I _{q_downlim} (step S5: NO), the q The shaft current command value I _q ^* is used as it is.
Next, the current command value generation unit 11 sets the d-axis current command value I _d ^* (negative value) according to the equation (7) (step S7). Further, the current command value generation unit 11 compares the d-axis current command value I _d ^* with the d-axis current upper limit value I _{d_limit} (negative value) of the above equation (8) (step S8). If the d-axis current command value I _d ^* is smaller than the d-axis current upper limit value I _{d_limit} (if the absolute value is large) (step S8: YES), the d-axis current command value I _d ^* is limited. . That is, the d-axis current upper limit value I _{d_limit} is substituted as the d-axis current command value I _d ^* for use in subsequent control (step S9). On the other hand, if the d-axis current command value I _d ^* is equal to or greater than the d-axis current upper limit value I _{d_limit} (step S8: NO), the d-axis current command value I _d ^* set according to the equation (7) is used as it is. .

Based on the q-axis current command value I _q ^* and the d-axis current command value I _d ^* thus obtained, the deviation calculation unit 19 obtains the d-axis current deviation δ _Id and the q-axis current deviation δ _Iq , respectively. Further, based on the current deviations δ _Id and δ _Iq , PI calculation processing by the PI control unit 12 is performed.
FIG. 7 is a diagram showing conditions imposed on the d-axis voltage command value V _d ^* and the q-axis voltage command value V _q ^* for the sine wave drive. In order for the voltage of each phase of the electric motor 3 to be a sine wave, the amplitude of each phase voltage is set to the power supply voltage E _d (voltage applied to the drive circuit 8; battery voltage may be substituted). The following equation (9) needs to be satisfied with respect to the d-axis voltage V _d and the q-axis voltage V _q .

√ (V _d ² + V _q ² ) ≦ E _d √3 / 2√2 = V _lim (9)
On the other hand, the q-axis voltage V _q and the d-axis voltage V _d in the steady state are expressed by the following equations (motor voltage equations in the dq coordinates).

これらを前記式(9)に代入すると、次の式(12)が得られる。

Substituting these into the equation (9) yields the following equation (12).

弱め磁束制御をしないとき、すなわち、ｄ軸電流指令値Ｉ_d ^*＝０で前記式(12)が満たされる条件は、次の式(13)で表される。

When the flux-weakening control is not performed, that is, the condition for satisfying the expression (12) with the d-axis current command value I _d ^* = 0 is expressed by the following expression (13).

これを変形すると、弱め磁束制御をしないときに正弦波駆動を行うための条件は、次の式(14)で与えられることがわかる。

If this is modified, it can be seen that the condition for performing sinusoidal drive when the flux-weakening control is not performed is given by the following equation (14).

一方、Ｉ_d ^*＝０では前記式(13)を満たさないとき、すなわち、下記式(15)の条件のときには、ｄ軸電流指令値Ｉ_d ^*≠０とすることにより前記式(12)を満たす必要がある。

On the other hand, when I _d ^* = 0 does not satisfy the above equation (13), that is, under the condition of the following equation (15), the above equation (12) is obtained by setting the d-axis current command value I _d ^* ≠ 0. It is necessary to satisfy.

そこで、前記式(12)をＩ_d ^*について解くと、下記式(16)を得る。

Therefore, when the equation (12) is solved for I _d ^* , the following equation (16) is obtained.

である。

It is.

Since the motor efficiency decreases as the absolute value of the d-axis current increases, the d-axis current command value I _d ^* may be set to a value having the minimum absolute value that satisfies the condition of the equation (16).
Since A> 0 is self-evident and C> 0 from the above equation (15), the following equations (17) and (18) are obtained.

前記式(18)のときは、ｄ軸電流指令値Ｉ_d ^*が正値となり、ｄ軸の界磁を強める方向に電流を流すことになるので、Ｂ＜０のときは、Ｉ_d ^*＝０とする。

In the case of the above equation (18), the d-axis current command value I _d ^* becomes a positive value, and the current flows in the direction of strengthening the d-axis field. Therefore, when B <0, I _d ^* = 0.

Here, if B ² −AC ≧ 0 is not satisfied, the value of the equation (17) becomes an imaginary number, and there is no I _d ^* that satisfies the equation (17). That is, no matter what q-axis current command value I _q ^* is set, it does not fall within the limit voltage range indicated by the circle in FIG. Therefore, when B ² -AC ≧ 0 is solved with respect to I _q ^* , the following equation (19) is obtained, and in order to make it within the limit voltage range, that is, to perform sinusoidal drive, the q-axis current command This is a condition imposed on the value I _q ^* .

である。

It is.

Therefore, the q-axis current command value I _q ^* is limited in advance to the range of the equation (19) (steps S3 to S6 in FIG. 6), and then the d-axis current command value I _d ^* is calculated by the equation (18). What is necessary is just to obtain | require (step S7 of FIG. 6). Thereby, the maximum output can be obtained within the limit voltage range.
On the other hand, when the sum of the second term and the third term of the equation (11), that is, ωL _d I _d ^* + ωφ is small, the change in the induced voltage of the motor 3 is small. The estimation accuracy becomes worse. Therefore, in this embodiment, the voltage threshold K is used to limit the d-axis current command value I _d ^* so that the following formula (20) is satisfied, thereby limiting the flux-weakening control. The voltage threshold value K is set to a value that can secure an induced voltage necessary for estimating the rotor rotational position with required accuracy.

ωL _d I _d ^* + ωφ ≧ K (20)
When this is solved for the d-axis current command value I _d ^* , the following equation (21) is obtained.
I _d ^* ≦ (K− _ωφ ) / (ωL _d ) = I _{d_limit} (21)
The processes in steps S8 and S9 in FIG. 6 are processes for applying such a restriction to the d-axis current command value I _d ^* .

As described above, according to this embodiment, when the flux-weakening control is performed, the d-axis current command value I _d ^* is set so that ωL _d I _d ^* + ω φ (induced voltage) is equal to or higher than the voltage threshold K. There are restrictions on this. As a result, it is possible to calculate the estimated rotational position θ ^ with high accuracy while increasing the output torque by performing flux-weakening control in the high rotational speed range. Therefore, it is possible to achieve both an increase in output torque and smooth control of the motor 3 in the high rotation speed range. In this way, it is possible to simultaneously obtain both effects of an increase in output torque due to the flux-weakening control and a cost reduction due to the adoption of the sensorless drive method.

  As mentioned above, although one Embodiment of this invention was described, this invention can also be implemented with another form. For example, the induced voltage of the motor 3 becomes small not only when the flux-weakening control is performed but also in the low rotational speed range, so that the position estimation accuracy by the position estimation unit 20 is low in the low rotational speed range. Therefore, it is preferable to use a motor having a large induced voltage constant φ so that a large induced voltage can be secured even in a low rotational speed range. In this case, the output torque in the high rotational speed region is reduced, but this decrease in output torque can be compensated by the flux weakening control. With such a configuration, the accuracy of the estimated rotational position θ ^ can be increased over the entire speed range, and a sufficient output torque can be obtained, so that the motor 3 can be driven smoothly and sufficiently. A steering assist force can be applied to the steering mechanism 2. In addition, it is not necessary to apply another algorithm for estimating the rotor rotational angular velocity in the low rotational speed region, so that the cost can be reduced.

In addition, for example, if the upper limit value I _{q_uplim} of the q-axis current command value I _q ^* when performing the flux-weakening control is set to a value between the characteristic L 1 ⁺ and the characteristic L ^{2 +} (maximum upper limit value) in FIG. The lower limit value I _{q_downlim} of the q-axis current command value I _q ^* when the flux-weakening control is also performed may be set to a value between the characteristic L1 ⁻ and the characteristic L2 ⁻ (minimum lower limit value) in FIG. . Thus, for example, characteristics L3 ⁺ indicated by the two-dot chain line in FIG. 5, L3 ^- characteristic L2 ⁺ As, L2 ^- than to the characteristics absolute value is set lower limit I _{Q_uplim} and lower limit value I _{Q_downlim} May be. Such characteristics L3 ⁺ and L3 ⁻ are, for example, in consideration of variations in characteristics of circuit elements such as the drive circuit 8 (manufacturing variations, temperature changes, secular changes, etc.) and fluctuations in input voltage (battery voltage). Regardless of these variations and fluctuations, the upper limit value I _{q_uplim} may _fall between the characteristics L1 ⁺ and L2 ⁺ and the lower limit value I _{q_downlim} may fall between the characteristics L1 ⁻ and L2 ⁻ . In this case, the characteristics L3 ⁺ and L3 ⁻ are obtained by multiplying the right sides of the equations (5) and (6) by a constant k (0 <k <1) as shown in the following equations (22) and (23). May be set.

また、前述の実施形態では、電動パワーステアリング装置の駆動源としてのモータ３に本発明が適用された例について説明したが、この発明は、電動パワーステアリング装置以外の用途のモータの制御に対しても適用が可能である。

In the above-described embodiment, the example in which the present invention is applied to the motor 3 as the drive source of the electric power steering apparatus has been described. However, the present invention is not limited to the control of the motor for applications other than the electric power steering apparatus. Is also applicable.

  In addition, various design changes can be made within the scope of the matters described in the claims.

1 is a block diagram for explaining an electrical configuration of an electric power steering device to which a motor control device according to an embodiment of the present invention is applied. FIG. It is an illustration figure for demonstrating the structure of a motor. It is a block diagram for demonstrating the structure of a position estimation part. It is a block diagram for demonstrating an example of a disturbance observer and the structure relevant to this. It is a figure for demonstrating operation | movement of an electric current command value production | generation part, and the upper limit value and lower limit value of q-axis electric current command value according to motor rotation angular velocity are shown. It is a flowchart for demonstrating operation | movement of the electric current command value production | generation part when performing flux-weakening control. It is a figure for demonstrating the conditions imposed on d-axis voltage command value and q-axis voltage command value for a sine wave drive.

Explanation of symbols

  5 ... Motor controller, 7 ... Microcomputer, 50 ... Rotor, 51, 52, 53 ... Stator winding

Claims (2)

A motor control device for controlling a motor including a rotor and a stator facing the rotor,
Motor induced voltage detection means for detecting a motor induced voltage generated by rotation of the rotor;
Position estimating means for determining the rotational position of the rotor based on the motor induced voltage detected by the motor induced voltage detecting means;
A weak magnetic flux control means for performing a weak magnetic flux control for weakening the field of the rotor and increasing the output of the motor;
A motor control device including restriction means for restricting the weak magnetic flux control by the weak magnetic flux control means. The motor control device controls the d-axis current and the q-axis current on the dq coordinate,
The motor control device according to claim 1, wherein the limiting means limits the d-axis current command value I _d ^* to a range satisfying the following expression (A).
I _d ^* ≧ (K−ωφ) ÷ (ωL _d ) (A)
Where K is a predetermined voltage threshold, ω is the rotational angular velocity of the rotor, φ is √ (3/2) of the maximum number of stator winding linkage magnetic fluxes due to the field, and L _d is the d-axis inductance.

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