JP2009033828A - Controller of motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To prevent an unsuitable current from flowing to cause unbalance of armature coil currents between phases while using a capacitive filter for preventing a high-frequency current component from flowing through the armature coil of an axial gap motor. <P>SOLUTION: The diplophase armature coils 6 and 7 of both stators 4 and 5 in an axial gap motor 1 are connected in series (or parallel) for each phase between a neutral 1N common to the armature coils 6 and 7 and a connection terminal 42 for conduction of each phase provided in the motor 1. One ends of capacitive filters 41 as many as phases of the armature coils 6 and 7 are interconnected while conducting to the neutral 1N, and the other end of each capacitive filter 41 is connected with the conduction line 23 of each phase for connecting an inverter circuit 38 and the motor 1. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a control device for an axial gap type electric motor.

2. Description of the Related Art An axial gap type electric motor including a rotor having a permanent magnet, two stators provided on both sides of the rotor in the axial direction of the rotor, and armature windings attached to the stators has been conventionally known. (For example, refer to Patent Document 1 and Patent Document 2). Such an axial gap type electric motor can generate a relatively high output torque while shortening the axial length of the rotor of the electric motor.
JP-A-10-271784 JP 2001-136721 A

  By the way, in the axial gap type electric motor, the magnetomotive force between the two stators can be increased and the output torque can be easily increased. On the other hand, the higher the magnetomotive force, the more the inverter circuit that energizes the armature winding. High-frequency current components such as ripple current resulting from the switching operation of each switch element (high-order frequency component current mainly composed of carrier frequency components such as triangular wave used in the inverter circuit) is the current of the armature winding Easy to superimpose on. Such a high-frequency current component increases the copper loss of the armature winding and the iron loss of the stator, and has the disadvantage that the energy efficiency of the motor is reduced. In general, an electric motor includes an inductive filter (generally a coil) connected in series to the armature winding in order to prevent a high-frequency current component from being superimposed on the armature winding. In many cases, the inductive filter further increases the copper loss. Furthermore, since the output voltage of the inverter circuit is applied to the armature winding with a voltage drop due to the inductive filter, there is also a disadvantage that the maximum value of the output that can be generated by the electric motor is lowered.

  In order to eliminate such inconvenience, a capacitive filter such as a capacitor is used, and a high-frequency current component of the output current of each phase of the inverter circuit is passed through the capacitive filter, so that the armature winding It can be considered that high-frequency current components do not flow.

  However, in this case, if the connection form between the capacitive filter and the inverter circuit or the armature winding is inappropriate, the resonance phenomenon that occurs between the capacitive filter and the armature winding or the capacitive filter of each phase. Unbalanced DC components are superimposed on the currents of the armature windings of each phase due to mutual variations in the characteristics (capacitance value, etc.) In addition, the center value (average value) of the current of each phase may cause an unbalanced offset between phases. When such an offset occurs, an eddy current is generated, which causes problems such as overheating of the stator, a decrease in energy efficiency, and demagnetization of the permanent magnet of the rotor.

  Also, due to the resonance phenomenon that occurs between the capacitive filter and the armature winding, the current flowing through the armature winding does not become a desired current, and consequently, in the operating range of the motor where the resonance phenomenon occurs, There is a fear that the motor cannot output a desired torque.

  The present invention has been made in view of such a background, and for an axial gap type electric motor, while using a capacitive filter for preventing a high frequency current component from flowing in the armature winding of the electric motor, Providing a control device that can prevent an inappropriate current from flowing, such as the current of the armature windings of the phase being unbalanced between the phases, and thus operating the motor satisfactorily with high energy efficiency The purpose is to do.

In order to achieve the above object, the motor control device of the present invention is equipped with a rotor having permanent magnets, two stators provided on both sides of the rotor in the axial direction of the rotor, and both the stators. A control device for energizing the armature windings of both stators of an axial gap type motor provided with a multi-phase armature winding, from an inverter circuit,
The multi-phase armature windings of the two stators are each between a neutral point common to the armature windings of the two stators and the connection terminals for energization of the respective phases provided in the motor. Each phase is connected in either a serial connection or a parallel connection, and the connection terminals for energization of each phase are connected to the output terminals of each phase of the inverter circuit via an energization line. ,
One ends of the same number of capacitive filters as the number of phases of the armature windings of both stators are electrically connected to the neutral point, and the other ends of the capacitive filters are connected to the energization of each phase. It is connected to a line (first invention).

  According to the first aspect of the invention, the high-frequency current component of the output current of each phase of the inverter circuit flows through the capacitive filter, so that the high-frequency current component flows through the armature windings of both stators. Is prevented. And in this case, each one end of the capacitive filter of the same number as the number of phases of the armature windings of both stators is electrically connected to the neutral point of the armature windings of both stators, and is connected to each other. The other end of the capacitive filter is connected to the energization line of each phase. That is, the potential at one end of each capacitive filter as a neutral point between the capacitive filters is set to be the same as the potential at the neutral point of the armature windings of both stators. As a result, it is possible to prevent a situation in which an interphase error occurs due to a deviation in amplitude and phase due to a resonance phenomenon due to an imbalance between the phases of the capacitive filter and the armature winding. It is possible to suppress the occurrence of an unbalanced offset between the phases of the current flowing through the armature winding.

  Thus, according to the first aspect of the present invention, the high-frequency current component is prevented from flowing through the armature windings of both stators by the capacitive filter, and the current flowing through the armature windings of both stators is unbalanced between the phases. Since it is possible to suppress the occurrence of such a situation, copper loss and iron loss in the electric motor can be reduced, and the energy efficiency of the electric motor can be increased. In addition, since the current flowing through the armature windings of both stators can be prevented from causing an unbalanced offset between phases, demagnetization of the permanent magnets of the rotor can be prevented.

  Further, according to the first invention, the armature windings of both stators are connected for each phase in either one of the connection forms of series connection and parallel connection. In addition, it is not necessary to provide separate capacitive filters and inverter circuits, and a simple and inexpensive control device can be configured.

  In the first invention, the current flowing through the armature windings of each phase of both stators is from the output current of the inverter circuit (here, the direction from the inverter circuit to the motor side is the positive direction of the output current) This is a current obtained by subtracting the current flowing through the capacitive filter corresponding to the phase (here, the direction from the energization line toward the capacitive filter is the positive direction of the current). Further, in order to output the target torque from the electric motor, it is necessary to pass a current corresponding to at least the target value of the output torque and the rotational speed of the rotor of the electric motor through the armature windings of both stators. In order to flow such a current through the armature windings of both starters, it is necessary to control the output current of the inverter circuit in consideration of the current flowing through the capacitive filter. In particular, in a situation where a resonance phenomenon occurs between the capacitive filter and the armature winding, the current flowing through the capacitive filter becomes a relatively large current. Need to compensate.

Therefore, in the first invention, a current sensor for detecting a current flowing through the energization line of each phase at a location between the capacitive filter and the inverter circuit;
Target armature current determination for determining a target armature current, which is a target value of the current flowing through the armature windings of both stators, according to at least the target value of the output torque of the motor and the rotational speed of the rotor of the motor Means,
The neutrality of each capacitive filter via the armature winding of the same phase as that of the energization line of the stator armature winding from one end on the energization line side of the capacitive filter Inverter target output current determination means for determining an inverter target output current that is a target value of the output current of the inverter circuit corresponding to the target armature current, based on a predetermined arithmetic expression including an impedance of a current path that reaches a point When,
It is preferable that the apparatus further comprises inverter control means for feedback-controlling the operation of the inverter circuit so that the output current of the inverter circuit detected via the current sensor matches the inverter target output current (second invention).

Alternatively, a current sensor for detecting a current flowing through the energization line of each phase at a location between the capacitive filter and the armature windings of the two stators;
Target armature current determining means for determining a target armature current, which is a target value of a current flowing through the armature windings of both stators, according to at least the target output torque of the motor and the rotational speed of the rotor of the motor; ,
Preferably, inverter control means for feedback controlling the operation of the inverter circuit so as to match the current of the armature windings of the two stators detected via the current sensor and the target armature current (third) invention).

  In the second aspect of the invention, since the current detected through the current sensor is the output current of the inverter circuit, both of the target armature currents for outputting the target output torque from the motor are obtained. In order to flow through the armature winding of the stator, the current obtained by adding the current flowing through the capacitive filter to the target armature winding is used as the target of the output current of the inverter circuit detected via the current sensor. It is necessary to control the inverter circuit.

  In this case, the impedance of each capacitive filter and the armature winding of the same phase as the energization line of the armature windings of the two stators from one end on the energization line side of the capacitive filter Based on a predetermined arithmetic expression representing the relationship between the impedance of the current path to the neutral point, the induced voltage of the armature winding, the current flowing in the armature winding, and the output current of the inverter circuit, An inverter target output current that is a target value of the output current of the inverter circuit corresponding to the target armature current can be determined. In this case, the impedance of each capacitive filter can be specified from the circuit constant value (capacitance value, etc.) of the elements constituting the capacitive filter and the rotational speed of the rotor of the motor. Similarly, the impedance of the current path can be specified from a circuit constant ground (such as an inductance or a resistance value) such as an armature winding provided in the current path and the rotational speed of the rotor of the motor. The induced voltage of the armature winding can be specified from the induced voltage constant of the motor and the rotational speed of the rotor.

  In the second invention, the operation of the inverter circuit is feedback-controlled so that the output current of the inverter circuit detected via the current detection sensor matches the inverter target output current thus determined. Thus, according to the second aspect of the present invention, the influence of the current flowing through the capacitive filter is compensated regardless of the occurrence of the resonance phenomenon between the capacitive filter and the armature winding, and the target output torque is compensated from the motor. Thus, an appropriate current (the target armature current) can be passed through the armature winding.

  On the other hand, in the third invention, the current detected through the current sensor is the current of the armature windings of the two stators, and the current does not depend on the current flowing through the capacitive filter. . In the third aspect of the invention, the operation of the inverter circuit is feedback controlled so that this current matches the target armature current. Thus, according to the third aspect of the present invention, the influence of the current flowing through the capacitive filter is compensated regardless of the occurrence of the resonance phenomenon between the capacitive filter and the armature winding, and the target output torque is Thus, an appropriate current (the target armature current) can be passed through the armature winding.

  In the present invention, an inductive filter may be connected in series to the armature winding for each phase of both stators.

  In the second invention or the third invention, the target armature current or the inverter target output current may be a target value of a current for each phase, but a so-called d-axis current (field field in dq vector control). Magnetic current) and q-axis current (torque current) target values may be used.

  A first embodiment of the present invention will be described with reference to FIGS. FIG. 1 is a block diagram showing the overall configuration of the motor control device of the present embodiment.

  With reference to FIG. 1, reference numeral 1 denotes an electric motor, and reference numeral 2 denotes a control device that performs operation control of the electric motor 1.

  The electric motor 1 is an axial gap type electric motor including a rotor 3, two stators 4, 5, and armature windings 6, 7 attached to the stators 4, 5. The electric motor 1 is mounted on the vehicle as a driving force generation source of an electric vehicle or a hybrid vehicle, for example, and can perform a power running operation and a regenerative operation.

  The structure of the rotor 3 and the stators 4 and 5 of the electric motor 1 will be described in more detail with reference to FIGS. 2 is a perspective view of the rotor 3 and the stators 4 and 5 of the electric motor 1, FIG. 3 is an exploded perspective view of the rotor 3, and FIG. 4 is a diagram showing an arrangement of magnetic poles when the rotor 3 is viewed in the axial direction. .

  The rotor 3 has a substantially annular shape, and as shown in FIG. 3, an annular main permanent magnet layer 9 having a plurality of fan-shaped main permanent magnets 8 and a plurality of rectangular plate-like sub-magnets. An annular first auxiliary permanent magnet layer 11 having a permanent magnet 10 and an annular second auxiliary permanent magnet layer 13 having a plurality of rectangular plate-like auxiliary permanent magnets 12 are both sub permanent permanent magnet layers 11. , 13 are attached to the outer circumferential surface and the inner circumferential surface of the annular magnetic structure 14 having a structure in which the main permanent magnet layer 9 is sandwiched between the magnetic cores 14 and 13, respectively. The outer cylinder frame 15 and the inner cylinder frame 16 are comprised. In the present embodiment, the number of main permanent magnets 8, sub permanent magnets 10, and sub permanent magnets 12 is the same (12 in the illustrated example). In addition, the outer cylinder frame 15a and the inner cylinder frame 15b include a plurality of (the main permanent magnet 8 and the sub permanent magnet 10) that extend radially between the inner circumferential surface of the outer cylinder frame 15a and the outer circumferential surface of the inner cylinder frame 15b. , And the same number as each of the secondary permanent magnets 12) are connected coaxially by the connecting portions 16, and the connecting portions 16 and the non-magnetic material are integrally formed.

  The plurality of fan-plate-shaped main permanent magnets 8 of the main permanent magnet layer 9 are arranged so as to be arranged in a ring shape in the circumferential direction of the rotor 3 with their thickness directions directed in the direction of the axis C of the rotor 3. Thus, the main permanent magnet layer 9 is formed.

  The plurality of rectangular plate-like sub permanent magnets 10 of the first sub permanent magnet layer 11 has a thickness direction in the circumferential direction of the rotor 3 (more specifically, a circle centered on a point on the axis C of the rotor 3). (Circumferential tangential direction) are arranged at equal angular intervals in the circumferential direction. Accordingly, when viewed in the direction of the axis C of the rotor 3, these secondary permanent magnets 10 are arranged radially as shown in FIG. In this case, the position (angular position) of each sub permanent magnet 10 in the circumferential direction of the rotor 3 is a position between the main permanent magnets 8 and 8 adjacent to each other in the circumferential direction of the rotor 3. In other words, the secondary permanent magnets 10 are arranged so that the secondary permanent magnets 10 are positioned at both ends of each primary permanent magnet 8 in the circumferential direction of the rotor 3.

  Further, the same shape as each main permanent magnet 8 is provided at a position between the sub permanent magnets 10 and 10 adjacent to each other in the circumferential direction of the rotor 3 (a position facing each main permanent magnet 8 in the thickness direction). A magnetic yoke 17 (fan-shaped) is mounted with its thickness direction directed toward the axis C of the rotor 3. Accordingly, in the first secondary permanent magnet layer 11, the secondary permanent magnets 10 and the magnetic yokes 17 are arranged so as to be alternately arranged in the circumferential direction of the rotor 3, and the first secondary permanent magnet layer 11 is configured by this arrangement. . The magnetic yokes 17 are fixed to the secondary permanent magnets 10 and 10 on both sides and superposed on the primary permanent magnet 8 between the secondary permanent magnets 10 and 10.

  The plurality of rectangular plate-like sub permanent magnets 12 of the second sub permanent magnet layer 13, similarly to the sub permanent magnet 10 of the first sub permanent permanent magnet layer 11, have their thickness directions directed in the circumferential direction of the rotor 3, They are arranged so as to be arranged at equiangular intervals in the circumferential direction. In this case, the position (angular position) of each secondary permanent magnet 12 in the circumferential direction of the rotor 3 is the same as that of the secondary permanent magnet 10 of the first secondary permanent magnet layer 11. Accordingly, the individual secondary permanent magnets 10 of the first secondary permanent magnet layer 11 and the individual secondary permanent magnets 12 of the second secondary permanent magnet layer 13 are arranged in pairs in the direction of the axis C of the rotor 3.

  Similarly to the first sub permanent magnet layer 11, each of the portions between the sub permanent magnets 12 and 12 adjacent to each other in the circumferential direction of the rotor 3 has the same shape as each main permanent magnet 8 (fan shape). The magnetic yoke 18 is mounted with its thickness direction directed toward the axial center C of the rotor 3. Accordingly, also in the second sub permanent magnet layer 13, the sub permanent magnets 12 and the magnetic yokes 18 are arranged so as to be alternately arranged in the circumferential direction of the rotor 3, and the second sub permanent magnet layer 13 is configured by this arrangement. . The magnetic yokes 18 are fixed to the secondary permanent magnets 12 and 12 on both sides and superposed on the primary permanent magnet 8 between the secondary permanent magnets 12 and 12.

  Thus, the magnetic structure 14 is configured by assembling the main permanent magnet 8, the sub permanent magnets 10 and 12, and the magnetic yokes 17 and 18 to each other. Then, the outer cylinder frame 15 is coaxially inserted into the magnetic structure 14 and fixed to the outer peripheral surface of the magnetic structure 14, and the inner cylinder frame 16 is coaxially inserted into the magnetic structure 14. The rotor 3 is configured by being fixed to the inner peripheral surface of the magnetic structure 14. In this case, each of the main permanent magnets 8 is positioned in a space between the connecting portions 16 and 16 adjacent to each other in the circumferential direction of the rotor 3, and the auxiliary permanent magnets 10 and 12 aligned in the axial direction of the rotor 3. The magnetic structure 14, the outer cylinder frame 15a, and the inner cylinder frame 15b are assembled so that the connecting portions 16 are sandwiched therebetween. The lengths of the outer cylinder frame 15 and the inner cylinder frame 16 in the axis C direction are substantially the same as the length (thickness) of the magnetic structure 14 in the axis C direction. The rotor 3 has a plane-symmetric structure with respect to a plane orthogonal to the axis C through the center in the direction of the axis C.

  Further, the magnetic poles of the main permanent magnets 8 and the sub permanent magnets 10 and 12 will be described. Each main permanent magnet 8 is magnetized in the thickness direction, and one surface in the thickness direction is an N pole and the other surface is an S pole. The main permanent magnets 8 and 8 adjacent to each other in the circumferential direction of the rotor 3 have their magnetization directions opposite to each other. Therefore, the arrangement of the magnetic poles of the main permanent magnet 8 on each surface of the main permanent magnet layer 9 in the direction of the axis C of the rotor 3, for example, the surface on the first sub permanent magnet layer 11 side is as shown in FIG. In the circumferential direction of the rotor 3, N poles and S poles are alternately arranged in the circumferential direction of the rotor 3. The same applies to the surface of the main permanent magnet layer 9 on the second sub permanent magnet layer 13 side. FIG. 4 shows the first sub permanent magnet layer 11 with the magnetic yoke 17 removed.

  Further, each sub permanent magnet 10 of the first sub permanent magnet layer 11 and each sub permanent magnet 12 of the second sub permanent magnet layer 13 are also magnetized in the thickness direction (the circumferential direction of the rotor 3), and the thickness thereof. One surface in the direction is the N pole, and the other surface is the S pole. In this case, as shown in FIG. 4 (see “N” and “S” without white squares), the magnetic poles on each surface in the thickness direction of each sub permanent magnet 10 are on the surface side. It is the same as the magnetic pole on the surface of the main permanent magnet 8 adjacent to the permanent magnet 10 on the first sub permanent magnet layer 11 side. Therefore, the magnetic poles of the mutually opposing surfaces of the secondary permanent magnets 10 and 10 adjacent to each other in the circumferential direction of the rotor 3 are the first secondary permanent magnet layer 11 side of the primary permanent magnet 8 between the secondary permanent magnets 10 and 10. It is the same as the magnetic pole on the surface. The same applies to the secondary permanent magnet 12 of the second secondary permanent magnet layer 13. In this case, since the magnetic poles on both surfaces in the thickness direction of each main permanent magnet 8 are different from each other, the magnetization directions of the sub permanent magnets 10 and 12 aligned in the direction of the axis C of the rotor 3 are opposite to each other.

  Thus, the rotor 3 having the main permanent magnet 8 and the sub permanent magnets 10 and 12 has a so-called Halbach type magnetic pole arrangement, and the magnetic flux density in the direction of the axis C of the rotor 3 is increased due to the Halbach effect. Can be.

  Each of the stators 4 and 5 has the same structure. As shown in FIG. 2, each of the stators 4 and 5 protrudes in the axial direction of the base body 19 from one surface of the annular base body 19 in the axial direction. The teeth 20 are arranged at equiangular intervals around the axis of the base body 19. The base 19 and the teeth 20 are integrally formed of a magnetic material. In the illustrated example, the number of teeth 20 of each of the stators 4 and 5 is 36.

  In each of the stators 4 and 5, armature windings 6 and 7 (not shown in FIG. 2) are accommodated in slots 21 which are grooves between adjacent teeth 20 and 20 in the circumferential direction. Armature windings 6 and 7 are attached. In the present embodiment, the armature windings 6 and 7 attached to the stators 4 and 5 are armature windings for three phases (U phase, V phase, W phase). Further, the mounting form of the armature windings 6 and 7 in each of the stators 4 and 5 is the same. For example, the armature winding 6 of each phase on the stator 4 side has a predetermined number of winding loops formed at equal angular intervals in the circumferential direction of the stator 4 when viewed in the axial direction of the stator 4. Mounted on the stator 4. The same applies to the armature winding 7 on the stator 5 side. Therefore, in the present embodiment, the configurations of the stator 4 and the armature winding 6 and the configurations of the stator 5 and the armature winding 7 are the same as each other.

  The stators 4 and 5 are arranged coaxially with the rotor 3 on both sides in the axial center C direction of the rotor 3 so that the rotor 3 is sandwiched between the stators 4 and 5 when the electric motor 1 is assembled. The motor 1 is fixed to a housing (not shown). In this case, the front end surfaces of the teeth 20 of the stators 4 and 5 are opposed to the rotor 3 in close proximity. Further, the gap between the rotor 3 and the stator 4 in the direction of the axis C of the rotor 3 (the distance between the surface of the rotor 3 on the side of the stator 4 and the tip surface of the teeth 20 of the stator 4), and the rotor and the stator 5 The gap between them (the distance between the stator 5 side surface of the rotor 3 and the tip surface of the teeth 20 of the stator 5) is equal, that is, the rotor 3 is positioned at the center between the stators 4 and 5. Further, the rotor 3 and the stators 4 and 5 are assembled to the electric motor 1. Further, in the present embodiment, when viewed in the axial direction of the rotor 3 in the assembled state of the electric motor 1, the positions of the teeth 20 of the stator 4 (angular positions around the axis) and the positions of the teeth 20 of the stator 5. The stators 4 and 4 are assembled to the electric motor 1 so that (angular position around the axis) matches. That is, the individual teeth 20 of the stator 4 and the individual teeth 20 of the stator 5 face each other in the direction of the axis C of the rotor 3. Then, the armature winding 6 of each phase of the stator 4 and the armature winding 7 of the stator 5 of the same phase as that of the stator 4 are, for each phase, a winding loop of the armature winding 6 on the stator 4 side. The winding loop of the armature winding 7 on the stator 5 side faces each other in the axial center C direction of the rotor 3 (when viewed in the axial center C direction of the rotor 3, the winding loop on the stator 4 side and the stator Each of the stators 4 and 5 is mounted so that the winding loop on the 5 side is at the same angular position. Accordingly, the magnetic circuit configuration from the rotor 3 to the stator 4 side of the electric motor 1 is the same as the magnetic circuit configuration from the rotor 3 to the stator 5 side.

  Further, the electric motor 1 is provided with an output shaft 1a (see a virtual line in FIG. 2) connected to the rotor 3, and the output shaft 1a passes through the stators 4 and 5 as shown in FIG. Thus, it is attached coaxially to the inner cylinder frame 16 of the rotor 3. The output shaft 1a is supported by the housing of the electric motor 1 (or the base 19 of the stators 4 and 5) via a bearing (not shown), and is rotatable together with the rotor 3.

  In the following description, when it is necessary to distinguish the phases (U phase, V phase, W phase) of the armature windings 6 and 7 or the components corresponding to the respective phases, subscripts are used respectively. u, v, and w are attached. For example, the U-phase, V-phase, and W-phase armature windings 6 of the stator 4 are denoted as an armature winding 6u, an armature winding 6v, and an armature winding 6w, respectively. When there is no need to distinguish each phase, the subscripts u, v, and w are often omitted.

  Returning to the description of FIG. 1, in the present embodiment, the armature winding 6 on the stator 4 side and the armature winding 7 on the stator 5 side are connected as follows. That is, each of the armature windings 6u, 6v, 6w of each phase of the stator 4 and each of the armature windings 7u, 7v, 7w of each phase of the stator 5 are connected in series for each phase. . The one end of each phase of the armature windings 7u, 7v, 7w of the stator 5 opposite to the armature windings 6u, 6v, 6w is the armature windings 6, 7 of both stators 4, 5. Are connected to a common (single) neutral point 1N. Therefore, the U-phase armature windings 6u and 7u connected in series, the V-phase armature windings 6v and 7v connected in series, and the W-phase armature windings 6w and 7w in series. Are connected by Y connection. One end of each side of the armature windings 6 u, 6 v, 6 w of each phase of the stator 4 opposite to the armature windings 7 u, 7 v, 7 w is for three phases provided on the outer surface of the motor 1. Are connected to the energization connection terminals 42u, 42v and 42w, respectively. Therefore, the armature winding 6 of the stator 4 and the armature winding 7 of the stator 5 are connected in series between the energizing connection terminal 42 and the single neutral point 1N for each phase. . The energization connection terminals 42u, 42v, and 42w are connected to an inverter circuit 38 of a power drive unit 31 (to be described later) of the control device 2 through energization lines (connection cables, bar-like conductors, etc.) 23u, 23v, and 23w, respectively. A current flows between the inverter circuit 38 and the armature windings 6 and 7 of each phase. Since the armature windings 6 and 7 are connected in series for each phase, the current flowing through the armature winding 6 and the current flowing through the armature winding 7 are always the same for each phase. Become.

  In the present embodiment, a resolver 24 serving as a rotation angle detection unit that detects the rotation angle of the rotor 3 is attached to the motor 1 for operation control of the motor 1.

  The above is the configuration related to the electric motor 1.

  Next, the control device 2 will be described with reference to FIGS. 1, 5 and 6. 5 is a diagram showing a general circuit configuration of the capacitive filter 41 provided in the control device 2, and FIG. 6 is a diagram showing an inverter circuit 38 of the power drive unit 31 provided in the control device 2 and a circuit configuration on the output side thereof. It is.

  Referring to FIG. 1, the control device 2 is configured by an electronic circuit unit including a microcomputer. The control device 2 is roughly divided into functional configurations, and a power drive unit 31 (hereinafter referred to as a PDU 31) to which the energization lines 23u, 23v, 23w for three phases are connected, and a control for controlling the operation of the PDU 31. PDU control unit 32 for executing processing, current sensors 33u, 33v, 33w for detecting output currents from PDU 31 to energization lines 23u, 23v, 23w of the respective phases, and outputs of current sensors 33u, 33v, 33w A band-pass characteristic filtering is performed to remove noise components to obtain the output current detection values Isu_s, Isv_s, Isw_s of each phase of the PDU 31, and the armature windings 6, 7 of each phase of the motor 1. Fill for preventing a high-frequency component current from flowing due to a switching operation of an inverter circuit 28 to be described later And a circuit 35. The output current detection values Isu_s, Isv_s, Isw_s are input from the BP filter 34 to the PDU control unit 32, and the rotation angle θm_s of the rotor 3 detected by the resolver 24 (hereinafter referred to as the rotor angle detection value θm_s). Is entered.

  Supplementally, since the sum of the output current detection values Isu_s, Isv_s, Isw_s is 0, any one of the current sensors 33u, 33v, 33w is omitted, and the output current detection value corresponding to the omitted current sensor. May be calculated from the other two detected output current values. For example, when the current sensor 33v is omitted, Isv_s can be calculated as Isv_s = −Isu_s−Isw_s.

  As shown in FIG. 6, the PDU 31 is formed by connecting in parallel three-phase arms 37u, 37v, and 37w each having a pair of switch elements 36 and 36 (IGBT in the illustrated example) connected in series. An inverter circuit 38 is provided. In the inverter circuit 38, a direct current voltage is applied from a direct current power supply (not shown) to a pair of input terminals 39a and 39b respectively connected to both ends of each arm 37. The inverter circuit 38 includes output terminals 40u, 40v, and 40w that are electrically connected to the midpoints (locations between the switch elements 36 and 36) of the arms 37u, 37v, and 37w of the respective phases. The energization lines 23u, 23v, and 23w are connected to 40u, 40v, and 40w, respectively. The PDU 31 having the inverter circuit 38 is connected to each switch of the inverter circuit 38 by PWM control according to an operation command (voltage command value of each phase of U phase, V phase, and W phase) given from the PDU control unit 32. The output current of the inverter circuit 38 is controlled by controlling on / off of the element 36. The output current of the PDU 31 detected by each current sensor 33 is an output current for each phase of the inverter circuit 38.

  The filter circuit 35 is a filter having a high-pass characteristic, and includes three-phase capacitive filters 41u, 41v, and 41w. Here, each capacitive filter 41 is a filter in which the imaginary part (reactance component) of each impedance is a negative value, and each impedance is substantially the same. In the present embodiment, each capacitive filter 41 is configured by a capacitor (capacitance element).

  However, each capacitive filter 41 does not necessarily need to be configured only by a capacitor. For example, as shown in FIG. 5, the capacitive filter 41 is configured by combining a resistance element 43a, a coil 43b (inductance element), and capacitors 43c and 43d. Also good. Thus, the circuit configuration of each capacitive filter 41 can have various forms.

  One end of each of the capacitive filters 41u, 41v, 41w is electrically connected to the neutral point 1N of the armature windings 6, 7 and connected to each other. That is, the capacitive filters 41u, 41v, 41w are connected by Y connection, and the neutral point of the Y connection is conducted to the neutral point 1N of the armature windings 6, 7. Further, the other ends of the capacitive filters 41u, 41v, 41w are respectively connected between the current sensors 33u, 33v, 33w and the electric motor 1, the U-phase energization line 23u, the V-phase energization line 23v, and the W-phase energization. It is connected to the line 23w. Accordingly, each of the capacitive filters 41u, 41v, 41w includes a series circuit of U-phase armature windings 6u, 7u, a series circuit of V-phase armature windings 6v, 7v, and a W-phase armature winding 6w, respectively. , 7w series circuit connected in parallel.

  In the present embodiment, each capacitive filter 41 is connected to the energization line 23 between the current sensor 33 and the electric motor 1 for each phase, so that the inverter circuit 38 detected by each current sensor 33 The output current is a combined current of the current flowing through the corresponding armature windings 6 and 7 and the current flowing through the capacitive filter 41 of that phase.

  Supplementally, the capacitive filter 41 and the current sensor 33 may be integrated with the inverter circuit 38 of the PDU 31.

  The PDU control unit 32 controls the energization current (phase current) of the armature windings 6 and 7 of each phase via the PDU 31. In the present embodiment, the PDU control unit 32 controls the phase currents of the armature windings 6 and 7 of the respective phases of the stators 4 and 5 of the electric motor 1 by so-called dq vector control. That is, the PDU control unit 32 combines the three-phase armature windings 6u, 6v, 6w of the stator 4 with the three-phase armature windings 7u, 7v, 7w of the stator 5, It is handled by converting it to an equivalent circuit in the dq coordinate system. The equivalent circuit has an armature winding on the d-axis (hereinafter referred to as d-axis armature winding) and an armature winding on the q-axis (hereinafter referred to as q-axis armature winding). In the dq vector control, the currents that flow through the armature windings 6 and 7 are the d-axis current that flows through the d-axis armature winding and the q-axis that flows through the q-axis armature winding. Treated as a combined current with current. The dq coordinate system rotates integrally with the rotor 3 of the electric motor 1 with the field direction by the main permanent magnet 8 and the sub permanent magnets 10 and 12 of the rotor 3 as the d axis and the direction orthogonal to the d axis as the q axis. This is a rotating coordinate system. Accordingly, the d-axis current and the q-axis current are currents that mean so-called field current and torque current, respectively.

  The PDU control unit 32 then causes the current flowing through each capacitive filter 40 so that the torque of the torque command value Tr_c given to the control device 2 from the outside as the target output torque of the electric motor 1 is output from the output shaft 1 a of the electric motor 1. The phase currents of the armature windings 6 and 7 of the stators 4 and 5 of the electric motor 1 are controlled via the PDU 31 while compensating for the effects of the above.

  The PDU control unit 32 that performs such control has, as its functional configuration, a rotor speed detection value ωm_s as a rotation speed detection value of the rotor 3 by differentiating the rotor angle detection value θm_s input from the resolver 24. Differential calculation unit 50 to calculate, d-axis current command value Imd_c which is a command value of d-axis current (field current) of armature windings 6 and 7, and q which is a command value of q-axis current (torque current) A current command determination unit 51 that determines the shaft current command value Imq_c, and the d-axis current command value Imd_c and the q-axis current command value Imq_c are armature currents that are complex expressions of the current command values of the armature windings 6 and 7. A complex current command conversion unit 52 for converting to a vector command value Im_co_c, and an inverter output current vector command value Is_co_c which is a complex expression of the command value of the output current of the inverter circuit 38 from the phase current vector command value Im_co_c and the like. The inverter output current command calculation unit 53 to be output and the inverter output current vector command value Is_co_c as the respective command values of the d-axis current component and the q-axis current component of the output current of the inverter circuit 38, are the inverter d-axis current command value Isd_c. And a dq command conversion unit 54 for converting into a set of inverter q-axis current command values Isq_c.

  The differential calculation unit 50 may calculate the angular velocity of the electrical angle of the rotor 3 by differentiating the electrical angle of the rotor 3 obtained by multiplying the detected rotor angle value θm_s by the number of pole pairs of the rotor 3.

  Supplementally, in the following description, reference numerals having the suffix “co”, such as the armature current vector command value Im_co_c and the inverter output current vector command value Is_co_c, mean a vector quantity in a complex expression. Shall.

  Further, the PDU control unit 32 uses the phase current detection values Isu_s, Isv_s, Isd_c input from the BP filter 34 as d detection values of the d-axis current component and the q-axis current component of the output current of the inverter circuit 38, respectively. A dq converter 55 for converting the detected shaft current value Isd_s and the detected q-axis current value Isq_s into a set, and the detected d-axis current value Isd_s and the detected q-axis current value Isq_s, respectively, the inverter d-axis current command value Isd_c and the inverter q Currents that determine the d-axis voltage command value Vsd_c and the q-axis voltage command value Vsq_c as the command values of the voltages of the d-axis armature winding and the q-axis armature winding so as to coincide with the shaft current command value Isq_c A set of the feedback control unit 56 and the d-axis voltage command value Vsd_c and the q-axis voltage command value Vsq_c is converted into voltage command values Vsu_c, Vsv_c, and Vsw_c for each phase of the U phase, V phase, and W phase. And a three-phase conversion unit 57. The voltage command values Vsu_c, Vsv_c, and Vsw_c are operation command values for the PDU 31.

  The processing of each functional unit of the PDU control unit 32 is sequentially executed as described below at a predetermined control processing cycle.

  The torque command value Tr_c of the electric motor 1 given from the outside to the control device 2 is sequentially input to the current command determination unit 51, and the rotor speed detection value ωm_s is sequentially input from the differentiation calculation unit 50. Then, the current command determination unit 51 sequentially determines the d-axis current command value Imd_c and the q-axis current command value Imq_c of the armature windings 6 and 7 from these input values according to a predetermined map. That is, Imd_c and Imq_c are feedforward command values determined according to the torque command value Tr_c and the rotor speed detection value ωm_s. In this case, basically, the q-axis current command value Imq_c is determined to be a value proportional to the torque command value Tr_c. Further, the d-axis current command value Imd_c has a magnitude of a combined vector of the d-axis voltage and the q-axis voltage determined according to the q-axis current command value Imq_c, the d-axis current command value Imd_c, and the rotor rotational speed ωm_s. It is determined so as not to exceed a predetermined value determined corresponding to one power supply voltage (input voltage to the inverter circuit 38). The d-axis current command value Imd_c basically causes a field current component in the field weakening direction to flow through the armature windings 6 and 7 in the operating region of the electric motor 1 where the rotor speed detection value ωm_s becomes high. Is the command value determined by

  The current command determining unit 51 corresponds to the target armature current determining means in the present invention, and the q-axis current command value Imq_c and the d-axis current command value Imd_c determined there correspond to the target armature current in the present invention. To do.

  The d-axis current command value Imd_c and the q-axis current command value Imq_c determined in this way are sequentially input to the complex current command conversion unit 52. In the present embodiment, it is assumed that the q axis and d axis of the dq coordinate system coincide with the real axis and imaginary axis of the complex coordinate system, respectively. Therefore, the complex current converter 52 determines a complex number having the input q-axis current command value Imq_c as a real part and the d-axis current command value Imd_c as an imaginary part as an armature current vector command value Im_co_c. That is, Im_co_c is determined by the following equation (1).

Im_co_c = Imq_c + j · Imd_c (1)
(Where j is the imaginary unit)

Im_co_c may be expressed in polar coordinates. In this case, Im_co_c is a vector quantity having √ (Imq_c ² + Imd_c ² ) as a radial component and tan ⁻¹ (Imd_c / Imq_c) as an angular component.

  The armature current vector command value Im_co_c determined in this way and the detected rotor speed value ωm_s are input to the inverter output current command calculation unit 53.

  Here, in this embodiment, as described above, the output current of the inverter circuit 38 detected by each current sensor 33 includes the current flowing through the corresponding armature windings 6 and 7 and the capacity of the phase. The combined current is combined with the current flowing through the filter 41. Therefore, in order to pass the current of the armature current vector command value Im_co_c (currents such that the d-axis current and the q-axis current match Imd_c and Imq_c, respectively) through the armature windings 6 and 7 of each phase, the armature It is necessary to output from the inverter circuit 38 a current obtained by adding the current flowing through the capacitive filter 41 to the current of the current vector command value Im_co_c. The inverter output current command calculation unit 53 converts the complex expression of the command value (command value of the output current for one phase) of the inverter circuit 38 to which the current flowing through the capacitive filter 41 is added in this way into the inverter output current. This is obtained as the vector command value Is_co_c.

  The processing of the inverter output current command calculation unit 53 will be described in detail with reference to FIG. Here, in the following description, the series circuit of the armature windings 6 and 7 of each phase is equivalent to a coil 70 having an inductance L, a resistance element 71 having a resistance value r, and a rotor 3 as shown in FIG. It is assumed that it is represented by a series circuit with a voltage source 72 that generates an induced voltage vector Vi_co proportional to the rotational speed. In this case, when the induced voltage vector Vi_co is the induced voltage constant of the electric motor 1 (total induced voltage constant including the armature windings 6 and 7) Ke when the angular velocity of the electrical angle of the rotor 3 is ω, FIG. As described in the above, Vi_co = Ke · ω. The induced voltage vector Vi_co is generated in the q-axis direction of the dq coordinate system. Therefore, when the q-axis and the d-axis coincide with the real axis and the imaginary axis of the complex coordinate system, respectively, The induced voltage Vi_co is a vector quantity having only a real part component (= Ke · ω). Supplementally, the inductance L is the average value of the inductance Ld of the d-axis armature winding and the inductance Lq of the q-axis armature winding when the armature windings 6 and 7 are handled in the dq coordinate system (= (Ld + Lq) / 2).

In general, an inductive filter (a filter in which the reactance component of the impedance has a positive value) connected in series to the armature winding of each phase may be incorporated in the electric motor. Therefore, in FIG. 6, in order to generalize the description of the processing of the inverter output current command calculation unit 53, inductivity expressed by a series circuit of a coil 73 having an inductance L ₂ and a resistance element 74 having a resistance value R ₂ . It is assumed that the filter 75 is connected in series to the series circuit of the armature windings 6 and 7 of each phase. However, the inductive filter 75 may be omitted. In this case, L ₂ = 0 and R ₂ = 0 may be set.

  In addition, the impedance of each capacitive filter 41 is set to Ze_co. In this case, as described in FIG. 6, the imaginary part Im (Ze_co) of the impedance Ze_co, that is, the reactance component is a negative value. In the present embodiment, each capacitive filter 41 is constituted by a capacitor. Therefore, if the capacitance value is C, Ze_co = 1 / (j · ω · C). However, the circuit configuration of the capacitive filter 41 has various forms as described above. Therefore, in FIG. 6, in order to generalize the description of the processing of the inverter output current command calculation unit 53, each capacitive filter 41 is expressed by a simple white square. In this case, Ze_co generally has a resistance component and a reactance component (<0).

  Based on the above, the processing of the inverter output current command calculation unit 53 will be described below. Focusing on any one of the U-phase, V-phase, and W-phase, for example, the U-phase, the output current vector of the U-phase of the inverter circuit 38 is Is_co, and the current vector flowing in the U-phase capacitive filter 41u is Ie_co , The current vector flowing in the U-phase armature windings 6 and 7 (hereinafter referred to as the armature current vector) is generated in Im_co, the entire series circuit of the U-phase armature windings 6u and 7u and the inductive filter 75. Let Vm_co be the voltage vector to be used. The voltage vector Vm_co is the same as the voltage vector generated in the U-phase capacitive filter 41. Further, the positive directions of Is_co, Ie_co, and Im_co are assumed to be directions of arrows in the drawing.

Also, the impedance of the U-phase armature windings 6u, 7u and the inductive filter 75 is Zm_co. In this case, if Zm_co≡Rm + j · ω · Lm, then Rm = R ₂ + r and Lm = L ₂ + L. More generally speaking, the impedance Zm_co means the impedance of the current path from one end of the capacitive filter 41 of each phase on the side of the energization line 23 to the neutral point 1N. In this case, if the resistance value of the energization line 23 is so large that it cannot be ignored, the resistance component of the energization line 23 may be included in the impedance Zm_co.

  At this time, as will be understood with reference to FIG. 6, the following expressions (2), (3), and (4) are established.

Is_co = Ie_co + Im_co (2)
Vm_co = Zm_co ・ Im_co + Vi_co (3)
Ie_co = Vm_co / Ze_co (4)

From these formulas (2) to (4), the following formula (5) is obtained.

Is_co = ((Zm_co · Im_co + Vi_co) / Ze_co) + Im_co
= Im_co · (1+ (Zm_co / Ze_co)) + (Vi_co / Ze_co) (5)

Furthermore, since Vi_co = Ke · ω as described above, the equation (5) becomes the following equation (6).

Is_co = Im_co · (1+ (Zm_co / Ze_co)) + (Ke · ω / Ze_co) (6)

These equations (5) and (6) are the impedances Ze_co and Zm_co for one phase, the induced voltage Vi_co (= Ke · ω) of the armature windings 6 and 7, the output current vector Is_co, This is an arithmetic expression representing the relationship with the child current vector Im_co.

  Here, the impedance Ze_co of the capacitive filter 41 is determined according to the circuit constant values (capacitance value, inductance value, resistance value) of the elements constituting the capacitive filter 41 and the angular velocity ω of the electrical angle of the rotor 3. In this case, since the circuit constant value of the element constituting the capacitive filter 41 can be a known value, Ze_co is uniquely determined according to the angular velocity ω of the rotor 3.

Similarly, the impedance Zm_co on the side of the motor 1 is the circuit constant value (values of R ₂ , L ₂ , r, L) of the series circuit of the inductive filter 75 and the armature windings 6 and 7 and the electrical angle of the rotor 3. It is determined according to the angular velocity ω. Since the circuit constant value can be a known value, Zm_co is uniquely determined according to the angular velocity ω of the rotor 3.

  Further, the induced voltage constant Ke of the electric motor 1 can be set to a known value by actual measurement or the like.

  Therefore, if the angular velocity ω of the rotor 3 and the armature current vector Im_co are determined, the output current vector Is_co of the inverter circuit 38 is uniquely determined by the equation (6).

  Therefore, the inverter output current calculation unit 53 calculates the inverter output current vector command value Is_co_c from the input armature current vector command value Im_co_c and the rotor speed detection value ωm_s based on the above equation (6).

  More specifically, since the rotor speed detection value ωm_s is the angular speed of the mechanical angle of the rotor 3, a value obtained by multiplying ωm_s by the number of pole pairs of the rotor 3 (this is the detected value of the angular speed of the electrical angle of the rotor 3). Is used as the value of ω on the right side of Equation (6). The armature current vector command value Im_co_c is used as the value of Im_co on the right side of the equation (6). Then, inverter output current calculation unit 53 calculates Is_co obtained by performing the calculation of the right side of equation (6) in this way as inverter output current vector command value Is_co_c.

  In this case, as a circuit constant value necessary for determining the impedances Ze_co and Zm_co, a value stored in a memory (not shown) in advance is used.

  The details of the processing of the inverter output current calculation unit 53 have been described above.

  In the description of the processing of the inverter output current calculation unit 53 described above, a generalized description has been given in consideration of the case where the inductive filter 75 is provided and the case where each capacitive filter 41 includes an element other than a capacitor. In the embodiment, the inductive filter 75 is actually omitted. Moreover, each capacitive filter 41 is comprised with the capacitor | condenser in this embodiment. In this case, the impedance Zm_co closer to the electric motor 1 than the capacitive filter 41 is Zm_co = r + j · ω · L. Further, as described above, the impedance Ze_co of each capacitive filter 41 is Ze_co = 1 / (j · ω · C).

  The inverter output current vector command value Is_co_c calculated by the inverter output current calculation unit 53 in this way is input to the dq command conversion unit 54. The dq command conversion b unit 54 converts the input inverter output current vector command value Is_co_c into a set of an inverter d-axis current command value Isd_c and an inverter q-axis current command value Isq_c. In this case, in this embodiment, as described above, the real and imaginary axes of the complex coordinate system are made to coincide with the q and d axes of the dq coordinate system, respectively, so that the real and imaginary parts of Is_co_c are respectively converted to inverters. The q-axis current command value Isq_c and the inverter d-axis current command value Isd_c are used. That is, if it is expressed as Isq_c = Isq + j · Isd, then Isq_c = Isq and Isd_c = Isd.

  As a result, the d-axis current component and q of the output current of the inverter circuit 38 required for flowing currents corresponding to the d-axis current command value Imd_c and the q-axis current command value Imq_c to the armature windings 6 and 7 of the motor 1. The command value of the shaft current component is determined in consideration of the current flowing through the capacitive filter 41.

  The inverter output current calculation unit 53 and the dq command conversion unit 54 together constitute an inverter target output current determination unit in the present invention. The inverter d-axis current command value Isd_c and the inverter q-axis current command value Isq_c correspond to the inverter target output current in the present invention.

  On the other hand, in parallel with the processing of the current command determination unit 51 and the like, the output current detection values Isu_s, Isv_s, Isw_s and the rotor angle detection value θm_s are input to the dq conversion unit 55, and the dq conversion unit 55 Processing is executed. The dq conversion unit 55 performs coordinate conversion of the output current detection values Isu_s, Isv_s, Isw_s according to the electrical angle of the rotor 3 to calculate the d-axis current detection value Isd_s and the q-axis current detection value Isq_s. In this case, the electrical angle of the rotor 3 is obtained by multiplying the detected rotor angle value θm_s by the number of pole pairs of the rotor 3.

  Next, the inverter d-axis current command value Isd_c and the inverter q-axis current command value Isq_c obtained by the dq command conversion unit 54, and the d-axis current detection value Isd_s and the q-axis current detection value obtained by the dq conversion unit 55. Isq_s is input to the current feedback control unit 56. The current feedback control unit 56 also receives the rotor speed detection value ωm_s.

  In this current feedback control unit 56, first, the processing of the calculation unit 58 for obtaining a deviation ΔIsd (= Isd_c−Isd_s) between the inverter d-axis current command value Isd_c and the d-axis current detection value Isd_s, and the inverter q-axis current command value Isq_c And a process of the calculation unit 59 for obtaining a deviation ΔIsq (= Isq_c−Isq_s) between the q-axis current detection value Isq_s.

  Next, the basic command value Vsd1_c of the d-axis voltage and the q-axis voltage so that the deviations ΔIsd and ΔIsq are brought close to 0 from the respective deviations ΔIsd and ΔIsq by a PI control law (proportional / integral control law) as a feedback control law. Processing of the PI control units 60 and 61 for calculating the basic command value Vsq1_c of each, and the correction amount Vsd2_c of the d-axis voltage and the correction of the q-axis voltage for canceling the speed electromotive force that interferes with each other between the d-axis and the q-axis Processing of the non-interference control unit 62 for obtaining the amount Vsq2_c is executed. The non-interference control unit 62 calculates the d-axis side correction amount Vd2_c from the inverter q-axis current command value Isq_c and the rotor speed detection value ωm_s, and calculates the q-axis side correction amount Vq2_c to the inverter d-axis current command value Isd_c. And the rotor speed detection value ωm_s.

  Next, the correction amount Vsd2_c is added to the basic command value Vsd1_c of the d-axis voltage, so that the calculation unit 63 obtains the final d-axis voltage command value Vsd_c, and the correction amount is added to the basic command value Vsq1_c of the q-axis voltage. By adding Vsq2_c, the processing of the calculation unit 64 for obtaining the final q-axis voltage command value Vsq_c is executed.

  By the processing of the current feedback control unit 56 described above, the PDU 31 is configured so that the d-axis current detection value Isd_s and the q-axis current detection value Isq_s are made to coincide with the inverter d-axis current command value Isd_c and the inverter q-axis current command value Isq_c, respectively. Thus, the d-axis voltage command value Vsq_c and the q-axis voltage command value Vsq_c are determined as operation commands for.

  The current feedback control unit 56 corresponds to the inverter control means in the present invention.

  The d-axis voltage command value Vsd_c and the q-axis voltage command value Vsq_c determined by the current feedback control unit 56 in this way and the rotor angle detection value θm_s are input to the three-phase conversion unit 57. The three-phase conversion unit 57 converts the d-axis voltage command value Vd_c and the q-axis voltage command value Vq_c according to the electrical angle of the rotor 3 obtained by multiplying the rotor angle detection value θm_s by the number of pole pairs of the rotor 3. Thus, the voltage command values Vsu_c, Vsv_c, and Vsw_c (hereinafter referred to as phase voltage command values Vsu_c, Vsv_c, and Vsw_c) for each phase are calculated. The three-phase conversion unit 57 inputs the calculated phase voltage command values Vsu_c, Vsv_c, and Vsw_c to the PDU 31.

  At this time, the PDU 31 controls on / off of each switch element 36 of the inverter circuit 38 by PWM control so that the voltages of the input phase voltage command values Vsu_c, Vsv_c, Vsw_c are generated on the output side of the inverter circuit 38. . As a result, the output current of each phase of the inverter circuit 38 is obtained by converting the current command value (Isd_c and Isq_c) of each phase corresponding to the inverter d-axis current command value Isd_c and the inverter q-axis current command value Isq_c It is controlled so as to coincide with the current command value of each phase obtained by coordinate conversion into three phases according to the angle. At this time, the inverter d-axis current command value Isd_c and the inverter q-axis current command value Isq_c take into account the current flowing through each capacitive filter 41. As a result, the armature 6 of each phase of the motor 1 is obtained. , 7 are controlled so as to coincide with the current command values of the respective phases corresponding to the d-axis current command value Isd_c and the q-axis current command value Isq_c determined by the current command determination unit 51. . As a result, the operation control of the electric motor 1 is performed so that the torque of the torque command value Tr_c is generated in the output shaft 1a of the electric motor 1.

  According to the first embodiment described above, the high-frequency current component included in the output current of the inverter circuit 38 flows through each capacitive filter 41, and thus the high-frequency current component flows through the armature windings 6 and 7. Can be prevented. In addition, since one end of each capacitive filter 41 connected to the neutral point 1N of the armature windings 6 and 7 is in the same potential as the neutral point 1N, the capacitive filter Suppressing the occurrence of an unbalanced offset between the currents flowing through the armature windings 6 and 7 due to the resonance phenomenon between the armature windings 6 and 7 and the armature windings 6 and 7 can do.

  For this reason, the copper loss and iron loss in the electric motor 1 can be reduced, and the energy efficiency of the electric motor 1 can be improved. Further, since it is possible to suppress the occurrence of an unbalanced offset between the currents flowing through the armature windings 6 and 7, the number of permanent magnets 8, 10, 12 provided in the rotor 3 can be reduced. Magnetism can be prevented. Further, since the output voltage of the inverter circuit 38 can be applied to the armature windings 6 and 7 of the electric motor 1 without being lowered by the capacitive filter 41, the maximum value of the output that can be generated by the electric motor 1 is prevented from being lowered. can do.

  Further, the inverter d-axis current command value Isd_c and the inverter q-axis current command value Isq_c as the command values of the output current of the inverter circuit 38 are determined in consideration of the current flowing through each capacitive filter 41, and these inverters d The output current of the inverter circuit 38 is feedback-controlled so that the d-axis current detection value Isd_s and the q-axis current detection value Isq_s coincide with the shaft current command value Isd_c and the inverter q-axis current command value Isq_c, respectively. Accordingly, the torque of the torque command value Tr_c is stably supplied to the electric motor 1 without depending on the current flowing through each capacitive filter 41 due to a resonance phenomenon between the capacitive filter 41 and the armature windings 6 and 7. Can be generated.

  Next, a second embodiment of the present invention will be described with reference to FIG. FIG. 7 is a block diagram showing the overall configuration of the motor control device of the present embodiment. In the description of the present embodiment, the same components as those in the first embodiment are denoted by the same reference numerals as those in the first embodiment, and detailed description thereof is omitted.

  Referring to FIG. 7, 1 is an electric motor and 80 is a control device. The electric motor 1 is the same as that of the first embodiment.

  The control device 80 is configured by an electronic circuit unit including a microcomputer and the like, like the control device 2 of the first embodiment. The control device 80 includes a power drive unit (PDU) 31, current sensors 33 u, 33 v, 33 w, a BP filter 34, and a filter circuit 35, which are the same as those in the first embodiment, as the functional configuration. However, in the present embodiment, the other ends of the capacitive filters 41u, 41v, and 41w of the filter circuit 35 that are opposite to one end that is electrically connected to the neutral point 1N of the electric motor 1 are current sensors 33u, 33v, 33 w and PDU 31 are connected to energization lines 23 u, 23 v, 23 w, and this is different from the first embodiment. Therefore, in the present embodiment, the current detected by each current sensor 33 is the same current that flows through the corresponding armature windings 6 and 7. In other words, the current is a current obtained by subtracting the current flowing through the capacitive filter 41 corresponding to the phase from the output current of each phase of the inverter circuit 38 of the PDU 31. Hereinafter, the current detection values of the respective phases obtained from the outputs of the current sensors 33u, 33v, and 33w via the BP filter 34 in this embodiment are referred to as armature current detection values Imu_s, Imv_s, and Imw_s, respectively.

  In addition, the control device 80 includes a PDU control unit 81 that is partly different in control processing from that of the first embodiment. Similar to the PDU control unit 32 of the first embodiment, the torque command value Tr_c and the rotor angle detection value θm_s by the resolver 24 are input to the PDU control unit 81, while the output current detection in the first embodiment is performed. The armature current detection values Imu_s, Imv_s, Imw_s are input instead of the values Isu_s, Isv_s, Isw_s.

  The PDU control unit 81 includes a differential calculation unit 50, a current command determination unit 51, a dq conversion unit 55, a current feedback control unit 56, and a three-phase conversion unit 57 that execute the same processing as that of the first embodiment. . However, in the present embodiment, the armature current detection values Imu_s, Imv_s, and Imw_s are input to the dq conversion unit 55 instead of the output current detection values Isu_s, Isv_s, and Isw_s in the first embodiment. The current feedback control unit 56 also includes a d-axis current command value Imd_c determined by the current command determination unit 51 instead of the inverter d-axis current command value Isd_c and the inverter q-axis current command value Isq_c in the first embodiment. The q-axis current command value Imq_c is input as it is. Thus, in the PDU control unit 81 of the present embodiment, some input values for the dq conversion unit 55 and the current feedback control unit 56 are different from the PDU control unit 32 of the first embodiment. Other than this, the functional configuration of the PDU control unit 81 is the same as that of the PDU control unit 32 of the first embodiment.

  The processing of the PDU control unit 32 in this embodiment is sequentially executed as described below at a predetermined control processing cycle.

  First, similarly to the first embodiment, the current command determination unit 51 determines the d-axis current command value Imd_c and the q-axis current command value Imq_c according to the torque command value Tr_c and the rotor speed detection value ωm_s. The armature current detection values Imu_s, Imv_s, Imw_s and the rotor angle detection value θm_s are input to the dq conversion unit 55, and the processing of the dq conversion unit 55 is executed. At this time, the dq conversion unit 55 performs coordinate conversion of the armature current detection values Imu_s, Imv_s, Imw_s according to the electrical angle of the rotor 3 corresponding to the rotor angle detection value θm_s, so that the d-axis current detection value Imd_s and The q-axis current detection value Imq_s is calculated. The d-axis current detection value Imd_s and the q-axis current detection value Imq_s obtained by the dq converter 55 are the actual d-axis current detection value and the q-axis current detection value of the armature windings 6 and 7, respectively. Meaningful.

  Next, the d-axis current command value Imd_c and the q-axis current command value Imq_c, the shaft current detection value Imd_s, and the q-axis current detection value Imq_s are input to the current feedback control unit 56. Note that the rotor speed detection value ωm_s is also input to the current feedback control unit 56.

  At this time, the current feedback control unit 56 performs the same processing as in the first embodiment. Specifically, first, a deviation ΔImd (= Imd_c−Imd_s) between the d-axis current command value Imd_c and the d-axis current detection value Imd_s is calculated by the calculation unit 58, and the q-axis current command value Imq_c and the q-axis current are calculated. A deviation ΔImq (= Imq_c−Imq_s) from the detected value Imq_s is calculated by the calculation unit 59.

  Next, the processing of the PI control units 60 and 61 is executed, and the basic command value Vmd1_c for the d-axis voltage and the basic command value Vmq1_c for the q-axis voltage are calculated from the deviations ΔImd and ΔImq, respectively, according to the PI control law. Further, the process of the non-interference control unit 62 is executed, and the d-axis voltage correction amount Vmd2_c and the q-axis voltage correction amount Vmq2_c for canceling the speed electromotive force that interferes with each other between the d-axis and the q-axis are obtained. The processes of the PI control units 60 and 61 and the non-interference control unit 62 are the same as those in the first embodiment.

  Next, the calculation unit 63 adds the correction amount Vmd2_c to the basic command value Vmd1_c for the d-axis voltage to obtain the final d-axis voltage command value Vmd_c, and at the calculation unit 64, the basic command value for the q-axis voltage. By adding the correction amount Vmq2_c to Vmq1_c, the final q-axis voltage command value Vmq_c is obtained.

  Thus, the d-axis voltage command value Vmd_c and the q-axis voltage command value Vmq_c determined by the current feedback control unit 56 and the rotor angle detection value θm_s are input to the three-phase conversion unit 57. Then, the three-phase conversion unit 57 performs coordinate conversion of the d-axis voltage command value Vd_c and the q-axis voltage command value Vq_c according to the electrical angle of the rotor 3 corresponding to the rotor angle detection value θm_s, so that the phase voltage command The values Vmu_c, Vmv_c, and Vmw_c are calculated. The three-phase conversion unit 57 inputs the calculated phase voltage command values Vmu_c, Vmv_c, and Vmw_c to the PDU 31.

  At this time, the PDU 31 controls on / off of each switch element 36 of the inverter circuit 38 by PWM control so that the voltages of the input phase voltage command values Vmu_c, Vmv_c, Vmw_c are generated on the output side of the inverter circuit 38. . As a result, the d-axis current detection value Imd_s and the q-axis current detection value Imq_s as detection values of the current flowing through the armature windings 6 and 7 coincide with the d-axis current command value Imd_c and the q-axis current command value Imq_c, respectively. Thus, the output current of the inverter circuit 38 is controlled. Accordingly, the currents flowing through the armatures 6 and 7 of the respective phases of the electric motor 1 are determined by the current command determination unit 51 without depending on the currents flowing through the respective capacitive filters 41. The d-axis current command values Isd_c and q The control is performed so as to match the current command value of each phase corresponding to the shaft current command value Isq_c. As a result, the operation control of the electric motor 1 is performed so that the torque of the torque command value Tr_c is generated in the output shaft 1a of the electric motor 1.

  According to the second embodiment described above, the high-frequency current component included in the output current of the inverter circuit 38 flows through each capacitive filter 41, and thus the high-frequency current component flows through the armature windings 6 and 7. Can be prevented. In addition, since one end of each capacitive filter 41 connected to the neutral point 1N of the armature windings 6 and 7 is in the same potential as the neutral point 1N, the capacitive filter Suppressing the occurrence of an unbalanced offset between the currents flowing through the armature windings 6 and 7 due to the resonance phenomenon between the armature windings 6 and 7 and the armature windings 6 and 7 can do.

  For this reason, the copper loss and iron loss in the electric motor 1 can be reduced similarly to 1st Embodiment, and the energy efficiency of the electric motor 1 can be improved. Further, since it is possible to suppress the occurrence of an unbalanced offset between the currents flowing through the armature windings 6 and 7, the number of permanent magnets 8, 10, 12 provided in the rotor 3 can be reduced. Magnetism can be prevented.

  Further, in the present embodiment, the detected value of the current in the armature windings 6 and 7 is added to the d-axis current command value Imd_c and the q-axis current command value Imq_c as the command values of the currents flowing through the armature windings 6 and 7. The output current of the inverter circuit 38 is feedback-controlled so that the detected d-axis current value Imd_s and the detected q-axis current value Imq_s match each other. Accordingly, the torque of the torque command value Tr_c is stably supplied to the electric motor 1 without depending on the current flowing through each capacitive filter 41 due to a resonance phenomenon between the capacitive filter 41 and the armature windings 6 and 7. Can be generated.

  In each of the embodiments described above, the armature windings 6 and 7 are connected in series for each phase. However, as shown in FIG. The armature windings 6 and 7 of each phase may be connected in parallel with the neutral point 1N common to the lines 6 and 7.

  Even in this case, the operation of the electric motor can be controlled by the same control process as in the first embodiment or the second embodiment. In the first embodiment, when the armature windings 6 and 7 of each phase are connected in parallel, the impedance Zm_co is equal to the parallel circuit of the armature windings 6 and 7 of each phase (or Impedance of the parallel circuit and the inductive filter), and its value is generally different from that of the first embodiment.

  In each of the above embodiments, the magnetic pole array of the rotor 3 is a Halbach-type magnetic pole array, but it is not always necessary to do so. For example, the permanent magnet provided in the rotor 3 may be the main permanent magnet 8 alone.

The block diagram which shows the whole structure of the control apparatus of the electric motor in 1st Embodiment of this invention. The perspective view of the rotor and stator of the electric motor in 1st Embodiment. FIG. 3 is an exploded perspective view of the rotor shown in FIG. 2. The figure which shows the arrangement | sequence of a magnetic pole when the rotor shown in FIG. 2 is seen in the axial center direction. The figure which generalizes and shows the capacitive filter with which the control apparatus of 1st Embodiment was equipped. The figure which shows the inverter circuit in 1st Embodiment, and the circuit structure of the output side. The block diagram which shows the whole structure of the control apparatus of the electric motor in 2nd Embodiment of this invention. The figure which shows the other example of the connection form of the armature winding of both the stators of an electric motor.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... Electric motor, 1N ... Neutral point, 2,80 ... Control apparatus, 3 ... Rotor, 4, 5 ... Stator, 6, 7 (6u, 6v, 6w, 7u, 7v, 7w) ... Armature winding, 8 , 10, 12 ... Permanent magnet, 23 (23u, 23v, 23w) ... Current line, 33 (33u, 33v, 33w) ... Current sensor, 38 ... Inverter circuit, 41 (41u, 41v, 41w) ... Capacitive filter, 42 (42u, 42v, 42w) ... connection terminals for energization, 51 ... current command determination unit (target armature current determination means), 53 ... inverter output current command calculation unit (inverter target output current determination means), 54 ... dq conversion (Inverter target output current determination means), 56... Current feedback control section (inverter control means).

Claims (3)

An axial gap type comprising a rotor having a permanent magnet, two stators provided on both sides of the rotor in the axial direction of the rotor, and multi-phase armature windings respectively mounted on the stators A control device for energizing an armature winding of both stators of an electric motor from an inverter circuit,
The multi-phase armature windings of the two stators are each between a neutral point common to the armature windings of the two stators and the connection terminals for energization of the respective phases provided in the motor. Each phase is connected in either a serial connection or a parallel connection, and the connection terminals for energization of each phase are connected to the output terminals of each phase of the inverter circuit via an energization line. ,
One ends of the same number of capacitive filters as the number of phases of the armature windings of both stators are electrically connected to the neutral point, and the other ends of the capacitive filters are connected to the energization of each phase. An electric motor control device connected to a line. The motor control device according to claim 1,
A current sensor for detecting a current flowing through the energization line of each phase at a location between the capacitive filter and the inverter circuit;
Target armature current determining means for determining a target armature current, which is a target value of a current flowing through the armature windings of both stators, according to at least the target output torque of the motor and the rotational speed of the rotor of the motor; ,
The neutrality of each capacitive filter via the armature winding of the same phase as that of the energization line of the stator armature winding from one end on the energization line side of the capacitive filter Based on a predetermined arithmetic expression representing the relationship between the impedance of the current path to the point, the induced voltage of the armature winding, the current flowing through the armature winding, and the output current of the inverter circuit, the target Inverter target output current determining means for determining an inverter target output current that is a target value of the output current of the inverter circuit corresponding to the armature current;
An electric motor control device comprising: inverter control means for performing feedback control of an operation of the inverter circuit so that an output current of the inverter circuit detected via the current sensor is matched with the inverter target output current. The motor control device according to claim 1,
A current sensor for detecting a current flowing through the energization line of each phase at a location between the capacitive filter and the armature windings of the two stators;
Target armature current determining means for determining a target armature current, which is a target value of a current flowing through the armature windings of both stators, according to at least the target output torque of the motor and the rotational speed of the rotor of the motor; ,
Inverter control means for feedback-controlling the operation of the inverter circuit so as to match the current of the armature windings of the stators detected by the current sensor and the target armature current. Electric motor control device.

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