JP2008268085A - Rotary sensor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a rotary sensor which eliminates the effects on the amplitudes of signals obtained from magnetoresistive elements due to the temperature, accompanying the rotation of a rotating body, and facilitates obtaining binarization signal. <P>SOLUTION: A constant current is supplied to the magnetoresistive elements 1, 2, and the signal is obtained as a voltage across both terminals and changed by resistances changed by a magnetic force changed due to the rotation of a rotor Rt. In a differential amplification means, comprising operational amplifiers 7, 8, 9 and resistors 10, 11, 12, 13, signals obtained by the elements 1, 2 and a reference voltage are inputted. The difference between the signals obtained by the elements 1, 2 is amplified and outputted. The value that corresponds to a difference-amplified signal level is compared with the threshold, and the binary signal is outputted, according to the relation of the magnitude. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a rotation sensor using a magnetoresistive element.

Conventionally, Patent Document 1 discloses a rotation sensor using a magnetoresistive element. In this rotation sensor, as shown in FIG. 11, two magnetoresistive elements 100 and 200 are arranged close to the rotor Rt. The two magnetoresistive elements 100 and 200 are connected in series, and a constant voltage is applied to the series circuit to obtain a midpoint voltage between the two magnetoresistive elements 100 and 200 as an output signal. Further, in the previous stage of obtaining a binarized signal based on this output signal, when the operational amplifier 15 amplifies, the output of the operational amplifier 15 is amplified by the comparators 32 and 34 and the amplitude center of the alternating current signal that is the output of the operational amplifier 15 is obtained. It is determined whether or not it is within a predetermined amplitude region sandwiching the signal, and if it is outside the region, the reference voltage (comparison voltage) VS is changed in a direction approaching the output of the operational amplifier 15 as shown in FIG. As a result, when the mounting position of the magnetoresistive element is shifted, the center of the amplitude of the signal is shifted as shown by a broken line in FIG. 12. In this case, the reference voltage (comparison voltage) VS is changed in a direction approaching the output of the operational amplifier 15. Therefore, it is avoided that the amplified signal sticks to the operation limit value.
JP-A-8-193841

  In recent years, there is a demand for obtaining a signal having a large amplitude and a small temperature characteristic in order to cope with a case where the distance between the rotor and the tip of the sensor (magnetoresistance element) is large. However, as shown in FIG. 11, the configuration in which the midpoint voltage is obtained as an output signal by applying the voltage to the bridge by constant voltage driving is a method of using a large temperature characteristic. For this reason, the circuit side corrects the temperature characteristic, etc. Ingenuity is required. Further, in the case of taking a differential using another set of the magnetoresistive elements 100 and 200 of FIG. 11 in order to increase the amplitude, in order to obtain a binary signal from the signal after differential amplification, a differential amplification signal is used. Must cross the threshold.

  The present invention has been made under such a background. The purpose of the present invention is to make the amplitude of the signal obtained by the magnetoresistive element accompanying the rotation of the rotating body less susceptible to temperature and the binarized signal. The object is to provide an easy-to-obtain rotation sensor.

  In order to solve the above-mentioned problem, in the invention according to claim 1, the rotating body is arranged close to the rotating body, a constant current is supplied from a constant current circuit, and a voltage between both terminals is used to rotate the rotating body. A first magnetoresistive element that obtains a signal that changes as a result of a change in resistance value due to a change in magnetic force, is disposed close to the rotating body, and is supplied with a constant current from a constant current circuit. As the voltage, the second magnetism is obtained by changing the resistance value due to the change of the magnetic force accompanying the rotation of the rotating body, and obtaining a signal having a phase opposite to the signal obtained by the first magnetoresistive element. A resistance element; differential amplification means for amplifying and outputting a difference between a signal obtained by the first magnetoresistance element and a signal obtained by the second magnetoresistance element; and an output signal of the differential amplification means Compare the value according to the level and the threshold And a binarizing unit that outputs a binarized signal according to the magnitude relationship, wherein the differential amplifying unit inputs a signal obtained by the first magnetoresistive element. An operational amplifier, a second operational amplifier for inputting a signal obtained by the second magnetoresistive element, and a third operational amplifier for inputting a reference potential; The first to fourth resistors are connected in series with the output terminal of the first operational amplifier, negative feedback is applied to the first operational amplifier via the first resistor, and the second operational resistor is connected between the second and third resistors. The output terminal of the second operational amplifier is connected, the second operational amplifier is negatively fed back through the third resistor, and the differential amplified signal is obtained at the output terminal of the first operational amplifier. To do.

  According to the first aspect of the present invention, a constant current is supplied to each of the first and second magnetoresistive elements, and each magnetoresistive element has a resistance due to a change in magnetic force accompanying rotation of the rotating body as a voltage between both terminals. A method of obtaining a signal that changes as the value changes (constant current drive) is used. Thereby, the amplitude of each signal obtained by the first and second magnetoresistive elements accompanying the rotation of the rotating body can be made less susceptible to temperature. That is, the difference in amplitude in each signal when the temperature changes can be reduced or the ratio can be made close to 1.

  In addition, as shown in FIG. 10, when the midpoint potential of the half bridge is used as an output and the midpoint potential of each bridge is obtained as an output signal using two halfbridges in order to obtain a differential, that is, a total of four magnetic When a resistor is used and a signal is obtained from between the magnetoresistive elements connected in series, the offset value of the midpoint potentials Va and Vb (the center value of the amplitude of the AC signal) can be made very close to the design value. . However, in the case of constant current driving, a total of two magnetoresistive elements are used, and the voltage between both terminals in a single magnetoresistive element is obtained as a signal, so that the signal obtained by the first magnetoresistive element and the second For signals obtained by the magnetoresistive element, it is difficult to bring the offset value of each signal (the center value of the amplitude of the AC signal) close to the design value. In the differential amplifying means according to the first aspect, a differential amplified signal in the vicinity of the reference potential, that is, a signal in the vicinity of an arbitrary level (AC signal) is simultaneously obtained during the differential amplification. As a result, when the binarizing means compares the value corresponding to the output signal level of the differential amplifying means with the threshold value and outputs the binarized signal according to the magnitude relationship, the binarizing means responds to the output signal level of the differential amplifying means. The value and the threshold value are likely to cross each other, and a binary signal can be easily obtained.

  As described in claim 2, in the rotation sensor according to claim 1, the resistance value of the second resistor is equal to the resistance value of the third resistor, and the resistance value of the first resistor is The resistance values of the four resistors should be equal.

  According to a third aspect of the present invention, in the rotation sensor according to the first or second aspect of the present invention, the rotation sensor has a differential gain exceeding an operating limit with respect to an amplified value of the AC signal from the differential amplification means, An amplifier for amplifying the difference between the signal from the amplifying means and the comparison voltage and outputting it to the binarizing means; a counting circuit; and a digital-to-analog conversion circuit for setting the comparison voltage according to the count value by the counting circuit; It is determined whether the output of the amplifier is within a predetermined amplitude region sandwiching the amplitude center of the AC signal that is the output of the amplifier, and when the output is out of the region, the count value of the counting circuit is changed. Count value changing means for changing the comparison voltage in a direction approaching the output of the amplifier. By adopting this configuration, the difference between the signal from the differential amplification means and the comparison voltage is amplified and output by the amplifier, and at this time, a limiter is applied to the amplified value exceeding the operation limit, That is, the operation limit value is maintained. Therefore, the signal near the center of the amplitude of the AC signal changes sharply and can be binarized with high accuracy. Further, the count value changing means determines whether or not the output of the amplifier is within a predetermined amplitude region sandwiching the amplitude center of the AC signal that is the output of the amplifier. Since the comparison voltage is changed so as to approach the output of the amplifier, the amplified signal is prevented from sticking to the operation limit value. Furthermore, the digital-analog converter circuit sets a comparison voltage according to the count value of the counting circuit, but the temperature varies with the amplitude of each signal obtained by the first and second magnetoresistive elements as the rotating body rotates. Since the ratio of the amplitude of each signal at this time can be made close to 1, the number of bits of the digital-analog conversion circuit can be reduced.

  4. The rotational sensor according to claim 1, wherein an anisotropic magnetoresistive element is used as the first magnetoresistive element and the second magnetoresistive element. And excellent durability and pairing.

  As described in claim 5, in the rotation sensor according to any one of claims 1 to 3, when a tunnel magnetoresistive element is used as the first magnetoresistive element and the second magnetoresistive element, Excellent temperature characteristics.

  As described in claim 6, in the rotation sensor according to any one of claims 1 to 5, when a polysilicon thin film resistor is used as the first to fourth resistors, the temperature sensor has excellent temperature characteristics. It becomes.

  As described in claim 7, when the rotation sensor according to any one of claims 1 to 6 is used in a vehicle, it is useful in an environment in which a change in temperature tends to be large. In particular, as described in claim 8, the rotation sensor according to claim 7 is useful when it is directly mounted on an engine because the temperature change tends to be large. Further, as described in claim 9, when the rotation sensor according to claim 7 is directly mounted on the transmission, the change in temperature tends to be large, which is useful.

DESCRIPTION OF EXEMPLARY EMBODIMENTS Hereinafter, an embodiment of the invention will be described with reference to the drawings.
FIG. 1 shows an overall circuit configuration of the rotation sensor in the present embodiment. In the binarizing means constituted by members 43 to 47 in the circuit configuration in FIG. 1, a value corresponding to the output signal level of the differential amplifying means to be described later is compared with a threshold value, and the magnitude relation thereof. A binarized signal whose level changes with the passage of teeth accompanying the rotation of the rotor Rt is output. FIG. 2 shows the preceding stage of the overall configuration of FIG. 1, that is, magnetoresistive elements 1 and 2, constant current circuits (constant current sources) 3 and 4, operational amplifiers 7, 8, and 9 and resistors 5, 6, and 10. , 11, 12, and 13 will be described. FIG. 3 is a diagram for explaining currents and potentials at respective parts in FIG.

  The rotation sensor of the present embodiment is for in-vehicle use. Specifically, the rotation sensor is a crank angle sensor or the like, and is directly mounted on the engine. Or it is a transmission input rotation sensor etc., and is directly mounted in a vehicle-mounted transmission.

  In FIG. 1, the sensor includes two magnetoresistive elements (MRE) 1 and 2. The two magnetoresistive elements 1 and 2 are arranged at positions close to the rotor Rt. As the magnetoresistive elements 1 and 2, anisotropic magnetoresistive elements (AMR) or tunnel magnetoresistive elements (TMR) are used. The rotor Rt has a plurality of teeth formed on the outer peripheral surface thereof. In FIG. 2, one end of the magnetoresistive element 1 is grounded, and the constant current circuit 3 is connected to the other end. Similarly, one end of the magnetoresistive element 2 is grounded, and a constant current circuit 4 is connected to the other end. The potential Va between the magnetoresistive element 1 and the constant current circuit 3 becomes an output signal. The potential Vb between the magnetoresistive element 2 and the constant current circuit 4 is an output signal. Each output signal becomes an AC signal as the rotor Rt rotates.

  Here, the magnetoresistive elements 1 and 2 are arranged with an inclination of approximately 90 ° in order to reverse the signals (Va and Vb). The constant current circuits 3 and 4 are constituted by current mirrors, and the same current flows through the magnetoresistive elements 1 and 2.

  In this way, the first magnetoresistive element 1 is disposed close to the rotor Rt as a rotating body, supplied with a constant current from the constant current circuit 3, and used as a voltage between both terminals to rotate the rotor Rt. A signal that changes as the resistance value changes due to a change in magnetic force is obtained. The second magnetoresistive element 2 is arranged close to the rotor Rt, supplied with a constant current from the constant current circuit 4, and changes its resistance value as a voltage between both terminals due to a change in magnetic force accompanying the rotation of the rotor Rt. As a result, a signal that changes and is opposite in phase to the signal obtained by the first magnetoresistive element 1 is obtained.

Further, a differential amplifying means is constituted by members indicated by reference numerals 5 to 13, and amplifies and outputs a difference between a signal obtained by the first magnetoresistive element 1 and a signal obtained by the second magnetoresistive element 2. To do.
In FIG. 2, a signal (Va) obtained by the magnetoresistive element 1 is input to the non-inverting input terminal of the operational amplifier 7. A signal (Vb) obtained by the magnetoresistive element 2 is input to the non-inverting input terminal of the operational amplifier 8.

  One end of the series circuit including the resistors 5 and 6 is grounded, and a constant voltage is applied to the other end. A potential (VM) between the resistors 5 and 6 is input to the non-inverting input terminal of the operational amplifier 9. Since the resistors 5 and 6 are divided resistors for generating the reference potential (VM), a polysilicon thin film resistor having a small temperature coefficient is used instead of the resistance of the magnetoresistive element.

  Four resistors 10, 11, 12, and 13 are connected in series between the output terminal of the operational amplifier 9 and the output terminal of the operational amplifier 7. The potential between the resistors 10 and 11 is input to the inverting input terminal of the operational amplifier 7. The output terminal of the operational amplifier 8 is connected between the resistors 11 and 12. The potential between the resistors 12 and 13 is input to the inverting input terminal of the operational amplifier 8. Negative feedback is applied to the output of the operational amplifier 9. As described above, the operational amplifiers 7, 8, 9 receive the signals (Va, Vb) and the reference potential (VM), respectively, and the operational amplifiers 7, 8, 9 are connected via the resistors 10, 11, 12, 13. ing. The resistors 10, 11, 12, and 13 are not thin film resistors of magnetoresistive elements, but are polysilicon thin film resistors having a small temperature coefficient.

  In this way, the differential amplifying means includes the first operational amplifier 7 for inputting the signal obtained by the first magnetoresistive element 1 and the second operational amplifier for inputting the signal obtained by the second magnetoresistive element 2. 8 and a third operational amplifier 9 for inputting a reference potential, and the first to fourth resistors are provided between the output terminal of the third operational amplifier 9 and the output terminal of the first operational amplifier 7 to which negative feedback is applied. 10, 11, 12, 13 are connected in series, negative feedback is applied to the first operational amplifier 7 via the first resistor 10, and the second operational amplifier 8 is connected between the second and third resistors 11, 12. An output terminal is connected, negative feedback is applied to the second operational amplifier 8 through the third resistor 12, and a differential amplified signal is obtained at the output terminal of the first operational amplifier 7. The resistance value of the second resistor 11 and the resistance value of the third resistor 12 are equal, and the resistance value of the first resistor 10 and the resistance value of the fourth resistor 13 are equal. Specifically, the resistance values of the resistors 11 and 12 are both 1 kΩ, and the resistance values of the resistors 10 and 13 are both the same predetermined value X kΩ.

Next, the efficiency of obtaining the amplitude of the signal (rotation detection signal) obtained by the magnetoresistive element will be described.
In this embodiment, a constant current driving method is adopted. In the case of FIG. 11, the constant voltage driving method is adopted.

FIG. 4 is an explanatory diagram of the constant voltage driving method, and FIG. 5 is an explanatory diagram of the constant current driving method.
As shown in FIGS. 4 and 5, in the comparative explanatory view of the efficiency of the constant voltage driving method and the constant current driving method, a constant voltage of 5 volts was applied to a half bridge composed of two resistors of 2.5 kΩ as shown in FIG. In this case, it is assumed that each resistance value change is changed by + 10% and −10%. In this case, the midpoint potential changes from 2.5 volts to 2.75 volts.

  In order to obtain the same change, when driven at a constant current, as shown in FIG. 5, if 1 mA is passed through one resistance of 2.5 kΩ, the same output change can be obtained with a 10% change in resistance value. In other words, considering the resistance value, it can be considered that the constant voltage drive + bridge has half the efficiency of the constant current drive.

Next, temperature characteristics regarding amplitude in a signal (rotation detection signal) obtained by the magnetoresistive element will be described.
FIG. 6 is a temperature resistance characteristic diagram of a single resistance. The horizontal axis in FIG. 6 represents the magnetic field strength, and the vertical axis represents the resistance value. FIG. 6 shows the change in resistance value when the magnetic field intensity is changed at each temperature of 130 ° C., 25 ° C., and −30 ° C., and the change ΔR at that time. Specifically, ΔR = 10Ω at 130 ° C., ΔR = 11Ω at 25 ° C., and ΔR = 12Ω at −30 ° C.

  As described above, the resistance value ΔR due to the magnetic field at each resistance temperature at 130 ° C., 25 ° C., and −30 ° C. varies greatly depending on the temperature. However, the resistance value variation ΔR due to the magnetic field changes in temperature. However, it turns out that it hardly changes.

  Based on the numerical values shown in FIG. 6, the constant voltage drive and the constant current drive are calculated by comparison. As shown in FIG. 7 (a), the constant voltage drive + half bridge and the constant current drive as shown in FIG. 7 (b). In comparison with + single resistance, it becomes as shown in FIGS.

  In FIG. 8, at each temperature (130 ° C., 25 ° C., −30 ° C.), the resistance value change of two magnetoresistive elements in FIG. 7A and one magnetoresistive element in FIG. Before and after values RA, RA ′, RB, RB ′, values VM (V), VM (V) ′ before and after the change of the midpoint potential in FIG. 7A at that time, and a change amount ΔVM (V) FIG. 7B shows values VM (I) and VM (I) ′ before and after the change of the voltage between both terminals of the magnetoresistive element in FIG. 7B and the change amount ΔVM (I).

  From the values in FIG. 8, in FIG. 9, as “−30 ° C. amplitude / 130 ° C. amplitude”, “ΔVM (V) at −30 ° C.” / “ΔVM (V) at 130 ° C.” “ΔVM (I) at −30 ° C.” / “ΔVM (I) at 130 ° C.” is obtained. In addition, as “−30 ° C. amplitude / 130 ° C. amplitude temperature coefficient”, {“ΔVM (V) at 130 ° C.” − “ΔVM (V) at −30 ° C.”} / “ΔVM (V at 130 ° C. ) ”And {“ ΔVM (I) at 130 ° C. ”−“ ΔVM (I) at −30 ° C. ”} /“ ΔVM (I) at 130 ° C. ”.

  In FIG. 9, the amplitude temperature coefficient of −30 ° C. amplitude / 130 ° C. is “−7083 ppm / ° C.” in the constant voltage drive + half bridge of FIG. 7A, but the constant current drive of FIG. The resistance is “−1250 ppm / ° C.”. Therefore, it can be seen that the constant voltage drive is a method of use having a large amplitude temperature characteristic. Therefore, the constant voltage driving method requires temperature characteristic correction on the circuit side. On the other hand, constant current driving is a method of use with small amplitude temperature characteristics, and the constant current driving method can eliminate the need for temperature characteristic correction on the circuit side.

  In FIG. 9, the −30 ° C. amplitude / 130 ° C. amplitude is “2.11 times” in the constant voltage drive + half bridge of FIG. 7A, but in the constant current drive + single resistance of FIG. 1.20 times ". Therefore, in the present embodiment, the constant temperature driving reduces the temperature coefficient for the amplitude. Thereby, the input range can be reduced to about 57% (= 1.20 / 2.11.). As a result, the input (previous output) range is reduced.

  In this way, in the present embodiment, the conventional constant voltage drive + bridge is used in this way. In this embodiment, a constant current is supplied to each of the first and second magnetoresistive elements 1, 2. In the magnetoresistive elements 1 and 2, a signal that changes as a resistance value changes due to a change in magnetic force accompanying rotation of the rotor Rt is obtained as a voltage between both terminals. Thereby, the amplitude of each signal (Va, Vb) obtained by the first and second magnetoresistive elements 1 and 2 accompanying the rotation of the rotor Rt can be made less susceptible to temperature. That is, the difference in amplitude in each signal when the temperature changes can be reduced or the ratio can be made close to 1 (“ΔVM at 130 ° C.” − “ΔVM at −30 ° C.” in FIGS. 8 and 9). Or “ΔVM at −30 ° C.” / “ΔVM at 130 ° C.” can approach 1).

Next, the crossing property with the threshold value for the differential amplification signal based on the signal (rotation detection signal) obtained by the magnetoresistive element will be described.
FIG. 10 shows a configuration example of constant voltage driving (constant voltage + bridge configuration example) for comparison.

  In FIG. 10, four magnetoresistive elements (MRE) 301, 302, 303, 304 are used. A magnetoresistive element 301 and a magnetoresistive element 302 are connected in series between the ground and the constant voltage terminal. Similarly, a magnetoresistive element 303 and a magnetoresistive element 304 are connected in series between the ground and the constant voltage terminal. Here, by arranging the magnetoresistive elements 301 and 304 and the magnetoresistive elements 302 and 303 to be inclined by 90 °, the midpoint potential (Va) of the magnetoresistive element 301 and the magnetoresistive element 302, the magnetoresistive element 303 and the magnetic At the midpoint potential (Vb) of the resistance element 304, a signal amplitude waveform having a substantially opposite phase is obtained. The signal at these two midpoints is differentiated in the operational amplifier. In this way, a signal is obtained near the midpoint in constant voltage driving.

  In a rotation sensor (proximity sensor) that detects the passage of teeth by placing a magnetoresistive element close to the rotor, this signal amplitude is binarized depending on whether it crosses a certain threshold value. It is necessary to come.

  In the present embodiment, as shown in FIG. 2, the magnetoresistive element 1 is connected to the constant current circuit 3, and the potential Va between the constant current circuit 3 and the magnetoresistive element 1 is taken out. Similarly, the magnetoresistive element 2 is connected to the constant current circuit 4, and the potential Vb between the constant current circuit 4 and the magnetoresistive element 2 is taken out. On the other hand, the resistor 5 and the resistor 6 are connected in series between the ground and the constant voltage terminal, and a midpoint potential (VM) between the resistor 5 and the resistor 6 is obtained.

  Here, as shown in FIG. 10, when the midpoint potential of the half bridge is used as an output and the midpoint potential of each bridge is obtained as an output signal using two halfbridges in order to obtain a differential, that is, a total of four When using the magnetoresistive elements 301, 302, 303, and 304 and obtaining a signal between the magnetoresistive elements connected in series, the offset values of the midpoint potentials Va and Vb (the center values of the amplitudes of the AC signals) are very low in design values. The value can be close to. However, in the case of constant current driving according to this embodiment (in the case of constant current + single resistance × 2), a total of two magnetoresistive elements 1 and 2 are used, and the voltage between both terminals of a single magnetoresistive element is signaled. Therefore, for the signal obtained by the first magnetoresistive element 1 and the signal obtained by the second magnetoresistive element 2, the offset value of each signal (the center value of the amplitude of the AC signal) is brought close to the design value. It becomes difficult. By combining the circuit (5 to 13) as shown in FIG. 2, the difference between the signal (Va) and the signal (Vb) (= Va−Vb) is changed to the resistance (10, 13) and the resistance (11, 12). ) Ratio X + 1 and the value added with the VM value is output. That is, the output V1 of the preceding stage is V1 = (Va−Vb) (1 + X) + VM.

  3, when the resistance value of the resistor 11 and the resistor 12 is 1 kΩ and the resistance value of the resistor 10 and the resistor 13 is XkΩ, the current I1 flowing through the resistors 12 and 13 is (Vb−VM) / X. It is. Therefore, the output portion potential VX of the operational amplifier 8 is VX = Vb + (Vb−VM) / X · 1. Further, the current I2 flowing through the resistors 10 and 11 = [Va− {Vb + (Vb−VM) / X · 1}] / 1 = Va−Vb−Vb / X + VM / X. Therefore, the output V1 = Va + (Va−Vb−Vb / X + VM / X) · X = Va + VaX−VbX−Vb + VM = (Va−Vb) (1 + X) + VM at the previous stage.

  Thus, in FIG. 2, 2.5 volts is set as the reference potential VM, and the difference between the signal (Va) and the signal (Vb) is added to the 2.5 volts, and the operational amplifier 7 outputs The output V1 becomes an AC signal centered on 2.5 volts as the rotor Rt rotates.

Therefore, even when the potentials Va and Vb deviate from the vicinity of the reference potential (2.5 volts), an output of about 2.5 volts (= VM) can be obtained at the time of the previous stage output.
As described above, in the differential amplifying means shown in FIG. 2 of the present embodiment, a differential amplified signal near the reference potential, that is, a signal near an arbitrary level (AC signal) is obtained simultaneously during differential amplification. As a result, when the binarizing means compares the value corresponding to the output signal level of the differential amplifying means with the threshold value and outputs the binarized signal according to the magnitude relationship, the binarizing means responds to the output signal level of the differential amplifying means. The value and the threshold value are likely to cross each other, and a binary signal can be easily obtained.

  Further, in the comparison between FIG. 2 and FIG. 10, in the present embodiment of FIG. 2, the magnetoresistive elements 1 and 2 may be half the resistance of the method of FIG. 10. Therefore, the element area of the magnetoresistive element can be halved. The fact that the element area is halved increases the degree of freedom of element arrangement. This makes it possible to arrange the signal at a location where the signal amplitude is larger than that of the method of FIG.

  Hereinafter, a case where the configuration shown in FIG. 1 is shown as a configuration in the subsequent stage with respect to FIG. 2 will be described. The subsequent circuit for the circuit of FIG. 2 may or may not perform amplification with a correction function on the differential amplification signal, but FIG. 1 employs an amplification system with a correction function.

  As shown in FIG. 1, the output terminal of the operational amplifier 7 is connected to the non-inverting input terminal of the operational amplifier 15 through the resistor 14. Negative feedback is applied to the output terminal of the operational amplifier 15 via a resistor 16. The operational amplifier 15 has a differential gain (500 times) exceeding the operation limit (lower limit 0 volt, upper limit 4 volt; amplitude value 4 volt) with respect to the amplified value (40 mV) of the AC signal from the differential amplification means (5-13). ) Is used.

  The output terminal of the operational amplifier 15 is connected to the non-inverting input terminal of the upper limit detection comparator 32 via the resistor 31. The output of the comparator 32 is positively fed back via a resistor 48 (100Ω in this embodiment). The output terminal of the operational amplifier 15 is connected to the non-inverting input terminal of the lower limit detection comparator 34 via the resistor 33. The output of the comparator 34 is positively fed back via a resistor 49 (100Ω in this embodiment). On the other hand, resistors 35, 36, and 37 are connected in series to a 5-volt power supply VDD. A connection point i between the resistors 35 and 36 is connected to an inverting input terminal of the upper limit detection comparator 32. That is, the upper limit detection comparator 32 compares the output voltage of the operational amplifier 15 with the divided voltage (= 3.5 volts) at the connection point i. The connection point j between the resistors 36 and 37 is connected to the inverting input terminal of the lower limit detection comparator 34. That is, the lower limit detection comparator 34 compares the output voltage of the operational amplifier 15 with the divided voltage (= 0.5 volts) at the connection point j.

  The output terminal of the upper limit detection comparator 32 is connected to one input terminal of the OR gate 38. The output terminal of the lower limit detection comparator 34 is connected to the other input terminal of the OR gate 38 via the inverter 39. The output terminal of the OR gate 38 is connected to the enable terminal of the clock generation CR oscillation circuit 40. When an H level signal is input from the OR gate 38, the clock generation CR oscillation circuit 40 becomes active and outputs a clock signal to the up / down counter 41. The up / down counter 41 is connected to the output terminal of the upper limit detection comparator 32. When the H level signal is input from the upper limit detection comparator 32, the count value is increased when the clock signal is input. Further, when the clock signal is input when the H level signal is not input from the upper limit detection comparator 32, the count value is decreased.

  The up / down counter 41 and the digital / analog conversion circuit 42 are connected by a signal line for each bit. A signal corresponding to the count value of the up / down counter 41 is sent to the digital / analog conversion circuit 42. The digital / analog conversion circuit 42 uses the voltage corresponding to the count value of the up / down counter 41 as a reference voltage (comparison voltage) VS as an operational amplifier. It outputs to 15 non-inverting input terminals.

  The output terminal of the operational amplifier 15 is connected to the non-inverting input terminal of the comparator 44 through the resistor 43. The output of the comparator 44 is positively fed back via a resistor 47 (100Ω in this embodiment).

  Resistors 45 and 46 are connected in series to a 5-volt power supply VDD. A connection point k between the resistors 45 and 46 is connected to an inverting input terminal of the comparator 44, and a divided voltage (= 2 volts) at the connection point k is set as a threshold value of the comparator 44.

  In this embodiment, the operational amplifier 15 constitutes an amplifier, the up / down counter 41 constitutes a counting circuit, and the upper limit detection comparator 32, the lower limit detection comparator 34, and the clock generation CR oscillation circuit 40 count values. Changing means is configured.

  In FIG. 13, the output of the operational amplifier 15 falls while the output of the previous stage rises, and the output is compared with the lower limit voltage value BS (= 0.5 volts) in the lower limit detection comparator 34. Is lower than the lower limit voltage value BS (timing t20 in FIG. 13), the output of the lower limit detection comparator 34 becomes H level, and a signal is output to the clock generation CR oscillation circuit 40 via the OR gate 38. By this signal input, a clock signal is output from the clock generation CR oscillation circuit 40 to the up / down counter 41, and the count value is decreased by "1". As a result, the reference voltage (comparison voltage) VS, which is the output of the digital-analog conversion circuit 42, decreases by a predetermined voltage. Therefore, the amplifier output increases by 0.5 volts to 1 volt.

  A similar operation is also performed at t21 in FIG. In the period from t22 to t23 in FIG. 13, the output of the operational amplifier 15 is lower than the lower limit voltage value BS, the output of the lower limit detection comparator 34 is always H level, and the clock generation CR oscillation circuit 40 ups and downs every predetermined time. A clock signal is output to the counter 41, the count value is reduced by the up / down counter 41, and the reference voltage (comparison voltage) VS is also lowered.

  On the other hand, in the state where the output of the front stage after the output of the front stage changes from increase to decrease, the output of the operational amplifier 15 is compared with the upper limit voltage value PS (= 3.5 volts) in the upper limit detection comparator 32. When the output of the operational amplifier 15 exceeds the upper limit voltage value PS (timing t24 in FIG. 13), the output of the upper limit detection comparator 32 becomes H level, and a signal is output to the clock generation CR oscillation circuit 40 via the OR gate 38. Is done. By this signal input, a clock signal is output from the clock generation CR oscillation circuit 40 to the up / down counter 41, and the count value is increased by "1". As a result, the reference voltage (comparison voltage) VS, which is the output of the digital-analog conversion circuit 42, increases by a predetermined voltage. Therefore, the amplifier output is reduced by 0.5 volts to 3 volts.

  A similar operation is performed at t25 and 26 in FIG. During the period from t27 to t28 in FIG. 13, the output of the operational amplifier 15 exceeds the upper limit voltage value PS, and the output of the upper limit detection comparator 32 is always at the H level. The clock signal is output to the counter 41, the count value is increased by the up / down counter 41, and the reference voltage (comparison voltage) VS is also increased.

  Further, the comparator 44 compares the amplifier output with the threshold voltage (= 2 volts), and the output is inverted at the timings t40, t41, 42, and 43 in FIG.

  Further, in FIG. 13, the output of the previous stage is stuck to the upper limit or the lower limit of the operation limit during the period of t51 to t52, t53 to t54, t55 to t56, t57 to t58.

  In this way, the differential gain exceeding the operation limit with respect to the amplified value of the AC signal from the differential amplifier means (5-13) has a differential gain, and the signal from the differential amplifier means (5-13) and the comparison voltage An operational amplifier 15 that amplifies the difference between them and outputs them to the binarization means (43 to 47), an up / down counter 41, a digital / analog conversion circuit 42 that sets a comparison voltage according to the count value of the up / down counter 41, It is determined whether or not the output of the operational amplifier 15 is within a predetermined amplitude region sandwiching the amplitude center of the AC signal that is the output of the operational amplifier 15, and when the output is out of the region, the count value of the up / down counter 41 is changed. Count value changing means (comparators 32 and 34, clock generation CR oscillation circuit 40) for changing the comparison voltage in a direction approaching the output of the operational amplifier 15.Then, the operational amplifier 15 amplifies and outputs the difference between the signal from the differential amplifying means (5-13) and the comparison voltage. At this time, a limiter is applied to the amplified value exceeding the operation limit. That is, the operation limit value is maintained. Therefore, the signal near the center of the amplitude of the AC signal changes sharply and can be binarized with high accuracy. Further, it is determined by the count value changing means (32, 34, 40) whether or not the output of the operational amplifier 15 is within a predetermined amplitude region across the amplitude center of the AC signal that is the output of the operational amplifier 15. When it is off, the count value of the up / down counter 41 is changed and the comparison voltage is changed in a direction approaching the output of the operational amplifier 15, so that the amplified signal is prevented from sticking to the operation limit value.

  As a problem when such an amplification method with a correction function is adopted, a sufficient number of digital-to-analog conversion stages (a sufficient number of bits in the digital-to-analog conversion circuit 42) that can cover the entire applied input (previous output) range is required. However, when the number of stages is increased, there is a problem that not only the circuit area increases, but also adjustment time is required and adjustment cannot catch up. Against this problem, the constant temperature drive reduces the amplitude temperature coefficient. Specifically, as explained in FIGS. 7, 8, and 9, it is 2.11 times at −30 ° C. and 130 ° C. As a result, the input range is reduced to about 57%. Therefore, by reducing the input (previous stage output) range, it is possible to reduce the number of bits of the digital-analog conversion circuit 42 and reduce the size.

  In this way, the digital / analog conversion circuit 42 sets the comparison voltage according to the count value of the up / down counter 41, and is obtained by the first and second magnetoresistive elements 1 and 2 accompanying the rotation of the rotor Rt. As for the amplitude of each signal, the ratio of the amplitude of each signal when the temperature changes can be made close to 1, so the number of bits of the digital-analog conversion circuit 42 can be reduced.

According to the above embodiment, the following effects can be obtained.
(1) In FIG. 1, by using constant current drive, the amplitude of each signal obtained by the first and second magnetoresistive elements 1 and 2 accompanying the rotation of the rotor Rt is less affected by temperature. Can do. That is, the difference in amplitude in each signal when the temperature changes can be reduced or the ratio can be made close to 1. In the case of FIG. 10, the offset values of the midpoint potentials Va and Vb (the center value of the amplitude of the AC signal) can be made very close to the design value. 2, it is difficult to bring the offset value (the center value of the amplitude of the AC signal) of each signal close to the design value. However, in the differential amplification unit of this embodiment, the differential amplification signal near the reference potential, that is, A signal in the vicinity of an arbitrary level (AC signal) is obtained at the same time during differential amplification, and when the binarizing means outputs a binarized signal, a value corresponding to the output signal level of the differential amplifying means and a threshold value are obtained. It is easy to cross and it is easy to obtain a binary signal.

  (2) When an anisotropic magnetoresistive element (AMR) is used as the first magnetoresistive element 1 and the second magnetoresistive element 2, it is excellent in durability and is particularly useful when directly mounted on an engine or the like. . An anisotropic magnetoresistive element (AMR) is excellent in pairing. That is, since it has a single layer structure, the difference in performance when two identical magnetoresistive elements are arranged can be reduced.

(3) When a tunnel magnetoresistive element (TMR) is used as the first magnetoresistive element 1 and the second magnetoresistive element 2, the temperature characteristics are excellent.
(4) When a polysilicon thin film resistor is used as the first to fourth resistors 10 to 13, the temperature characteristics are excellent.

  (5) Since the rotation sensor is for in-vehicle use, the rotation sensor is useful in an environment where the temperature change is likely to be large. In particular, when mounted directly on the engine, the change in temperature is likely to be large, which is useful. Further, when directly mounted on the transmission, the change in temperature is likely to be large, which is useful.

The circuit block diagram of the whole rotation sensor in this embodiment. The circuit block diagram of the front | former stage (constant current + two single resistances) of the rotation sensor in this embodiment. FIG. 3 is an explanatory diagram of output in each part of FIG. 2. Efficiency explanatory drawing of a constant voltage drive system. Efficiency explanatory drawing of a constant current drive system. Single resistance thermomagnetic characteristics diagram. It is a figure for comparing a signal amplitude temperature characteristic, (a) is a circuit block diagram of constant voltage drive + half bridge, (b) is a circuit block diagram of constant current drive + single resistance. It is a figure for comparing a signal amplitude temperature characteristic, Comprising: The figure which shows resistance value and voltage value in each temperature. It is a figure for comparing a signal amplitude temperature characteristic, Comprising: The figure which shows the ratio and temperature coefficient of the amplitude accompanying a temperature change. FIG. 3 is a circuit configuration diagram of constant voltage + bridge. The circuit block diagram of the whole rotation sensor in background art. Output waveform diagram. The wave form diagram in a rotation sensor.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... Magnetoresistance element, 2 ... Magnetoresistance element, 3 ... Constant current circuit, 4 ... Constant current circuit, 5 ... Resistance, 6 ... Resistance, 7 ... Operational amplifier, 8 ... Operational amplifier, 9 ... Operational amplifier, 10 ... Resistance, 11 ... Resistor 12 ... Resistor 13 ... Resistor 14 ... Resistor 15 ... Operational Amplifier 16 ... Resistor 31 ... Resistor 32 ... Comparator 33 ... Resistor 34 ... Comparator 35 ... Resistor 36 ... Resistor 37 ... Resistor DESCRIPTION OF SYMBOLS 38 ... OR gate, 39 ... Inverter, 40 ... Clock generation CR oscillation circuit, 41 ... Up / down counter, 42 ... Digital-analog conversion circuit, 43 ... Resistor, 44 ... Comparator, 45 ... Resistor, 46 ... Resistor, 47 ... Resistor, 48 ... resistance, 49 resistance, Rt ... rotor.

Claims (9)

A constant current is supplied from the constant current circuit (3) and is placed close to the rotating body (Rt), and the resistance value changes as a voltage between both terminals due to a change in magnetic force accompanying the rotation of the rotating body (Rt). A first magnetoresistive element (1) from which a signal that changes can be obtained;
A constant current is supplied from the constant current circuit (4) and is placed close to the rotating body (Rt), and a resistance value is generated as a voltage between both terminals due to a change in magnetic force accompanying the rotation of the rotating body (Rt). A second magnetoresistive element (2) that is changed by changing and that can obtain a signal having a phase opposite to that of the signal obtained by the first magnetoresistive element (1);
Differential amplification means (5-13) for amplifying and outputting a difference between a signal obtained by the first magnetoresistive element (1) and a signal obtained by the second magnetoresistive element (2);
Binarizing means (43 to 47) for comparing a value corresponding to the output signal level of the differential amplifying means (5 to 13) with a threshold value and outputting a binarized signal according to the magnitude relationship;
A rotation sensor comprising:
The differential amplifying means inputs a first operational amplifier (7) for inputting a signal obtained by the first magnetoresistive element (1) and a signal obtained by the second magnetoresistive element (2). A second operational amplifier (8) and a third operational amplifier (9) for inputting a reference potential are provided, and an output terminal of the third operational amplifier (9) subjected to negative feedback and an output of the first operational amplifier (7). The first to fourth resistors (10, 11, 12, 13) are connected in series with the terminal, and negative feedback is applied to the first operational amplifier (7) through the first resistor (10). The output terminal of the second operational amplifier (8) is connected between the second and third resistors (11, 12), and negative feedback is applied to the second operational amplifier (8) through the third resistor (12). And a differential amplifier signal obtained at the output terminal of the first operational amplifier (7). Support. The rotation sensor according to claim 1,
The resistance value of the second resistor (11) and the resistance value of the third resistor (12) are made equal, and the resistance value of the first resistor (10) and the resistance value of the fourth resistor (13) are A rotation sensor characterized by equality. The rotation sensor according to claim 1 or 2,
It has a differential gain that exceeds the operating limit with respect to the amplified value of the AC signal from the differential amplifier means (5-13), and amplifies the difference between the signal from the differential amplifier means (5-13) and the comparison voltage And an amplifier (15) for outputting to the binarization means (43 to 47),
A counting circuit (41);
A digital-to-analog conversion circuit (42) for setting the comparison voltage according to the count value by the counting circuit (41);
It is determined whether the output of the amplifier (15) is within a predetermined amplitude region sandwiching the amplitude center of the AC signal that is the output of the amplifier (15). 41) count value changing means (32, 34, 40) for changing the comparison voltage in a direction approaching the output of the amplifier (15) by changing the count value;
And a rotation sensor. The rotation sensor according to any one of claims 1 to 3,
A rotation sensor using anisotropic magnetoresistive elements as the first magnetoresistive element (1) and the second magnetoresistive element (2). The rotation sensor according to any one of claims 1 to 3,
A rotation sensor using a tunnel magnetoresistive element as the first magnetoresistive element (1) and the second magnetoresistive element (2). In the rotation sensor according to any one of claims 1 to 5,
A rotation sensor using a polysilicon thin film resistor as the first to fourth resistors (10 to 13). The rotation sensor according to any one of claims 1 to 6,
A rotation sensor characterized by being used in a vehicle. The rotation sensor according to claim 7, wherein
A rotation sensor that is directly mounted on an engine. The rotation sensor according to claim 7, wherein
A rotation sensor that is mounted directly on a transmission.

Images