JP2008245397A - Motor drive unit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To enable sensorless sine wave motor driving by V/f control. <P>SOLUTION: The direct-current power of a direct-current power supply 2 is converted into alternating-current power through an inverter circuit 3; the motor 4 is thereby driven to rotationally drive a motor load 5; the peak value of a motor current or a motor current corresponding to a rotating magnetic field is detected by a current detecting means 6; the output voltage of the inverter circuit 3 is controlled to a predetermined value by a controlling means 7; and this output voltage is stabilized by a frequency correcting means 76. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a motor drive device, and more particularly to a motor control means by V / f control of a permanent magnet motor.

Conventionally, this type of motor drive device detects an inverter circuit output current, performs coordinate conversion of the motor current according to a motor applied voltage phase, and performs V / f control so that the motor current after the coordinate conversion becomes a predetermined value. (For example, refer to Patent Document 1).
JP 2000-236694 A

  However, since the conventional motor drive device instantaneously detects the motor current corresponding to the motor applied voltage phase, and converts the coordinate to a DC component, the high-speed A / D conversion means or the high-speed current detection means and the high-speed coordinates are converted. Conversion means is required, and there is a problem that a processor such as a microcomputer for controlling the inverter circuit is high speed and high price. Further, since the control is performed by the effective current vector after the coordinate conversion, there is a problem that current setting becomes complicated in control of a no-load or a fan or a pump whose load changes according to the rotation speed.

  Further, since the frequency correction is performed regardless of the set frequency, there is a problem that it becomes easy to distort when the frequency is high.

  The present invention solves the above-described conventional problems. Basically, stable motor control is possible by V / f control without coordinate conversion, and the torque varies according to the load with large torque variation and the rotation speed. It is an object of the present invention to realize a motor drive device that can stably control a motor without a complicated load or load, and can drive a sensorless sine wave with a low-speed and inexpensive processor and a simple control program.

  In order to solve the above-described conventional problems, the motor driving device of the present invention drives a permanent magnet motor with a sensorless sine wave by an inverter circuit that converts DC power into AC power, and corresponds to a peak value of motor current or a rotating magnetic field. The inverter circuit output voltage and motor drive frequency are controlled so that the motor current is detected and the set value is obtained, the frequency is corrected by the error signal between the motor current set value and the detected motor current, and the frequency correction gain is changed according to the drive frequency. By doing so, it is possible to operate stably.

  The motor driving device of the present invention detects the motor current corresponding to the peak value of the motor current or the rotating magnetic field and controls the inverter circuit output voltage and the output frequency so as to obtain the set value. Regardless of the polarity or non-saliency motor, stable control can be performed without turbulence, and also stable control can be easily performed in lead angle control, and control is possible without high-speed A / D conversion means and high-speed calculation means, so that it is inexpensive. A motor drive device capable of sensorless sine wave drive can be realized with a processor and a simple control program. In addition, a simple and inexpensive current sensor can be used, and the control program can be simplified, so that an inexpensive and highly reliable motor drive device can be realized. Furthermore, since the processor is lightly loaded, a single processor can control multiple motors simultaneously. And a system capable of simultaneously driving a plurality of motors can be easily configured.

  A first invention is a DC power supply, an inverter circuit that converts DC power of the DC power supply into AC power, a permanent magnet motor driven by the inverter circuit, a load driven by the motor, Current detection means for detecting a drive current; and control means for controlling the inverter circuit according to an output signal of the current detection means to drive the motor in a sine wave. The control means is a peak current or rotation of the motor. Current setting means for setting a drive current corresponding to a magnetic field; motor current detected by the current detection means; or current comparison means for comparing a drive current signal corresponding to a rotating magnetic field with a setting signal of the current setting means; Voltage control means for controlling the output voltage of the inverter circuit from an output signal of the current comparison means; and output of the inverter circuit It comprises frequency setting means for setting the wave number and frequency correction means for correcting the output frequency of the frequency setting means from the output signal of the current comparison means, and the frequency correction means changes the control gain according to the motor drive frequency. Therefore, it is possible to prevent turbulence in V / f control, operate from the maximum load to no load, simplify the current detection means and motor control program, and realize an inexpensive and highly reliable motor drive device. .

  In the second invention, the frequency correction means for correcting the output frequency in the first invention is such that the control gain increases as the inverter circuit output frequency increases, and the control gain can be increased as the rotational speed increases. Therefore, it is possible to prevent turbulence in a high speed region and to control stable rotation even at high speed. In particular, this is effective for stabilization control during high-speed operation of the salient pole permanent magnet motor (IPMSM).

  In the third invention, the frequency correction means for correcting the output frequency in the first invention is such that the control gain is set to zero when the motor is started or is not corrected, and the frequency is not corrected when the motor is started at a low speed. Since current control or V / f control is performed, there is no frequency fluctuation at the start of the motor, so that a sine wave current flows through the motor and stable start is possible.

  According to a fourth invention, the current detection means in the first invention detects the peak value of the voltage of at least one shunt resistor connected between the inverter circuit and the DC power supply, and the at least one shunt Since the current can be controlled by the resistor, the control circuit is simplified, and an inexpensive and highly reliable motor drive device can be realized.

(Embodiment 1)
FIG. 1 shows a block diagram of a motor drive apparatus according to Embodiment 1 of the present invention.

  In FIG. 1, a DC power source 2 is constructed by applying AC power from an AC power source 1 to a DC power source circuit composed of a rectifier circuit, and DC power is converted into three-phase AC power by a three-phase full-bridge inverter circuit 3. 4 is driven. In the DC power supply 2, capacitors 21a and 21b are connected in series to the DC output terminal of the full-wave rectifier circuit 20, and the connection point of the capacitors 21a and 21b is connected to one terminal of the AC power supply input to constitute a voltage doubler rectifier circuit. The voltage applied to the inverter circuit 3 is increased to reduce the current, thereby reducing the inverter circuit loss. The motor 4 drives a motor load 5 such as a fan or a pump. By connecting the current detection means 6 to the negative voltage side of the inverter circuit 3 and detecting the current flowing through the inverter circuit 3, it corresponds to the output current of the inverter circuit 3, that is, the peak current Ip of the motor 4 or the rotating magnetic field. Drive current is detected.

  The current detection means 6 is a so-called one-shunt method current detection method, and is a shunt resistor 60 connected to the emitter terminal side of the lower arm transistor of the inverter circuit 3 and a current detection circuit that detects a current flowing through the shunt resistor 60. 61. The current detection circuit 61 includes a signal level conversion circuit and a peak current detection circuit for A / D conversion and current detection by a processor such as a microcomputer. Details will be described later.

  In the 1 shunt method, when the carrier frequency is high or the modulation degree becomes large, a current undetectable region appears. Therefore, the 3 shunt method is more suitable for detecting an instantaneous current corresponding to each phase. Although excellent, in the present invention, since the peak value of the motor sine wave current or the current corresponding to the rotating magnetic field is detected, the circuit configuration is simpler with the one shunt method. Of course, there is no problem with the three-shunt method.

  The control means 7 controls the output voltage of the inverter circuit 3 so that the peak current Ip of the motor 4 or the drive current corresponding to the rotating magnetic field becomes a set value, and the current setting means for setting the motor drive current I. The output signal Δi of the current comparison means 71 that compares the output signal ips 70 and the output signal ip of the current detection means 6 is added to the output voltage control means 72 to control the inverter output voltage control signal Vδ. The inverter output voltage control signal Vδ is applied to the inverter circuit control means 73, and the three-phase output voltage of the inverter circuit 3 is PWM controlled to drive a sine wave.

  The frequency setting means 74 sets a motor drive current frequency, and is set to a drive frequency f corresponding to the number of poles p and the number of rotations n of the rotor. The applied voltage Va of the motor 4 may be substantially equal to the motor induced voltage Vm or a voltage slightly higher than the motor induced voltage Vm. Therefore, the V / f control means 75 causes the motor rotation speed n, that is, the inverter to be applied. A voltage Vf (Vf = Ke × f) substantially proportional to the circuit drive frequency f is applied to the output voltage control means 72, and the inverter output voltage control signal Vδ is controlled in accordance with the motor speed. Vδ is obtained from Equation 1.

  That is, the motor voltage control signal Vδ is obtained by adding the proportional integral control value of the current error signal Δi to the V / f control voltage Vf, and is feedback controlled so that the motor drive current I becomes the set value Ips. If the proportional gain Kp is increased, the noise is weakened and oscillation tends to occur. Therefore, the noise resistance performance is improved by decreasing the proportional gain Kp and increasing the integral gain Ki.

  The output signal Δi of the current comparison means 71 and the output signal f of the frequency setting means 74 are applied to the frequency correction means 76, and the output signal f1 of the frequency correction means 76 is applied to the phase generation means 77.

  The phase generation unit 77 generates a phase signal θ corresponding to the output signal f1 of the frequency correction unit 76, and the phase θ is obtained from the integral value of the angular frequency ω or the product ωt of the time t and the angular frequency from the zero phase. And a phase signal is added to the inverter circuit control means 73 to perform PWM control. The motor phase control signal is obtained from Equation 2.

  The frequency correction means 76 corrects and changes the inverter drive frequency in proportion to the current error signal Δi and performs stabilization control, and performs a turbulence prevention operation. That is, according to Equation 3, when the motor current ip increases from the set value ips (because Δi becomes negative), the drive frequency is decreased, and when the motor current ip decreases from the set value ips (because Δi becomes positive), the reverse occurs. Increase the drive frequency.

  The frequency correction unit 76 includes a proportional integration unit 76a and an addition unit 76b. The proportional integration unit 76a obtains the product of the current error signal Δi and the output signal f of the frequency setting unit 74 by a proportional constant k, and adds the addition unit 76b. Adds the output signal of the proportional integration unit 76a to the output signal f of the frequency setting means 74 to obtain the inverter drive frequency f1. Kf shown in Equation 3 is equal to the product of the output signal f of the frequency setting means 74 and the proportionality constant k (Kf = f · k). Formula 3 means that the frequency control gain Kf increases in proportion to the set frequency f.

  FIG. 2 is a control vector diagram of the surface magnet synchronous motor showing the present invention, and shows the relationship between the magnet axis dq coordinates of the motor and the motor applied voltage γ-δ coordinates. The motor applied voltage coordinate (γ-δ coordinate) is advanced by the load angle δ from the dq coordinate, the motor applied voltage Va is equal to the δ-axis voltage, and only the δ-axis is controlled, so Va = Vδ and Vγ = 0. Therefore, the reverse coordinate transformation is unnecessary. The motor induced voltage Em is on the q axis, and the vector of the motor current I is set to be substantially equal to the q axis current Iq at the rated load. In FIG. 2, the motor current vector I is displayed with a phase γ delay from the q axis. The phase of the motor applied voltage Va and the current I is indicated by φ.

  The present invention controls the motor current vector I to a set value, and means to control the peak values of the motor phase currents Iu, Iv, Iw. Further, since the magnetic flux Ψ of the rotating magnetic field is a product of the inductance L and the current I, that is, Ψ = L · I, controlling the current I means controlling the rotating magnetic flux Ψ constant. It is obvious that the effective values are not limited to the peak values of the motor phase currents Iu, Iv, and Iw.

  In the conventional method, that is, when the δ-axis current Iδ is controlled to a predetermined value, the δ-axis current Iδ fluctuates due to load fluctuations, so it is necessary to change the Iδ setting value according to the load state. When the motor current (or the motor peak current Ip) is controlled to be constant, the load angle δ and the phase φ change automatically according to the load, so there is a feature that it is not necessary to change the current value.

  Even when the reactive current component Iγ is controlled to be constant, the load angle δ automatically changes from the rated load to no load, and it does not need to change the reactive current setting value, but operates stably, but the current vector I or current peak value The method of controlling Ip is easy to stabilize. In particular, in the reactive current Iγ constant control, in the advance control in which the load increases and the current vector I increases, the reactive current value Iγ decreases and the reactive current component change rate corresponding to the torque fluctuation decreases. It becomes difficult. However, in the present invention for controlling the current vector I or the current peak value Ip, the effect of torque fluctuation or load angle fluctuation appears directly as fluctuations in the current vector I or peak current Ip, which is excellent for detecting load fluctuations. It has a feature that is excellent in stabilization against load fluctuations. In particular, in the advance angle control in which the current vector I is advanced from the q axis, the I control is more advantageous than the Iδ control and the Iγ control because the torque fluctuation becomes more remarkable.

  If the current setting value ips is set according to the load, high-efficiency operation can be performed. Therefore, it may be changed according to the load torque and the rotational speed. However, according to the present invention, if the set value of the current setting means 70 is made constant from the time of start-up, the start-up torque can be increased, and further, stable operation can be achieved even if the motor current is controlled constant up to the advance angle control region.

  To further explain the frequency correction operation, if the motor current ip increases from the set value ips, the error signal Δi becomes a negative value, and the frequency correction signal Δf (Δf = Kf × Δi) also becomes a negative value, so that f1 = The corrected frequency f1 is lowered by the control of f + Δf, the γ-δ axis approaches the dq axis, the load angle δ decreases, and the motor current I decreases, so that the constant current control operation is performed. Turbulence occurs when there is torque fluctuation in V / f control, but by adding frequency control, the disturbance is suppressed and the rotational speed fluctuation is extremely reduced. Is done. Furthermore, since the frequency correction signal Δf increases in proportion to the set frequency f, the control gain increases as the frequency increases. Since the motor coil voltage vector ωLI in the vector diagram of FIG. 2 is proportional to the frequency, it is considered that the present invention corrects the motor drive frequency in proportion to the motor coil voltage vector. In other words, the motor coil voltage vector ωLI increases as the drive frequency increases, and the motor coil voltage vector ωLI and the load angle δ are correlated, so that the load angle fluctuation, that is, the turbulence phenomenon is also proportional to the motor coil voltage vector ωLI. . Therefore, it can be seen that the turbulence can be reduced by correcting the frequency in proportion to the motor coil voltage vector ωLI.

  FIG. 3 shows a startup control method of the motor applied voltage and frequency and the frequency correction gain.

  So-called V / f control is performed to linearly increase the applied voltage and drive frequency from the start of operation to the target rotational speed, and the frequency correction gain Kf is also increased in proportion to the drive frequency. The motor current set value may be constant. In the case of a fan or pump load, high-efficiency operation control can be performed by changing the current set value according to the drive frequency. The startup time ts until the target rotational speed is reached can be reduced by changing the start time ts according to the moment of inertia of the load. That is, as the moment of inertia increases, the turbulence can be reduced by increasing the startup time ts.

  By changing the frequency correction gain Kf in accordance with the drive frequency, the motor rotation speed fluctuation at the low start-up speed can be reduced, and the start-up current can be increased by bringing the motor current close to a sine wave. As shown in Kfa of FIG. 3, the control gain is not increased linearly in proportion to the time, the control gain in the initial stage of startup is set to almost zero, and the control gain is increased during the startup as shown in Kfb. Good. Note that if the proportional control gain is increased, oscillation easily occurs and noise is weakened. Therefore, a low-pass filter and a limiter may be appropriately provided.

  FIG. 4 shows a detailed embodiment of a single shunt type peak current detection circuit according to the present invention.

  Fig. 4 shows the detection of the motor peak current using an operational amplifier and a peak hold circuit. If the single shunt method is used, the number of parts for current detection is small, and only one high-speed A / D conversion unit is required. Furthermore, the setting of A / D conversion timing is flexible.

  The peak value of the voltage vn generated in the shunt resistor 60 corresponds to the peak value of each phase output current of the inverter circuit 3. Since an A / D conversion circuit built in a processor such as a microcomputer operates within a predetermined DC voltage range, it needs to be amplified and level-shifted so as to change within the DC voltage range. In other words, the motor current signal peak value may be set to change within the input dynamic range of the A / D conversion circuit. Hereinafter, the peak current detection circuit 61a will be described in detail.

  The capacitor 600 is connected in parallel with the shunt resistor 60, the shunt resistor 60 and the resistors 601 and 602 are connected in series, and the resistor 601 is pulled up to the DC power source (Vcc) of the peak current detection circuit 61a. A connection point between the resistor 601 (resistance value R1) and the resistor 602 (resistance value R2) is connected to the + input terminal of the high-speed operational amplifier 603, and a feedback resistor 604a (resistance value) is connected between the output terminal and the −input terminal of the high-speed operational amplifier 603. R4) is connected, and a resistor 604b (resistance value R3) is connected between the negative input terminal and the ground potential and used as a non-inverting amplifier. The terminal voltage vn of the shunt resistance resistance Ro is vn = Ro × I where I is the current, the voltage dividing ratio k of the resistors 601 and 602 is k = R2 / (R1 + R2), and the feedback gain K is K = R4. / R3, where the output diode voltage drop of the high-speed operational amplifier 603 is vd, the output voltage vp of the current detection circuit 61 is expressed by Equation 4.

  Here, if the product of the voltage division ratio k and the feedback amplification factor K, that is, k × K = vd / Vcc, the diode voltage drop vd can be canceled and converted to a DC voltage signal vp corresponding to the motor peak current. Is done.

  The output diode 605, the output capacitor 606, and the parallel resistor 607 of the high-speed operational amplifier 603 constitute a peak hold circuit, and the parallel resistor 607 is for setting integration time. The diodes 608a and 608b are connected for overvoltage protection of the A / D conversion circuit of the processor connected to the output of the current detection circuit 61. In order to reduce the voltage drop of the output diode 605 or the influence of temperature characteristics, it is better to use a Schottky diode. Furthermore, in order to reduce the influence of the temperature characteristic of the output diode 605, a temperature compensating diode may be connected in series with the resistor 601 or 602.

  When the peak current detection circuit 61a shown in FIG. 4 is used, it is possible to detect the motor current peak value without being affected by the inverter switching noise by detecting the current at the peak or valley of the carrier signal. There is no need to detect current several times within one cycle of the carrier signal, and the current detection algorithm is simplified. In particular, the current detection timing control becomes complicated in the simultaneous driving of a plurality of motors, and in some cases, there is a problem of being affected by the switching noise of other inverter circuits at the time of current detection. Setting to the peak or valley of the carrier signal eliminates the influence of switching noise.

  As described above, the present invention controls the motor current corresponding to the motor peak current or the rotating magnetic flux to a predetermined value in order to drive the permanent magnet motor by sensorless sine wave by V / f control. The constant output current is controlled by controlling the motor applied voltage and the motor drive frequency according to the error signal, and the motor drive frequency control gain is changed according to the set frequency to prevent and stabilize the turbulence. According to the present invention, stabilization control is possible regardless of a saliency motor or a non-saliency motor, and advance angle control is also facilitated. Even if the output current setting value is kept constant from the start, the stable operation from the start to the rating is performed, and the saliency motor has a feature that it automatically changes from the delay angle to the advance angle and operates stably.

  Hereinafter, a second embodiment of the present invention will be described with reference to FIGS.

(Embodiment 2)
FIG. 5 shows a detailed example of the current detection circuit of the motor drive apparatus according to Embodiment 2 of the present invention.

  FIG. 5 shows a current detection level conversion circuit 61b, which is obtained by removing the peak hold circuit from the peak current detection circuit 61a shown in FIG. An embodiment will be described in which the output signal of the level conversion circuit 61b is added to an A / D converter with a built-in processor such as a microcomputer, and the motor current signal after A / D conversion is detected as a peak current by a program of the processor.

  6 and 7 show the shunt resistance voltage waveform and current detection timing during two-phase modulation.

  6 and 7, c represents a triangular wave carrier signal, and vu and vv represent u-phase and v-phase modulated signals, respectively. Since the w-phase lower arm transistor is forced to conduct, the w-phase modulation signal is not shown.

  There are two types of patterns in which the motor peak current appears in the two-phase modulation as shown in FIGS. 6 and 7. In FIG. 6, the motor current peak value appears at the peak timing (zero vector) t3 of the carrier signal c. It can be detected. Therefore, it is preferable to compare the current signals after A / D conversion at the carrier signal peak time t3 during one cycle of the motor driving frequency and set the maximum value as the motor current peak value. However, since the carrier signal peak detection method can obtain data only once in one cycle, there is a problem that the reliability of data is lowered when the carrier frequency is low or low.

  If not only the carrier signal peak but also the peak value of the current pattern shown in FIG. 7 is detected, the current peak value detection accuracy is improved. That is, after the carrier signal peak time t3, the current may be detected at a timing t5 between t4 and t6 when the PWM control level comparator operates.

  FIG. 8 shows a flowchart of a peak current detection program according to the present invention.

  From step 100, the carrier signal interrupt processing program starts in synchronization with the peak t3 of the carrier signal c, and in step 101, A / D conversion input processing is executed. Next, the process proceeds to step 102 where the A / D converted current signal is stored in the data memory I. Next, the process proceeds to step 103 where the current signal I is compared with the maximum value Ip. Let the signal I be the maximum value Ip. Next, the routine proceeds to step 105 where a comparison interrupt signal between the carrier signal c and the modulation signal vu or vv is permitted. Next, the routine proceeds to step 106, and if there is a comparison interrupt signal, the routine proceeds to step 107, where a predetermined time of several microseconds is delayed, and then the routine proceeds to step 108 where A / D conversion is performed to detect a current. The processor is configured to generate a comparison interrupt signal at time t4 in FIG. Also, a processor that automatically performs A / D conversion processing with a predetermined time delay from time t4 can be realized. Next, the process proceeds to step 109 to store the A / D converted signal in the data memory I. In step 110, the current value I is compared with the maximum value Ip. If the current value I is greater than the maximum value Ip, the step Proceeding to 111, the current value I is replaced with the maximum value Ip. The above operation is executed for each carrier signal, the obtained maximum value Ip is compared with the set value Ips, and the applied voltage and the drive frequency are changed.

  The maximum value may be obtained and updated every cycle of the driving frequency.

  As described above, since peak current detection can be obtained from program processing, there is a problem that high-speed A / D conversion is required. However, the number of components is larger than that of the external circuit method including the peak holder shown in FIG. There is a feature that detection accuracy is improved and detection accuracy is improved. The two-phase modulation has been described above, but the same applies to the three-phase modulation.

  As described above, according to the present invention, the motor applied voltage is controlled so that the motor peak current or the motor drive current corresponding to the motor rotating magnetic field becomes the set value. Accordingly, the phase relationship between the γ-δ axis and the dq axis becomes a predetermined value corresponding to the load, and sensorless sine wave driving is possible. In particular, even if the motor load fluctuates greatly from no load to the rated load, it is possible to operate with only constant current control, so the control is very simple. In addition, by changing the frequency correction gain for correcting the drive frequency by the current error signal according to the drive frequency, the high-speed region irregularity is suppressed, and the rotational speed variation becomes extremely small. In addition, since there is no need to perform coordinate conversion or vector decomposition of the motor current, there is a feature that the control program is simplified and the motor control is easy even with an 8-bit microcomputer because almost no calculation is required.

  The present invention uses V / f control that does not estimate the rotor position and uses almost no motor parameters. Furthermore, since the rotation speed is open loop control, fluctuations in the rotation speed are very small, the control method is simple, and current detection is simple and low. Noise, low cost, high reliability motor drive device can be realized. In particular, control is possible regardless of saliency motors and non-saliency motors, the advance angle control is easy, the motor control program and current detection are simplified, and the burden on the processor is reduced. It can be applied to a compressor motor, a washing motor, and a drying fan motor simultaneous sine wave drive system, and an inexpensive and highly reliable multiple motor simultaneous drive device can be realized.

  As described above, the motor drive device of the present invention detects the motor current corresponding to the peak value of the motor current or the rotating magnetic field by driving the permanent magnet motor with the inverter circuit that converts the DC power into the AC power. The inverter circuit output voltage and the motor drive frequency are controlled so as to be set to the set value, so that it can be applied to most motor drive devices that drive permanent magnet motors. The present invention can be applied to a motor driving device for a vacuum cleaner, a motor driving device for a vacuum cleaner, a fan motor driving device such as a ventilation fan or a combustor, and a compressor motor driving device for an air conditioner or a refrigerator. Furthermore, the present invention can also be applied to a multiple motor simultaneous drive system such as a heat pump washer / dryer or an air conditioner.

The block diagram of the motor drive device in an embodiment of the invention Motor control vector diagram of motor drive device according to the present invention The graph of the starting control method of the motor drive unit in an embodiment of the invention Peak current detection circuit diagram of motor drive device in the embodiment of the present invention Current detection circuit diagram of motor drive device according to second embodiment of the present invention Graph of shunt resistance voltage waveform and current detection timing in the second embodiment of the present invention Graph of shunt resistance voltage waveform and current detection timing in the second embodiment of the present invention Flowchart of current detection program in the second embodiment of the present invention

Explanation of symbols

2 DC power supply 3 Inverter circuit 4 Motor 5 Motor load 6 Current detection means 7 Control means 70 Current setting means 71 Current comparison means 72 Output voltage control means 74 Frequency setting means 76 Frequency correction means

Claims (4)

DC power source, inverter circuit for converting DC power of the DC power source into AC power, a permanent magnet motor driven by the inverter circuit, a load driven by the motor, and a current for detecting a driving current of the motor And a control means for controlling the inverter circuit by an output signal of the current detection means to drive the motor in a sine wave. The control means is a drive current corresponding to a peak current of the motor or a rotating magnetic field. Current setting means for setting the motor peak current detected by the current detection means, or current comparison means for comparing the drive current signal corresponding to the rotating magnetic field with the setting signal of the current setting means, and the output of the current comparison means Voltage control means for controlling the inverter circuit output voltage from the signal, and setting the output frequency of the inverter circuit The motor driving system comprises a frequency setting unit and a frequency correcting unit for correcting the output frequency of the frequency setting unit from the output signal of the current comparing unit, and the frequency correcting unit changes the control gain according to the motor driving frequency. apparatus. 2. The motor driving apparatus according to claim 1, wherein the frequency correction means for correcting the output frequency increases the control gain as the inverter circuit output frequency increases. The motor drive apparatus according to claim 1, wherein the frequency correction means for correcting the output frequency sets the control gain to zero or does not correct the frequency when the motor is started. 2. The motor driving apparatus according to claim 1, wherein the current detecting means detects a peak value of a voltage of at least one shunt resistor connected between the inverter circuit and the DC power source.

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