JP2008104275A - Constant current controlled dc-dc converter circuit with function for interrupting no-load oscillation - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To inexpensively achieve a function for interrupting a no-load oscillation which interrupts a switching operation when the load is disconnected and an output is abnormally increased in a constant current controlled flyback DC-DC converter circuit. <P>SOLUTION: When a primary current of a transformer T1 is turned off, an increase in a voltage of an output capacitor C5 is reflected on an increase in a voltage of a capacitor C23 for a control power supply by charging the capacitor C23 for the control power supply in an oscillation control circuit IC1 from a secondary flyback winding n4 through a diode D71, and an overvoltage protection circuit embedded in an integrated circuit for composing the oscillation control circuit IC1 interrupts an oscillation. While the oscillation is interrupted in the oscillation control circuit IC1, an input DC power supply Vdc continuously supplies latching power to the capacitor C23 for the control power supply so as to prevent a protection operation from being released in the overvoltage protection circuit. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a constant current control type DC-DC converter circuit having a no-load oscillation stop function, and is used for dimming control of an LED module according to a dimming signal.

  FIG. 6 is a circuit diagram of an LED lighting device capable of phase control. A series circuit of an AC power supply 4 and a phase control type dimmer 3 is connected between the AC input terminals of the power supply device 1. The phase control dimmer 3 includes a phase control element 31 such as a triac, a phase control circuit 32 that starts the phase control element 31 at a predetermined phase angle from the zero cross point, and a variable control of the ignition phase angle. A light control operation unit 33 is provided. After firing at a predetermined firing phase angle during a half cycle, the phase control element 31 continues to conduct for a predetermined phase angle until the half cycle ends. In this specification, the period from the start of the half cycle until the phase control element 31 is fired is referred to as the firing phase angle, and the conduction period (on period) from when the phase control element 31 is fired until the half cycle is completed. Is called the conduction phase angle.

  The power supply device 1 outputs a direct current having a magnitude corresponding to the conduction phase angle of the phase-controlled AC voltage between the DC output terminals. For example, the LED module 2 is connected between the DC output terminals. The LED module 2 is a module in which a plurality of LEDs are connected in series. The direct current output from the power supply device 1 is variably controlled by the phase control type dimmer 3 so that the output light of the LED module 2 is dimmed. Is done.

  As a conventional example of the LED lighting device capable of phase control, Patent Document 1 (Japanese Patent Laid-Open No. 2004-273267) discloses a phase control in which a conduction phase angle is controlled by a phase control dimmer connected in series to an AC power source. A rectifier circuit that rectifies an AC voltage and converts it into a pulsating DC voltage, a smoothing capacitor that smoothes the output of the rectifier circuit, and a smoothing capacitor A DC power supply circuit that steps down the DC voltage smoothed by the switching element to a DC voltage of a desired voltage value, and an LED lighting circuit that lights a plurality of LEDs using the output of the DC power supply circuit as a power source, By making the output of the DC power supply circuit substantially the same voltage as the ON voltage of the plurality of LEDs, and detecting the AC voltage phase-controlled by the phase control element Detection means for detecting the ON period of the phase control element, and blocking means for cutting off current supply from the DC power supply circuit to the LED lighting circuit during the OFF period of the phase control element based on the detection result of the detection means An LED lighting device is disclosed. According to Patent Document 1, it has been proposed to use a one-stone type switching power source configured using an insulating transformer as the DC power source circuit.

  Although the LED lighting device of Patent Document 1 has a simple and inexpensive configuration, it can supply a stable voltage to the LED lighting circuit over the entire phase angle of the phase control element, so that the LED is controlled using a phase control type dimmer. There is an advantage that light can be dimmed stably.

  However, the technique of Patent Document 1 has a problem that the output light flickers because the LED blinks every half cycle of the AC power supply. Moreover, as a DC power supply circuit for supplying power to the LED lighting circuit, a one-stone switching power supply configured using an insulating transformer is used, but it is necessary to intermittently output current of the switching power supply every half cycle of AC voltage. Therefore, as the output control method, constant voltage control is adopted instead of constant current control. Therefore, there is a possibility that the lamp current varies with respect to the LED on-voltage variation.

  Therefore, if the ON period of the phase control element is detected every half cycle of the AC voltage, and the switching power supply is controlled at a constant current according to the detected output, the lamp current can be made constant, so that flickering of the output light can be prevented. It is considered that the lamp current can be controlled with high accuracy even with respect to the variation in the on-voltage of the LED.

  As a conventional example of a switching power supply for controlling a lamp current flowing in an LED lighting circuit at a constant current, Patent Document 2 (Japanese Patent Laid-Open No. 2004-39288) discloses a circuit as shown in FIGS. Has been. This conventional example includes a switching element Tr1 that is switched at a high frequency, a control circuit 5 that controls ON / OFF of the switching element Tr1 at a high frequency, and an inductance element L1 that is connected to an input DC power supply Vdc via the switching element Tr1. A diode D10 connected in series with the inductance element L1 with a polarity that conducts when the switching element Tr1 is OFF, and a smoothing capacitor C5 connected to the inductance element L1 and the DC power source Vdc via the diode D10, A current detection resistor R35 inserted between the capacitor C5 and the LED module 2 is provided, and the ON duty of the switching element Tr1 by the control circuit 5 is set so that the detection value of the current detection resistor R35 approaches the target value. Feedback control As shown in FIG. 7B, the control circuit 5 is a PWM control circuit including an error amplifier (E / A) 5a, an oscillator 5b, and a comparator 5c. The switching element Tr1 is switched at a high frequency (several tens to several hundreds kHz) so that the terminal voltage is constant.

  In order to limit the upper limit value of the output voltage at no load, the input terminal of the control circuit 5 is connected to the high potential side of the output capacitor C5 via the Zener diode ZD6 and the resistor R39. When the LED is disconnected, no current flows through the current detection resistor R35, so the current detection signal fed back to the input terminal of the control circuit 5 through the diode D11 and the resistor R13 becomes zero, and the control circuit 5 increases the output. As described above, the ON duty of the switching element Tr1 is increased. Therefore, although the output voltage rises, the Zener diode ZD6 is turned on when the output voltage becomes equal to or higher than the Zener voltage, so that the output voltage is limited to approximately the Zener voltage.

Thus, in the technique of Patent Document 2, it is possible to eliminate variations in light output due to variations in the forward voltage of the LEDs by controlling the constant current to flow through the LEDs using a switching power supply, When the LED is disconnected, the output voltage can be increased within a predetermined value with a simple configuration.
JP 2004-273267 A JP 2004-39288 A

  In Patent Document 2 described above, although the output voltage can be limited to a predetermined value when the LED is disconnected, a state in which a no-load secondary voltage higher than the normal output voltage is output continues. In addition, the switching element Tr1 is continuously turned on and off, and the output capacitor C5 is continuously applied with a higher voltage than usual. Therefore, when such a no-load state continues for a certain period of time or longer, it is desirable to stop the operation of the switching power supply to avoid power loss and reduce stress on the circuit elements.

  In the technique of Patent Document 2, since a battery for automobiles is assumed as the DC power supply Vdc, a boost chopper circuit is used as a switching circuit. However, when an AC voltage controlled in phase as shown in FIG. 6 is used as an input power supply, it is desired to employ a step-down DC-DC converter using an insulating transformer, as shown in Patent Document 1. . In that case, since the switching element is disposed on the primary side of the insulating transformer, a means for transmitting an abnormal voltage rise generated on the secondary side of the insulating transformer to the primary side is required. If a photocoupler is used, an abnormal signal can be transmitted. However, the photocoupler transmits the detected value of the output current from the secondary side to the primary side, or the detected value of the conduction phase angle of the AC voltage from the primary side to the secondary side. In addition, providing a dedicated photocoupler for LED disconnection detection complicates the configuration of the device and increases the product cost.

  The present invention has been made in view of the above-described points, and uses a characteristic unique to a flyback type DC-DC converter to add a no-load oscillation stop function without increasing the cost. This is the issue.

  In order to solve the above-mentioned problem, the invention according to claim 1 includes a switching element Q8 (see FIG. 3) that is switched at a high frequency and a high-frequency switching element Q8 as shown in FIGS. The oscillation control circuit IC1 that performs ON / OFF control at the time, the transformer T1 in which the primary windings n1 and n2 are connected to the input DC power source (smoothing capacitor C2) via the switching element Q8, and the switching element Q8 when the switching element Q8 is OFF A first diode D1 connected in series with the first secondary winding n5 of the transformer T1 with a conducting polarity, and the first secondary winding n5 of the transformer T1 via the first diode D1 The connected first capacitor C5, the current detection circuit 23 inserted between the first capacitor C5 and the DC load 2, and the detection value of the current detection circuit 23 are close to the target value. In a constant current control type flyback DC-DC converter circuit comprising a feedback control circuit 22 that variably controls the ON duty of the switching element Q8 by the oscillation control circuit IC1 in the direction to which the switching element Q8 is turned off The second diode D71 connected in series with the second secondary winding n4 of the transformer T1, and the second secondary winding n4 of the transformer T1 via the second diode D71. The second capacitor C23 and a protection circuit (the comparator 55 and the latch circuit 56 in FIG. 3) for holding the oscillation operation of the oscillation control circuit IC1 in a stopped state when an abnormal increase in the voltage Vcc of the second capacitor C23 is detected. It is characterized by comprising.

  According to a second aspect of the present invention, in the first aspect of the present invention, the second capacitor C23 is a smoothing capacitor for supplying control power to the oscillation control circuit IC1, and the protection circuit is an integrated circuit constituting the oscillation control circuit IC1. It is an overvoltage protection circuit built in.

The invention according to claim 3, characterized in that in the invention of claim 2, the integrated circuit constituting the oscillation control circuit IC1 is unprotect operation of the overvoltage protection circuit and the voltage of the second capacitor C23 is less than the set voltage V _UL Means (comparator 59 in FIG. 3), and the second capacitor C23 is supplied with power from the input DC power supply (smoothing capacitor C2) so as to be equal to or higher than the set voltage _VUL during the oscillation stop period of the oscillation control circuit IC1. It is characterized by being.

  According to the first aspect of the present invention, the flyback DC-DC converter is characterized in that when the switching element is OFF, the voltage of the first flyback winding is clamped to the voltage of the first capacitor via the first diode. By taking out the voltage obtained at the second flyback winding to the second capacitor through the second diode, the abnormal voltage rise on the secondary side can be detected from the primary side. Therefore, it is possible to add a no-load oscillation stop function with a simple configuration, and to realize a safe and low-cost constant current power supply.

  According to the second aspect of the present invention, since the time until the oscillation stops at the time of no load can be set by utilizing the delay of the voltage rise due to the capacity of the smoothing capacitor for supplying the control power, the oscillation stop function is inadvertently caused by noise or the like It can be prevented from operating.

  According to the third aspect of the present invention, it is possible to prevent repetition of oscillation stop and oscillation resumption when there is no load, and the stress of the circuit element can be reduced.

  A detailed circuit configuration of the first embodiment of the present invention is shown in FIGS. FIG. 1 shows a circuit configuration on the primary side, FIG. 2 shows a circuit configuration on the secondary side, and FIG. 3 shows an internal configuration of an integrated circuit IC1 (Shindengen MR1722) incorporating a switching element and a control circuit.

  The primary side circuit shown in FIG. 1 and the secondary side circuit shown in FIG. 2 are electromagnetically coupled at the transformer T1, and are optically coupled at the photocouplers PC1 and PC2, thereby preventing noise. The capacitors C3 and C4 are coupled at a high frequency. Therefore, the primary side ground line G1 and the secondary side ground line G2 are insulated in terms of direct current. The photocoupler PC1 is an insulating element for feeding back a feedback control signal from the secondary-side output control circuit 22 to the primary-side control integrated circuit IC1, and the photocoupler PC2 is detected by the primary-side phase detection circuit 16 It is an insulating element for transmitting the conducted phase angle to the arithmetic circuit 21 on the secondary side.

  The primary side circuit is provided with an input connector CON1, and the secondary side circuit is provided with an output connector CON2. As shown in FIG. 6, a series circuit of an AC power supply 4 and a phase control element 31 is connected to the input connector CON1 and a phase-controlled AC voltage is applied. A direct current load such as the LED module 2 is connected to the output connector CON2, and a direct current corresponding to the conduction phase angle of the phase-controlled alternating voltage is output.

  An AC input terminal of the full-wave rectifier DB in the rectifying / smoothing circuit 12 is connected to the input connector CON1 through the filter circuit 11. A smoothing capacitor C2 is connected to the DC output terminal of the full-wave rectifier DB via an inrush current prevention circuit 13. A smoothed DC voltage Vdc is obtained at the smoothing capacitor C2.

  The inrush current prevention circuit 13 includes a parallel circuit of a thermistor PTH and a thyristor Q5, and immediately after the power is turned on, the inrush current is prevented by charging the smoothing capacitor C2 via the thermistor PTH. When excitation of the primary windings n1 and n2 of the transformer T1 is started, the capacitor C17 is charged via the diode D8 and the resistors R21, R43, and R42 by the voltage of the primary winding n1, and the thyristor Q5 is turned on. As a result, power loss due to the thermistor PTH does not occur during normal operation.

  The filter circuit 11 includes a line filter LF1 and a noise preventing capacitor C1, and also includes a non-linear resistance element ZNR for preventing overvoltage and a fuse FS for interrupting overcurrent. In this circuit, the noise preventing capacitor C1 is connected by a resistor near the zero cross of the AC voltage so that the phase advance current flowing through the noise preventing capacitor C1 does not hinder the extinction of the phase control element 31 connected in series with the AC power supply 4. A phase correction circuit 17 for short-circuiting is provided. Further, a phase detection circuit 16 is provided for detecting the conduction phase angle of the phase-controlled AC voltage. Detailed configurations and operations of the phase correction circuit 17 and the phase detection circuit 16 will be described later.

  The DC voltage Vdc of the smoothing capacitor C2 is converted into a voltage by a DC-DC converter using the transformer T1 and the control integrated circuit IC1, and a DC output voltage Vout is obtained at the smoothing capacitor C5 of the secondary circuit (FIG. 2). In this DC-DC converter, the primary current supplied from the smoothing capacitor C2 to the primary windings n1 and n2 of the transformer T1 is intermittent at high frequency, and electromagnetic energy is accumulated in the transformer T1 during the period when the primary current is flowing. And a so-called flyback type DC-DC converter that charges the smoothing capacitor C5 via the diode D1 by the back electromotive voltage obtained in the secondary winding n5 of the transformer T1 during the period when the primary current is cut off. .

  Hereinafter, the circuit configuration and operation of the flyback type DC-DC converter will be described in detail. The positive electrode of the smoothing capacitor C2 is connected to the drain terminal (5th pin) of the integrated circuit IC1 through the primary windings n1 and n2 of the transformer T1. As shown in FIG. 3, the drain terminal and the source terminal of the switching element Q8 made of MOSFET are connected to the drain terminal (5th pin) and the ground terminal (3rd pin) of the integrated circuit IC1, respectively. The negative electrode of the smoothing capacitor C2 is connected to the ground line G1 of the primary circuit.

  The switching element Q8 is ON / OFF driven at a high frequency by a control circuit built in the integrated circuit IC1. When the switching element Q8 is turned ON, the path from the positive electrode of the smoothing capacitor C2 to the primary windings n1 and n2 of the transformer T1, the drain terminal / source terminal of the switching element Q8, the primary side ground line G1, and the negative electrode of the smoothing capacitor C2 Next current flows. The primary current increases linearly, and electromagnetic energy is accumulated in the transformer T1.

  The ON width of the switching element Q8 is determined by comparing the voltage of the ON width timer 52 with the voltage of the feedback terminal (second pin) by the comparator 51 in the integrated circuit IC1, and the latch circuit 53 is reset by the output of the comparator 51. Then, the switching element Q8 is turned off.

  When the switching element Q8 is turned OFF, the electromagnetic energy accumulated in the transformer T1 is released, and the smoothing capacitor C5 of the secondary side circuit is charged via the diode D1 by the counter electromotive voltage induced in the secondary winding n5. At this time, the smoothing capacitor C23 for supplying the control power supply voltage Vcc via the diode D71 is charged by the back electromotive voltage induced in the secondary winding n4.

  During the period when the switching element Q8 is OFF, the secondary current of the transformer T1 decreases linearly, and when the release of the electromagnetic energy accumulated in the transformer T1 is completed, the induced voltage of the secondary winding n4 disappears. To do. The induced voltage of the secondary winding n4 is input to the zero cross detection terminal (first pin) of the integrated circuit IC1 through the resistor R22. A capacitor C22 between the terminal and the ground is a capacitor for preventing malfunction. As shown in FIG. 3, a zero current detection circuit 54 is connected to the zero cross detection terminal (first pin) of the integrated circuit IC1, and when the disappearance of the secondary current of the transformer T1 is detected, the latch circuit 53 is detected. Is set and the switching element Q8 is turned on again.

  Thus, since the DC-DC converter of this circuit operates in what is called a boundary mode, the power conversion efficiency is increased. Further, a partial resonance capacitor C20 is connected in parallel between the drain terminal (5th pin) and the ground terminal (3rd pin), and the switching element Q8 is switched at a timing when the drain-source voltage of the switching element Q8 becomes substantially zero. Since the constant is set to turn ON, high efficiency is realized by reducing switching loss.

  Although the secondary windings n3 and n4 are secondary windings and are connected to the primary circuit, in this specification, the primary winding and the secondary winding of the transformer T1 are respectively input windings. It means a line and an output winding. When a primary side circuit or a secondary side circuit is used, a circuit connected to the ground line G1 on the input connector CON1 side is connected to the primary side circuit or the output connector CON2 side. A circuit connected to the ground line G2 is referred to as a secondary circuit.

  A power supply voltage Vcc for control is supplied from the smoothing capacitor C23 to the power supply terminal (4th pin) of the integrated circuit IC1. Three types of control power supply circuits for supplying the control power supply voltage Vcc to the smoothing capacitor C23 are provided. Of these, the first control power supply circuit is connected to the secondary winding n4 via the diode D71. Is a circuit that charges the smoothing capacitor C23 with the output of. The smoothing capacitor C23 is an electrolytic capacitor, and the capacitor C21 connected in parallel thereto is a small-capacitance capacitor for bypassing the high-frequency current from the secondary winding n4 through the diode D71. When the output of the DC-DC converter is large, the electromagnetic energy accumulated in the transformer T1 increases, so that the induced voltage of the secondary winding n4 also increases. Therefore, the control power supply voltage Vcc is supplied by the first control power supply circuit when the LED module 2 as the load is fully lit.

  The second control power supply circuit 14 is a circuit that charges the smoothing capacitor C23 through the diode D5, the transistor Q4, and the diode D7 by the output of the secondary winding n3. The number of turns of the secondary winding n3 is set larger than that of the secondary winding n4, and the induced voltage is large. The diode D5 connected in series to the secondary winding n3 is connected so as to have a polarity that conducts during the period when the primary current flows through the transformer T1, and the secondary winding n4 serves as a flyback winding. In contrast, the secondary winding n3 functions as a feedforward winding. Here, the secondary winding n4 which is a flyback winding charges the smoothing capacitor C23 with a voltage reflecting the output voltage Vout of the secondary side circuit, whereas the secondary winding which is a feedforward winding. n3 can charge the smoothing capacitor C23 with a voltage reflecting the input voltage Vdc (charging voltage of the smoothing capacitor C2) of the primary circuit. Therefore, even if the charging voltage of the smoothing capacitor C23 by the secondary winding n4 that is the flyback winding decreases during dimming of the LED module 2 that is the load, the secondary winding n3 that is the feedforward winding The control power supply voltage Vcc can be supplied by charging the smoothing capacitor C23.

  The second control power supply circuit 14 includes a cutoff circuit that cuts off the supply of the control power supply voltage from the secondary winding n3 while the control power supply voltage can be secured from the secondary winding n4. This cutoff circuit includes a transistor Q4, a Zener diode ZD1, and a bias resistor R9. The voltage Vcc of the smoothing capacitor C23 charged from the secondary winding n4 via the diode D71 is the Zener voltage of the Zener diode ZD1. When the voltage is higher, the base Q-emitter of the transistor Q4 is reverse-biased so that the transistor Q4 is cut off, and charging of the smoothing capacitor C23 by the second control power supply circuit 14 is prevented. On the other hand, when the voltage Vcc of the smoothing capacitor C23 becomes lower than the Zener voltage of the Zener diode ZD1 and the base-emitter of the transistor Q4 and the diode D7 are forward-biased, the transistor Q4 becomes conductive, and from the secondary winding n3. The smoothing capacitor C23 is charged through the diode D5, the transistor Q4, and the diode D7. The capacitor C29 is a small-capacitance capacitor for bypassing the high-frequency current from the secondary winding n3 via the diode D5.

  As described above, when the LED module 2 as a load is dimmed, the control power supply voltage is supplied from the second control power supply circuit 14, but the dimming becomes deeper and the phase-controlled alternating current is supplied. As the voltage conduction phase angle becomes narrower, the DC voltage Vdc of the smoothing capacitor C2 decreases. Then, the secondary winding n3, which is a feedforward winding, charges the smoothing capacitor C23 with a voltage reflecting the input voltage Vdc of the primary circuit, so that the control power supply voltage Vcc is also applied to the second control power supply circuit 14. There may be cases where sufficient supply is not possible.

  Therefore, in this circuit, a third control power supply circuit 15 that supplies a control power supply voltage from the DC output terminal of the rectifying and smoothing circuit 12 is provided. The third control power supply circuit 15 supplies the control power supply voltage from the DC output terminal of the rectifying and smoothing circuit 12 while the control power supply voltage can be secured from the secondary winding n3 that is the feedforward winding. It has a cut-off circuit to cut off. The cutoff circuit includes a transistor Q10, a Zener diode ZD4, and bias resistors R44 and R52. The Zener voltage of the Zener diode ZD4 is higher than the Zener voltage of the Zener diode ZD1 of the second control power supply circuit 14. It is set low. During the period when the control power supply voltage Vcc is supplied from the second control power supply circuit 14, the base Q and the emitter of the transistor Q10 are reverse-biased so that the transistor Q10 is cut off, and the third control power supply The smoothing capacitor C23 is prevented from being charged by the circuit 15. On the other hand, when the voltage Vcc of the smoothing capacitor C23 becomes lower than the Zener voltage of the Zener diode ZD4 and the base-emitter of the transistor Q10 and the diode D2 are forward-biased, the transistor Q10 becomes an unsaturated conductive state, and the rectifying and smoothing circuit The smoothing capacitor C23 is charged from the 12 smoothing capacitors C2 through the collector-emitter resistance of the transistor Q10. When the conduction phase angle of the phase-controlled AC voltage changes, the DC voltage Vdc of the smoothing capacitor C2 of the rectifying and smoothing circuit 12 varies greatly, but the DC voltage Vdc and the control power supply voltage Vcc are controlled by the collector-emitter resistance of the transistor Q10. Is absorbed, the control power supply voltage Vcc stabilized with reference to the Zener voltage of the Zener diode ZD4 is supplied.

  The third control power supply circuit 15 is also used to supply an initial power supply voltage to the control integrated circuit IC1 at the start of operation immediately after power-on. That is, before the operation of the control integrated circuit IC1, the input current does not flow through the transformer T1, and therefore the first control power supply circuit (diode D71) and the second control power supply circuit 14 are used as the control power supply. The voltage Vcc cannot be supplied. In this case, when the third control power supply circuit 15 supplies the initial power supply voltage, the control integrated circuit IC1 starts oscillating operation, and when the input current flows through the transformer T1, the second control power supply circuit 14 Start power supply. As a result, when the third control power supply circuit 15 is cut off, and when the first control power supply circuit (diode D71) starts supplying power, the second control power supply circuit 14 is cut off.

  The third control power supply circuit 15 is also used as a latch power supply for holding the control integrated circuit IC1 in the operation stop state after the protective operation by no-load detection. The protection operation by this no-load detection will be described later in connection with the constant current control described later, and the detection of the conduction phase angle of the phase-controlled AC voltage that is the basis of the constant current control will be described first.

  4A and 4B are waveform diagrams for explaining the operation of the phase detection circuit 16. FIG. 4A is a waveform of a phase-controlled AC voltage supplied to the input connector CON1, and FIG. 4B is a detection voltage Vo of the phase detection circuit 16. It is a waveform. In the figure, in the cycle from t1 to t4, the conduction start timing of the phase control element 31 is early, and the conduction period is long. On the other hand, in the cycle from t4 to t7, the conduction start timing of the phase control element 31 is late and the conduction period is short. Here, if the phase angle occupied by the conducting period (t2 to t3, t5 to t6) is referred to as the conducting phase angle as described above, the load current is controlled to increase as the conducting phase angle increases. Shall.

  The phase detection circuit 16 detects the conduction phase angle for one half wave of the phase-controlled AC voltage and does not detect the conduction phase angle for the other half wave. This is related to the fact that a high-precision arithmetic circuit 21 (see FIG. 2) including the microcomputer IC4 is arranged at the subsequent stage of the phase detection circuit 16. Under this assumption, the conduction phase angle for both positive and negative half-waves. Is detected, the light output fluctuates finely every half wave of the AC voltage, causing flickering of the light output, particularly at low output. As described above, when the high-precision arithmetic circuit 21 including the microcomputer IC4 is used, the optical output can be adjusted to, for example, 256 steps within a range of 5 to 100% with respect to the maximum output. However, especially in the region where the light output is low, even if there is a difference in the dimming level in one step, it is easily perceived by the human eye, and if the detected value of the conduction phase angle fluctuates every positive and negative half cycles, The light output will flicker.

  Originally, the conduction phase angle of the phase-controlled AC voltage should be controlled to be equal for each positive and negative half cycle. However, there are problems with the electrical characteristics of the phase control circuit 32 and the sensitivity of the phase control element 31. In some cases, the positive and negative conduction phase angles do not exactly match. Of course, it is conceivable to average the detected values a plurality of times by the averaging process of the microcomputer IC4. However, in this case, the averaging is performed even when the conduction phase angle is changed by the dimming operation. There is a risk of causing. Therefore, in this circuit, the conduction phase angle is detected for one half wave of the phase-controlled AC voltage, and the same conduction phase angle is detected for the other half wave until the next conduction phase angle is detected. By controlling the microcomputer IC 4 to keep the same detection value, the flickering of the light output is prevented.

  In order to detect the conduction phase angle with high accuracy, the phase detection circuit 16 uses a voltage detection circuit IC5 made of a reset IC (Rohm BD4943G) with high voltage detection accuracy. In the voltage detection circuit IC5, when the voltage applied between the voltage input terminal VDD (No. 2 pin) and the ground terminal (No. 3 pin) exceeds the reference voltage, the output terminal Vo (No. 1 pin) becomes H level. When the voltage is lower than the reference voltage, the output terminal Vo (the 1st pin) operates so as to become L level.

  One AC input terminal of the full-wave rectifier DB is connected to the anode of a diode D9, and the cathode of the diode D9 is connected to a resistance voltage dividing circuit composed of a series circuit of resistors R15, R18, and R19. The voltage dividing circuit is connected to the DC output terminal on the negative side of the full-wave rectifier DB via the ground line G1 of the primary circuit. A zener diode ZD2 for preventing overvoltage and a capacitor C11 for preventing noise are connected in parallel to the resistor R19. When an AC voltage is applied to the forward polarity of the diode D9, a voltage obtained by dividing the input AC voltage is applied to the resistor R19, and the output terminal Vo becomes H level in a period longer than the reference voltage of the voltage detection circuit IC5. The switching element Q2 made of MOSFET is turned on. Only during this period, a current flows from the control power supply voltage Vcc through the light emitting element of the photocoupler PC2, the resistor R33, and the switching element Q2, and the light receiving element of the photocoupler PC2 becomes conductive. Therefore, a rectangular wave signal having a pulse width corresponding to the conduction phase angle can be transmitted to the secondary side circuit.

  Next, in the secondary side circuit, a series circuit of the light receiving element of the photocoupler PC2 and the resistor R53 is connected between the control power supply voltage Vdd on the secondary side and the ground line G2 on the secondary side. A connection point between the light receiving element of the photocoupler PC2 and the resistor R53 is connected to a terminal TIO10 (16th pin) which is an input port of the microcomputer IC4 (NEC µPD78F9212) constituting the arithmetic circuit 21. Since the terminal TIO10 of the microcomputer IC4 is at the L level while the light receiving element of the photocoupler PC2 is conducting, the conduction phase angle can be measured by measuring the time at the L level. The microcomputer IC4 has a memory table that determines an output current to the LED module 2 according to the measured conduction phase angle, and outputs a set value of the output current read from the memory table to an output control circuit. The PWM signal is output to the terminal TOH1 (the 1st pin) to be supplied to the terminal 22. This PWM signal is a rectangular wave voltage signal of about 1 kHz, for example, and the capacitor C18 is charged / discharged by turning on / off the switching element Q6 of the output control circuit 22 with a duty ratio corresponding to the set value of the output current. A target value of output current is given as an analog potential of C18. A secondary control power supply voltage Vdd is supplied between the power supply terminal (5th pin) and the ground terminal (4th pin) of the microcomputer IC4.

  The output control circuit 22 includes an integrated circuit IC3 (New Japan Radio NJM2146) for constant current control. A secondary-side control power supply voltage Vdd is supplied between the power supply terminal (8th pin) and the ground terminal (4th pin) of the integrated circuit IC3. This integrated circuit IC3 includes two operational amplifiers having differential inputs between the 2-3rd pin and the 5-6th pin, respectively, and gives the detection value of the current detection circuit 23 to the former to provide a constant current. While controlling, the detection value of the voltage detection circuit 24 is given to the latter, and the overvoltage is detected. The integrated circuit IC3 includes a reference voltage generation circuit therein, and a reference voltage is output from the 7th pin. When the switching element Q6 is OFF, the capacitor C18 is charged via the resistors R4 and R56. When the switching element Q6 is ON, the capacitor C18 is discharged by the series circuit of the resistors R23 and R11 and the parallel circuit of the resistor R56. With this time constant circuit, a target value for constant current control is input to the input terminal (second pin) of the integrated circuit IC3.

  Of the two operational amplifiers built in the integrated circuit IC3, the pair of input terminals of the operational amplifier for constant current control are connected to the 2nd and 3rd pins, and the output terminal is connected to the 1st pin. The detection value of the current detection circuit 23 is input to the input terminal (third pin) of the integrated circuit IC3 via the input resistor R13, and feedback is provided to the output terminal (first pin) of the integrated circuit IC3. A series circuit of a resistor R6 and a capacitor C27 is connected as an impedance. The current detection circuit 23 is a current detection resistor R35 inserted in series between the output connector CON2 and the ground line G2, and corresponds to the current output from the smoothing capacitor C5 to the external LED module 2 via the output connector CON2. Generate voltage.

  The voltage detection circuit 24 includes a voltage dividing circuit composed of a series circuit of resistors R39, R40, and R41, and the divided voltage is compared with the reference voltage at the 7th pin by an operational amplifier between the 5th and 6th pins of the integrated circuit IC3. When the output voltage Vout becomes excessive, the output is controlled to be suppressed. The voltage dividing ratio of the voltage detection circuit 24 can be switched by the transistor Q7, and the output voltage Vout is set by switching the transistor Q7 to the ON state or the OFF state by the output port (14th pin) of the microcomputer IC4 of the arithmetic circuit 21. Can be changed.

  The operational amplifier between the 2nd and 3rd pins of the integrated circuit IC3 is connected to the output terminal (1st pin) of the integrated circuit IC3 in the direction in which the target value of constant current control set in the capacitor C18 and the detection value of the current detection circuit 23 match. ) Is variably controlled. As a result, a current flows from the control power supply voltage Vdd of the secondary circuit to the light emitting element of the photocoupler PC1 via the resistor R17 in accordance with the potential of the output terminal (first pin) of the integrated circuit IC3. The resistance value of the light receiving element of the photocoupler PC1 changes according to the value.

  In the primary circuit, the light receiving element of the photocoupler PC1 is connected to the feedback terminal (No. 2 pin) of the control integrated circuit IC1, and as shown in FIG. 3, the built-in constant current source passes through the No. 2 pin. As a result, a voltage value as a control target is generated in the parallel circuit of the resistor R29, the capacitor C24, and the light receiving element of the photocoupler PC1. In accordance with this voltage value, the control integrated circuit IC1 variably controls the ON width of the switching element Q8. Thus, the output current detected by the current detection circuit 23 is controlled to be a constant current determined according to the conduction phase angle of the AC voltage.

  By the way, the LED module 2 may come off due to poor contact of the output connector CON2 or disconnection of wiring. Further, since the LED module 2 has a built-in series circuit of a plurality of LEDs, one of the LEDs may break down and a lead wire may be disconnected. Then, the detection value of the current detection circuit 23 becomes zero, but since the output current is controlled to converge to the target value, the control integrated circuit IC1 tends to increase the ON width of the built-in switching element Q8. The voltage of the smoothing capacitor C5, which is controlled and is the output capacitor of the DC-DC converter, is abnormally boosted.

In order to prevent the abnormal boosting state due to such a no-load state from continuing, this circuit uses the overvoltage protection circuit of the control integrated circuit IC1. As described above, since the control integrated circuit IC1 is supplied with the control power supply voltage Vcc from the secondary winding n4 that is a flyback winding, the voltage of the smoothing capacitor C5 that is the output capacitor of the secondary side circuit. If the voltage rises abnormally, the voltage of the secondary winding n4 also rises reflecting the voltage rise, and as a result, the control power supply voltage Vcc rises. As shown in FIG. 3, the comparator 55 of the overvoltage protection circuit built in the integrated circuit IC1 compares the control power supply voltage Vcc of the 4th pin with the reference voltage V _OVP for overvoltage protection. When Vcc exceeds the reference voltage V _OVP for overvoltage protection, the output of the comparator 55 becomes H level. Thereby, the latch circuit 56 is set through the OR gate, and the switching element Q8 can be held in the OFF state by closing the gate 58 with a signal obtained by inverting the Q output by the inverting circuit 57.

The comparator 59 for releasing the latch compares the control power supply voltage Vcc and the reference voltage _VUL . When the control power supply voltage Vcc becomes lower than the reference voltage _VUL , the output of the comparator 59 becomes H level, and the latch circuit 56 It is reset and the overvoltage protection latch is released.

In this circuit, after the latch circuit 56 is set by the overvoltage protection circuit, as higher is retained than the control power supply voltage Vcc is the reference voltage V _UL, zener zener diode ZD4 of third control power supply circuit 15 The voltage is set. That is, when the latch circuit 56 is set by the overvoltage protection circuit and the oscillation of the DC-DC converter is stopped, the power supply by the first control power circuit (diode D71) and the second control power circuit 14 is Since it is stopped, the control power supply voltage Vcc, which has been overvoltage, decreases. At this time, if the output voltage of the third control power supply circuit 15 is too low, the oscillation operation is resumed when the control power supply voltage Vcc becomes equal to or lower than the reference voltage _VUL. Will repeat. To prevent this, in this circuit, by leaving a voltage of the third control power supply circuit 15 is set higher than the reference voltage V _UL unlatching, during power-up period so that the oscillation is unsuspended I have to.

  If it is a conventional fluorescent lamp inverter circuit, when the fluorescent lamp comes off or the filament breaks, it automatically detects when no load is detected and the inverter circuit oscillates and the user replaces the fluorescent lamp. The oscillation operation of the inverter circuit was resumed. This eliminates the need to turn on the power again, improving the user-friendliness. However, since the lifetime of LED modules is overwhelmingly longer than that of fluorescent lamps, it is not assumed that the user will replace the LED modules. Even if the LED module breaks down, the repairer will perform the replacement work after turning off the AC power supply. In this circuit, make sure that the overvoltage protection latch is not released until the AC power supply is turned off. Yes.

The time required for releasing the overvoltage protection latch is the time from when the AC power is turned off until the smoothing capacitor C2 is discharged, and thereby the charging voltage of the smoothing capacitor C23 becomes equal to or lower than the reference voltage _VUL. However, since it is considered that the time required for replacing the LED module is shorter, there is no problem in actual use.

  Next, the phase correction circuit 17 that is a feature of this circuit will be described. This phase correction circuit 17 prevents the noise prevention capacitor C1 near the zero cross of the AC voltage so that the phase advance current flowing in the noise prevention capacitor C1 does not hinder the extinction of the phase control element 31 connected in series with the AC power supply 4. Is a circuit that short-circuits with a resistor. As shown in FIG. 1, a noise prevention capacitor C1 is connected to the filter circuit 11 in parallel with the input connector CON1. Although the phase reactive current always flows through the capacitor C1, since it is a small-capacitance capacitor, if the current to the resistive load, which is an effective current, always flows, the influence of the reactive current can be ignored. . Therefore, it is conceivable to connect a resistive load in parallel with the capacitor C1, but in this case, a power loss due to the resistive load always occurs, resulting in a reduction in efficiency.

  If the current drawn into the full-wave rectifier DB always flows, it can be considered that the resistive load is substituted. However, in this circuit, since the capacitor input type smoothing circuit using the smoothing capacitor C2 is used, the phase-controlled AC voltage is used. Input current flows only near the peak. Then, in the vicinity of the zero cross of the AC voltage, the phase advance current flowing through the noise prevention capacitor C1 becomes dominant, and when the half cycle of the AC voltage is completed, the phase advance current of the next half cycle has already started to flow. If this phase advance current exceeds the holding current of the phase control element, the phase control element may not be extinguished, and in that case, the phase control element will be conductive for the entire period of the next half cycle. The light output increases rapidly during dimming output.

  Therefore, in this circuit, when the phase-controlled AC input voltage becomes smaller than the set voltage, the AC input terminals of the full-wave rectifier DB are short-circuited by resistors. In this way, since the resistive load is not connected during a period in which the phase-controlled AC input voltage is higher than the set voltage, power loss can be reduced, circuit efficiency can be increased, and the capacitor C1 flows. Since the resistive load is always connected in parallel with the capacitor C1 during the end of the half cycle in which the phase advance current is a problem, the phase control element 31 can be reliably extinguished.

  The phase correction circuit 17 connects the series circuit of the switching element Q1 and resistors R1, R25, and R27 to the output terminal of the filter circuit 11 via the diodes D3 and D4 and the negative-side diode (not shown) of the full-wave rectifier DB. Thus, a resistor R10 for supplying a gate bias voltage from the control power supply voltage Vcc is provided so that the switching element Q1 can be kept ON even near the zero cross of the AC voltage. In addition, a transistor Q3 for turning off the switching element Q1 and a gate charge discharging resistor R51 are provided, and a Zener diode ZD5 and resistors R7, R8, and R16 for turning on the transistor Q3 are provided.

5A and 5B are diagrams for explaining the operation of the phase correction circuit 17. FIG. 5A shows the output voltage of the full-wave rectifier DB, FIG. 5B shows the ON / OFF state of the transistor Q3, and FIG. 5C shows the ON / OFF state of the switching element Q1. , (D) is the current of the resistor R1. Here, conceptually ignoring the voltage drop of the diodes D3 and D4 and the full-wave rectifier DB, the output voltage of the full-wave rectifier DB is the Zener voltage V _ZD5 of the Zener diode _ZD5 + the base-emitter on-voltage of the transistor Q3. When V _BE3 is exceeded, the transistor Q3 is turned on and the switching element Q1 is turned off. During this period, the resistors R1, R25, and R27 are disconnected from the AC input terminal of the full-wave rectifier DB, so that no power loss due to the resistors occurs.

On the other hand, when the output voltage of the full-wave rectifier DB is lower than the base-emitter ON voltage V _BE3 of the Zener voltage V _ZD5 + transistor Q3 of the Zener diode ZD5, the transistor Q3 is turned OFF, the switching element Q1 is turned ON. In the vicinity of the zero cross of the AC voltage, a power source for maintaining the ON state of the switching element Q1 cannot be obtained on the AC input terminal side of the full-wave rectifier DB. However, in this circuit, the control power supply voltage Vcc for the DC-DC converter is By utilizing this, the ON state of the switching element Q1 is maintained. Therefore, in the vicinity of the zero crossing of the AC voltage where the phase advance current of the noise preventing capacitor C1 is a problem, the switching element Q1 is turned on, and a series circuit of resistors R1, R25, and R27 is connected between the AC input terminals of the full-wave rectifier DB. By short-circuiting, the phase advance current can be canceled and the phase control element 31 can be surely extinguished.

  Since the total of the effective current flowing through the resistors R1, R25, and R27 and the reactive current flowing through the capacitor C1 needs to be set to be lower than the holding current of the phase control element 31 in the vicinity of the zero cross, it is connected in parallel with the power source. It is necessary to set a relatively low resistance value as the resistor, but in this circuit, since the AC voltage is connected only during a low period, power loss does not increase. In the prototype, when the capacitance of the noise prevention capacitor C1 is 0.01 μF, it was confirmed that the resistor R1, R25, and R27 operate without problems by using 4.3 kΩ × 3.

  Finally, the control power supply circuit of the secondary side circuit will be described. The secondary side circuit has a high-precision arithmetic circuit 21 including a microcomputer IC4 and also requires a high-precision power supply voltage for analog signal transmission by the photocoupler PC1, so that the three-terminal regulator IC2 is more stable. A control power supply voltage Vdd is generated. The smoothing capacitor C26 on the input side of the three-terminal regulator IC2 is charged by the fourth control power supply circuit 25 or the fifth control power supply circuit 26. The fourth control power supply circuit 25 is a circuit that charges the smoothing capacitor C26 from the secondary winding n6 that is the feedforward winding of the transformer T1 via the resistor R31 and the diode D61. Used for dimming with relatively large output. The fifth control power supply circuit 26 is a circuit that charges the smoothing capacitor C26 through the transistor Q9 and the diode D6 from the charging voltage of the smoothing capacitor C5 that is the output capacitor of the DC-DC converter, and the dimming with a relatively small output. Sometimes used.

  The fifth control power supply circuit 26 includes a cut-off circuit that cuts off the supply of current through the diode D6 during a period when the fourth control power supply circuit 25 can sufficiently secure the charging current of the smoothing capacitor C26. This cutoff circuit includes a Zener diode ZD3, a transistor Q9, and a bias resistor R5. When the charging voltage of the smoothing capacitor C26 is higher than the Zener voltage of the Zener diode ZD3, the base-emitter of the transistor Q9 is reverse-biased. Transistor Q9 is turned off. Further, when the charging voltage of the smoothing capacitor C26 becomes lower than the Zener voltage of the Zener diode ZD3, and the diode D6 is forward-biased between the base and the emitter of the transistor Q9, the transistor Q9 becomes an unsaturated conductive state, and its collector- A charging current flows from the smoothing capacitor C5 to the smoothing capacitor C26 via the resistance between the emitters.

  The power supply device 1 of this circuit may be housed in the housing of the lighting fixture together with the LED module 2, or only the LED module 2 is housed in the housing of the lighting fixture, and the power supply device 1 is separate from the fixture body. It may be external.

It is a circuit diagram which shows the structure of the primary side of Embodiment 1 of this invention. It is a circuit diagram which shows the structure of the secondary side of Embodiment 1 of this invention. It is a circuit diagram which shows the internal structure of the integrated circuit used for Embodiment 1 of this invention. It is a wave form diagram for operation | movement description of the phase detection circuit of Embodiment 1 of this invention. It is a wave form diagram for operation | movement description of the phase correction circuit of Embodiment 1 of this invention. It is a circuit diagram which shows the use condition of Embodiment 1 of this invention. It is a circuit diagram of a conventional example, (a) is a circuit diagram of a constant current control type switching power supply circuit, (b) is a circuit diagram of the control circuit.

Explanation of symbols

C2 smoothing capacitor (for input DC power supply)
Q8 Switching element (IC1 built-in)
T1 transformer n5 first secondary winding D1 first diode C5 first capacitor (output capacitor)
n4 Second secondary winding D71 Second diode C23 Second capacitor (for control power supply)
IC1 Integrated circuit for control 22 Output control circuit 23 Current detection circuit 55 Comparator 56 Latch circuit

Claims (3)

A switching element that is switched at a high frequency, an oscillation control circuit that controls ON / OFF of the switching element at a high frequency, a transformer having a primary winding connected to an input DC power source via the switching element, A first diode connected in series with the first secondary winding of the transformer with a polarity that conducts when OFF, and a first diode connected to the first secondary winding of the transformer via the first diode 1 capacitor, a current detection circuit inserted between the first capacitor and the DC load, and the ON duty of the switching element by the oscillation control circuit is varied in a direction in which the detection value of the current detection circuit approaches a target value. In a constant current control type flyback DC-DC converter circuit comprising a feedback control circuit to control,
A second diode connected in series with the second secondary winding of the transformer with a polarity that conducts when the switching element is OFF, and the second secondary winding of the transformer via the second diode A no-load oscillation stop function comprising: a connected second capacitor; and a protection circuit that holds the oscillation operation of the oscillation control circuit in a stopped state when an abnormal rise in the voltage of the second capacitor is detected. Constant current control type flyback DC-DC converter circuit. 2. The second capacitor according to claim 1, wherein the second capacitor is a smoothing capacitor for supplying control power to the oscillation control circuit, and the protection circuit is an overvoltage protection circuit built in an integrated circuit constituting the oscillation control circuit. A constant current control type flyback DC-DC converter circuit with a no-load oscillation stop function. 3. The integrated circuit constituting the oscillation control circuit according to claim 2, further comprising means for canceling the protection operation of the overvoltage protection circuit when the voltage of the second capacitor becomes lower than a set voltage, and the second capacitor includes the oscillation control circuit. A constant current control type flyback DC-DC converter circuit with a no-load oscillation stop function, wherein power is supplied from an input DC power supply so as to be equal to or higher than the set voltage during an oscillation stop period of the circuit.

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