JP2008028530A - Wireless receiver provided with variation correction circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a wireless receiver provided with a correction function for correcting variations in a cut-off frequency of an anti aliasing filter for eliminating an aliasing noise at a pre-stage of an ADC (Analog-Digital Converter) and a channel filter for limiting the band to eliminate the effect of a received signal on adjacent channels particularly with respect to the wireless receiver in a wireless communication system. <P>SOLUTION: A wireless receiving circuit of the wireless receiver wherein an orthogonal demodulator comprising multipliers, an oscillator, and a phase shifter separates a received signal into I and Q phases and thereafter analog filters (anti aliasing filter, channel filter), the ADC next to them, and digital filters next to the ADC apply processing to the signal, includes: a means for directly inputting an output of the oscillator to the analog filters; and a monitor/control section that monitors outputs of the analog filters converted into digital by the ADC and controls parameters such as coefficients of the digital filters depending on a result of the outputs to correct a frequency/phase characteristic of the analog filters. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a wireless receiver of a wireless communication system, and in particular, an anti-aliasing filter that removes aliasing noise before an ADC (Analog-Digital Converter), and an influence on an adjacent channel of a received signal by limiting a band. The present invention relates to a radio receiver including a correction circuit that corrects variation in cutoff frequency of a channel filter.

FIG. 1 shows a basic configuration example of a radio receiver (portable terminal radio unit 1) in a conventional radio communication system.
In FIG. 1, a radio signal received by an antenna (ANT) 11 is amplified by a low noise amplifier (LNA) 12 and received in an I-system reception path (multiplier I (13), LPFI (17), ADCI (19), DTFI (DTF; DigiTal Filter (21)) and Q-system reception paths (multiplier Q (14), LPFQ (18), ADCQ (20), DTFQ (22)) receive the I-system respectively. The signal and the signal are demodulated as a Q-system received signal whose phase is shifted by 90 degrees.

  As shown in FIG. 1, analog filters (LPFI (17) and LPFQ (18)) are used in the front stage (RFLSI) of the receiver 1. Here, 1) a channel filter that limits a band so that a signal does not affect adjacent channels, and 2) an anti-aliasing filter function that removes aliasing noise before the reception AD converter.

  Variations in the cut-off frequency of the filter cause deterioration of the immunity to interference waves and S / N deterioration due to aliasing noise. In addition, since the group delay of the filter directly affects the demodulation accuracy, the specifications required for the filter have become very strict in recent years when the wireless communication system is increased in speed and accuracy.

  When the filters (LPFI (17) and LPFQ (18)) are constituted by analog circuits, individual variations occur in filter characteristics due to process variations and relative errors of elements. It is necessary to provide a means for correcting the above.

FIG. 2 shows a configuration example (referred to as a conventional circuit 1) of a conventional variation correction circuit (filter replica circuit). Here, only the configuration of the I system is shown, and the Q system is the same as the I system. FIGS. 3 and 4 show one implementation example of an integrator (filter replica circuit) for variation correction.
Conventionally, an integrator (filter replica circuit) having the same circuit configuration as the filter is provided separately from the main signal filter (LPFI (17) or LPFQ (18)), and the time constant of the filter replica circuit is measured. Thus, the process state of the element is grasped, and the element of the replica circuit is feedback-controlled so that the time constant falls within a certain range by comparison with the reference value. At the same time, by controlling the elements of the main signal filter with the control signal, correction is performed so that the cutoff frequency of the filter is within the allowable design value.

As an operation principle, as shown in FIG. 3, the integrator is operated so that the time constant τ of the integrator coincides with the external clock frequency f _ex . That is, the charge transfer amount ΔQ ₁ (charge charge amount) from the reference voltage Vref to the capacitor C ₀ via the resistor R and the charge transfer amount ΔQ ₂ from the capacitor C ₀ to the reference voltage −Vref amount C during one clock cycle ( The value of the resistance R is adjusted so that the discharge charge amount becomes equal.

In the initial state, the resistance value is reset to the maximum level (R _MAX ), and the integrator is reset to the charge accumulation value “0” state. The charge from Vref is compared as positive, the charge from −Vref is compared as negative, and the arrival level difference is integrated a certain number of times. If the integration result Vo is higher than SG, the correction is completed, but if it is lower, the resistance value is lowered by one step and the differential integration operation is performed again (see FIG. 4). The resistance control signal is shared with the main signal filter (LPFI 17), and the optimum time constant is also realized in the main signal filter. For example, in the case of a resistor, the element value can be adjusted by connecting a resistor and a switch element connected in parallel and switching the switch element ON / OFF to change the total resistance value.

  As another means (referred to as the conventional circuit 2), as a means for suppressing the variation of the analog filter, the filter in front of the ADC is a loose filter that satisfies the cutoff frequency <fs / 2 only for the purpose of removing the aliasing noise, A method of digitally configuring a filter for channel separation is disclosed (see Patent Document 1).

Japanese Patent No. 3032268

  However, the above-described conventional circuit 1 is an adjustment for turning ON / OFF the switch element in the analog circuit, and normally, since the resistance and capacity variations are corrected only by switching the resistance elements, the capacity variations cannot be corrected. There was a problem. In addition, since there is a variation in the ON resistance of the analog switch element, there is a problem that there is a limit to the correction accuracy of the resistance. Furthermore, since this example is based on the premise that the process state of the main signal filter and the replica circuit is the same, if there is a relative error between the main signal filter and the replica circuit, the variation cannot be corrected. there were.

  On the other hand, as shown in the operation example of the conventional circuit 2 in FIG. 5, in the conventional circuit 2, the analog filter in the previous stage of the ADC is an anti-aliasing filter (AAF) only for the purpose of removing aliasing noise. In such a filter, since the interference wave adjacent to the desired wave cannot be completely removed, the ADC needs a dynamic range that does not distort the interference wave, and requires an expensive ADC having a wide input dynamic range. there were.

  In view of the above problems, an object of the present invention is to directly filter an output signal of an oscillator for a quadrature demodulator that is normally input directly to a multiplier or input to a multiplier after a phase shift of 90 degrees. To measure the frequency characteristics of the filter and to reset the coefficient of the digital filter on the analog filter and digital base band (DBB) side to the optimum value according to the result. An object of the present invention is to provide a radio receiver having a variation correction function for correcting a variation of an analog circuit in a total of analog + digital by setting an optimum digital filter based on the measured variation result.

  According to the present invention, a received signal is separated into an I phase and a Q phase by a quadrature demodulator composed of a multiplier, an oscillator, and a phase shifter, and then an analog filter, an AD converter that follows it, and a digital filter that follows the analog filter. In a radio reception circuit for processing a signal, the means for directly inputting the output of the oscillator to the analog filter and the output of the analog filter converted into digital by the AD converter are monitored, and the output result is determined according to the output result. Thus, there is provided a radio receiver comprising a monitor / control unit for correcting the frequency / phase characteristics of the analog filter by controlling parameters such as coefficients of the digital filter.

  The analog filter further includes means for correcting the analog frequency characteristic, and the monitor / control unit controls the correction means according to the measurement result of the frequency characteristic by the output monitor of the analog filter. Then, the analog frequency characteristic is corrected. The direct input means includes a path changeover switch, and the monitor / control unit controls the path changeover switch to determine the normal connection state and the frequency / phase characteristics during the reception operation of the radio receiver. Switches between the characteristic measurement states to be monitored. Furthermore, the monitor / control unit sweeps the oscillation output frequency from the oscillator within the measurement range in the characteristic measurement state.

  Further, according to the present invention, the received signal is separated into the I phase and the Q phase by the quadrature demodulator including the multiplier, the oscillator, and the phase shifter, and then the first analog filter, the subsequent AD converter, and the subsequent signal. In a radio reception circuit that processes an I-phase signal with a digital filter, a second analog filter, an AD converter that follows it, and a Q-phase signal with a digital filter that follows, the output of the oscillator is changed to the first or second output. In the first and second AD converters, means for directly inputting to any one of the analog filters, means for directly inputting the output of the oscillator to either the second or first AD converter, The output of the first or second analog filter converted to digital is compared with the output of the first or second AD converter, and the first and second digital filters are compared according to the difference. There is provided a wireless receiver having a monitor / control unit for controlling parameters such as a filter coefficient.

  According to the present invention, the output signal of the oscillator for the quadrature demodulator is directly input to the filter, the frequency characteristic of the filter is measured, and the coefficient of the digital filter on the digital base band (DBB) side according to the result By providing a means for resetting to the optimum value and setting the optimum digital filter based on the measured variation result, it is possible to correct the analog circuit variation in a total of analog + digital. It is also possible to perform correction with an analog filter alone.

  These series of correction operations are performed at least once after the system power is turned on until communication is started. In a system that performs a burst operation such as TD-CDMA, for example, It is also possible to perform it at a timing before the reception operation starts.

  Further, according to the present invention, analog circuit variations can be corrected in a total of analog + digital without requiring control from the outside of a measuring instrument or portable terminal, and correction accuracy can be further improved.

  Furthermore, according to the present invention, not only the frequency characteristic of the filter gain but also the phase characteristic can be measured by comparing the relative characteristics of the I phase and the Q phase.

FIG. 6 shows an embodiment of a variation correction circuit according to the present invention.
In FIG. 6, a switch S1 (41) is provided between the oscillator 15 and the multiplier 13, a switch S2 (43) is provided between the multiplier 13 and the LPF 17, and a switch S3 (42) is provided between the oscillator 15 and the LPF 17. During normal communication, the switches S1 and S2 are turned on and the switch S3 is turned off, the output of the transmitter 15 is connected to the multiplier 13, and the frequency conversion of the received signal is performed. In this example, only the I system is shown for ease of explanation.

  In order to measure the frequency characteristics of the LPF 17, the switches S1 and S2 are turned OFF and the switch S3 is turned ON, and the output of the oscillator is directly input to the LPF 17, and the output signal frequency of the oscillator 15 is swept to several KHz to several tens of MHz. . The frequency characteristics of the filter LPF 17 can be known by monitoring the LPF output signal converted into a digital signal by the AD converter 19. In accordance with the difference between the frequency characteristic and the design value, the coefficient of the digital filter DTF 21 on the DBB side is reset to an optimum value by the control circuit (CONT 44).

  Note that the frequency characteristics of the LPF 17 may be directly adjusted using the resistance adjuster shown in FIG. 4 instead of the DTF 21. As a result, the ADC 19 may have a dynamic range including only a desired wave, and an expensive ADC having a wide input dynamic range including an interference wave is not necessary (see FIG. 5). In addition, the control circuit CONT44 performs the control of the switches S1 to S3, the frequency sweep control of the transmitter 15, the resistance selection of the resistance adjuster, and the monitoring of the LPF output signal in this example. This control circuit CONT44 can be mounted on a portable terminal in the same manner as other components by using, for example, a DSP (Digital Signal Processor), and thus requires control from the outside of the measuring instrument or portable terminal. Therefore, it is possible to measure the filter characteristics and set the optimum filter constants.

FIG. 7 shows a filter cutoff frequency measurement method in the embodiment of FIG.
After selecting the resistance of the resistance adjuster, a signal having a sufficiently low frequency is first output from the transmitter 15 and input to the LPF 17. After the output signal of the LPF 17 is converted to a digital signal by the ADC 19, the amplitude of the signal is measured and set to 0 dB ((a) in FIG. 7). Next, the same measurement is repeated while increasing the frequency, and the relationship between the input signal frequency and the ADC output signal amplitude attenuation is measured (FIG. 7B).

FIG. 8 shows another embodiment of the variation correcting circuit according to the present invention.
In FIG. 8, in a radio receiver composed of an I phase and a Q phase, a switch S1I (41), a multiplier I (13), and a filter LPFI are provided between the transmitter 15 and the I phase multiplier I (13). Switch S2I (42) during (17), switch S3I (43) between oscillator 15 and filter LPFI (17), S4I (45) between LPFI (17) and AD converter ADCI (19), transmission S5I (46) is provided between the converter 15 and the AD converter ADCI (19).

  Similarly, a 90-degree phase shifter (16) and a switch S1Q (51) are provided between the transmitter 15 and the Q-phase side multiplier Q (14), and a switch is provided between the multiplier Q (14) and the filter LPFQ (18). S2Q (52), switch S3Q (53) between the oscillator 15 and the LPFQ (18), S4Q (55) between the LPFQ (18) and the AD converter ADCQ (20), the transmitter 15 and the AD converter ADCQ ( S5Q (56) is provided during 20).

  During normal communication, the switches S1I (41), S2I (42), S4I (45), S1Q (51), S2Q (52), S4Q (55) are ON, and the switches S3I (43), S5I (46), S3Q (53), S5Q (56) is turned OFF, and the output of the transmitter 15 is connected to the multiplier Q (14) via the multiplier I (12) and the 90-degree phase shifter (16) to convert the frequency of the received signal. Is done.

  When measuring the characteristics of the I-phase side filter LPFI (17), the switches S1I (41), S2I (42) and S5I (46) are turned OFF, the switches S3I (43) and S4I (45) are turned ON, and transmission is performed. The output signal of the device 15 is directly input to the filter LPFI (17). At the same time, S1Q (51), S2Q (52), S3Q (53), and S4Q (55) are turned off and S5Q (56) is turned on, so that the output of the AD converter ADCQ (20) on the Q side is filtered. A signal not passing through the LPFQ (18) can be obtained.

  Here, the frequency characteristic of the filter LPFI (17) can be known by sweeping the output signal frequency of the transmitter 15 and monitoring the output signal of the AD converter ADCI (19) by the same method as in FIG. . Further, by measuring the phase difference between the AD converter output signals of the I / Q phases, the phase characteristics of the I-phase side filter LPFI (17) can be measured together.

  Similarly, when measuring the characteristics of the Q-phase side filter LPFQ (18), the switches S1Q (51), S2Q (52), and S5Q (56) are turned off, and the switches S3Q (53) and S4Q (55) are turned on. The output signal of the transmitter 15 is directly input to the filter IPFQ (18). At the same time, S1I (41), S2I (42), S3I (43), and S4I (45) are turned off and S5I (46) is turned on, so that the output of the AD converter ADCI (19) on the I side is filtered. A signal not passing through the LPFI (17) can be obtained.

  Here, the frequency characteristic of the filter LPFQ (18) can be known by sweeping the output signal frequency of the transmitter 15 and monitoring the output signal of the AD converter ADCQ (20) by the same method as in FIG. . Moreover, the phase characteristic of the Q-phase side filter can be measured by measuring the phase difference between the AD converter output signals of the I / Q phases.

  Also in the above embodiment, as in FIG. 6, the control circuit CONT44 performs control of the switches S1I to S4I and S1Q to S4Q, frequency sweep of the oscillator, and monitoring of the filter LPFI and LPFQ output signals. This control circuit CONT44 can be mounted on a portable terminal in the same manner as other components by using, for example, a DSP, thereby allowing a filter without requiring control from the outside of the measuring instrument or portable terminal. It is possible to measure characteristics and set an optimum filter constant.

FIG. 9 shows a method for measuring the cutoff frequency and phase characteristic of the filter in the embodiment of FIG. 8 according to the present invention.
First, a sufficiently low frequency signal is output from the transmitter, and is input to the Q-phase filter LPFQ18 and the I-phase AD converter ADCI19. After the output signal of the Q-phase filter LPFQ18 is converted into a digital signal by the AD converter ADCQ20, the amplitude of the signal is measured and set to 0 dB. Similarly, when the amplitude of the output signal of the I-phase AD converter ADCI 19 is also measured, if there is no relative error in the I / Q phase, the amplitude is also 0 dB.

  Next, the same measurement is repeated while increasing the frequency of the transmitter 15, and the relationship between the input signal frequency and the output signal amplitude attenuation amount of the Q-phase side AD converter ADCQ20 with respect to the output signal amplitude of the I-phase side AD converter ADCI is shown. taking measurement. At the same time, the delay time of the Q-phase signal with respect to the I-phase signal at each frequency is set as the phase characteristic of the Q-phase filter LPFQ20.

  In order to measure the characteristics of the I-phase filter LPFI 17, the output signal of the transmitter 15 may be input to the I-phase filter LPFI 17 and the Q-phase AD converter ADCQ20 and the same measurement may be performed.

  In the method of FIG. 8, since the relative gain and phase change of the I phase and the Q phase are measured, for example, even when the output signal amplitude of the transmitter changes according to the frequency, the frequency of the filter It is possible to measure properties. The above measurement may be performed after selecting the resistance of the resistance adjuster.

FIG. 10 shows a configuration example of a digital filter (DTF) and a characteristic correction method thereof.
The filter in FIG. 10 is a general FIR filter, and is configured with n tap stages. The response characteristic of the FIR filter changes by changing the coefficient (h _{0 to} h _n ).

  As shown in FIG. 11, the coefficient may be adjusted so that the characteristic of the digital filter becomes the difference between the analog filter characteristic measured earlier and the ideal filter characteristic. The optimum coefficient value corresponding to the characteristics of the analog filter may be retrieved from a table stored in advance or may be calculated each time. Adjustment of the digital filter coefficient is also performed by the control circuit CONT 44 as described in the embodiment of FIGS. In this example, the variation correction characteristics 1 and 2 are created by DTF so that ideal characteristics can be obtained for each of the ADC output variation characteristics 1 and 2, and the total of analog + digital correction is performed. In addition, the analog circuit (LPF) is first corrected roughly (anti-aliasing filter correction and channel filter correction) by selecting the resistance of the resistance adjuster (see FIG. 4), and the analog + digital total and accurate by DTF. May be corrected.

It is the figure which showed the basic structural example of the radio receiver in the conventional radio | wireless communications system. It is the figure which showed one structural example of the conventional dispersion | variation correction circuit (filter replica circuit). FIG. 5 is a diagram illustrating an example of realizing an integrator (filter replica circuit) for variation correction. FIG. 10 is a diagram showing another example of realizing an integrator (filter replica circuit) for variation correction. It is the figure which illustrated the problem of the conventional circuit schematically. It is the figure which showed one Example of the dispersion | variation correction circuit by this invention. It is the figure which showed the cut-off frequency measuring method in FIG. It is the figure which showed another Example of the dispersion | variation correction circuit by this invention. It is the figure which showed the measuring method of the cut-off frequency and phase characteristic of the filter in FIG. It is the figure which showed the structural example of the digital filter (DTF), and its characteristic correction method. It is the figure which showed the coefficient adjustment example of the analog filter characteristic by this invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 11 Antenna 12 Low noise amplifier 13, 14 Multiplier 15 Oscillator 16 90 degree phase shifter 17, 18 Analog filter 19, 20 AD converter 21, 22 Digital filter 41-43, 45-46 Switch 51-53, 55-56 Switch 44 control unit

Claims (5)

Wireless reception that separates the received signal into I and Q phases by a quadrature demodulator consisting of a multiplier, oscillator, and phase shifter, and then processes the signal with an analog filter, followed by an AD converter, and then a digital filter In the circuit
Means for directly inputting the output of the oscillator to the analog filter;
The output of the analog filter converted into digital by the AD converter is monitored, and the frequency / phase characteristics of the analog filter are controlled by controlling parameters such as the coefficient of the digital filter according to the output result. A radio receiver comprising a monitor / control unit for correction. The analog filter further includes means for correcting the analog frequency characteristic,
2. The radio according to claim 1, wherein the monitor / control unit corrects the analog frequency characteristic by controlling the correcting unit according to a measurement result of the frequency characteristic by an output monitor of the analog filter. Receiving machine. The means for direct input comprises a path switch.
The monitor / control unit controls the path changeover switch to switch between a normal connection state during a reception operation of the wireless receiver and a characteristic measurement state for monitoring and measuring the frequency / phase characteristic. The wireless receiver according to claim 1.   4. The radio receiver according to claim 3, wherein the monitor / control unit sweeps the oscillation output frequency from the oscillator within the measurement range in the characteristic measurement state. The quadrature demodulator consisting of a multiplier, oscillator, and phase shifter separates the received signal into I and Q phases, and then the I-phase signal is output by the first analog filter, the subsequent AD converter, and the subsequent digital filter. , A second analog filter, followed by an AD converter, followed by a digital receiver that processes the Q-phase signal,
Means for directly inputting the output of the oscillator to either the first or second analog filter;
Means for directly inputting the output of the oscillator to the second or first AD converter;
The output of the first or second analog filter digitally converted in the first and second AD converters;
A monitor / control unit that compares outputs of the first and second AD converters and controls parameters such as coefficients of the first and second digital filters in accordance with a difference between the outputs. And wireless receiver.

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