JP2007097113A - Band path filter, high frequency module, and radio communication device using the same - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a band path filter and a high frequency module with which the large amount of attenuation is obtained in a narrow band, used for ultrawideband (UWB: Ultra Wide Band) with large pass band width, small-sized dimension and excellent low-loss characteristics, and to provide a radio communication device using the same. <P>SOLUTION: The band path filter is constituted such that two or more conductive patterns and dielectric layers are laminated alternately. It comprises: N (N≥2) pieces of resonators at least a part of which is arranged in an overlapped manner when seen from this laminating direction so as to be electromagnetically connected mutually; and an input part and an output part which are connected with each other of two resonators selected from the N pieces of resonators. When one end (grounding end) of each of the N pieces of resonators is grounded, the length of the signal propagation direction of each of the N pieces of resonator is λ/4 fundamentally if the propagation wavelength inside the dielectric layer in a substantial center frequency of the pass band is represented as λ. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  TECHNICAL FIELD The present invention relates to a bandpass filter and a high-frequency module having a broadband and steep attenuation characteristic, which are preferably used for an ultra-wideband UWB (Ultra Wide Band) in the field of wireless communication, and a wireless communication device using the same. is there. UWB is used as a data transmission medium for PC peripheral devices such as PC adapters, external storage devices, printers, scanners, and hubs, or as a data transmission medium for digital home appliances such as digital televisions, projectors, 5.1ch speaker systems, and video cameras. Is expected to be used.

  In recent years, attention has been paid to an ultra wide band (UWB) as a new communication means.

  UWB realizes large-capacity data transfer with a pass band of 3.1 GHz to 10.6 GHz.

  When this UWB is compared with a wireless local area network (hereinafter referred to as W-LAN) used as one of data communication means, the communication distance and the data transfer rate are different. In the W-LAN, the communication distance is 30 to 100 m, the transmission power is 500 mW, and the communication speed is about 11 Mbps, whereas in the UWB, the communication distance is as short as 10 m, but the transmission power is 100 mW and the power consumption is low. Is capable of high-speed data communication of 100 Mbps at a communication distance of about 10 m and 480 Mbps at a communication distance of 2 m or less.

  According to the US FCC regulations, there are some rules for the frequency band used for UWB, but a wide band from 3.1 GHz to 10.6 GHz is used.

  Thus, one feature of UWB is the use of a wide band. The ratio band (bandwidth / center frequency) is required to be 40% or more, and in some cases 108%.

  Further, the average transmission power density of UWB is defined as a low value of less than -41.3 dBm / MHz. Here, −41.25 dBm / MHz corresponds to radiated power that generates an electric field strength of 54 dBμV = 500 μV / m at a distance of 3 m from the wave source.

  Thus, another feature of UWB is low transmission power.

  An example of the spectrum mask under the outdoor environment specified by the FCC is as follows. The transmission power in the passband from 3.16 GHz to 4.75 GHz is set as a reference (0 dB), and −20 dB at 3.1 GHz and −1.61 GHz at −1. It is specified to be 30 dB. Moreover, under substantial use conditions, it is necessary to consider the influence with W-LAN (802.11.a), and attenuation of 5.15 GHz is required.

Therefore, another feature of UWB is that the transmission power spectrum is required to be steeply attenuated in a short band adjacent to the pass band.
“Ultra-wideband bandpass filter using microstrip-CPW broadside coupling structure” March 2005 C-2-114 p.147

  From the above, in a UWB wireless communication device, a filter inserted into a transmission / reception signal passage path is required to have a wide band, low loss, and high attenuation near the pass band.

  In recent years, SAW filters and BAW filters based on quartz and piezoelectric ceramics that can obtain high Q values have been used as narrowband filters that require low loss and high attenuation in the communication field. Is 3 to 4% or less at a center frequency of 2 GHz, and the bandwidth is 0.06 to 0.08 GHz, which is two orders of magnitude narrower than the bandwidth of UWB. These bandwidths are determined by the electromechanical coupling coefficient of the crystal or the piezoelectric substrate, and it is difficult to increase the bandwidth from the viewpoint of materials.

  On the other hand, a planar circuit filter is used as a commonly used filter.

  FIG. 45 is a perspective view showing a planar circuit filter in which two microstrip lines 31 and 32 are arranged on a dielectric substrate 33.

  Of the two microstrip lines 31, 32, one is used for input and the other is used for output. They are arranged side by side on the same wiring layer, and the long sides of both lines are connected close to each other. ing. In this way, coupling by arranging two resonators side by side on the same plane is so-called “edge coupling”. This coupling causes resonance to realize a narrow band filter.

  However, in the planar circuit filter, since the two microstrip lines 31 and 32 are arranged side by side on a dielectric substrate, strong coupling cannot be obtained, and a wideband filter with a relative bandwidth of 110% is obtained. Is difficult to realize. In addition, it is difficult to obtain steep attenuation characteristics. If the attenuation pole is formed to improve the attenuation characteristic, the circuit configuration becomes complicated.

  Therefore, in recent years, a wideband filter is known by a band rejection filter (planar circuit filter) that attenuates using a notch on the low frequency side and the high frequency side. However, this has the problem that the dimension becomes large.

  From the viewpoint of improving dielectric packaging density and increasing functionality, there is a growing demand for downsizing and low profile for parts such as wideband bandpass filters, so the above structure is a small bandpass filter for UWB applications. Is not suitable.

  The present invention provides a wideband bandpass filter and a high-frequency module that have a wide passband width, a small size and low loss, and can obtain high attenuation in a narrow band, and a radio communication device using the same. Objective.

  The band-pass filter according to the present invention includes N (N ≧ 2) pieces in which a plurality of conductor patterns and dielectric layers are alternately laminated, and are arranged at least partially overlapping when viewed from the lamination direction and are electromagnetically coupled to each other. And an input unit and an output unit coupled to each of two resonators selected from N resonators, and one end (ground end) of each of the N resonators is grounded. The length of the N resonators in the signal propagation direction is basically λ / 4 where λ is the propagation wavelength inside the dielectric layer at the approximate center frequency of the passband. .

  The bandpass filter having this configuration can realize surface coupling (broadside coupling) at a portion where N resonators are arranged to overlap each other. For this reason, the amount of coupling becomes strong, and a low-loss pass characteristic in the wide band and a steep attenuation characteristic outside the band can be obtained.

  As each of the N resonators, for example, a structure including a strip line, a microstrip line, or a coplanar line can be adopted, and one end of the resonator including these lines is short-circuited to correspond to λ / 4. Can be of length.

  The ground end of each resonator may be on the same end of the resonator as viewed from the stacking direction, or may be on the opposite end of the previous-stage resonator. These are appropriately determined depending on the required pass band. In particular, when they are present at the opposite end, the coupling between the resonators becomes stronger, and a wider band can be achieved.

  The input unit and the output unit may have a configuration including a capacitor element or an inductor element coupled to a resonator. In this case, by setting the constant of the element to a predetermined value, a strong coupling amount can be obtained at the time of input / output of the signal in the input unit and the output unit, so that the pass loss of the band pass filter can be reduced.

  The input unit and the output unit may include a configuration including an input line and an output line coupled to a resonator. In this case, the input / output lines can be routed on the substrate and connected to other circuits, which is advantageous for reducing the height of the bandpass filter.

  Here, the input direction to the input line and the output direction from the output line may be different or the same. These are appropriately determined depending on the required pass band. In particular, when they are the same, the coupling between the resonators becomes stronger, and a wider band can be achieved.

  Further, the non-grounded end of the resonator may be grounded via a capacitor element formed by a lumped constant or a pattern. In such a structure, each of the N resonators is formed in a rectangular shape when viewed from the stacking direction, and only one end (grounding end) of the rectangular resonator is grounded. It is preferable that a ground conductor is provided on the same plane as surrounding the resonator. With this structure, it is not necessary to use a via for the grounded portion of the resonator, and manufacturing variations can be reduced.

  Furthermore, it is preferable that a capacitance is added between a non-ground end as the other end with respect to the ground end and a portion of the ground conductor adjacent to the non-ground end. Specifically, the first conductor is provided close to the upper side or the lower side of the non-grounded end in the resonator, and the via conductor that connects the first conductor and the ground conductor is provided. Preferably, the capacitance is formed between the resonator and the first conductor. Thereby, since the length of the resonator in the signal propagation direction can be further shortened, the longitudinal dimension of the bandpass filter can be reduced, and the bandpass filter can be mounted at a higher density. Further, the higher order mode can be moved to the high frequency side, and the out-of-band characteristics can be improved.

  The ground end side region of the resonator may be an inductor element formed with a lumped constant or a pattern. For example, each of the N resonators is preferably formed so as to be narrowed stepwise or continuously toward the ground end side when viewed from the stacking direction. Also in this case, since the length of the resonator in the signal propagation direction can be further shortened, the longitudinal dimension of the bandpass filter can be reduced, and the bandpass filter can be mounted at a higher density. Similarly, the higher order mode can be moved to the high frequency side, and the out-of-band characteristics can be improved.

  In the band-pass filter of the present invention, it is preferable that at least one of a capacitance and an inductance is added as an electromagnetic coupling between the input line and the output line.

  Specifically, a second conductor is provided on the same plane as the input line, a third conductor is provided on the same plane as the output line, and the second conductor and the third conductor are connected to each other. It is preferable that the capacitance or the inductance is added between the input line and the output line by providing a via conductor to be connected. Also, a second conductor is provided in proximity to the upper side or lower side of the input line, a third conductor is provided in proximity to the upper side or lower side of the output line, and the second conductor and the first conductor are provided. It is preferable that the capacitance or the inductance is added between the input line and the output line by providing a via conductor that connects the three conductors. Accordingly, a new attenuation pole can be formed outside the pass band and in the vicinity of the boundary between the pass band and the out band, and a steep skirt characteristic can be obtained. The second conductor and the third conductor are formed so as to be narrowed stepwise or continuously toward the portion connected to the via conductor when viewed from the stacking direction. This is preferable from the viewpoint of providing the attenuation pole at the low frequency side.

  In the band-pass filter of the present invention, it is preferable that at least one of capacitance and inductance is added as electromagnetic coupling between any two resonators selected from the N resonators. .

  Specifically, a fourth conductor is provided on the same plane as any one of the N resonators, and on the same plane as a resonator different from the one resonator. Is provided with a fifth conductor and a via conductor connecting the fourth conductor and the fifth conductor is provided to add the capacitance or the inductance between the two arbitrary resonators. It is preferable. Further, a fourth conductor is provided adjacent to an upper side or a lower side of any one of the N resonators, and an upper side of a resonator different from the arbitrary one of the resonators or A fifth conductor is provided adjacent to the lower side, and a via conductor that connects the fourth conductor and the fifth conductor is provided, so that the capacitance or between the arbitrary two resonators is provided. The inductance is preferably added. As a result, a new attenuation pole can be formed between the input line and the output line outside the pass band and in the vicinity of the boundary between the pass band and the out band, and a steep skirt characteristic can be obtained.

  Furthermore, at least one resonator is provided between the fourth conductor or the fifth conductor and the input unit or the output unit, and the resonator sees the input unit and the output unit from the input unit or the output unit. The fourth conductor and the fifth conductor are preferably covered. Thereby, it can prevent that a 4th conductor or a 5th conductor raise | generates unnecessary coupling | bonding with an input part or an output part, and can resonate, and can suppress the unnecessary resonance peak outside a zone | band.

  The fourth conductor and the fifth conductor are formed so as to be narrowed stepwise or continuously toward the portion connected to the via conductor when viewed from the stacking direction. This is preferable from the viewpoint of setting the attenuation pole to a low frequency side.

  For the band-pass filter of the present invention, a plurality of attenuation poles can be formed at the same time as long as the above-described structure relating to pole formation permits. Further, if necessary, an attenuation pole can be generated in the band.

  Further, the resonators arranged in a stacked manner can be configured in two rows horizontally. In this case, the signal direction can be turned back by arranging the coupling conductors for coupling the resonators on the lowermost surface across two rows. The length of the coupling conductor in the signal propagation direction is basically ½ wavelength.

  With this configuration, the same effect as when the number of stages N is doubled can be obtained without changing the height of the resonator, and the bandpass filter can be reduced in height. In addition, by comprising two or more rows, it is possible to further increase the number of stages without changing the height of the resonator.

  In the case of using an input line and an output line, the width of the input line or the output line is preferably formed in a step shape at the end of the portion that overlaps the resonator as viewed from the stacking direction. Thereby, the attenuation pole can be controlled, and the attenuation outside the band can be made steep.

  Moreover, a high frequency module can be manufactured using the band pass filter of the present invention.

  In addition, a wireless communication device that can be miniaturized can be manufactured by mounting the band-pass filter or the high-frequency module. According to this wireless communication device, it is possible to improve reception sensitivity, wideband communication, low power consumption, and prevention of mutual interference with a wireless LAN.

  Hereinafter, embodiments of the present invention will be described with reference to the accompanying drawings.

  FIG. 1 is an explanatory diagram showing a circuit configuration of a bandpass filter of the present invention.

  The band-pass filter includes N (N ≧ 2) resonators 1 and 2 (two resonators shown in FIG. 1) stacked one above the other at a predetermined interval. This resonator is formed by alternately laminating a plurality of conductor patterns and dielectric layers by, for example, laminating a plurality of dielectric layers having a conductor pattern formed on the upper surface, and a strip line, microstrip line or coplanar line. It is comprised by the structure containing. Here, in the case where the resonators 1 and 2 include a strip line or a microstrip line, the ground (not shown) constituting the line is, for example, above the input line 3 shown in FIG. 1 and / or the output line. 4 having a structure in which the resonator 1 and the resonator 2 share the ground.

  The two resonators 1 and 2 include conductors having the same dimensions (the length in the signal propagation direction is basically λ / 4 where λ is the propagation wavelength inside the dielectric layer at the approximate center frequency of the pass band). It is a structure, and at least a part, preferably almost all, is overlapped when viewed from the stacking direction (upward direction in FIG. 1). Due to this overlap, the two resonators 1 and 2 are electromagnetically coupled to each other (indicated by M in FIG. 1). This coupling is referred to as so-called “broadside coupling”, and is a method in which the main surfaces of the upper and lower two resonators 1 and 2 are opposed to each other. Here, normally, when designing a narrow band filter, the center frequency and the resonance frequency are made substantially equal. However, in the design of a broadband bandpass filter using broadside coupling, since the coupling is strong, the center frequency of the filter and the resonance frequency of the resonator do not necessarily match. Therefore, it is necessary to set the resonance frequency of the resonator to be slightly higher than the center frequency of the filter, and the approximate center frequency here includes a frequency shift up to the resonance frequency.

  Further, one end of each of the two resonators 1 and 2 (when one end is coupled to the input unit and the output unit, the end on the side opposite to the coupled side (the right side in FIG. 1)) Both are grounded (referred to as grounding end).

  Also, in the figure, the input direction to the input unit and the output direction from the output unit, that is, the signal propagation direction is represented by “F”, and the length along the signal propagation direction F of the resonator (resonator of the resonator shown in the figure). The length in the longitudinal direction is basically λ / 4, where λ is the propagation wavelength inside the dielectric layer at the approximate center frequency of the passband. Here, basically, in order to obtain overall filter characteristics such as coupling adjustment between resonators, it is necessary to slightly shift (finely adjust) the length of the resonator from λ / 4. As will be described later, the length of the resonator can be shortened to less than λ / 4 by using a capacitor element or an inductor element.

  1 is electromagnetically coupled to the input electrode IN and the output electrode OUT via the capacitor elements C1 and C2, respectively. These electromagnetically coupled portions are referred to as “input unit” and “output unit”. That is, the capacitor elements C1 and C2 are included in the input unit and the output unit and constitute a part thereof. Here, as a coupling element coupled to the resonator, an inductor element can be used in addition to the capacitor elements C1 and C2. In addition, it is not always necessary to use a lumped constant element, and the input unit and the output unit may be configured using distributed constant lines. Further, as described later, the entire surface may be coupled (broadside coupling) using an input line and an output line.

  With such a structure, strong coupling can be obtained between the two resonators 1 and 2, and the passband can be widened. In addition, the bandpass filter can be downsized.

  On the other hand, FIG. 2 is an explanatory diagram showing another circuit configuration of the bandpass filter. This circuit configuration is different from that of FIG. 1 in that the input direction F and the output direction F are in the same direction (the input end and the output end are located on the opposite side when viewed from the stacking direction). ing. Therefore, the ground ends of the resonators 1 and 2 are present at the opposite ends of the previous-stage resonator as viewed from the stacking direction, and are on the opposite sides.

  As in FIG. 1, this structure can achieve a wider passband and a smaller size, and has a structure in which the positions of the grounding ends are staggered. The coupling between the devices 1 and 2 can be strengthened, which is advantageous for realizing a wider band (to be described later by comparing the data in FIGS. 35 and 36).

  FIG. 3 is an explanatory diagram showing still another circuit configuration of the band-pass filter. One end (input end and output end) of each resonator 1 and 2 is electromagnetically coupled to the input electrode IN and output electrode OUT via the capacitor elements C1 and C2, respectively, and the input end and output end ( A structure in which the non-ground end) is grounded via the capacitor elements C3 and C4 is shown.

  In this circuit configuration, the input direction F and the output direction F of the resonators 1 and 2 are different (the input end and the output end (non-grounded ends) are located on the same side when viewed from the stacking direction, and the grounded end is (It is located at the end on the same side of the resonator as viewed from the stacking direction). Further, it differs from that of FIG. 1 in that the non-grounded end is grounded via the capacitor elements C3 and C4. Thereby, a part of the effective length of the resonators 1 and 2 is replaced by the capacitor elements C3 and C4, and the length of the resonators 1 and 2 (the length in the signal propagation direction) is the length of the resonator in FIG. It can be made shorter than a certain λ / 4.

  FIG. 4 is an explanatory diagram showing still another circuit configuration of the band-pass filter. The non-grounded ends of the resonators 1 and 2 are electromagnetically coupled to the input electrode IN and the output electrode OUT via the capacitor elements C1 and C2, respectively, and these non-grounded ends are connected via the capacitor elements C3 and C4. The grounded structure is shown.

  In this configuration, the non-grounded ends of the resonators 1 and 2 are positioned on the opposite sides of the resonator 1 and the resonator 2. This is the same structure as that of FIG. 2 in which the non-grounded end is grounded via the capacitor elements C3 and C4. Also in this configuration, since a part of the effective length of the resonators 1 and 2 is replaced by the capacitor elements C1 and C2, the length of the resonators 1 and 2 (the length in the signal propagation direction) is shorter than λ / 4. can do. Therefore, it is possible to reduce the size as in FIG. 2, and to achieve a wider band than the structure of FIG. Furthermore, according to this structure, as shown in the data of an example described later, the higher-order mode can be moved to the high frequency side, and the out-of-band characteristics can also be improved.

  1 to 4 described so far, the input / output electrodes IN and OUT are connected to the input end and output end (non-ground end) of the resonators 1 and 2 via the capacitor elements C1 and C2 or the inductor element. However, instead of the capacitor element or the inductor element, an input line and an output line coupled to the resonators 1 and 2 may be used.

  As the input line and output line, a structure including a strip line, a microstrip line, or a coplanar line can be used. These lines are broadside coupled to the resonators 1 and 2.

  FIG. 5 is an explanatory diagram showing a cross section of a band-pass filter using a structure including an input line 3 and an output line 4 coupled to the resonators 1 and 2 as an input part and an output part.

  Thus, according to the structure using the input line 3 and the output line 4, it is necessary to arrange on the upper surface of the dielectric substrate as in the case of using a chip component as a lumped element as a capacitor element or an inductor element, for example. The number of parts can be reduced, and the height of the bandpass filter can be reduced. Further, when the other conductor pattern is formed on the dielectric layer, the input line 3 and the output line 4 can be formed on the dielectric layer simultaneously with the other conductor pattern, which increases the number of manufacturing processes. Absent.

  FIG. 6 is an explanatory diagram showing a cross section of another form of bandpass filter using a structure including the input line 3 and the output line 4 coupled to the resonators 1 and 2 as the input part and the output part. The difference between this configuration and the configuration shown in FIG. 5 is that the input direction to the input line 3 is in the direction toward the non-grounded end of the resonator 1 as opposed to the input direction shown in FIG. The output direction from the output line 4 is also in the direction toward the ground end of the resonator 2, contrary to FIG. 5.

  FIG. 7 is an explanatory view showing a cross section of still another embodiment of a band-pass filter using a structure including an input line 3 and an output line 4 coupled to resonators 1 and 2 as an input part and an output part. This is different from FIGS. 5 and 6 in that the ground ends of the resonators 1 and 2 are both arranged on the same side of the resonators 1 and 2 (right side in the figure). Further, the input direction to the input line 3 and the output direction from the output line 4 are opposite to each other.

  FIG. 8 is an explanatory view showing a cross section of still another embodiment of a band-pass filter using a structure including an input line 3 and an output line 4 coupled to the resonators 1 and 2 as an input part and an output part. In the configuration of FIG. 8, similarly to FIG. 7, the ground ends of the resonators 1 and 2 are both arranged on the same side of the resonators 1 and 2 (right side in the figure). The difference between this configuration and the configuration of FIG. 7 is that in FIG. 7, the input end IN and the output end OUT of the bandpass filter are on the same side as the ground ends of the resonators 1 and 2. In FIG. 8, it is on the opposite side. Therefore, the signal input / output terminals IN and OUT and the ground terminal can be arranged on the opposite side.

  As described above, in the embodiments shown in FIGS. 5 to 8, since a line such as a strip line coupled to the resonators 1 and 2 can be used for signal input / output, a lumped element is not used. Therefore, it is possible to reduce the size and facilitate the manufacture. At the same time, the input end, output end and grounding position of the resonator can be selected as shown in FIG. 5 to FIG.

  When a line such as a strip line is used for signal input / output, the width of the input line or the output line is the end of the portion that overlaps the resonators 1 and 2 when viewed from the stacking direction (T in FIG. 5). It is desirable to form in a step shape. The specific shape of the bandpass filter described above will be described in detail later with reference to FIG.

  The N (N ≧ 2) resonators in the above bandpass filter are formed by alternately laminating a plurality of conductor patterns and dielectric layers. For example, a plurality of dielectric layers having a predetermined conductor pattern are stacked on the upper surface.

  Each dielectric layer is made of, for example, low temperature co-fired ceramics (LTCC), and the conductor pattern formed on each dielectric layer is made of a low resistance conductor such as copper or silver. The In particular, the band-pass filter can be reduced in size by using a dielectric having a high dielectric constant.

  A multilayer substrate in which a plurality of conductor patterns and dielectric layers are alternately laminated as described above is formed by a well-known multilayer ceramic technology. For example, a conductive paste is applied to the surface of a ceramic green sheet to form each resonator. Each of the conductor patterns constituting the film is formed and then laminated, thermocompression-bonded under a required pressure and temperature, and fired. In each dielectric layer, via-hole conductors necessary for connecting the upper and lower conductor patterns are appropriately formed across a plurality of layers.

  FIG. 9 is a cross-sectional view specifically showing a structure in which the bandpass filter shown in FIG. 2 is formed inside a dielectric multilayer substrate, and FIG. 10 shows an arrangement of conductor patterns of this bandpass filter. -It is the perspective view which looked at the A end surface (illustration of the dielectric layer is omitted).

  Of the dielectric layers formed in multiple layers (three or more layers), conductor patterns A1 and A2 constituting the resonators 1 and 2 are formed in two adjacent layers, respectively. Grounding patterns E1 and E2 which are ground conductors for grounding the ends of the two are formed. It should be noted that the layer in which the conductor patterns A1 and A2 of the resonators 1 and 2 are formed and the layer in which the ground patterns E1 and E2 are formed are not necessarily adjacent to each other (two or more layers). May be separated).

  The ground patterns E1 and E2 and the conductor patterns A1 and A2 constituting the resonators 1 and 2 are connected at the ground ends of the resonators 1 and 2 by via-hole conductors 5 and 6 that penetrate the dielectric layer. As a result, the resonators 1 and 2 are grounded at the ground end.

  The input end of the resonator 1 (or the input end of the resonator 2) is connected to a pad formed on the uppermost dielectric layer (that is, the main surface of the dielectric substrate) via the via-hole conductor 8 (7). Has been. A chip-like lumped constant capacitor element C1 (C2) is connected to this pad.

  Similarly to the input end, the output end of the resonator 2 (or the output end of the resonator 1) is also connected to pads 10 and 11 formed on the main surface of the uppermost dielectric layer via the via-hole conductor 7 (8). Then, a chip-like lumped constant capacitor element C2 (C1) is connected to the pads 10 and 11.

  In the above description, the band-pass filter shown in FIG. 2 is assumed and described using a cross-sectional view and a perspective view. However, the band-pass filter of FIG. Basically, the same structure can be taken only by different positions. 3 and 4, the capacitor elements C3 and C4 are connected to the resonators 1 and 2, and the connection structure between the capacitor elements C3 and C4 and the ground patterns E1 and E2 is also shown in the above diagram. 9. Similar to that described with reference to FIG. 10, this can be done via via-hole conductors.

  Next, a specific example of a structure in which a band pass filter using the input / output lines shown in FIGS. 5 to 8 is formed inside a dielectric multilayer substrate will be described.

  5 to 8 uses a structure including an input line 3 and an output line 4 coupled to the resonators 1 and 2 as an input unit and an output unit. In the bandpass filter shown in FIGS. 5 to 8, it is necessary to ground the ends of the resonators 1 and 2 sandwiched between the input line 3 and the output line 4.

  Therefore, ground conductors (grounding patterns) for grounding the ends of the resonators 1 and 2 are provided on the upper and lower dielectric layers of the conductor patterns constituting the resonators 1 and 2 and the input / output lines 3 and 4. Alternatively, a structure in which the resonator is provided in the same layer as the dielectric layer on which the resonators 1 and 2 are formed can be employed.

  FIGS. 11 to 14 show specific examples in which ground conductors (ground patterns E1 and E2) for grounding the ends of the resonators 1 and 2 are provided in the same dielectric layer as the resonators 1 and 2. FIG.

  FIG. 11 shows a dielectric layer (one layer) provided with the input line 3, dielectric layers (two layers, three layers) provided with the resonators 1 and 2, and a dielectric layer provided with the output line 4 (two layers and three layers). It is a top view which decomposes | disassembles and shows (4 layers). The laminated structure of the bandpass filter corresponds to the bandpass filter described with reference to FIG.

  According to this structure, a rectangular resonator 1, 2 is formed by providing a conductor pattern in which a “U” -shaped air gap is formed in part in the second and third layers. The ground conductors (ground patterns E1, E2) are formed so as to surround the resonators 1, 2 so that only one end (ground end) of the resonators 1, 2 is grounded. Here, the “U” -shaped air gap is an air gap formed around the resonators 1 and 2 and in a portion other than the ground end.

  The width W of the input line 3 provided in one layer, the width of the resonator 1 provided in two layers, the width of the resonator 2 provided in three layers, and the width W of the output line 4 provided in four layers are Both are substantially the same. The width W of the input line 3 provided in one layer is narrowed stepwise at the end (signal input end) T of the portion overlapping the resonators 1 and 2 when viewed from the stacking direction. The narrowed width is indicated by Wa. Further, the output line 4 is provided in the four layers, but as in the first layer, the width W thereof is the end (signal output end) T of the portion overlapping with the resonators 1 and 2 when viewed from the stacking direction. In FIG. As a result, the attenuation pole can be controlled. Therefore, it is advantageous for improving the damping characteristic.

  As described above, according to the structure shown in FIG. 11, the resonators 1 and 2 and the ground conductors (ground patterns E1 and E2) are provided in the same layer. Even if they are not connected with vias, the ends of the resonators 1 and 2 are grounded. Thereby, there exists an advantage that a conductor pattern can be formed in one process, and a manufacturing process does not increase and it is easy to manufacture.

  12 shows a laminated structure of the bandpass filter corresponding to FIG. 6, FIG. 13 shows a laminated structure of the bandpass filter corresponding to FIG. 7, and FIG. 14 shows a laminated structure of the bandpass filter corresponding to FIG. Is shown. Similarly to the structure shown in FIG. 11, this structure is also provided with a conductor pattern in which a “U” -shaped air gap is formed in a part of the second and third layers. 2 and ground patterns E1 and E2 are formed so as to surround the resonators 1 and 2 so that only one end (ground end) of the resonators 1 and 2 is grounded. In any structure, since the resonators 1 and 2 and the ground patterns E1 and E2 are provided in the same layer, there is an advantage that they can be formed at a time and the manufacturing process is not increased and the manufacturing is easy. Further, the width of the input line 3 or the output line 4 is formed in a step shape on the end face T of the portion overlapping the resonators 1 and 2 when viewed from the stacking direction. become able to. Therefore, it is advantageous for improving the damping characteristic.

  11 to 14, if the width of the gap (distance between the resonator and the ground pattern) is wide, the non-ground end as the other end with respect to the ground end of the resonator and the non-ground end in the ground conductor. There is no need to take into account the capacitance between the adjacent parts. However, if the width of the gap is narrow, a capacitance is formed between them, which corresponds to FIG. 15 described later. However, as shown in FIG. 16 and FIG. 17 as a concrete form, FIG. 15 is further provided between the non-grounded end as the other end with respect to the grounded end of the resonator and the portion near the non-grounded end of the ground conductor. It is explanatory drawing supposing the structure which adds a capacitance.

  15 uses a structure including an input line 3 and an output line 4 coupled to the resonator 1 as an input unit and an output unit, and a capacitance is added between the non-grounded end of the resonator 1 and the ground conductor. It is sectional drawing which shows the structure of a band pass filter.

  In the configuration of FIG. 15, the non-grounded end of the resonator 1 is further grounded via the capacitor element C in the structure of FIG. 5. With this configuration, part of the effective length of the resonator 1 is replaced by the capacitor element C, and the length of the resonator 1 in the signal propagation direction can be shortened to less than λ / 4. Further, as shown by data of an example described later, the higher order mode can be moved to the high frequency side, and the out-of-band characteristics can also be improved. Specifically, as shown in FIG. 38, S21 is not attenuated due to the higher order mode (in this case, 3λ / 4 mode) at about 14 GHz, but the higher order mode is shifted to the high frequency side by using this structure. It can be moved to improve out-of-band S21 characteristics. Note that this method of adding capacitance is applicable to the structures shown in FIGS.

  Specifically, as a form in which this capacitance is added, as shown in FIG. 16, a ground pattern is formed in a step shape whose width changes stepwise, and the ground ends around the resonators 1 and 2 are formed. Among the gaps formed in the other portions (the “U” -shaped gap), the width of the region near the non-grounding end is narrowed. According to this structure, capacitance is generated in a region where the width of the gap is narrowed, and the length of the resonators 1 and 2 in the signal propagation direction can be further shortened compared to the structure shown in FIG. Further, the higher order mode can be moved to the high frequency side, and the out-of-band characteristics can be improved. Furthermore, with this step-like structure, the Q value of the resonator can be improved by separating the width of the gap between the ground end side and the ground pattern in the resonator.

  Moreover, as a form to which capacitance is added, as shown in FIG. 17, a structure in which capacitance is formed in the stacking direction with respect to the bandpass filter stacking structure of FIG.

  Specifically, the first conductor 91 is provided close to the upper side or the lower side of the non-ground end in the resonator 1, and the via conductor that connects the first conductor 91 and the ground conductor (ground pattern E1). By providing 51, a capacitance is formed between the resonator 1 and the first conductor 91. According to this structure, a capacitance larger than the capacitance formed on the same plane can be obtained, and the length of the resonator 1 in the signal propagation direction can be further shortened. Further, the higher order mode can be moved further to the high frequency side, and the out-of-band characteristics can be improved.

  FIG. 18 is a cross-sectional view showing a band-pass filter formed by adding an inductance to resonators 1 and 2 using a structure including input line 3 and output line 4 coupled to resonator 1 as an input part and an output part. is there. In the configuration of FIG. 18, the ground end of the resonator 1 is formed in the structure of FIG. 5, for example, by making the ground end side of the resonator 1 narrower than the non-ground end side as viewed from the stacking direction. An inductance is added to the side (grounded through the inductor element L). Thereby, a part of the effective length of the resonator 1 is replaced by the inductor element L, and the length of the resonator 1 in the signal propagation direction can be shortened as compared with the structure shown in FIG. Further, similar to the method of adding capacitance shown in FIG. 15, the higher order mode can be moved to the high frequency side, and the out-of-band characteristics can be improved. Note that this method of adding inductance is applicable to the structures shown in FIGS.

  Specifically, each resonator is formed so as to be narrowed stepwise or continuously toward the ground end when viewed from the stacking direction, and the one shown in FIG. It is formed (in stages). According to this structure, a large inductance can be obtained, and the length of the resonator 1 in the signal propagation direction can be shortened compared to the structure shown in FIG. Further, the higher order mode can be moved further to the high frequency side, and the out-of-band characteristics can be improved.

  The structure for adding capacitance and the structure for adding inductance may be used in combination at the same time.

  Further, it is preferable that a capacitance or an inductance is added between the input unit and the output unit as an electromagnetic coupling means, in other words, as a circuit. For example, in a structure including an input line, an N-stage (three stages in the figure) resonator, and an output line as a bandpass filter, FIG. 20 includes a capacitive jump coupling between the input line and the output line. FIG. 21 shows a structure in which an inductance jumping coupling is provided between an input line and an output line.

  In this way, by adding capacitance and inductance between the input line and the output line to cause electromagnetic coupling, a new attenuation pole is formed outside the passband and near the boundary between the passband and the outband. It can be formed, and a steep skirt characteristic can be obtained.

  As the specific structure described above, as shown in FIG. 22, the second conductor 92 is provided on the same plane as the input line 3, the third conductor 93 is provided on the same plane as the output line 4, A structure in which a capacitance or inductance is added between the input line 3 and the output line 4 by providing the via conductor 51 that connects the second conductor 92 and the third conductor 93 can be given. With this structure, the input line and the output line are electromagnetically coupled, and have both the capacitance shown in FIG. 20 and the inductance shown in FIG. Thus, in the case of the structure in which the edge coupling is performed on the same plane, there is an advantage that it is easy to obtain a weak coupling and it is easy to form the attenuation pole on the high frequency side. The simulation results for this structure are shown in FIG.

  23 and 24, a second conductor 92 is provided close to the lower side of the input line 3, and a third conductor 93 is provided close to the upper side of the output line 4. By providing the via conductor 51 that connects the conductor 92 and the third conductor 93, a structure in which capacitance or inductance is added as an electromagnetic coupling means between the input line 3 and the output line 4 is shown. ing. Here, the second conductor 92 is coupled to the narrow region 3 a of the input line 3, and the third conductor 93 is coupled to the narrow region 4 a of the output line 4.

  In the structure shown in FIG. 23, the second conductor 92 and the third conductor 93 are connected to the ground pattern E1 as the ground conductor. In the structure shown in FIG. 24, the second conductor 92 and the third conductor No. 93 is not connected to the ground pattern E1 as the ground conductor, but both are outside the high-pass band and can form a new attenuation pole in the vicinity of the boundary between the pass band and the non-band. Skirt characteristics can be obtained. In particular, in the case of FIG. 24, more attenuation poles can be formed compared to FIG. 23, which is effective in improving the skirt characteristics and the out-of-band characteristics.

  25 and 26, a second conductor 92 is provided close to the upper side of the input line 3, a third conductor 93 is provided close to the lower side of the output line 4, and the second A structure in which a capacitance or inductance is added between the input line 3 and the output line 4 by providing the via conductor 51 that connects the conductor 92 and the third conductor 93 is shown. Here, the second conductor 92 is coupled to the wide region of the input line 3, and the third conductor 93 is coupled to the wide region of the output line 4. In this case, unlike the structure shown in FIGS. 23 and 24, there is an advantage that it is easy to form the attenuation pole on the low frequency side.

  The second conductor 92 and the third conductor 93 are formed so as to be narrowed stepwise or continuously toward the portion connected to the via conductor 51 when viewed from the stacking direction. It is desirable from the viewpoint that the position of the attenuation pole is set on the low frequency side.

  Further, it is preferable that a capacitance or inductance is added as an electromagnetic coupling means between any two resonators. For example, in a structure including an input line, a three-stage resonator, and an output line as a band-pass filter, FIG. 27 shows a structure in which a capacitive jump coupling is provided between any two resonators. 28 shows a structure in which an inductive jump coupling is provided between any two resonators. In this way, by adding a capacitance or inductance between any two resonators for electromagnetic coupling, capacitance or inductance is added as an electromagnetic coupling means between the input unit and the output unit. Similarly to the above, it is possible to form a new attenuation pole outside the pass band and in the vicinity of the boundary between the pass band and the out band, and to obtain a steep skirt characteristic.

  The capacitive coupling and the inductive coupling shown in FIGS. 27 and 28 can exist simultaneously. Further, when the number of stages of the band pass filter is increased, the number of combinations is also increased. For example, when considering a five-stage filter, the first-stage resonator and the fifth-stage resonator may be coupled, or the second-stage resonator and the fourth-stage resonator may be coupled. May be allowed. In some cases, capacitance or inductance may be added as an electromagnetic coupling means between the input line and the resonator or between the output line and the resonator. For example, the input line and the second-stage resonator can have a jump coupling, and the output line and the fourth-stage resonator can have a jump coupling.

  As the specific structure described above, the fourth conductor 94 is provided on the same plane as any one of the resonators, and the same plane as a resonator different from the one arbitrary resonator. Is provided with a fifth conductor 95, and a via conductor 51 is provided to connect the fourth conductor 94 and the fifth conductor 95, so that the capacitance or the capacitance between the two arbitrary resonators is provided. A structure to which inductance is added can be mentioned. Such a structure corresponds to the structure shown in FIG. 24 which has already been described as a structure in which capacitance or inductance is added between the input line 3 and the output line 4. That is, the second conductor 92 and the third conductor 93 in the structure shown in FIG. 24 have a structure in which an electromagnetic coupling of capacitance and inductance is added between the second conductor 92 and the third conductor 93 disposed on the same plane. It has become.

  Further, as shown in FIG. 29, a fourth conductor 94 is provided in the vicinity of the upper side or the lower side of any one of the resonators, and the resonance is different from that of the arbitrary one resonator. A fifth conductor 95 is provided close to the upper side or the lower side of the container, and a via conductor 51 that connects the fourth conductor 94 and the fifth conductor 95 is provided. And a structure in which the capacitance or the inductance is added between the resonators. With this structure, any resonator is electromagnetically coupled, and has both the capacitance shown in FIG. 27 and the inductance shown in FIG. In this case, like the structure shown in FIGS. 25 and 26, there is an advantage that it is easy to form the attenuation pole on the low frequency side.

  The fourth conductor 94 and the fifth conductor 95 are formed so as to be narrowed stepwise or continuously toward the portion connected to the via conductor 51 when viewed from the stacking direction. It is desirable from the viewpoint that the position of the attenuation pole is set on the low frequency side.

  As an example of the combination of the configurations described so far, FIG. 30 shows an example in which the configurations of FIGS. 19, 22, and 29 are combined. In this device, each resonator is formed in a step shape (stepwise) toward the ground end side when viewed from the stacking direction. The second conductor 92 is provided on the same plane as the input line 3, the third conductor 93 is provided on the same plane as the output line 4, and the second conductor 92 and the third conductor 93 are connected to each other. By providing the via conductor 51, a capacitance or inductance is added between the input line 3 and the output line 4. Further, a fourth conductor 94 is provided in the vicinity of the upper side or the lower side of any one of the resonators, and the upper side or the lower side of the resonator different from the arbitrary one resonator. A fifth conductor 95 is provided adjacently, and a via conductor 51 that connects the fourth conductor 94 and the fifth conductor 95 is provided, so that the capacitance is between the two arbitrary resonators. Alternatively, the inductance is added.

  Here, in the embodiment shown in FIG. 30, one resonator 1 is provided as the second layer between the first layer input line 3 and the third layer fourth conductor 94. Through the gap between the resonator 1 and the ground pattern E1 as the ground conductor, the input line 3 and the fourth conductor 94 cause unnecessary coupling to cause resonance, which is unnecessary outside the band. A resonance peak may occur. This also occurs between the ninth-layer output line 4 and the seventh-layer fifth conductor 95.

  Therefore, as shown in FIG. 31, the fourth conductor 94 is covered from the input line 3 by the second-layer resonator 1, and the fifth conductor 95 is covered from the output line 4 by the eighth-layer resonator 1. It is preferable to be formed as described above. Here, being covered means that the fourth conductor 94 in the third layer is exposed through a gap between the resonator 1 in the second layer and the ground pattern E1 as the ground conductor as viewed from the input line 3. And the fifth layer in the seventh layer through the gap between the resonator 1 in the eighth layer and the ground pattern E1 as the ground conductor as viewed from the output line 4. It means that the conductor 95 is not formed at a position where it is exposed. At this time, the fourth conductor 94 is connected to one end of a line-like sixth conductor 96 formed independently in the resonator 1 of the fourth layer by a via conductor, and the fifth conductor 95 has six layers. A line-shaped seventh conductor 97 formed independently inside the resonator of the eye is connected to one end of the seventh conductor 97 by a via conductor, and the other ends of the sixth conductor 96 and the seventh conductor 97 are connected to each other by a via conductor. Is done. By doing so, unnecessary coupling between the input line 3 and the fourth conductor 94 and between the output line 4 and the fifth conductor 95 can be suppressed. Note that “independently” means that the sixth conductor 96 or the seventh conductor 97 is separated from the resonator 1 by a slit formed around the sixth conductor 96 or the seventh conductor 97.

  Further, as shown in FIG. 32, the sixth conductor 96 in the fourth layer is not exposed through the gap between the resonator 1 in the second layer and the ground pattern E1 as the ground conductor as viewed from the input line 3. Thus, the shape of the ground pattern E1 of the third layer shown in FIG. 31 is changed to the shape provided with the protrusion E11 shown in FIG. 32 so that the sixth conductor 96 is covered with this protrusion E11 and the output As shown in FIG. 31, the seventh conductor 97 in the sixth layer is not exposed through the gap between the resonator 1 in the eighth layer viewed from the line 4 and the ground pattern E1 as the ground conductor. It is preferable that the shape of the ground pattern E1 is changed to a shape provided with the protrusion E11 shown in FIG. 32 so that the seventh conductor 97 is covered with the protrusion E11.

  With such a structure, it is possible to realize out-of-band characteristics and formation of attenuation poles on the low-frequency side and the high-frequency side that are superior to the structure shown in FIG.

  Next, another embodiment in which the resonators 1 and 2 arranged in a stacked manner are configured in two horizontal rows will be described.

  FIG. 33 is an explanatory view showing a band-pass filter having this structure, and FIG. 34 shows a dielectric layer (one layer) provided with the input line 3 and the output line 4 as a specific example of FIG. 33, a resonator 1a. , 2a, a dielectric layer (two layers) provided with resonators 1b and 2b, and a dielectric layer (four layers) provided with coupling conductors in an exploded view. FIG.

  According to this structure, the input line 3 and the output line 4 are formed in one layer, and the widths W of the input line 3 and the output line 4 are narrowed stepwise at the signal input end and the signal output end.

  In the two layers, a ground pattern E1 is formed on the entire surface, and resonators 1a and 2a are formed by providing "U" -shaped gaps in portions corresponding to the respective columns of the ground pattern E1. . In addition, the ground pattern E2 is formed on the entire surface of the three layers, and resonators 1b and 2b are formed by providing "U" -shaped gaps in portions corresponding to the respective columns of the ground pattern E2. . The “U” -shaped gap refers to a gap formed around the resonators 1 and 2 and at a portion other than the grounded end.

  In the four layers, a square frame-shaped gap is formed in the ground pattern E3 provided on the entire surface of the dielectric layer. The inside of this groove becomes the coupling conductor 12. The coupling conductor 12 has a structure that is not connected to any conductor. The length of the coupling conductor 12 in the signal propagation direction is basically λ / 2, where λ is the propagation wavelength inside the dielectric layer at the approximate center frequency of the passband. The coupling conductor 12 has a function of coupling the resonators 1b and 2b on the bottom surface.

  As described above, band-pass filters similar to those in FIG. 5 are configured in two horizontal rows so that signals are folded back by the coupling conductor 12.

  Therefore, since the height of the bandpass filter can be halved, it is possible to realize a reduction in the height of the wireless communication apparatus in which the bandpass filter is mounted. At the same time, since the input line 3 and the output line 4 can be formed on the same dielectric, input / output signals can be easily input and output.

  Finally, FIG. 35 shows a configuration example of a wireless communication device equipped with the bandpass filter described above.

  According to FIG. 35, the wireless communication device includes a baseband IC 25 that processes baseband signals, an RFIC 24 that processes high-frequency signals, a balun 23 that converts balanced signals and unbalanced signals, the bandpass filter 22 of the present invention, and transmission / reception. It consists of a high-frequency switch 21 for switching and an antenna.

  The RFIC 24 performs frequency conversion and high frequency amplification of a transmission signal acquired from the baseband IC 25, and performs low noise amplification of a reception signal. The high-frequency switch 21 is a switch that temporally switches the transmission and reception paths.

  The band-pass filter 22 functions as a band-pass filter that passes the band of the UWB transmission / reception signal and sharply attenuates the signal outside the band. With this function, mutual interference with other systems can be prevented without attenuating transmitted / received signals.

  Although the embodiments of the present invention have been described above, the present invention is not limited to the embodiments described so far, and various modifications can be made within the scope of the present invention.

  With the configuration in which ground conductors are provided above and below the filter of FIG. 1, the pass characteristic S21 and the reflection characteristic S11 of the bandpass filter were calculated using simulation software. The calculation conditions are dielectric constant 9.4, capacitor C1 = C2 = 0.6 pF, length L of resonators 1 and 2 = 4 mm, distance S between resonators 1 and 2 = 0.06 mm The distance D between the ground conductors was 0.9 mm, and the widths W of the resonators 1 and 2 were 0.1 mm (see FIG. 10 for the dimensions of each part).

  The result is shown in the graph of FIG. The horizontal axis of the graph represents frequency, and the vertical axis represents attenuation.

  FIG. 36 shows that the pass loss is less than 1.5 dB within the pass band of about 1.5 GHz from 3.16 GHz to 4.75 GHz. An attenuation pole is generated in the vicinity of about 6 GHz. Furthermore, even at 5.75 GHz or more, it is shown that the attenuation amount is 8 dB over 10 GHz.

  Therefore, it can be understood that the present invention can realize a filter having a low loss and a steep attenuation characteristic in a wide band of 1.5 GHz. Further, since the filter thickness D is sufficiently low, 0.9 mm, it can be mounted on a low-profile wireless communication device.

  FIG. 37 is a graph showing the pass characteristic S21 and the reflection characteristic S11 when the band-pass filter having the input / output end of FIG. 2 on the opposite side is sandwiched between the upper and lower ground conductors. The calculation conditions are: dielectric constant 9.4 of the dielectric, capacitor C1 = C2 = 3 pF, length L of the resonators 1 and 2 = 4 mm, distance S between the resonators 1 and 2 = 0.06 mm, upper and lower ground The distance D between the conductors was 0.9 mm, and the width W of the resonators 1 and 2 was 0.1 mm.

  According to the graph of FIG. 37, the pass band is further wider than that of FIG. It is shown that the pass loss is less than 1.5 dB within the passband. A sufficient amount of attenuation outside the passband is also obtained. The reason why the broadband is realized in this way is considered to be that the amount of coupling between the resonators is increased by arranging the ground ends of the resonators 1 and 2 to be staggered.

  FIG. 38 shows a band-pass filter in which the input / output ends of FIG. 3 are on the same side and the ungrounded ends of the resonators 1 and 2 are grounded via the lumped constant capacitor elements C1 and C2. It is a graph of the transmission characteristic S21 and the reflection characteristic S11 when sandwiched between two. The calculation conditions are: dielectric constant 9.4, capacitor C1 = C2 = 0.8 pF, capacitor C3 = C4 = 0.2 pF, distance S = 0.06 mm between resonators 1 and 2, upper and lower ground conductors The distance D between them was 0.9 mm, and the width W of the resonators 1 and 2 was 0.1 mm. However, the length L of the resonators 1 and 2 was set to a value 3.5 mm shorter than 4 mm.

  According to the graph of FIG. 38, a wider band can be realized as compared with FIG. Moreover, since the length L of the resonators 1 and 2 can be shortened, it is considered that it is further advantageous for miniaturization of a wireless communication device in which this bandpass filter is mounted.

  In FIG. 39, the input line 3 and the output line 4 coupled to the resonators 1 and 2 are used as the signal input unit and output unit, and the ground patterns E1 and E2 are used to form the resonators 1 and 2. FIG. 12 shows graphs of transmission characteristics S21 and reflection characteristics S11 of a band-pass filter formed on a dielectric layer, that is, a band-pass filter having the configuration shown in FIGS. The calculation conditions are as follows: the relative permittivity of the dielectric material 2.2, the length L of the resonators 1 and 7 = 7 mm, the distance between the input / output line 3 and the resonator 1 of 0.51 mm, and the distance S between the resonators 1 and 2 0.51 mm, distance between resonator 2 and input / output line 4 0.51 mm, distance D between upper and lower input / output lines 3, 4 is 1.55 mm, width W of input / output lines 3, 4 and resonators 1, 2 = 3.8 mm and Wa = 1.6 mm.

  According to the graph of FIG. 39, it can be seen that a wide band pass characteristic is obtained although there is a pass attenuation amount of about 3 dB in the pass band.

  FIG. 40 is a graph of the pass characteristic S21 and the reflection characteristic S11 of the bandpass filter in which the resonators shown in FIGS. 33 and 34 are arranged in two rows in the upper and lower stages. The calculation conditions are as follows: the relative permittivity of the dielectric material 2.2, the length L of the resonators 1 and 7 = 7 mm, the distance between the input / output line 3 and the resonator 1 of 0.51 mm, and the distance S between the resonators 1 and 2 0.51 mm, the distance between the resonator 2 and the input / output line 4 is 0.51 mm, the distance D between the upper and lower input / output lines 3 and 4 is 1.55 mm, and the width W of the input / output lines 3 and 4 and the resonators 1 and 2 = 3.8 mm and Wa = 1.6 mm.

  According to the graph of FIG. 40, it can be seen that although the loss of the pass band is reduced, a wide band pass characteristic is obtained, and at the same time, the attenuation characteristic outside the band increases the steepness.

In FIG. 41, the input line 3 and the output line 4 coupled to the resonator 1 are used as the signal input unit and the output unit, and the ground pattern E1 is formed on the dielectric layer on which the resonator 1 is formed. A graph of the pass characteristic S21 of a band-pass filter in which inductance is added to each resonator, that is, a five-stage filter having the configuration shown in FIGS. Here, the configuration shown in FIG. 19 is such that each resonator is formed in a step shape (stepwise) toward the ground end as viewed from the stacking direction. For comparison, a calculation result of a five-stage filter having a structure shown in FIG. 11 (a structure having no electrode step in FIG. 19) in which no new attenuation pole is formed is also shown. The calculation conditions are as follows: the relative permittivity of the dielectric material 2.2, the length of the conventional resonator 1 = 7 mm, the length L of the step-forming resonator L = 5.5 mm (inner step portion W ₀ = 0.6 mm, L ₀ = 0.4 mm), the distance between the input line 3 and the resonator 1 is 0.2 mm, the distance between the resonators S = 0.65 mm, the distance between the resonator and the output line 4 is 0.2 mm, the input / output lines 3, 4 and The width 1 of the resonator 1 was set to 3.2 mm, Wa = 0.6 mm, and length L = 5.5 mm.

  With the structure of FIG. 19, the resonator length can be reduced by 1.5 mm compared to the structure shown in FIG. Moreover, according to the graph of FIG. 41, the out-of-band characteristic on the high frequency side is greatly improved. For example, in this case, the attenuation range of −20 dB or less is improved to 19 GHz by adopting the structure of FIG. 19 while the structure shown in FIG. 11 is up to 15 GHz.

  In FIG. 42, an input line 3 and an output line 4 coupled to the resonator 1 are used as a signal input unit and an output unit, a ground pattern E1 is formed so as to surround the resonator 1, and the input line 3 and the output line 4 is formed of conductors on the same plane and connected with via conductors 51 to couple the input and output, that is, the pass characteristic S21 of the five-stage bandpass filter having the configuration shown in FIG. A graph is shown. Here, for comparison, the calculation result of the five-stage filter of the structure shown in FIG. 13 (the structure without the second conductor 92, the third conductor 93, and the via conductor in FIG. 22) in which no new attenuation pole is formed. Are also shown. The calculation conditions are as follows: the relative permittivity of the dielectric material 2.2, the length L of the resonator 1 = 7 mm, the distance between the input / output line 3 and the resonator 1 0.2 mm, the distance S between the resonators S = 0.75 mm, Output lines 3 and 4 and resonator 1 width W = 3.4 mm, Wa = 0.6 mm, electrode 9 width W = 0.8 mm, length L = 0.7 mm, electrode 9 and input line 3 and output line 4 and the distance g = 0.7 mm.

  According to the graph of FIG. 42, a new attenuation pole is formed on the high frequency side in the structure shown in FIG. 22, and a steep skirt characteristic can be achieved as compared with the structure shown in FIG. By using this method, it is possible to obtain a steep skirt characteristic with a small number of steps, which is effective for miniaturization and low loss.

  In FIG. 43, the input line 3 and the output line 4 coupled to the resonator 1 are used as a signal input unit and an output unit, a ground pattern E1 is formed so as to surround the resonator 1, and an upper side of an arbitrary resonator. Alternatively, a conductor is formed close to the lower side and a conductor is formed close to the upper side or the lower side of a resonator different from this resonator, and these are connected by the via conductor 51, that is, FIG. The graph of the pass characteristic S21 of the five-stage band pass filter of the structure is shown. Here, for comparison, calculation of a five-stage filter having a structure shown in FIG. 13 (a structure without the fourth conductor 94, the fifth conductor 95, and the via conductor 51 in FIG. 25) in which no new attenuation pole is formed. The results are also shown. The calculation conditions are as follows: the relative permittivity of the dielectric material 2.2, the length L of the resonator 1 = 7 mm, the distance between the input / output line 3 and the resonator 1 0.2 mm, the distance S between the resonators S = 0.75 mm, Output lines 3 and 4 and resonator 1 width W = 3.4 mm, Wa = 0.6 mm, electrode 9 width W = 3.4 mm, length L = 6.6 mm, electrode 9 and fourth and seventh layers The distance d from the eye resonator 1 was set to 0.2 mm.

  According to the graph of FIG. 43, a new attenuation pole is formed on the low frequency side in the structure shown in FIG. 29, and a steep skirt characteristic can be achieved as compared with the structure shown in FIG. By using this method, it is possible to obtain a steep skirt characteristic with a small number of steps, which is effective for miniaturization and low loss.

  Further, by making the structure of FIG. 22 and the structure of FIG. 29, attenuation poles can be formed on the high frequency side and the low frequency side, and steep skirt characteristics can be obtained.

Furthermore, the pass characteristic S21 of the bandpass filter having the pattern structure shown in FIG. 30 was calculated using simulation software. The calculation conditions are as follows: relative permittivity of dielectric: 2.2, length L of step-forming resonator 1 = 5.2 mm, width W = 3.2 mm (inner step portion width W ₀ = 1.1 mm, length) L ₀ = 0.4 mm), the distance between the input line 3 and the resonator 1 is 0.2 mm, the distance between the resonators S = 0.65 mm, the distance between the resonator and the output line 4 is 0.2 mm, and the input / output lines 3 and 4 The width W = 3.2 mm, the length L = 5.4 mm, and the width Wa of the input / output lines 3a, 4a = 0.6 mm. The lengths Lg of the electrodes 92 and 93 were 0.7 mm, the width Wg was 1.0 mm, and the distance g between the input line 3 and the output line 4 was 0.5 mm. The fourth conductor 94 and the fifth conductor 95 have a width W = 3.0 mm and a length L = 4.4 mm, and the line 2 has W = 0.1 mm and a length L = 1.2 mm. 94, and the distance d between the fifth conductor 95 and the resonators 1 in the fourth and sixth layers was d = 0.2 mm.

On the other hand, the pass characteristic S21 of the bandpass filter having the structure shown in FIG. 32 of the present invention was calculated using simulation software. The calculation conditions are as follows: relative permittivity of dielectric: 2.2, length L of step-forming resonator 1 = 5.2 mm, width W = 3.2 mm (inner step portion width W ₀ = 1.2 mm, length) L ₀ = 0.4 mm), the distance between the input line 3 and the resonator 1 is 0.2 mm, the distance between the resonators S = 0.65 mm, the distance between the resonator and the output line 4 is 0.2 mm, and the input / output lines 3 and 4 The width W = 3.2 mm, the length L = 5.4 mm, and the width Wa of the input / output lines 3a, 4a = 0.6 mm. The length Lg of the second conductor 92 and the third conductor 93 is 0.6 mm, the width Wg is 1.0 mm, and the distance g between the input line 3 and the output line 4 is 0.5 mm. The width W of the fourth conductor 94 and the fifth conductor 95 is 3.0 mm, the length L is 3.6 mm, and the sixth conductor 96 in the fourth layer and the seventh conductor 97 in the sixth layer are W = 0.2 mm, length L = 2.0 mm, distance d between the fourth conductor 94 and the fifth conductor 95 and the resonators 1 in the fourth and sixth layers was set to 0.2 mm. Moreover, the size of the protrusion E11 of the third layer and the seventh layer was set such that the width W = 1.2 mm and the length L = 0.8 mm.
The calculation results are shown in FIG.

  According to the graph of FIG. 44, attenuation poles are formed on the low-frequency side and the high-frequency side regardless of which structure of FIGS. 30 and 32 is used, and a steep skirt characteristic can be achieved. By using this method, it is possible to obtain a steep skirt characteristic with a small number of steps, which is effective for miniaturization and low loss. However, in the structure shown in FIG. 30, a sharp resonance peak is generated outside the band, and the out-of-band characteristics are slightly deteriorated. On the other hand, in the structure shown in FIG. 32, this resonance peak is suppressed, and good out-of-band characteristics are achieved over a wide range.

It is explanatory drawing which shows an example of the circuit structure of the band pass filter of this invention. It is explanatory drawing which shows the other circuit structure of the band pass filter of this invention. It is explanatory drawing which shows other circuit structure of the band pass filter of this invention. It is explanatory drawing which shows the further another circuit structure of the band pass filter of this invention. It is sectional drawing which shows the structure of the band pass filter of this invention using the input line 3 and the output line 4 as an input part and an output part. It is sectional drawing which shows the other structure of the band pass filter of this invention using the input line 3 and the output line 4 as an input part and an output part. It is sectional drawing which shows other structure of the band pass filter of this invention using the input line 3 and the output line 4 as an input part and an output part. It is sectional drawing which shows other structure of the band pass filter of this invention using the input line 3 and the output line 4 as an input part and an output part. It is sectional drawing which shows the structure which formed the band pass filter shown in FIG. 2 inside the dielectric multilayer substrate. FIG. 10 is a perspective view of the bandpass filter shown in FIG. 9. It is explanatory drawing which shows the conductor pattern of each layer of the band pass filter shown in FIG. It is explanatory drawing which shows the conductor pattern of each layer of the band pass filter shown in FIG. It is explanatory drawing which shows the conductor pattern of each layer of the band pass filter shown in FIG. It is explanatory drawing which shows the conductor pattern of each layer of the band pass filter shown in FIG. It is explanatory drawing which shows an example of the circuit structure for size reduction and the out-of-band characteristic improvement of the band pass filter of this invention. It is explanatory drawing which shows an example of the conductor pattern of each layer of the band pass filter shown in FIG. FIG. 16 is an explanatory diagram illustrating another example of the conductor pattern of each layer of the bandpass filter illustrated in FIG. 15. It is explanatory drawing which shows the other example of the circuit structure for size reduction and out-of-band characteristic improvement of the band pass filter of this invention. It is explanatory drawing which shows an example of the conductor pattern of each layer of the band pass filter shown in FIG. It is explanatory drawing which shows an example of the circuit structure for forming an attenuation pole in the band pass filter of this invention. It is explanatory drawing which shows the other example of the circuit structure for forming an attenuation pole in the band pass filter of this invention. It is explanatory drawing which shows an example of the conductor pattern of each layer for forming an attenuation pole in the band pass filter shown in FIG. 20 and FIG. It is explanatory drawing which shows the other example of the conductor pattern of each layer for forming an attenuation pole in the band pass filter shown in FIG. 20 and FIG. FIG. 22 is an explanatory diagram showing still another example of a conductor pattern of each layer for forming an attenuation pole in the bandpass filter shown in FIGS. 20 and 21. FIG. 22 is an explanatory diagram showing still another example of a conductor pattern of each layer for forming an attenuation pole in the bandpass filter shown in FIGS. 20 and 21. FIG. 22 is an explanatory diagram showing still another example of the conductor pattern of each layer for forming an attenuation pole in the bandpass filter shown in FIGS. 20 and 21. It is explanatory drawing which shows the further another example of the circuit structure for forming an attenuation pole in the band pass filter of this invention. It is explanatory drawing which shows the further another example of the circuit structure for forming an attenuation pole in the band pass filter of this invention. It is explanatory drawing which shows an example of the conductor pattern of each layer for forming an attenuation pole in the bandpass filter shown in FIG. 27 and FIG. FIG. 30 is an explanatory diagram showing an example of a conductor pattern of each layer having a structure in which the bandpass filters shown in FIGS. 19, 22 and 29 are combined. It is explanatory drawing which shows an example of the structure which improved the band pass filter shown in FIG. It is explanatory drawing which shows the other example of the structure which improved the band pass filter shown in FIG. It is explanatory drawing which shows the band pass filter which comprised the resonator 2 rows horizontally. It is explanatory drawing which shows the conductor pattern of each layer of the band pass filter shown in FIG. It is a block diagram which shows the structural example of the radio | wireless communication apparatus carrying the band pass filter of this invention. It is a graph which shows the result of having calculated the passage characteristic S21 and the reflection characteristic S11 of the bandpass filter shown in FIG. 1 using simulation software. It is a graph which shows the result of having calculated the passage characteristic S21 and the reflection characteristic S11 of the bandpass filter shown in FIG. 2 using simulation software. It is a graph which shows the result of having calculated the passage characteristic S21 and the reflection characteristic S11 of the bandpass filter shown in FIG. 3 using simulation software. It is a graph which shows the result of having calculated the passage characteristic S21 and the reflection characteristic S11 of the bandpass filter shown in FIG. 5 using simulation software. FIG. 34 is a graph of a pass characteristic S21 and a reflection characteristic S11 of a band-pass filter configured by horizontally arranging resonators arranged in two upper and lower stages shown in FIG. It is a graph which shows the result of having calculated the pass characteristic S21 of the band pass filter shown in FIG. 19 using simulation software. It is a graph which shows the result of having calculated the passage characteristic S21 of the band pass filter shown in FIG. 22 using simulation software. It is a graph which shows the result of having calculated the pass characteristic S21 of the band pass filter shown in FIG. 29 using simulation software. It is a graph which shows the result of having calculated and compared the pass characteristic S21 of the band pass filter shown in FIG. 30 and the band pass filter shown in FIG. 32 using simulation software. It is a perspective view which shows the conventional planar circuit filter which arranged two microstrip lines on the dielectric substrate.

Explanation of symbols

1, 2: Resonator 3: Input line 4: Output lines 5, 6, 7, 8: Via hole conductor 51: Via conductor 91: First conductor 92: Second conductor 93: Third conductor 94: Fourth Conductor 95: fifth conductor 96: sixth conductor 97: seventh conductor 10, 11 pad 21 high frequency switch 22 band pass filter 23 balun 24 RFIC
25 Baseband IC
A1, A2 Conductor pattern E1, E2 Ground pattern

Claims (25)

N (N ≧ 2) resonators in which a plurality of conductor patterns and dielectric layers are alternately stacked, at least partially overlapping when viewed from the stacking direction and electromagnetically coupled to each other;
An input unit and an output unit coupled to each of the two resonators selected from the N resonators;
One end (ground end) of each of the N resonators is grounded,
A bandpass characterized in that the length of the signal propagation direction of each of the N resonators is basically λ / 4 where λ is a propagation wavelength inside the dielectric layer at a substantially center frequency of a pass band. filter. The band-pass filter according to claim 1, wherein each of the N resonators has a structure including a strip line, a microstrip line, or a coplanar line. The band-pass filter according to claim 1 or 2, wherein a ground end of each of the N resonators is present at an end portion on the same side of the resonator as viewed from the stacking direction. 3. The band-pass filter according to claim 1, wherein a ground end of each of the N resonators is present at an end portion on the opposite side of the previous-stage resonator as viewed from the stacking direction. The band pass filter according to claim 1, wherein the input unit and the output unit include a capacitor element or an inductor element coupled to the resonator. The band-pass filter according to claim 1, wherein the input unit and the output unit include an input line and an output line coupled to the resonator. The band-pass filter according to claim 6, wherein an input direction to the input line is different from an output direction from the output line. The bandpass filter according to claim 6, wherein an input direction to the input line and an output direction from the output line are the same. Each of the N resonators is formed in a rectangular shape when viewed from the stacking direction, and the resonance is on the same plane as each resonator so that only one end (ground end) of the rectangular resonator is grounded. The band pass filter according to any one of claims 6 to 8, wherein a ground conductor is provided surrounding the periphery of the vessel. 10. The bandpass filter according to claim 9, wherein a capacitance is added between a non-grounded end as the other end with respect to the grounded end and a portion of the ground conductor adjacent to the non-grounded end. . A first conductor is provided close to an upper side or a lower side of a non-grounded end of the resonator, and a via conductor that connects the first conductor and the ground conductor is provided, whereby the resonator The band-pass filter according to claim 10, wherein the capacitance is formed between the first conductor and the first conductor. The band-pass filter according to claim 11, wherein each of the N resonators is formed to be narrowed stepwise or continuously toward the ground end side when viewed from the stacking direction. The bandpass filter according to any one of claims 6 to 12, wherein a capacitance or an inductance is added between the input line and the output line. A second conductor is provided on the same plane as the input line, a third conductor is provided on the same plane as the output line, and a via conductor connecting the second conductor and the third conductor is provided. The bandpass filter according to claim 13, wherein the capacitance or the inductance is added between the input line and the output line by being provided. A second conductor is provided in proximity to the upper side or lower side of the input line, a third conductor is provided in proximity to the upper side or lower side of the output line, and the second conductor and the third conductor are provided. 14. The bandpass filter according to claim 13, wherein the capacitance or the inductance is added between the input line and the output line by providing a via conductor connecting the conductor. The second conductor and the third conductor are formed so as to be narrowed stepwise or continuously toward a portion connected to a via conductor when viewed from the stacking direction. The band pass filter according to 14 or 15. The band pass filter according to any one of claims 1 to 16, wherein a capacitance or an inductance is added between any two resonators selected from the N resonators. A fourth conductor is provided on the same plane as any one of the N resonators, and a fifth conductor is provided on the same plane as a resonator different from the one resonator. The capacitance or the inductance is added between the two arbitrary resonators by providing a via conductor that connects the fourth conductor and the fifth conductor. The band-pass filter according to claim 17. A fourth conductor is provided adjacent to an upper side or a lower side of any one of the N resonators, and an upper side or a lower side of a resonator different from the arbitrary one resonator. A fifth conductor is provided adjacent to the first conductor, and a via conductor is provided to connect the fourth conductor and the fifth conductor, whereby the capacitance or the inductance is provided between the two arbitrary resonators. The bandpass filter according to claim 17, further comprising: At least one resonator is provided between the fourth conductor or the fifth conductor and the input unit or the output unit, and the fourth conductor viewed from the input unit and the output unit by the resonator. The bandpass filter according to claim 19, wherein the conductor and the fifth conductor are covered. The fourth conductor and the fifth conductor are formed so as to be narrowed stepwise or continuously toward a portion connected to a via conductor when viewed from the stacking direction. The band pass filter according to 18-20. The laminated structure composed of the N resonators is configured in two horizontal rows,
The number of stacked resonators constituting each row is the same,
The resonator coupled to the input unit and the resonator coupled to the output unit are disposed on the top surface of each row,
The band-pass filter according to any one of claims 1 to 21, wherein coupling conductors for coupling the resonators on the bottom surface are arranged in two rows. 23. The width of the input line or the output line is formed in a step shape when viewed from the stacking direction at an end of a portion overlapping with the resonator. Bandpass filter. A high-frequency module comprising the band-pass filter according to claim 1. A wireless communication device using the bandpass filter according to claim 23 or the high-frequency module according to claim 23.

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