JP2007057340A - Oscillation circuit and angular velocity sensor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To realize an oscillation circuit which widely deals with characteristic variations in oscillators and is excellent in stability, and to provide a precise angular velocity sensor employing the oscillation circuit. <P>SOLUTION: The oscillation circuit comprises a crystal oscillator 2 having drive electrodes 3, 4, a current-to-voltage converting circuit 7 for converting output current of the drive electrodes of the crystal oscillator 2 into an AC voltage, a variable gain amplifier 10 for amplifying an AC signal output from the current-to-voltage converting circuit 7, a fixed gain amplifier 11 for amplifying the AC signal output from the current-to-voltage converting circuit 7, and an AGC circuit 9 for controlling the output current of the crystal oscillator so as to be constant. The fixed gain amplifier 11 is equipped with a switch circuit for switching at least two gains. The stable oscillation circuit is thereby realized. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to an oscillation circuit that stably oscillates a piezoelectric vibrator such as a quartz crystal vibrator with varying characteristics, and an angular velocity sensor using the oscillation circuit.

  Conventionally, an angular velocity sensor using a piezoelectric vibrator or the like is an indispensable sensor for controlling the attitude of a car navigation system or robot, and as the demand for the car navigation system or robot increases, the operating temperature range is increased. However, there is an increasing demand for a wide and highly accurate angular velocity sensor. In order to meet these demands, a vibration control device is disclosed as an angular velocity sensor including an automatic gain control circuit that controls the vibration amplitude of a vibrator to be constant even when the ambient temperature changes (for example, Patent Document 1). reference).

  Hereinafter, Patent Document 1 as a conventional technique will be described with reference to the drawings. FIG. 10 is a block diagram of a conventional vibration control device that detects angular velocity. 50 is a vibration control device, and the output signal P50 of the built-in drive device 60 is connected to the non-inverting input terminals of the feedback amplifiers 51 and 52, respectively, and the inverting input terminals of the feedback amplifiers 51 and 52 are formed in the vibrator 53. Connected to one of the electrodes of the piezoelectric elements 53a and 53b. The other electrodes of the piezoelectric elements 53a and 53b are connected to the inverting input terminal of the sum amplifier 54 having the feedback resistor 54a, and the product of the combined current passing through the piezoelectric elements 53a and 53b from the output terminal and the feedback resistor 54a. The self-excited signal P51 corresponding to is output and applied to the input terminal 60a of the driving device 60, and the vibrator 53 is self-excited.

  FIG. 11 shows an exemplary configuration of the driving device 60. The driving device 60 includes an automatic gain control circuit (hereinafter abbreviated as AGC circuit) 61, a non-inverting amplifier 64 as a gain fixing amplifier, and an inverting amplifier 65. The AGC circuit 61 has a comparator 62 and an amplitude controller 63. The comparator 62 converts the self-excited signal P51 from the input terminal 60a into a direct current and compares it with a reference level, and generates a comparison signal P52 according to the comparison result. Then, it is supplied to the amplitude controller 63.

  The amplitude controller 63 receives the self-excited signal P51 from the input terminal 60a, controls the amplitude of the self-excited signal P51 based on the comparison signal P52 from the comparator 62, and outputs an AGC signal P53. The AGC signal P53 is amplified to a predetermined magnitude by the non-inverting amplifier 64, then inverted by the inverting amplifier 65, and output as the output signal P50 from the output terminal 60b.

  Thus, the vibrator 53 continues self-excited vibration while the amplitude is controlled by the driving device 60, and current values flowing through the equivalent resistances of the piezoelectric elements 53a and 53b at the inverting input terminals of the feedback amplifiers 51 and 52, respectively. Therefore, only a current corresponding to the Coriolis force flows through the feedback resistors 51a and 52a of the feedback amplifiers 51 and 52. As a result, by amplifying the outputs of the feedback amplifiers 51 and 52 by the differential amplifier 55, the phase component corresponding to the input angular velocity can be amplified, so that a signal corresponding to the detected angular velocity can be output from the output terminal 56.

  With the above configuration, even if the equivalent resistance of the vibrator 53 fluctuates due to a change in ambient temperature, the combined current passing through the piezoelectric elements 53a and 53b operates so as to be kept constant, so that the vibrator 53 has a constant amplitude. As a result, the angular velocity can be detected with little influence of the ambient temperature change.

JP-A-8-14913 (page 4, Fig. 1)

  However, the vibrator 53 has a variation in equivalent resistance value due to a variation in product shape and the like. The variation amount of the equivalent resistance value varies depending on the manufacturing lot of the vibrator 53, and there is a small variation lot. There are also lots that vary greatly. Here, when the variation amount of the equivalent resistance value of the vibrator 53 is small, it is absorbed by the operation of the AGC circuit 61. However, when the variation amount is large, the gain of the amplitude controller 63 of the AGC circuit 61 is variable. Since there is a limit to the range, a situation occurs that cannot cope with variations in equivalent resistance value. In such a case, the vibrators 53 are sorted in advance, and only vibrators having a small variation in equivalent resistance value are used.

  However, particularly when the resonator is a crystal resonator, crystal impedance (hereinafter abbreviated as CI value), which is equivalent resistance, varies greatly due to slight variations in shape. A large amount of crystal resonators are generated, and the mass production efficiency of the crystal resonators is remarkably lowered, resulting in a great problem of a significant cost increase of the crystal resonators.

  Further, if the variable range of the amplification factor of the amplitude controller 63 is maximized, the number of vibrators to be discarded can be reduced to some extent. However, if the variable range of the amplification factor is increased, the stability of the amplitude controller 63 deteriorates. In addition, there is a new problem that noise cannot be reduced and maintained.

  SUMMARY OF THE INVENTION An object of the present invention is to solve the above-described problems, to realize an oscillation circuit that is widely compatible with variations in characteristics of vibrators and has excellent stability, and to provide a highly accurate angular velocity sensor using this oscillation circuit. .

  In order to solve the above problems, the oscillation circuit and the angular velocity sensor of the present invention employ the following configurations.

  The oscillation circuit of the present invention has a drive electrode, a piezoelectric vibrator that is excited by a drive voltage applied to the drive electrode, and a current / current that converts an output current from the drive electrode of the piezoelectric vibrator into an AC voltage. A voltage conversion circuit, a gain variable amplifier capable of amplifying an AC signal from the current / voltage conversion circuit to vary an amplification factor, and a gain fixed amplifier in which the AC signal from the current / voltage conversion circuit is amplified and the amplification factor is fixed A rectifier circuit that rectifies an output current from the drive electrode, and a comparison that compares a signal output from the rectifier circuit with a predetermined reference signal and outputs a control voltage in accordance with fluctuations in the output signal of the rectifier circuit A drive voltage applied to the drive electrode of the piezoelectric vibrator based on the control voltage output from the comparison circuit, and the output current from the piezoelectric vibrator is kept constant. And an automatic gain control circuit for controlling the gain, wherein the gain-fixed amplifier has at least two gains, and switching means for switching the at least two gains according to a predetermined situation is provided. It is characterized by that.

  As a result, by switching the gain of the fixed gain amplifier according to the situation, it is not necessary to maximize the variable range of the variable gain amplifier, and a stable oscillation circuit with less noise can be realized. In addition, by switching the gain of the fixed gain amplifier, the piezoelectric vibrators that have been discarded up to now can be used, so that mass production of piezoelectric vibrators and cost reduction by increasing production efficiency can be realized.

  The switching means includes a resistance element and a switch.

  Accordingly, the gain of the fixed gain amplifier can be easily and reliably switched by switching the resistance element by the switch.

  Further, the switching means includes a capacitor and a switch.

  As a result, the capacitor built in the IC has good relative accuracy and excellent temperature characteristics. Therefore, it is easy to integrate the switching means with the oscillation circuit into an IC, and it has high accuracy and excellent temperature characteristics. An oscillation circuit can be realized.

  The switch is a MOS transistor.

  As a result, the switch of the switching means can be incorporated as a MOS transistor in the IC of the oscillation circuit having the CMOS structure, and a small and highly reliable oscillation circuit can be realized.

  The piezoelectric vibrator is a quartz crystal vibrator.

  As a result, the crystal resonator has an extremely high Q value and excellent temperature characteristics as compared with a resonator using ceramic or the like, and thus a high-performance oscillation circuit with excellent stability can be realized.

  An angular velocity sensor according to the present invention detects a stress caused by Coriolis force generated in the piezoelectric vibrator by the oscillation circuit described above and a detection electrode provided on the piezoelectric vibrator as a charge, and a current / voltage conversion circuit of the oscillation circuit. And a detection circuit that outputs a detection signal corresponding to the angular velocity by performing synchronous detection with an AC signal from the above or a drive voltage applied to a drive electrode of the piezoelectric vibrator.

  As a result, Coriolis force generated in the piezoelectric vibrator is detected by the detection electrode provided in the piezoelectric vibrator of the oscillation circuit, and the synchronous detection is efficiently performed, so that a highly accurate angular velocity sensor with excellent stability can be realized. .

  As described above, the oscillation circuit of the present invention switches the gain of the fixed gain amplifier according to the situation, so that it is not necessary to maximize the variable range of the variable gain amplifier, and a stable oscillation circuit can be realized. In addition, by switching the gain of the fixed gain amplifier, it is possible to handle a wide range of variations in piezoelectric vibrators, so it is possible to use piezoelectric vibrators that have been discarded up to now, and mass production and production of piezoelectric vibrators Cost can be reduced by increasing efficiency.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. FIG. 1 is a block diagram showing an outline of the oscillation circuit of the present invention. FIG. 2 is a cross-sectional view showing the electrode structure of the crystal resonator of the oscillation circuit of the present invention. FIG. 3 is a graph for explaining the amplification factor characteristic of the variable gain amplifier of the oscillation circuit of the present invention. FIG. 4 is a circuit diagram showing an example using a resistor and a switch of a fixed gain amplifier constituting the oscillation circuit of the present invention. FIG. 5A is a table showing an example of the switching operation range of the gain fixed amplifier of the oscillation circuit of the present invention. FIG. 5B is a table showing an example of the operation range of a conventional gain-fixed amplifier.

FIG. 6 is a circuit diagram showing another example using resistors and switches of a fixed gain amplifier constituting the oscillation circuit of the present invention. FIG. 7 is a circuit diagram showing an example using a capacitor and a switch of a fixed gain amplifier constituting the oscillation circuit of the present invention. FIG. 8 is a circuit diagram showing another example using a capacitor and a switch of a fixed gain amplifier constituting the oscillation circuit of the present invention. FIG. 9 is a block diagram showing an outline of the angular velocity sensor of the present invention.

  The configuration of the oscillation circuit of the present invention will be described with reference to FIG. In FIG. 1, reference numeral 1 denotes an oscillation circuit according to the present invention. Reference numeral 2 denotes a quartz crystal vibrator as a piezoelectric vibrator, which includes a pair of drive electrodes 3 and 4. Reference numeral 7 denotes a current-voltage conversion circuit (hereinafter abbreviated as I / V conversion circuit), which inputs an output current Iout flowing out from one drive electrode 4 of the crystal resonator 2 and outputs an AC signal V1.

  Reference numeral 8 denotes a low-pass filter (hereinafter abbreviated as LPF), which receives an AC signal V1 and outputs a filter output signal V2. Reference numeral 9 denotes an automatic gain control circuit (hereinafter abbreviated as AGC circuit), which includes a rectifier circuit 9a, a reference signal source 9b, and a comparison circuit 9c. The rectifier circuit 9a receives the AC signal V1 and outputs a DC signal V3, and the comparator circuit 9c inputs the DC signal V3 and a predetermined reference signal V4 output from the reference signal source 9b and outputs a control voltage V5.

  A variable gain amplifier 10 receives the filter output signal V2 from the LPF 8 and outputs an amplified signal V6 based on the control voltage V5. Reference numeral 11 denotes a gain-fixed amplifier, which includes an amplifier 12 and a switching circuit 13 as switching means for switching the gain. The amplifier 12 receives the amplified signal V6 from the variable gain amplifier 10, switches the amplification factor based on the switching signal V7 from the switching circuit 13, outputs the driving voltage Vout, and applies it to the driving electrode 3 of the crystal resonator 2. .

  Reference numeral 14 denotes a storage circuit composed of a nonvolatile memory or the like, which outputs a switching control signal V8 and inputs it to the switching circuit 13. The switching control signal V8 may be 1 bit or a plurality of bits of 2 bits or more depending on the specification of the oscillation circuit. Reference numeral 15 denotes a phase circuit composed of an all-pass filter circuit, which receives an AC signal V1 that is an output voltage of the I / V conversion circuit 7 to perform phase adjustment and outputs a detection control signal V9. The phase circuit 15 may input the drive voltage Vout.

  Next, an example of the electrode structure of the crystal unit 2 will be described based on the cross-sectional view of FIG. In FIG. 2, the crystal resonator 2 is a resonator having a resonance frequency of several tens of KHz, and is a three-pronged resonator having three branches of two drive branches 2a and 2b and one detection branch 2c. is there. A pair of drive electrodes 3 and 4 are formed on the drive branches 2a and 2b. The drive electrode 3 includes drive electrodes 3a and 3b formed on two opposing surfaces of the drive branch 2a and a drive branch. The driving electrodes 3c and 3d are formed on two opposing surfaces of 2b.

  Further, the drive electrode 4 is constituted by drive electrodes 4a and 4b formed on the other two surfaces facing the drive branch 2a and drive electrodes 4c and 4d formed on the other two surfaces opposed to the drive branch 2b. Become. The drive electrodes 3a, 3b, 3c, and 3d are electrically connected and connected to the outside as the drive electrode 3, and the drive electrodes 4a, 4b, 4c, and 4d are also electrically connected and connected to the outside as the drive electrode 4. Is done.

  A pair of detection electrodes 5 and 6 is formed on the detection branch 2c, and the detection electrode 5 includes detection electrodes 5a and 5b formed on a part of the opposing surface of the detection branch 2c. The electrode 6 is also composed of detection electrodes 6a and 6b formed on part of the opposing surface of the detection branch 2c. The detection electrodes 5a and 5b are electrically connected and connected to the outside as the detection electrode 5, and the detection electrodes 6a and 6b are also electrically connected and connected to the outside as the detection electrode 6. Note that the structure of the crystal resonator 2 is not limited to the three-pronged vibrator as shown in FIG.

Next, the operation of the oscillation circuit 1 of the present invention will be described with reference to FIGS. The crystal resonator 2 is excited by an alternating drive voltage Vout applied to the drive electrode 3, and an alternating output current Iout flows out from the drive electrode 4. The I / V conversion circuit 7 converts an output current Iout of several tens of KHz from the crystal resonator 2 into an AC voltage and outputs the AC voltage V1. The LPF 8 adjusts the phase of the AC signal V1 to establish the oscillation phase condition and remove the high frequency component.

  The rectifier circuit 9a of the AGC circuit 9 performs full-wave rectification or half-wave rectification of the AC signal V1, and outputs a DC signal V3 according to the magnitude of the AC signal V1. The comparison circuit 9c compares the DC signal V3 with a reference signal V4 having a predetermined magnitude, and outputs a control voltage V5 according to the fluctuation of the DC signal V3. The variable gain amplifier 10 varies the amplification factor based on, for example, about 1 to 5 based on the control voltage V5, and variably amplifies the filter output signal V2 based on the AC signal V1 from the crystal unit 2.

  The storage circuit 14 stores information based on the CI value of the crystal resonator 2 to be used, for example, at the manufacturing stage of the oscillation circuit 1, and outputs the information to the gain fixed amplifier 11 as the switching control signal V8. The switching circuit 13 of the fixed gain amplifier 11 operates an internal switch based on the switching control signal V8, outputs a switching signal V7 in order to switch the gain of the amplifier 12, and at least two gains of the amplifier 12 are gained. Switch. The detailed configuration and operation of the gain fixed amplifier 11 will be described later.

  With the above operation, the crystal resonator 2 is driven by the drive voltage Vout which is the output of the fixed gain amplifier 11, and self-oscillation is continued at a frequency equal to the resonance frequency. When the crystal resonator 2 continues to oscillate, the driving branches 2a and 2b shown in FIG. 2 vibrate in the arrow X direction, and the detection branch 2c also synchronizes with the driving branches 2a and 2b in the arrow X direction. Vibrate.

  Here, if the CI value of the crystal unit 2 increases due to a temperature change or the like, the output current Iout flowing through the crystal unit 2 decreases, so the AC signal V1 that is the output of the I / V conversion circuit 7 decreases. . As a result, the level of the DC signal V3 that is the output of the rectifier circuit 9a is lowered, and the level of the control voltage V5 that is the output of the comparison circuit 9c that compares the DC voltage V3 with the reference signal V4 is raised. As a result, the gain of the variable gain amplifier 10 is increased, and the amplified signal V6 that is output from the gain amplifier 10 is increased. Therefore, the drive voltage Vout is also increased, and the increase in the CI value of the crystal resonator 2 is compensated. A decrease in the flowing output current Iout is prevented and the output current Iout is kept constant.

  Similarly, if the CI value of the crystal unit 2 decreases due to a temperature change or the like, the output current Iout flowing through the crystal unit 2 increases, so that the AC signal V1 that is the output of the I / V conversion circuit 7 is large. Become. As a result, the level of the DC signal V3 that is the output of the rectifier circuit 9a increases, and the level of the control voltage V5 that is the output of the comparison circuit 9c that compares the DC voltage V3 and the reference signal V4 decreases. As a result, the gain of the variable gain amplifier 10 is reduced, and the amplified signal V6 as an output thereof is reduced, so that the drive voltage Vout is also reduced, and the decrease in the CI value of the crystal unit 2 is compensated for. An increase in the flowing output current Iout is prevented, and the output current Iout is kept constant.

  FIG. 3 shows an amplification factor characteristic in which the amplification factor of the variable gain amplifier 10 is varied by the control voltage V5. FIG. 3 shows how the amplification factor of the variable gain amplifier 10 varies depending on the fluctuation of the CI value of the crystal resonator. An example of the transition to will be described. Here, it is assumed that the variable gain amplifier 10 has an amplification factor of about 1 when the control voltage V5 is 0 volts, and an amplification factor of about 5 when the control voltage V5 is the maximum Vmax.

  Here, for example, oscillation is continued using a crystal resonator having a standard CI value, and the oscillation circuit 1 operates stably with the gain of the variable gain amplifier 10 at that time being G1 (that is, about 3 times). And At this time, if the CI value of the crystal resonator increases due to a temperature change or the like, as described above, the control voltage V5 from the AGC circuit 9 increases, so that the gain of the variable gain amplifier 10 is G2 (approx. 4 times). )

  Similarly, if the CI value of the crystal resonator decreases due to a temperature change or the like, as described above, the control voltage V5 from the AGC circuit 9 decreases, so that the gain of the variable gain amplifier 10 is G3 (about twice). Approach). As described above, even if the CI value of the crystal resonator 2 fluctuates due to a temperature change or the like, the gain of the variable gain amplifier circuit 10 is varied by the AGC circuit 9, and the output current Iout flowing through the crystal resonator 2 is kept constant. Therefore, the vibration amplitude of the crystal resonator 2 is always constant, and stable oscillation can be continued.

  As described above, the variable gain amplifier circuit 10 can vary the gain within a range of about 1 to 5 times. However, the linearity is poor with respect to the control voltage V5 in a region where the gain is low or high. In addition, since the AGC circuit 9 has a longer lock time and noise is likely to be generated, the gain of the variable gain amplifier circuit 10 is about 2 to 4 times as shown in FIG. The rate range is the recommended operating range.

  Next, a detailed circuit configuration of the fixed gain amplifier 11 will be described with reference to FIG. In FIG. 4, an amplifier 12 includes a differential amplifier circuit 12a having a positive input terminal and a negative input terminal, and a feedback resistor Rf as a resistance element connecting the negative input terminal and the output terminal of the differential amplifier circuit 12a. And a divided resistor R1 as a resistance element having one end connected to the negative input terminal of the differential amplifier circuit 12a. The switching circuit 13 includes a MOS transistor SW1 as a switch and a dividing resistor R2 as a resistance element connecting the drain terminal D1 and the source terminal S1 of the MOS transistor SW1.

  The drain terminal D1 of the MOS transistor SW1 is connected to the other end of the dividing resistor R1 of the amplifier 12 as the switching signal V7, the source terminal S1 of the MOS transistor SW1 is connected to the circuit GND, and the gate of the MOS transistor SW1. The terminal G1 is connected to the switching control signal V8 from the memory circuit 14. Further, the positive input terminal of the differential amplifier circuit 12a receives the amplified signal V6 from the variable gain amplifier 10, and the output terminal of the differential amplifier circuit 12a outputs the drive voltage Vout.

  Next, the operation of the fixed gain amplifier 11 will be described. The switching control signal V8 is a digital signal composed of logic “0” or logic “1”. For example, when the switching control signal V8 is logic “0”, the MOS transistor SW1 is turned off. The resistance value between the negative input terminal 12a and GND is R1 + R2, and the gain G of the amplifier 12 is expressed by Equation 1 shown in FIG. When the switching control signal V8 is logic “1”, the MOS transistor SW1 is turned on, so that the dividing resistor R2 is electrically in a shoot state, and the resistance value between the negative input terminal of the differential amplifier circuit 12a and GND is The gain G of the amplifier 12 becomes the equation 2 shown in FIG.

  As an example, if the feedback resistor Rf is 40 KΩ, the dividing resistor R1 is 10 KΩ, and the dividing resistor R2 is 30 KΩ, the gain (Equation 1) when the MOS transistor SW1 is OFF is doubled, and the MOS transistor SW1 When ON is ON, the gain (Equation 2) is 5 times.

  With the above operation, the fixed gain amplifier 11 can switch between the two gains according to the logic of the switching control signal V8. It is a main object of the present invention to realize a stable oscillation circuit by absorbing variations in the CI value of the crystal resonator 2 by switching the gain of the fixed gain amplifier 11.

Next, based on FIG. 5A, the relationship between the gain switching of the fixed gain amplifier 11 and the CI value selection range of the crystal resonator 2 will be described. Here, the crystal resonator 2 measures individual CI values in the manufacturing stage. For example, crystal resonators in regions where the CI value is extremely small are grouped in groups A and C.
Quartz resonators with slightly smaller I values are classified into four groups, group B, crystal resonators with slightly larger CI values as group C, and quartz resonators with extremely large CI values as group D. .

  The crystal oscillators of the group A having an extremely small CI value and the group D having an extremely large CI value are discarded because the operation of the oscillation circuit is unstable and noise increases. The small group B crystal resonators and the group C crystal resonators having slightly larger CI values are dealt with by switching the gain of the fixed gain amplifier circuit 11.

  Specifically, when a crystal resonator of group B having a slightly smaller CI value is used for the oscillation circuit 1, the logic “0” is stored in the memory circuit 14 so that the switching control signal V 8 becomes the logic “0”. To do. As a result, the MOS transistor SW1 of the switching circuit 13 of the fixed gain amplifier 11 is turned off, so that the gain of the fixed gain amplifier 11 is selected to be small (for example, twice).

  When a crystal resonator of group C having a slightly large CI value is used for the oscillation circuit 1, the logic “1” is stored in the memory circuit 14 so that the switching control signal V8 becomes the logic “1”. As a result, the MOS transistor SW1 of the switching circuit 13 of the fixed gain amplifier 11 is turned on, so that a large gain (for example, five times) is selected as the gain of the fixed gain amplifier 11.

  As a result, even if either the group B or the group C is used as the crystal resonator, the difference in the CI value can be apparently canceled by switching the gain of the fixed gain amplifier 11. The operating point is substantially equal to the amplification factor G1 shown in FIG. 3, and even if the CI value of the crystal resonator changes due to a temperature change or the like, the amplification factor of the variable gain amplifier 10 remains within the recommended operating range, and stable oscillation is achieved. You can continue.

  Next, FIG. 5B illustrates an example of a selection range of the CI value of the crystal resonator when a conventional gain-fixed amplifier is used. In this case, since the gain fixed amplifier cannot switch the gain, if the variation range of the CI value of the crystal resonator is wide, it exceeds the recommended operation range of the variable gain amplifier 10 shown in FIG. Problems such as instability occur. For this reason, the crystal resonator can use only products with a narrow CI value variation range. For example, as shown in FIG. 5B, the CI value range can be used only for the group B. , Group A, C, and Group D must be discarded, resulting in a large number of crystal resonators to be discarded.

  As described above, by adopting the oscillation circuit of the present invention, the gain of the fixed gain amplifier is switched according to the CI value of the crystal resonator using the gain, so that even if the CI value fluctuates due to a temperature change or the like, the gain can be varied. The amplification factor of the amplifier 10 can be operated within the recommended operating range. As a result, the value of the output current Iout for driving the crystal unit 2 is always constant without being influenced by the ambient temperature or the like. The oscillation amplitude of 2 can always maintain stable oscillation without fluctuation.

  In addition, by switching the gain of the fixed gain amplifier, it is possible to deal with a wide range of variations in the CI value of the crystal unit, so it is possible to use a crystal unit that has been discarded until now, and mass production of crystal units is possible. It is easy and can reduce the cost of crystal units by increasing production efficiency. Since the switch of the switching circuit 13 uses a MOS transistor, a one-chip IC can be realized by incorporating the switching circuit 13 of the fixed gain amplifier 11 in the IC of the oscillation circuit of the CMOS structure, and is small and excellent in reliability. An oscillation circuit can be provided. Furthermore, since a quartz vibrator having an extremely high Q value is used as the piezoelectric vibrator, a high-performance oscillation circuit having good temperature characteristics and excellent stability can be realized.

  Next, another embodiment of the fixed gain amplifier will be described with reference to FIG. The gain-fixed amplifier 11 shown in FIG. 6 has two MOS transistors as switches of the switching circuit 13, and is characterized in that the gain can be switched in three stages. It should be noted that the same elements as those of the fixed gain amplifier of FIG.

  In FIG. 6, the amplifier 12 has the same configuration as that shown in FIG. The switching circuit 13 has two MOS transistors SW1 and SW2 as switches. The drain terminal D1 and the source terminal S1 of the MOS transistor SW1 are connected to both ends of a dividing resistor R3 as a resistance element, and the source terminal S1 is connected to the GND of the circuit.

  The drain terminal D2 of the MOS transistor SW2 is connected to one end of the dividing resistor R2 and one end of the dividing resistor R1 of the amplifier 12, and the source terminal S2 is connected to the GND of the circuit. Here, a line connecting the drain terminal D2 and one end of the dividing resistor R1 functions as the switching signal V7. The other end of the dividing resistor R2 is connected to the drain terminal D1 of the MOS transistor SW1.

  As a result, the dividing resistors R1, R2, and R3 are connected in series between the negative input terminal of the differential amplifier circuit 12a of the amplifier 12 and GND. Note that the gate terminals G1 and G2 of the MOS transistors SW1 and SW2 are connected to switching control signals V8a and V8b having a 2-bit configuration, which are outputs of the memory circuit 14, respectively.

  Next, the operation of the fixed gain amplifier 11 according to another embodiment will be described with reference to FIG. Here, when the two switching control signals V8a and V8b are both logic "0", the two MOS transistors SW1 and SW2 are both turned OFF, and therefore the resistance between the negative input terminal of the differential amplifier circuit 12a and the GND. The value is R1 + R2 + R3, and the gain G of the amplifier 12 is Equation 3 shown in FIG. When the switching control signal V8a is logic "1" and the switching control signal V8b is logic "0", the MOS transistor SW1 is turned on and the MOS transistor SW2 is turned off, so that only the dividing resistor R3 is in a short state. Thus, the resistance value between the negative input terminal of the differential amplifier circuit 12a and GND is R1 + R2, and the gain G of the amplifier 12 is expressed by Equation 4 shown in FIG.

  When the switching control signal V8b is logic “1”, the MOS transistor SW2 is turned on, so that the dividing resistors R2 and R3 are both in a shoot state and between the negative input terminal of the differential amplifier circuit 12a and GND. The resistance value is the dividing resistor R1, and the gain G of the amplifier 12 is expressed by Equation 5 shown in FIG. When the switching control signal V8b is logic “1”, the switching control signal V8a may be either logic “0” or logic “1”.

  As an example, if the feedback resistor Rf is 50 KΩ, the dividing resistors R1 and R2 are 10 KΩ, and the dividing resistor R3 is 30 KΩ, the gain (Equation 3) when the MOS transistors SW1 and SW2 are both OFF is doubled. When the MOS transistor SW1 is ON and the MOS transistor SW2 is OFF, the gain (Equation 4) is 3.5 times, and when the MOS transistor SW2 is ON, the gain (Equation 5) is 6 times.

  With the above operation, the gain-fixed amplifier 11 shown in FIG. 6 can switch three gains according to the logic of the 2-bit switching control signals V8a and V8b, so that the variation in the CI value of the crystal resonator can be absorbed more greatly. Therefore, it is possible to further reduce the number of crystal units to be disposed of and to further improve the production efficiency of crystal units.

Next, another embodiment of the fixed gain amplifier will be described with reference to FIG. The gain-fixed amplifier 11 shown in FIG. 7 is characterized in that a switching circuit 13 as switching means switches gain by combining a capacitor and a MOS transistor. It should be noted that the same elements as those of the fixed gain amplifier of FIG. In FIG. 7, an amplifier 12 includes a differential amplifier circuit 12a having a positive input terminal and a negative input terminal, a feedback capacitor Cf connecting the negative input terminal and the output terminal of the differential amplifier circuit 12a, and a difference. The dividing capacitor C1 is formed by connecting the negative input terminal of the dynamic amplifier circuit 12a and the GND of the circuit.

  The switching circuit 13 includes a MOS transistor SW1 as a switch, and a dividing capacitor C2 that connects the drain terminal D1 of the MOS transistor SW1 and the negative input terminal of the differential amplifier circuit 12a. A line connecting the division capacitor C2 and the negative input terminal of the differential amplifier circuit 12a functions as the switching signal V7. The source terminal S1 of the MOS transistor SW1 is connected to the circuit GND, and the gate terminal G1 of the MOS transistor SW1 is connected to the switching control signal V8 from the memory circuit 14.

  Next, the operation of the fixed gain amplifier 11 according to another embodiment will be described with reference to FIG. When the switching control signal V8 is logic “0”, the MOS transistor SW1 is turned off, so that the capacitance value between the negative input terminal of the differential amplifier circuit 12a and GND is only the dividing capacitor C1, and the gain G of the amplifier 12 is Equation 6 shown in FIG. 7 is obtained. Further, when the switching control signal V8 is logic “1”, the MOS transistor SW1 is turned on, so that the capacitance value between the negative input terminal of the differential amplifier circuit 12a and GND becomes the division capacitor C1 + C2, and the gain G of the amplifier 12 Is represented by Equation 7 shown in FIG.

  As an example, if the feedback capacitor Cf is 100 pF, the dividing capacitor C1 is 100 pF, and the dividing capacitor C2 is 300 pF, the gain (Equation 6) when the MOS transistor SW1 is OFF is doubled, and the MOS transistor SW1 The gain (equation 7) when is ON is 5 times.

  With the above operation, the fixed gain amplifier 11 can switch between the two gains according to the logic of the switching control signal V8. Therefore, the variation in the CI value of the crystal resonator is absorbed similarly to the fixed gain amplifier 11 shown in FIG. A stable oscillation circuit can be realized. In addition, since the capacitor built in the IC has good relative accuracy and excellent temperature characteristics, a fixed gain amplifier with high gain accuracy and excellent temperature characteristics can be built in the oscillation circuit to form an IC. It can be easily realized.

  Next, another embodiment of the fixed gain amplifier will be described with reference to FIG. The gain-fixed amplifier 11 of FIG. 8 is characterized in that the switching circuit has two capacitors and two MOS transistors, and the gain can be switched in three stages. It should be noted that the same elements as those of the fixed gain amplifier of FIG. In FIG. 8, the amplifier 12 has the same configuration as that shown in FIG.

  The switching circuit 13 includes MOS transistors SW1 and SW2 as switches, a dividing capacitor C2 connecting the drain terminal D1 of the MOS transistor SW1 and the negative input terminal of the differential amplifier circuit 12a, and a drain terminal D2 of the MOS transistor SW2. And a dividing capacitor C3 connecting the negative input terminal of the differential amplifier circuit 12a. Two lines connecting the divided capacitors C2 and C3 and the negative input terminal of the differential amplifier circuit 12a function as the switching signal V7.

  The source terminals S1 and S2 of the MOS transistors SW1 and SW2 are connected to the GND of the circuit, and the gate terminals G1 and G2 of the MOS transistors SW1 and SW2 have a 2-bit configuration that is an output from the memory circuit 14. The switching control signals V8a and V8b are connected respectively.

  Next, the operation of the fixed gain amplifier 11 according to another embodiment will be described with reference to FIG. Here, when the two switching control signals V8a and V8b are both logic "0", the two MOS transistors SW1 and SW2 are both turned OFF, and therefore the capacitance between the negative input terminal of the differential amplifier circuit 12a and the GND. The value is only the dividing capacitor C1, and the gain G of the amplifier 12 is expressed by Equation 8 shown in FIG. When the switching control signal V8a is logic "1" and the switching control signal V8b is logic "0", the MOS transistor SW1 is turned on and the MOS transistor SW2 is turned off, so that the negative input of the differential amplifier circuit 12a. The capacitance value between the terminal and GND is the dividing capacitor C1 + C2, and the gain G of the amplifier 12 is given by Equation 9 shown in FIG.

  When the switching control signals V8a and V8b are both logic "1", the MOS transistors SW1 and SW2 are both turned on, so that the capacitance value between the negative input terminal of the differential amplifier circuit 12a and GND is the divided capacitor C1 + C2 + C3. Thus, the gain G of the amplifier 12 is expressed by Equation 10 shown in FIG.

  As an example, assuming that the feedback capacitor Cf is 100 pF, the dividing capacitor C1 is 100 pF, the dividing capacitor C2 is 150 pF, and the dividing capacitor C3 is 250 pF, the gain when both the MOS transistors SW1 and SW2 are OFF (Formula 8) Is doubled, the gain when the MOS transistor SW1 is ON and the MOS transistor SW2 is OFF (Equation 9) is 3.5 times, and the gain when both the MOS transistors SW1 and SW2 are ON (Equation 10) Becomes 6 times.

  With the above operation, the fixed gain amplifier 11 shown in FIG. 8 can switch three gains according to the logic of the 2-bit switching control signals V8a and V8b. Therefore, the fixed gain amplifier 11 by switching the resistance shown in FIG. Similarly, it is possible to absorb the variation in the CI value of the crystal resonator more greatly, further reducing the number of crystal resonators to be disposed of, and further improving the production efficiency of the crystal resonator.

  The MOS transistors SW1 and SW2 of the switching circuit 13 shown in FIGS. 4, 6, 7, and 8 may have any form as long as they are electrical switches, for example, bipolar transistors or mechanical switches. Or, a pattern cut of a printed board may be used. The number of switches and the number of dividing resistors and dividing capacitors may be arbitrarily increased or decreased depending on the specifications of the oscillation circuit. Further, the memory circuit 14 that outputs the switching control signal V8 is not limited to a non-volatile memory or the like, and any circuit that stores logic can be used.

  Next, an angular velocity sensor using the oscillation circuit of the present invention will be described with reference to FIG. Reference numeral 20 denotes an angular velocity sensor of the present invention, which includes an oscillation circuit 1 and a detection circuit 30 that detects Coriolis force from the detection electrodes 5 and 6 of the crystal resonator 2. Since the configuration and operation of the oscillation circuit 1 have been described above, they will be omitted, and the configuration and operation of the detection circuit 30 will be described below.

  In FIG. 9, reference numerals 31 and 32 denote two I / V conversion circuits, which are connected to the detection electrodes 5 and 6 of the crystal resonator 2, respectively, and input detection currents I1 and I2 to detect detection voltages V10 and V11. Is output. Reference numeral 33 denotes a differential amplifier circuit which receives the detection voltages V10 and V11 from the I / V conversion circuits 31 and 32 and outputs a differential output V12. Reference numeral 34 denotes a synchronous detection circuit, which receives the differential output V12, performs synchronous detection based on the detection control signal V9 from the oscillation circuit 1, and outputs a detection output V13. Reference numeral 35 denotes an LPF, which receives the detection output V13 and outputs an angular velocity detection signal V14.

Next, the angular velocity detection operation of the angular velocity sensor 20 will be described with reference to FIGS. Here, as described above, the crystal resonator 2 continues to oscillate at a constant amplitude by the oscillation circuit 1, and at this time, if the crystal resonator 2 is rotated at the angular velocity ω, the X direction shown in FIG. Coriolis force F proportional to the angular velocity ω is applied in the Z direction perpendicular to the vibration.

  This Coriolis force F is expressed by F = 2 · m · ω · V, m is an equivalent mass of the driving branches 2a, 2b, or the detecting branch 2c, and V vibrates at the resonance frequency f0 (Hz). Speed. The crystal resonator 2 is excited by the stress caused by the Coriolis force F at a frequency equal to the resonance frequency in the Z direction shown in FIG. 2, and this vibration causes the detection electrodes 5 and 6 formed on the detection branch 2c to be piezoelectric. Electric charges are generated due to the distortion effect.

  Due to the generated charges, detection currents I1 and I2, which are minute reverse-phase currents, flow through the detection electrodes 5 and 6, respectively. The I / V conversion circuits 31 and 32 of the detection circuit 30 convert the detection currents I1 and I2 into current voltages and output detection voltages V10 and V11, respectively, and the differential amplifier 33 calculates the difference between the detection voltages V10 and V11. Amplify and output differential output V12. The synchronous detection circuit 34 receives the differential output V12, efficiently performs synchronous detection in accordance with the timing of the detection control signal V9 output from the oscillation circuit 1, and outputs the detection output V13 converted into direct current. The LPF 35 cuts the AC component of the detection output V13 and outputs an angular velocity detection signal V14 that is a DC voltage corresponding to the angular velocity.

  Here, in order to increase the detection accuracy of the angular velocity detection signal V14, since the Coriolis force F described above depends on the vibration velocity V of the crystal resonator, it is extremely important to stabilize the vibration velocity V. For this reason, the oscillation circuit 1 of the present invention is switched according to the CI value of the crystal resonator using the gain of the fixed gain amplifier 11, so that even if the CI value fluctuates due to a temperature change or the like, the variable gain amplifier 10 Since the amplification factor can be operated within the recommended operation range, the output current Iout flowing through the crystal resonator 2 can be always kept constant.

  As a result, the vibration amplitude of the crystal resonator 2 is always constant, and a stable vibration speed V can be realized. As a result, the angular velocity sensor using the oscillation circuit of the present invention can detect a highly accurate angular velocity excellent in stability. The block diagrams and circuit diagrams shown in the embodiments of the present invention are not limited, and may be arbitrarily changed as long as the functions are satisfied.

It is a block diagram which shows the outline of the oscillation circuit of this invention. It is sectional drawing which shows the electrode structure of the crystal oscillator of the oscillation circuit of this invention. It is a graph explaining the gain characteristic of the gain variable amplifier of the oscillation circuit of this invention. It is a circuit diagram which shows an example using the resistor and switch of the gain fixed amplifier which comprise the oscillation circuit of this invention. It is a table | surface which shows an example of the switching operation range of the gain fixed amplifier of the oscillation circuit of this invention. It is a table | surface which shows an example of the operating range of the conventional gain fixed amplifier. It is a circuit diagram which shows the other example using resistance and the switch of the gain fixed amplifier which comprises the oscillation circuit of this invention. It is a circuit diagram which shows an example using the capacitor | condenser and switch of the gain fixed amplifier which comprise the oscillation circuit of this invention. It is a circuit diagram which shows the other example using the capacitor | condenser and switch of the gain fixed amplifier which comprise the oscillation circuit of this invention. It is a block diagram which shows the outline of the angular velocity sensor of this invention. It is a block diagram of the vibration control apparatus which detects the conventional angular velocity. It is a block diagram of the drive device which drives the vibrator included in the conventional vibration control device.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Oscillation circuit 2 Crystal oscillator 2a, 2b Drive branch 2c Detection branch 3, 4 Drive electrode 5, 6 Detection electrode 7, 31, 32 I / V conversion circuit 8, 35 LPF
9, 61 AGC circuit 9a Rectifier circuit 9b Reference signal source 9c Comparison circuit 10 Gain variable amplifier 11 Gain fixed amplifier 12 Amplifier 12a, 33 Differential amplifier circuit 13 Switching circuit 14 Storage circuit 15 Phase circuit 20 Angular velocity sensor 30 Detection circuit 34 Synchronous detection Circuit 50 Amplitude control device 51, 52 Feedback amplifier 53 Oscillator 54 Summing amplifier 55 Differential amplifier 60 Drive device 62 Comparator 63 Amplitude controller 64 Non-inverting amplifier 65 Inverting amplifier Rf Feedback resistor R1, R2, R3 Dividing resistor Cf Feedback Capacitor C1, C2, C3 Split capacitor SW1, SW2 MOS transistor Iout Output current I1, I2 Detection current V1 AC signal V2 Filter output signal V3 DC signal V4 Reference signal V5 Control voltage V6 Amplification signal V7 Switching signal V8 Switching control signal V9 Detection Control signal V10, V11 Detection voltage V12 Differential output V13 Detection output V14 Angular velocity detection signal Vout Drive voltage

Claims (6)

A piezoelectric vibrator having a drive electrode and excited by a drive voltage applied to the drive electrode;
A current / voltage conversion circuit for converting an output current from the drive electrode of the piezoelectric vibrator into an AC voltage;
A variable gain amplifier capable of amplifying an AC signal from the current / voltage conversion circuit to vary the gain;
A gain-fixed amplifier that amplifies the AC signal from the current / voltage conversion circuit and has a fixed amplification factor;
A rectifier circuit that rectifies an output current from the drive electrode, a comparison circuit that compares a signal output from the rectifier circuit with a predetermined reference signal, and outputs a control voltage in accordance with a change in the output signal of the rectifier circuit; And controlling the drive voltage applied to the drive electrode of the piezoelectric vibrator based on the control voltage output from the comparison circuit, and controlling the output current from the piezoelectric vibrator to be constant An automatic gain control circuit;
An oscillation circuit having
An oscillation circuit characterized in that the gain-fixed amplifier has at least two gains, and provided with switching means for switching the at least two gains according to a predetermined situation. The oscillation circuit according to claim 1, wherein the switching unit includes a resistance element and a switch. The oscillation circuit according to claim 1, wherein the switching unit includes a capacitor and a switch. 4. The oscillation circuit according to claim 2, wherein the switch is a MOS transistor. The oscillation circuit according to claim 1, wherein the piezoelectric vibrator is a crystal vibrator. The oscillation circuit according to any one of claims 1 to 5;
A stress caused by Coriolis force generated in the piezoelectric vibrator is detected as a charge by a detection electrode provided on the piezoelectric vibrator, and an AC signal from the current / voltage conversion circuit of the oscillation circuit or a drive electrode of the piezoelectric vibrator is detected. A detection circuit that performs synchronous detection with an applied drive voltage and outputs a detection signal corresponding to the angular velocity;
An angular velocity sensor comprising:

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