JP2006516369A - Spread spectrum transmission and reception using continuous waveforms in harmonic relations - Google Patents

Abstract

Systems and methods for transmitting and receiving spread spectrum signals using continuous waveforms are described. A method of transmitting a spread spectrum signal is to generate a plurality of generally continuous waveforms, integrate the plurality of generally continuous waveforms, modulate the plurality of generally integrated continuous waveforms to generate a spread spectrum signal, and then spread the spectrum. Including transmitting a signal. A method of receiving a spread spectrum signal is to generate a plurality of generally continuous waveforms, integrate the plurality of generally continuous waveforms, modulate the plurality of generally integrated continuous waveforms using a code, and perform the modulated integration. The method includes mixing a plurality of generally continuous waveforms using a spread spectrum signal to generate a baseband signal, integrating the baseband signal, and detecting data.

Description

(Technical field)
This patent application generally relates to communication devices, and more particularly to ultra-wideband (UWB) high speed data communication (HDR) communication devices.

(background)
With the widespread use of wireless communication devices in the unlicensed spectrum and the increasing demand for higher data bandwidths, these frequency spectrum bands are severely distorted. Examples of unlicensed bands include the 915 MHz, 2.4 GHz industrial, scientific and medical (ISM) bands, and the 5 GHz unlicensed National Information Infrastructure (UNII) band. New equipment and new standards constantly appear. For example, IEEE802.11b, IEEE802.11a, IEEE802.15.3, HiperLAN / 2 standards. Creating and accepting standards will continually place additional burdens on these frequency bands. Coexistence with many systems competing with the wireless frequency (RF) spectrum will become increasingly important as consumer products become more popular.

  Those skilled in the art have found that the bandwidth available in the unlicensed band (and the generally available RF spectrum) is putting pressure on the available data bandwidth of the wireless system. Also, the data rate throughput capability varies in proportion to the available bandwidth, but is only logarithmically proportional to the available signal to noise ratio. Therefore, transmission at increasingly higher data rates over constrained bandwidth requires the use of complex communication systems using advanced signal modulation methods.

  Complex communication systems typically require very high signal-to-noise ratios, making faster data rate systems even more vulnerable to interference from other users of the spectrum and from multipath interference. It becomes easy to be affected. The high sensitivity to interference further exacerbates the above-mentioned coexistence problem. In addition, legal restrictions in any RF band impose a limit on the data rate throughput of the communication system, as it squeezes the maximum available signal to noise ratio. Therefore, there is a need for a high-speed data communication apparatus and system that can coexist without difficulty with other existing wireless communication systems and can firmly support relatively high-speed data communication in a multipath environment.

(Overview)
One aspect of the present invention relates to a communication device such as a communication transmission device or a communication reception device. In one exemplary embodiment, an RF transmitter according to the present invention includes a reference signal generator, a signal generator, and a mixer.

  The reference signal generator generates a reference signal having a predetermined or desired frequency. The signal generator generates an operation signal according to the selection signal. The operating signal has a frequency equal to the frequency of the reference signal multiplied by a number. If desired, in some embodiments, this number may consist of an integer, and in other embodiments, this number may consist of a non-integer number. The mixer mixes the operating signal with another signal to generate a transmission signal.

  In another exemplary embodiment, an input receiver according to the present invention includes two mixers, a first mixer and a second mixer. The receiver further includes an integrator / sampler and a signal generator.

  The first mixer receives the input RF signal and the second input signal as its inputs. The first mixer mixes the input signals and generates a mixed signal. The integrator / sampler receives the mixed signal and processes it to produce an output signal. The signal generator generates an operation signal according to the selection signal. The operating signal has a frequency equal to the reference signal frequency multiplied by a number. If desired, some embodiments may comprise this number from an integer, and other embodiments may comprise this number from a non-integer. The second mixer mixes the operating signal with the template signal to generate a second input signal for the first mixer.

(Detailed explanation)
The present invention examines high-speed data communication devices and corresponding methods. The communication apparatus according to the present invention solves the problems of coexisting communication systems and provides relatively fast data communication. It should be noted that a wireless or wireless communication system according to the present invention provides relatively fast data communication in a “poor” propagation environment such as a multipath environment. Moreover, as described below, the novel idea described here can be applied to a land line communication system. For example, a communication system using a coaxial cable or the like.

  In one exemplary embodiment according to the present invention, a high speed data communication UWB data transmission system employs carrier frequency two phase displacement modulation (BPSK) modulation well known to those skilled in the art who understand the advantages of the present description. A high power spectral density (PSD) is obtained at the frequency f of such a system.

ここで、ｆ_ｃは基準クロック周波数を表し、ｎはキャリアサイクル１チップ当たりの数を表す。すなわち、

Here, f _c represents a reference clock frequency, n represents represents the number per carrier cycle one chip. That is,

図１１Ａまたは図１１Ｂに示されているように、チップは信号要素をいう。別の言い方では、チップは、送信信号を生成するために用いる要素シーケンスにおける１つの要素をいう。送信信号は、チップのシーケンス（チップシーケンス）を、拡散コード、すなわち、相対的に広帯域に拡散する送信信号に広がるコードで乗算することにより得られる。１ビット当たりの所望のエネルギーレベルに比例する複数のチップが、各データビットを符号化する。

As shown in FIG. 11A or 11B, a chip refers to a signal element. In other words, a chip refers to one element in an element sequence used to generate a transmission signal. The transmission signal is obtained by multiplying a sequence of chips (chip sequence) with a spreading code, ie a code that spreads over a transmission signal that is spread over a relatively wide band. Multiple chips that are proportional to the desired energy level per bit encode each data bit.

  In the present embodiment, the modulation chipping rate is balanced with the carrier frequency. In other words, n is a relatively small number. In another embodiment, for example, n is a value less than 10, such as 3 or 4. In other illustrative embodiments, n can be used in the range of 1 to 500, or 1 to 42. Using the value of n in the latter range, it is also possible to achieve a UWB bandwidth of 500 MHz or higher, and a frequency limit of about 10.6 GHz as defined by Federal Communications Commission (FCC) Part 15 regulations.

  Understanding the benefits of the description of the invention As will be appreciated by those skilled in the art, other positive integer values may be used for n if desired. In general, the choice of the value of n depends on the respective ultra-wideband definition. Depending on the desired bandwidth, an appropriate value of n can be selected if desired.

  The value of n (rounded up to an integer value) roughly corresponds to the desired center operating frequency divided by half the desired bandwidth. That is,

または

Or

ここで、ｆ_０および△ｆはそれぞれ、中心動作周波数と所望の帯域幅とを表す。例えば、ＵＷＢに関するＦＣＣの定義の上記の例は、ｎの値は１〜４２の範囲になる。すなわち、現行のＦＣＣパート１５の制限周波数である１０．６ＧＨｚを下回って動作する５００ＭＨｚ幅ＵＷＢシステム（帯域幅の半分）では、次のようになる。

Here, f ₀ and Δf represent the center operating frequency and the desired bandwidth, respectively. For example, in the above example of the FCC definition for UWB, the value of n is in the range of 1-42. That is, in a 500 MHz wide UWB system (half the bandwidth) operating below 10.6 GHz, which is the limit frequency of the current FCC part 15, the following occurs.

または

Or

ＦＣＣはまた、２２〜２９ＧＨｚの周波数範囲で少なくとも５００ＭＨｚの帯域幅のＵＷＢ信号を許可している。これは、上の値がｎ＝１１６，０００に対応する。従って、本発明の説明による利点を理解している当業者は、任意の応用に対する性能および設計仕様や要件により、事実上任意の範囲でｎの値を選択することができる。おおむね信号帯域幅は、ｎの値の逆に変化することに留意されたい。

The FCC also allows UWB signals with a bandwidth of at least 500 MHz in the frequency range of 22-29 GHz. This corresponds to an upper value of n = 16,000. Accordingly, those skilled in the art who understand the benefits of the description of the present invention can select the value of n in virtually any range, depending on the performance and design specifications and requirements for any application. Note that generally the signal bandwidth varies inversely with the value of n.

  FIG. 1 shows several PSD profiles for various values of n (number of carrier cycles per chip). The PSD profile 11 corresponds to n = 1, and the PSD profile 12 and the PSD profile 13 correspond to n = 2 and n = 3, respectively. Note that as the value of n increases, the bandwidth of the modulated signal decreases. In UWB systems that want to suppress a specified maximum PSD (eg, a specified PSD characteristic by a supervisory authority), try to obtain a spectrum that is as flat as possible to minimize the total transmitted power in a given bandwidth. Note further.

  Similarly, such systems attempt to select a transmission bandwidth regardless of the modulation rate in order to maximize the total transmission power. As will be appreciated by those skilled in the art, in a conventional BPSK system, even with the highest bandwidth, the PSD profile is not flat and n = 1. As is clear from the parameter n, the bandwidth depends on the chip rate. The dependence of the bandwidth on the parameter n is undesirable for a variety of reasons, such as it being difficult or impossible to meet predetermined regulations or design specifications.

  For illustration purposes, FIG. 2 shows various signals corresponding to the BPSK transmission system. The carrier signal 21 includes only the fundamental frequency. Alternatively, rather than a continuous sine wave signal, the carrier signal 21 may include other waveforms as described below. FIG. 2 also shows a pseudo-random noise (PN) sequence 22. Note that the waveforms shown in FIG. 2 correspond to a communication system with one chip per RF cycle (ie, n = 1) and four chips per data bit.

  The third waveform shown in FIG. 2 corresponds to data bit 23. The PN sequence 22 encodes the data bits 23 to begin at time 27 and end at time 28. The signal 24 is generated by encoding the data bits 23. The signal 24 modulates the carrier 21 to generate a modulated signal 25. Signal 26 operates as a gate signal. In other words, the communication system transmits the modulated signal 25 while the gate signal 26 is active (during the active portion of the signal 26). The modulated signal 25 has substantially the same spectrum as the spectrum 11 shown in FIG. 1 (that is, when the parameter n has one value).

  The data rate or data throughput of the communication system can be determined from various system parameters. For example, assume that the carrier signal has a frequency of 4 GHz and the system operates with one chip per RF cycle (ie, n = 1) and 4 chips per data bit. Given these optional parameters, those skilled in the art who understand the benefits of the description of the invention will readily understand that this system results in a data rate of 1 gigabit per second (Gb / s). it can.

  One exemplary embodiment of a high speed data communication UWB system according to the present invention includes a high speed data communication UWB transmitter and a high speed data communication UWB receiver. FIG. 3 shows an exemplary embodiment of a high speed data communication UWB transmitter 4 according to the present invention.

  The transmitter 4 includes a reference clock 41 (reference clock generator), a timing controller 42, a data buffer 43, a PN generator 45 (pseudo random noise sequence generator), a data / PN combiner 46, a mixer 47, and an antenna 48. And a harmonic generator 49. The reference clock 41 generates a signal having a desired frequency. The frequency of the reference clock 41 corresponds to the carrier frequency of the transmitter 4. Accordingly, the frequency of the reference clock 41 corresponds to a desired carrier frequency. The reference clock 41 can be implemented in many ways and various techniques that are within the knowledge of those of ordinary skill in the art who understand the benefits of the present description.

The reference clock 41 is connected to the harmonic generator 49. Based on the clock signal, the harmonic generator 49 receives from the reference clock 41 and generates one or more harmonics of the carrier frequency (frequency of the clock reference 41). For example, assuming an arbitrary clock frequency f _c, as those skilled in the art will appreciate the benefit of the description of the invention, the second harmonic signal at the output of the harmonic generator 49 has a frequency 2 · f _c such . The harmonic generator 49 is tuned to the reference clock to generate one or more harmonics (ie, one or more harmonics are synchronized with the reference clock).

  It should be noted that the harmonic generator 49 can be implemented in many ways, as will be appreciated by those skilled in the art who understand the advantages of the present description. For example, a comb generator. As another example, a phase-locked loop can be used if desired. As another example, an oscillator followed by a digital divider circuit can be used. One or more harmonics can be obtained by dividing a signal of an arbitrary frequency by various integers. For this implementation, a fractional N synthesizer can be used if desired.

  Various circuits and techniques can also be used to synchronize one or more harmonics with the reference clock. Such circuits and techniques are within the knowledge of one of ordinary skill in the art who understands the advantages of the description of the invention. As an example, the comb generator can synchronize one or more harmonics with a reference clock.

  The mixer 47 receives one or more harmonics from the harmonic generator 49. The mixer 47 mixes one or more harmonics of the carrier frequency with the signal received from the data / PN combiner 46 (further described below). The mixer 47 supplies the obtained signal to the antenna 48. The antenna 48 propagates these signals to the transmission medium. In the illustrative embodiment, antenna 48 constitutes a broadband antenna.

  Examples of wideband antennas include the following patent documents. U.S. Patent No. 6,091,374, filed Sep. 27, 2000 U.S. Patent Application Serial No. 09/670, 792, filed Jan. 2, 2001 U.S. Patent Application Serial No. 09 / 753,244, Jan. 2001 US patent application serial number 09/753, 243, filed February 15, 2002, US patent application serial number 09/077, 340, US patent application serial number 09 / 419,806. All are assigned to the assignee of the present application. If desired, a broadband horn antenna and a ridge horn antenna can also be used. As yet another example, a differential drive wire segment can be used as a simple and efficient broadband radiator. Other suitable broadband antennas may also be used, as is well known to those skilled in the art who understand the advantages of the description of the invention.

  Note that some antennas are of the type “having constant gain in frequency”, resulting in a system with frequency dependent propagation characteristics. Other antennas, such as horn antennas, have a “constant aperture” variety and produce frequency-dependent propagation behavior. To use harmonics at relatively high frequencies, the exemplary embodiment according to the present invention uses an antenna “having a constant aperture in frequency”, but is well known to those skilled in the art who understand the advantages of the present description. As described above, other types of antennas can be used.

  The reference clock 41 is also connected to the timing controller 42. The timing controller 42 clocks data in the data buffer 43. Note that the timing signal from the timing controller 42 also clocks the clock PN generator 45. The data buffer 43 receives the input data from the data port 44. The PN sequence from the PN generator 45 modulates the data from the data buffer 43 using the data / PN combiner 46 in a manner well known to those skilled in the art who understand the benefits of the present description. The PN encoded data from the data / PN combiner 46 modulates one or more harmonics in the mixer 47. In the illustrative embodiment of the present invention, the data / PN combiner 46 constitutes an exclusive-or (XOR) gate, but others known to those skilled in the art who understand the benefits of the description of the present invention are well known. A suitable circuit can also be used.

  In the illustrative embodiment, a filter may be used at the output of the harmonic generator 49 to adjust the amplitude of one or more harmonics to be approximately the same value. However, it should be noted that in other embodiments according to the invention, unequal amplitudes may be used if desired. By using unequal amplitudes, it is also possible to control the amount of energy of the transmitted signal at a particular frequency or frequency band.

The unequal amplitude affects the amount of energy in the various parts of the corresponding PSD profile. For example, reduced (or removed) amplitude will reduce energy in the corresponding frequency band (FIG. 16 shows an example of such a system. Coexistence with a wireless system operating in that band. (In order to improve, the radiant energy of the band 267 is further reduced.)
FIG. 4 shows exemplary waveforms corresponding to the high speed data communication UWB transmitter 4. The signal 421 corresponds to the output of the harmonic generator 49. Signal 422 corresponds to a relatively short PN sequence of 4 chips per data bit. Signal 423 corresponds to a relatively short data sequence. A signal 429 indicates more detailed timing of the transmitter 4 and constitutes an output signal of the reference clock 41.

  Those skilled in the art who understand the advantages of the description of the invention will understand that a chip sequence longer than 4 chips per bit is desirable depending on the application. For example, such a chip sequence can be used when the transmission medium constitutes an RF channel having a large multipath or when it is desired to increase the energy per data bit (at the expense of the data throughput rate). .

  In general, if desired, one chip per bit can be made as small as possible to obtain the maximum data rate. Also, tens of thousands of chips can be used per bit to obtain “integration” gain at the expense of data rate. Thus, as will be appreciated by those skilled in the art, the range of chips per bit can vary greatly depending on the design and performance specifications for a particular application, if desired. For example, in the illustrative embodiment of the present invention, approximately 1 to 200 chips per bit can be used if desired. As another example, an IEEE 802.15 compliant embodiment typically has a high data rate of 480 Mb / s corresponding to several chips per bit and 11 Mb / s corresponding to about several hundred chips per bit. s and a slow data rate are desirable.

  Those skilled in the art who understand the advantages of the description of the present invention will recognize that the number of PN chips per data bit is a multi-gate effective coding gain measure against interference and multipath failure. Thus, a mechanism is provided that reduces the effects of interference and multipath by using multiple chips per data bit.

  As described above, the data / PN combiner 46 can be implemented using an exclusive OR gate. Signal 424 indicates the result of the exclusive OR operation on signals 422 and 423. The modulated RF signal 425 is obtained by combining the signal 421 with the signal 424 by the mixer 47. Timing signal 426 indicates the transmission time of the sequence of data bits 423.

  FIG. 5 shows an exemplary embodiment of a high speed data communication UWB receiver 5 according to the present invention. The receiver 5 includes a reference clock 53, a tracking loop 52, an integrator / sampler 51, a PN generator 55, a data / PN combiner 56, a mixer 57, an antenna 58, and a harmonic generator 59. Blocks and components that are similarly labeled in the receiver 5 have the same structure and operation as the corresponding blocks and components of the transmitter 4 shown in FIG.

  Referring to FIG. 5, in the high-speed data communication UWB receiver 5, the receiving antenna 58 is connected to the transmission medium by the received modulated signal 425 (as shown in FIG. 3, the signal has the waveform shown in FIG. 4 as an example). ) To the mixer 57. The mixer 57 supplies the output signal to the integrator / sampler 51. The integrator / sampler 51 integrates the output signal of the mixer 57 and transmits the restored data bit signal 563 as the data output 54.

  Mixer 57 also receives template signal 567. The data / PN combiner 56 generates a template signal 567 from the outputs of the PN generator 55 and the harmonic generator 59. In the illustrative embodiment of the present invention, the data / PN combiner 56 constitutes an exclusive OR (XOR) gate, as is well known to those skilled in the art who understand the advantages of the present invention. A suitable circuit can also be used. The harmonic generator 59 operates in a manner similar to the harmonic generator 49 shown in FIG. 3 and may have the same structure or circuit.

  A tracking loop 52 known in the art controls a reference clock 53 and a PN generator 55. As known to those skilled in the art who appreciate the benefits of the present description, the tracking loop 52 controls the timing of the PN generator 55 to ensure proper signal acquisition and tracking. The reference clock 53 supplies a reference clock signal 569 to the PN generator 55 and the harmonic generator 59.

  Note that tracking loop 52 can be implemented in various ways, if desired. As is well known to those skilled in the art, the choice of implementation method depends on a number of factors such as design and performance specifications and characteristics. As is well known to those skilled in the art to understand the advantages of the present description, the tracking loop 52 operates in conjunction with the template signal 567 to form a locking mechanism (template receiver or suitable template receiver) that receives the transmitted signal.

  Mixer 57 mixes the signal received from antenna 58 with template signal 567 to generate signal 568. The integrator / sampler 51 integrates the signal 568 to generate a restored data signal 563. The integrator / sampler 51 controls signal acquisition and tracking in the high-speed data communication UWB receiver 5 that drives the tracking loop 52.

  FIG. 6 shows exemplary waveforms corresponding to the high speed data communication UWB receiver 5. Signal 562 constitutes the output of PN generator 55. Signal 561 corresponds to the output of harmonic generator 59 and signal 567 corresponds to the output signal of the data / PN combiner. The signal 568 constitutes an output signal of the mixer 57 and is supplied to the integrator / sampler 51. A signal 563 is an output signal of the integrator / sampler 51. Finally, in order to show the timing of the receiver 5 in more detail, the signal 569 constitutes the output signal of the reference clock 53.

  FIG. 7 further shows details of the timing relationship of various signals in the high-speed data communication UWB transmitter 4. Waveform 75 corresponds to a signal in the transmission medium (ie, propagated from antenna 48). A waveform 76 indicates a transmission period, that is, a time during which the transmitter 4 transmits. Finally, waveform 73 shows data bit stream 73 during transmission period 76. Waveform 79 shows a timing reference clock check mark for the other waveforms shown in FIG.

  In another embodiment according to the present invention, the high-speed data communication UWB transmitter 4 can be operated in one of two modes according to selected or predetermined parameters. In each mode, a specific or predetermined PSD profile can be generated by using a specific or predetermined harmonic order (ie, the carrier harmonic is selected and used for each mode). By selecting a specific mode, the transmitter 4 is operated to generate an output signal that conforms to a specific PSD profile or that meets a predetermined condition (as described above, eg, by a supervisory body such as an FCC). be able to.

  FIG. 8 shows two exemplary desired or predetermined PSD profiles corresponding to the two modes of operation in such an embodiment. The transmitter according to the present invention can produce an output according to a selected predetermined PSD amplitude profile mask 80 and a predetermined PSD amplitude profile mask 81. In such a transmitter embodiment, the frequency of the reference clock (ie, the frequency of the reference clock 41 shown in FIG. 3) is about 1.8 GHz. Accordingly, the second and third harmonics appear at approximately 3.6 GHz and 5.4 GHz, respectively.

  In the first mode of operation according to PSD amplitude profile mask 80, a 3.6 GHz carrier (second harmonic of the reference clock frequency) is modulated with one chip per two RF carrier cycles. Also, the 5.4 GHz carrier (third harmonic of the reference clock frequency) is modulated with one chip per three RF cycles. In this mode of operation, the transmitter has a chipping rate of 1.8 gigachips per second. This transmitter generates a transmission PSD profile 83. Note that the transmit PSD profile 83 is generally flat and matches the PSD mask 80 (ie, what remains under the PSD mask 80).

  In the second mode of operation, while modulating the third harmonic of the 1.80 GHz clock (ie, the harmonic appearing at 5.6 GHz) at the rate of one chip per four RF cycles, the second harmonic Suppress the wave. As a result, the transmitter will have a chipping rate of 1.35 gigachips per second.

  It should be noted that embodiments according to the invention including more than two modes of operation can be implemented if desired. For example, a UWB device including m operation modes can be configured. m is an integer greater than 1. Such systems can be implemented in a variety of ways, as is well known to those skilled in the art who appreciate the benefits of the present description. For example, any combination of one or more harmonics can be selected using a selectable harmonic filter bank (ie, the order of the harmonics used can be selected). Such UWB wireless devices can selectively avoid interference from other wireless system operations in the same band or multiple bands or with other wireless system operations. It should be noted that in the illustrative embodiment according to the present invention, in addition to polarity modulation (ie, BPSK modulation), “one or more m harmonics” can be considered as a certain modulation format.

  Although the above description refers to second and third harmonics, those skilled in the art who understand the advantages of the description of the invention will also recognize that other harmonics may be used if desired. In other words, in each operation mode, harmonics exceeding the third harmonic can also be used. Further use of harmonics increases the total transmit power while adapting to each respective mask at the same time (ie, remaining under the PSD mask).

  FIG. 9 shows the PSD profile of an exemplary embodiment of the invention using higher order harmonics. The transmitted PSD profile 91 corresponds to the modulated third and fourth harmonics of the 1.1-GHz reference clock. PSD profile 91 assumes modulation at a rate of 1.1 gigachips per second.

  If more transmission power is desired, third to seventh harmonics can be used. As a result, the transmission PSD profile 93 is obtained. Note that both the PSD profile 92 and the PSD profile 93 are generally flat in shape. Note further that both PSD profile 92 and PSD profile 93 match a predetermined or desired PSD amplitude profile mask 90. Thus, if desired, the communication system can be implemented with a specific output power profile that matches a given PSD profile by using many harmonics of a reference clock frequency with the appropriate order.

  Note that the appropriate clock reference frequency and associated harmonics can be used to coexist with other devices that use a particular RF band or spectrum. For example, in another embodiment according to the present invention, clock reference parameters and harmonic carriers are combined so that PSD high-speed data UWB transmission coexists with wireless devices operating in the 2.4 GHz ISM band and the 5 GHz UNII band. select.

  That is, in such an embodiment, the reference clock has a frequency of about 1.1 GHz. Also, the transmitter uses both the third and fourth harmonics of the reference clock frequency (ie, 3.3 GHz and 4.4 GHz, respectively) as the carrier frequency modulated at a reference clock speed of approximately 1.1 GHz.

  FIG. 10 shows an exemplary PSD profile for such an embodiment of the present invention. The transmit PSD profile 101 fits between the 2.4 GHz ISM band 102 and the 5 GHz UNII band 103 to meet the desired level of coexistence. Note that the communication system can also support relatively high speed data communication. For example, if one PN chip is used to configure one data bit, the resulting data rate is 110 megabits per second (Mb / s).

The signal harmonics can also add the desired design freedom that can be selected for the relative phase of the carrier. For example, in a communication system according to the present invention using third and fourth harmonics, the time signal x (t), that is, the sum of carrier harmonics is generally expressed as follows.
x (t) = sin (2π · 3 · _fr t) + sin (2π · 4 · _fr t + φ)
Here, fr represents the reference clock frequency, and φ represents a selectable or predetermined phase angle between 0 and 2π radians. It should be noted that in an exemplary embodiment according to the present invention, the phase angle can also be implemented using a filter, as is well known to those skilled in the art who understand the advantages of the present description.

  It should be noted that various values of φ can be used, if desired, in an exemplary embodiment according to the present invention, with 0 ≦ φ ≦ 2π. FIG. 11A shows an exemplary output signal 121A for one cycle of a transmitter in a UWB communication system according to the present invention. The signal 121A corresponds to φ = π. The start point 122 and the end point 123 coincide with the chip boundary. For example, a signal 421, a chip signal 422, and an output signal of the PN generator shown in FIG.

If desired, the output signal x (t) can also be expressed using a cosine. That is,
x _i (t) = cos (2π · 3 · _fr t) + cos (2π · 4 · _fr t + φ)
Here, f _r represents the reference clock frequency, phi represents a predetermined phase angle between the selectable or 0~2π radians (including the endpoints). FIG. 11B shows another exemplary output signal 121B for one cycle of a transmitter in a UWB communication system according to the present invention. The output signal 121B has a start point 122B and an end point 123B.

Those skilled in the art who understand the advantages of the description of the present invention will understand that the signals x (t) and x _i (t) constitute an orthogonal signal. Accordingly, as described below, quadrature phase shift modulation (QPSK) modulation can be performed using signals x (t) and x _i (t).

  Signals 121A and 121B have relatively small signal levels at their respective start points (ie, 122A and 122B) and at their respective end points (ie, 123A and 123B). In the exemplary embodiment according to the invention, these relatively small signal levels switch the signal on and off. It should be noted that this avoids switching transitions due to incomplete switching that would change the resulting spectrum to an unwanted one.

In the illustrative embodiment according to the invention, it is also possible to represent a harmonic carrier with a composite signal S which constitutes the integration of a sine wave signal and / or a cosine signal.
S (t) = Σsin {2π · n · _fr · (ts)}
Here, the integration extends over the range of the desired harmonic n (ie, from the lowest order to the highest order of the desired harmonic). In other words, the composite signal S constitutes the sum of the harmonic carriers over the selected range n. It should also be noted that an orthogonal UWB communication device can be implemented with cosine harmonics.

  As described above, in some embodiments, n can be in the range of 3-4 (corresponding to a UWB communication device operating in the desired frequency range of 3.1 GHz to 5.2 GHz). FIG. 12 shows the timing relationship between several signals of n = 3 in an embodiment according to the present invention. Signal 139 includes a reference clock signal so that timing relationships between the various signals can be easily represented. The signal 131 corresponds to the composite signal S described above. The signal 132 represents a sine wave signal and is a harmonic that becomes the composite signal 131. Reference clock signal 139 corresponds to the zero crossing in the sexual direction of sinusoidal signal 132.

  Note that the time displacement s cancels the chipping signal from the carrier signal. That is, the time displacement s appears as an offset between the reference clock signal 139 (or sine wave signal 132) and the chipping signal.

  FIG. 12 shows signals corresponding to several values of the time displacement s. Each time displacement s represents an offset between the reference clock signal 139 (or sinusoidal signal 132) and one of each of the chipping signal 133, the chipping signal 134, and the chipping signal 135. In particular, the chipping signal 133 corresponds to a zero time displacement s. Chipping signal 134 and chipping signal 135 represent time displacements s of 0.25 and 0.5, respectively.

  Those skilled in the art who understand the advantages of the description of the present invention will appreciate that, due to symmetry, a negative value of s results in the same result as a positive value of s. Therefore, in the description of the present invention, the size s, that is, | s | is described. Further, FIG. 12 shows a chipping sequence “101” as an example for explanation. However, as is well known to those skilled in the art who understand the advantages of the description of the present invention, a desired PN sequence can be used. Please keep in mind.

  FIG. 13 shows several PSD profiles of an illustrative embodiment according to the present invention. PSD profile 143 shows the power spectral density of signal 131 multiplied by PN chipping sequence 133. Similarly, PSD profile 144 corresponds to the power spectral density of signal 131 multiplied by PN chipping sequence 134. Finally, PSD profile 145 shows the power spectral density of signal 131 multiplied by PN chipping sequence 135.

  FIG. 13 also shows a boundary 146 for the 2.4 GHz ISM band and a boundary 147 for the UNII band. For the two harmonics, the time displacement value | s | = 0.25 produces a generally flat PSD profile 144. A person skilled in the art who understands the advantages of the description of the invention, however, uses third order and fourth order, if desired, using time displacement values (s) in the range of about 0.1 to about 0.9. It will be appreciated that approximately the same PSD profile of harmonics can be generated. In a similar manner, if desired, a communication system having a desired or predetermined PSD profile can be implemented using other time displacement s values and appropriate harmonics.

  As an example, FIG. 14 shows some of the PSD profiles corresponding to the exemplary embodiment of the present invention using more and more harmonics. FIG. 14 includes a PSD profile 151, a PSD profile 152, and a PSD profile 153. A generally flat PSD profile 151 corresponds to a signal from the fundamental frequency to the seventh harmonic using the time displacement value | s | = 0.375. Similarly, PSD profile 152 relates to signals containing second to seventh harmonics using time displacement values | s | 0.375. Finally, PSD profile 153 corresponds to a signal containing third to seventh harmonics with a time displacement value | s | = 0.375.

  Note that values of time displacement s between about 0.1 and about 0.9 produce a generally flat PSD profile similar to the PSD profile shown in FIG. As described above, it matches the PSD profile (ie, constrained by the maximum PSD value), but using multiple harmonics will increase the total transmitted or radiated power.

  The time displacement s can be generated and implemented in various ways, as is well known to those skilled in the art who understand the advantages of the description of the invention. For example, s may be implemented as a digitally derived clock shift in the timing controller 42 of the transmitter 4 and the PN generator 55 of the receiver 4. As another example, the desired time shift can be implemented by using a physical delay line in the mixer 47 digital transmitter path of the transmitter 4 and the receiver 5 mixer 57.

  By calculating the Fourier transform of the composite signal S, the illustrated spectrum can be obtained. That is, if the data pulse is generally rectangular and not filtered (eg, the chipping signal 422 shown in FIG. 4), the PSD can be obtained as follows.

ここで、ｆ_ｒはチッピングクロック周波数を表し、ｎ_１およびｎ_２は用いる高調波の次数（すなわち、用いる高調波の範囲の上下のバウンダリ）に対応する。所望ならば、ｎ_１〜ｎ_２の範囲内の選択した高調波を除外してさらにスペクトラムを整形することもできることに留意されたい。図１５は、この技術を適用した例を示す。

Here, f _r represents the chipping clock frequency, n ₁ and n ₂ harmonic order of used (i.e., the upper and lower boundary of the range of harmonics used) corresponding to. Note that if desired, the spectrum can be further shaped to exclude selected harmonics in the range of n ₁ -n ₂ . FIG. 15 shows an example in which this technique is applied.

  Referring to FIG. 15, PSD profile 161 shows the power spectral density of an embodiment of a communication system according to the present invention using third to seventh harmonics that are 1.1 GHz clocks. In contrast, PSD profile 162 corresponds to a system using fifth to seventh harmonics. As a result, the PSD energy in the latter system is approximately over 5 GHz.

  As a third example, the PSD profile 163 corresponds to a system using third-order, fourth-order, sixth-order, and seventh-order harmonics. By excluding the fifth harmonic from this system, a gap is formed in the vicinity of 5 GHz to 6 GHz. As a result, this system can effectively coexist with a system operating in the 5 GHz UNII band. Note that filtering can easily remove the sidelobe energy shown in FIG.

  The PSD profile shown in FIG. 15 corresponds to an embodiment for explaining a communication system according to the present invention. By successfully using the selected harmonics along with the selected clock frequency, a wide variety of communication systems having a predetermined PSD profile can be designed and implemented in a flexible manner. The choice of design parameters (eg, clock frequency, component, harmonic order) is based on the desired design and performance specifications and is within the knowledge of those skilled in the art who understand the advantages of the description of the invention.

  FIG. 16 shows a PSD profile of another exemplary embodiment of a communication system or apparatus according to the present invention. These embodiments are consistent with PSD masks where 3.1 GHz radiation is at least -10 dB from the peak (labeled 265 as shown in FIG. 16). This mask also specifies that 10.6 GHz radiation is at least -10 dB from the peak (labeled 266 in FIG. 16).

  The UNII band 267 extends from 5.15 GHz to about 5.9 GHz. FIG. 16 shows four PSD profiles (denoted as profiles 261, 262, 263, 264, respectively) corresponding to selecting the harmonic orders used separately. All four PSD profiles correspond to a baseband chipping reference clock frequency of 1.4 GHz. Also, the PSD profile is approximately 0.375 time displacement between the reference clock signal and the chipping sequence (see FIG. 13 and accompanying description of time displacement s and its effect on the PSD profile). Assume s.

  As mentioned above, PSD profiles 261, 262, 263, 264 show various choices for the harmonic order used. The PSD profile 261 corresponds to a communication system that uses the third to seventh harmonics of the chipping reference clock. Thus, such a system effectively occupies an authorized bandwidth between 3.1 GHz and 10.6 GHz.

  PSD profile 262 corresponds to a system that uses the third, fifth, sixth, and seventh harmonics of the chipping reference clock. That is, unlike the system corresponding to the PSD profile 261, the fourth harmonic that partially overlaps the UMI band 267 is excluded.

  The system corresponding to the PSD profile 263 uses the third to sixth harmonics of the chipping reference clock. Therefore, this system excludes relatively higher frequencies by not using higher order harmonics.

  The PSD profile 264 relates to a communication system using third, fifth, and sixth harmonics of the chipping reference clock. In this system, the fourth harmonic that partially overlaps with the UNII band 267 is excluded. In this system, its operating mode can be switched between PSD profile 261 and PSD profile 262 or between PSD profile 263 and PSD profile 264, as described in detail below.

  Table 1 below summarizes the harmonics used in the system corresponding to PSD profiles 261, 262, 263, 264.

上述のように、本発明の例示の実施の形態による通信システムは、マルチモード動作を含む。このようなシステムでは、所望のあるいは所定の条件または刺激に基づき、ある動作モードから別の動作モードに切り換えることができる。図３を参照すると、コントローラ入力信号４０により、トランスミッタ４におけるモード切換が可能になる。コントローラ入力信号４０、トランスミッタ４、すなわち、タイミングコントローラ４２の状態により、本発明の説明の利点を理解する当業者に明らかな方法で、基準クロックサイクルに対するチッピング持続時間を決定する。

As mentioned above, the communication system according to the exemplary embodiment of the present invention includes multi-mode operation. In such a system, it is possible to switch from one operating mode to another based on a desired or predetermined condition or stimulus. Referring to FIG. 3, the controller input signal 40 allows mode switching in the transmitter 4. The state of the controller input signal 40, transmitter 4, ie timing controller 42, determines the chipping duration for the reference clock cycle in a manner apparent to those skilled in the art who understand the advantages of the description of the invention.

  With the communication system according to the present invention, mode switching can be performed in response to virtually any stimulus, if desired. For example, the mode switching is performed by the system user manually selecting the mode. As another example, mode switching can be performed automatically in accordance with, for example, a predetermined or selected system event.

  As another example, mode switching can be performed semi-automatically, but the user manually selects according to the flagged event or the event that attracts the user's attention. In other embodiments, mode switching is controlled by internal or external variables or quantities, such as time. Alternatively, the operation mode can be switched by a remote signal received by the communication system.

  As yet another example, modes can be switched by various communication systems and devices according to embodiments of the present invention in response to detecting radio signal energy in one or more desired bands. For example, in response to detecting the presence of radio signal energy in the UNII band (between 5.15 GHz and 5.85 GHz), the UWB communication device or system according to the present invention causes the transmission to have predetermined spectrum data. In addition, the operation mode can be switched. The new mode of operation corresponds to a PSD profile that eliminates, reduces and minimizes interference from all devices operating in that particular band. For example, the new PSD profile constitutes the PSD profile 163 shown in FIG.

  Thus, the mode switching stimulus in such a system is to detect the presence of an RF signal from a device such as a UNII band device operating within a specific band or at a specific frequency or frequencies. The presence of a communication system or device is switched mode so as to eliminate or minimize interference, e.g. by eliminating harmonic components that result in transmission energy in the affected frequency range or band. Configure. Such a feature further provides means for coexisting with a device operating in an existing radio frequency band, such as a UMI radio device.

  Note that the above example only constitutes a sampling of how to switch between operating modes. Other techniques and mechanisms for mode switching can be used as desired by those skilled in the art who understand the advantages of the description of the invention, depending on the desired design and performance specifications. If desired, these techniques can also be applied to various embodiments of the communication system and apparatus according to the present invention.

  FIG. 17 shows an exemplary embodiment according to the present invention for a communication system incorporating mode switching. High speed data communication UWB communication system 11 includes a transceiver 111 having an internal power source 112 (eg, a battery or other power source). System 11 also includes a second transceiver 113 having an internal power source 114 (eg, a battery or other power source). Mode switching in the system 11 is performed depending on whether the system is operating with an internal power source or an external power source (not explicitly shown in FIG. 17).

  When system 11 uses internal power source 112 and internal power source 114, it can operate in a mode that matches a particular PSD profile. For example, the PSD mask 81 shown in FIG. This mode corresponds to, for example, indoor system operation. Because the system 11 has less potential interference with other systems while operating indoors, the PSD mask 81 corresponding to indoor operation may have more relaxed requirements.

  Conversely, if the system 11 uses external power (provided via port 115 to transceiver 111 or via port 116 to transceiver 113), the system 11 will not match another PSD profile. Can operate in any mode. For example, the PSD mask 82 shown in FIG. The second mode corresponds to, for example, outdoor system operation. Therefore, by switching the operation mode, the UWB communication system according to the present invention can be adapted to a stricter outdoor PSD mask and moreover match a more relaxed PSD mask during indoor operation.

  To switch modes, the system 11 detects the application of external power and provides a trigger signal to the transmitter controller input 40 (see FIG. 3). In response, as described above with reference to FIG. 8, the timing controller 42 and the harmonic generator 49 adjust predetermined timing parameters to generate a desired PSD profile. Similar operation occurs in the receiver circuit of the transceiver. The companion transceiver or corresponding transceiver also adjusts its transmitter and receiver parameters according to the particular PSD profile received by the receiver circuit.

  Although FIG. 17 shows a pair of transceivers, it should be noted that other systems may include a transceiver and receiver, or a transmitter or receiver, if desired. As is well known to those skilled in the art who understand the advantages of the present description, mode switching in such systems is performed using similar techniques and mechanisms as described above.

  Another aspect of the present invention relates to pulse shaping in the chipping signal 422 (reproduced in FIG. 18 for convenience). The chipping signal 422 includes a generally rectangular pulse. As a result, the spectrum in the frequency domain of the chip which is roughly given by the following known sine function can be obtained.

（チップは、信号４２２の縦セグメントの間、あるいは信号２２２のゼロ交差の間の時間における距離に対応する。）ミキサ４７での乗算演算により、そのスペクトラムを周波数領域内に移動させ、スペクトラムのコピーを信号４２１に存在する各高調波信号（高調波ジェネレータ４９の出力信号）で調心する。

(The chip corresponds to the distance in time between the vertical segments of signal 422 or the zero crossing of signal 222.) The multiplication operation at mixer 47 moves the spectrum into the frequency domain and copies the spectrum. Is aligned with each harmonic signal (output signal of the harmonic generator 49) present in the signal 421.

  While the above description assumes a chipping signal having a generally rectangular pulse (eg, chipping signal 422), other pulse shapes may be used if desired. For example, the pulse may have a more “round” shape.

  An example of a more “round” pulse shape is a Gaussian impulse. Mathematically, the Gaussian impulse s (t) is expressed as follows:

ここで、ｔは時間を表し、τはパルス幅を定義するパラメータを表す。フーリエ変換ｓ（ｔ）を用いることにより、周波数領域内のスペクトラムの形状を得ることができる。数学的に、フーリエスペクトラムｓ（ｔ）を次のように表すことができる。

Here, t represents time and τ represents a parameter defining the pulse width. By using the Fourier transform s (t), the shape of the spectrum in the frequency domain can be obtained. Mathematically, the Fourier spectrum s (t) can be expressed as:

上記の関係を用いて、周波数ｆ_Ｂ（例えば１．１ＧＨｚ）に対応するパルス幅を設計することができる。大きさＳ（ｆ）は、所望の量（例えば、１０ｌｏｇ［Ｓ（ｆ_Ｂ）］＝−１０ｄＢ）により、基準値を下回る。この技術により、τに対する設計値を生成し、次にｓ（ｔ）を評価できるようになる。

Using the above relationship, a pulse width corresponding to the frequency f _B (eg, 1.1 GHz) can be designed. The size S (f) falls below the reference value by a desired amount (for example, 10 log [S (f _B )] = − 10 dB). This technique makes it possible to generate a design value for τ and then evaluate s (t).

  Note that FIG. 18 shows an exemplary Gaussian impulse. Other shapes can be used if desired, as is well known to those of ordinary skill in the art who understand the benefit of the description of the invention. For example, if desired, a trapezoidal chipping signal 133, a chipping signal 134, and a chipping signal 135 shown in FIG. 12 can be used.

  Also note that by shaping or filtering the pulses before mixing with signals having higher frequencies (harmonic signals), avoiding pulse design or shaping at these higher frequencies. . For the filtered signal, the PSD can be obtained as follows.

ここで、ｐ（ｔ）はデータ信号を濾波したベースバンドを表す。一例として、図１８に示す１つのチップのチッピングシーケンス２２２等のガウス濾波信号がある。また、複数の高調波を用いることにより、整形パルスを周波数領域内に移動して、移動したものを所望の高調波キャリアで調心することができることに留意されたい。

Here, p (t) represents a baseband obtained by filtering the data signal. An example is a Gaussian filtered signal such as the single chip chipping sequence 222 shown in FIG. It should also be noted that by using multiple harmonics, the shaped pulse can be moved into the frequency domain and the moved can be centered with the desired harmonic carrier.

  Although FIG. 18 shows a chipping sequence 422 and a chipping sequence 222 having +1 and −1 amplitude swings, other swings can be used as is well known to those skilled in the art who understand the benefits of the description of the present invention. For example, if desired, a chipping sequence using a swing with an amplitude of +1.0 or -1 can be implemented.

  If desired, various modulation methods and techniques in the communication system and apparatus according to the present invention can be used. For example, in an exemplary embodiment of the invention, techniques similar to conventional quadrature phase shift modulation (QPSK) systems can be used. In other exemplary embodiments according to the present invention, techniques similar to offset QPSK (OQPSK) may be used.

  In particular, in the embodiment using QPSK, two harmonic carriers requiring two degrees of freedom are used so that signals related as two sets of harmonics have an orthogonal relationship. In particular, if there is a phase difference between the two reference clocks and a further phase lag in one of the harmonic generator lines, two desired degrees of freedom are formed. A QPSK-like UWB system according to the present invention using two harmonic carriers has the desired characteristic of a data rate twice that of the BPSK-like system, but is consistent with a given or desired standard. It also has a basically flat PSD profile.

  An OQPSK system is formed by further providing a half chip length offset between the two data streams modulating the quadrature harmonic carrier. Such OQPSK systems have the desired PSD spectrum or profile characteristics smoothed to the PSD profile of the QPSK system.

  FIG. 20 shows an example of a waveform of a set of OQPSKKUWB signals in the illustrative embodiment. Signal 2110 includes sinusoidal harmonics, such as the signal shown in FIG. 11A, while signal 2130 includes cosine harmonics, such as the signal shown in FIG. 11B. Regardless of the data signal 2120, the data stream 2120 changes the polarity of the signal 2110 and the data stream 2140 changes the polarity of the signal 2130.

  Also, the signal 2130 moves in time to the right side of the signal 2110 so that the maximum envelope value 2135 of the signal 2130 roughly corresponds to the minimum envelope value 2115 of the signal 2110. Also, to maintain orthogonality, the zero crossing of signal 2110 corresponds to the individual signal peak of signal 2130. Conversely, the zero crossing of signal 2130 corresponds to each peak of signal 2110.

  Signal 2150 represents the sum of quadrature signals 2110 and 2130. Those skilled in the art who understand the benefits of the description of the present invention will appreciate that the peak-to-average value of the composite signal is less than the peak-to-average value of either signal 2110 or signal 2130. This characteristic results in a smoother PSD profile and allows RF transmission at a power level with less 'safety' margin with respect to regulatory restriction levels.

  In other embodiments according to the present invention, a differential phase shift modulation (DPSK) method may be used. As is well known to those skilled in the art who understand the advantages of the description of the present invention, the transmitter according to the present invention can be changed, for example, to the transmitter 4 shown in FIG. 3 to generate the DPSK signal. The transmitter 4 generates a DPSK signal as follows. Referring to FIG. 3, transmitter 4 receives data at data input 44. Similar to conventional DPSK, the transmitter 4 encodes data differentially. That is, the transmitter 4 encodes the data to change within the data stream.

  For example, assume that the sequence starts with a binary “1” bit. When the next bit is “1”, it indicates that the transmitter 4 has transmitted “0” before that (no change). On the other hand, if “0” follows the original “1”, the transmitter 4 encodes “1”. Therefore, the transmitter 4 encodes the change from 1 to -1 (or -1 to 1) as binary "1". Conversely, the transmitter 4 does not encode a bit-to-bit change (eg, 1 followed by 1 or −1 followed by −1) as binary “0”. As can be seen from the above description, to transmit m bits, m + 1 bits (m bits of data follow the start bit) are transmitted. This is because data is encoded by changes in input data bits.

  Referring to FIG. 3, the data buffer 43 performs the above-described differential encoding. The PN generator 45 generates a chip sequence associated with a delay or time D equal to the number of chips for one data bit. The time delay D may be one chip time in one exemplary embodiment and may constitute an encoded sequence of bits in another illustrative embodiment (eg, D is one It can be the number of chips associated with a data bit). In other words, a time delay D per chip (time between the start of two chips) or per bit (time between the start of two bits) can be used. Regardless of the choice of time delay D, the system keeps D constant.

  In an exemplary embodiment according to the present invention, a chip sequence is generated by using a Barker code or sequence. Each chip sequence is equal in length to one of the well-known Barker sequences. Preferably, the transmitter 4 uses a 13, 11, or 7 length barker sequence, but as is well known to those skilled in the art to understand the description of the invention, other barker sequences may be used if desired. A sequence can also be generated. Table 2 below lists the well-known Barker codes.

当業者が理解するように、表２のコードシーケンスを逆にしたものもやはりバーカコードを構成する。また、記載のコードシーケンスを逆にしたもの（すなわち、１を−１で置換したり、この逆で得たりしたコードシーケンス）がバーカコードである。

As will be appreciated by those skilled in the art, the reverse of the code sequence in Table 2 also constitutes a Barker code. Further, a bar code is obtained by reversing the described code sequence (that is, a code sequence obtained by replacing 1 with -1 or obtained by reversing the code sequence).

  It should be noted that other types of codes can be used rather than using Barker codes, as is well known to those skilled in the art who understand the advantages of the present description. For example, if desired, a scissor cord can be used. Other examples include Hadamard codes, Walsh codes, and codes with low cross-correlation characteristics.

  The PN generator 45 multiplies each bit obtained from the data buffer 43 using the Barker sequence. Thus, the signal 424 (data / PN combiner 46 output signal) constitutes either a Barker sequence or the inverse of the Barker sequence (ie, obtained by multiplying the code sequence of Table 2 by -1). To do. For example, assuming that the PN generator uses a length 11 Barker code, the time or delay D is equal to the length of 11 chips. As another example, FIG. 12 shows one chip time. This is associated with the barker chip (signal 1562) shown in FIG. 6 and associated with a length 4 barker code).

  FIG. 19 shows an exemplary embodiment 19 of a differential receiver according to the present invention suitable for receiving a DPSK signal. The receiver 19 includes an antenna 910, a mixer 916, an integrator 918, a sample and hold 920, and an analog-digital converter (ADC) 922. The receiver 19 can optionally include an amplifier 912 and an amplifier 914.

  The antenna 910 receives a differentially encoded signal. Amplifier 912 amplifies the received signal and provides the resulting signal to one of the inputs of mixer 916 and amplifier 914. Through delay element 916, the output signal of amplifier 916 (if used) is connected to another input of mixer 916.

  The time delay D supplied by the delay element 916 is equal to 1 bit time. Therefore, the mixer 916 multiplies the reception signal by the reception signal delayed by time D. Due to the differential encoding of the signal (described above), when the receiver 19 receives binary “1”, the bit sign of the delayed received signal changes.

  The output from the mixer 916 is supplied to the integrator 918. The output of the mixer 916 constitutes the +1 Barker sequence of Table 2 multiplied by the reverse Barker sequence. Therefore, a voltage in the negative direction over the length of the Barker code is obtained at the output of the integrator 918. Sample and hold 920 samples the output signal of integrator 918 when the signal crosses the threshold. The sampled signal is supplied to the ADC 922 by the sample and hold 920. The ADC 922 generates output data bits.

  Note that in the illustrative embodiment, the integral of length is time D. However, based on the design and performance specifications, longer or shorter time periods may be used, as is well known to those skilled in the art who understand the advantages of the description of the invention.

  The optional amplifier 912 and the optional amplifier 914 can constitute either a linear amplifier or a limiting amplifier if desired. Furthermore, the amplifier 914 can be used to compensate for the loss of the delay element 916. It should also be noted that the amplifier 912 shown in FIG. 19 can be placed after the delay element 916.

  The delay element 916 can be implemented in a variety of ways, as will be appreciated by those skilled in the art who understand the advantages of the present description. For example, a relatively simple delay element includes a transmission line length of electrical length D. The length of the coaxial line, print strip line, or microstrip can be used in various ways to implement such an apparatus.

  By implementing the amplifier 912 and the amplifier 914 as limiting amplifiers, design requirements for the mixer 916 are eased. The mixer 916 can have a variety of structures and circuits, as is well known to those skilled in the art who understand the advantages of the present description. For example, if desired, the mixer 916 can comprise a passive ring diode mixer or a four quadrant multiplier.

  In a conventional DPSK system, the data bits constitute a length D equal to the length of one data bit. In such a system, the phase of the carrier (0 or π / 2 radians) is modulated at the bit rate. In contrast, a communication system or apparatus according to the present invention uses a harmonic wavelet Barker encoding sequence (eg, as shown in FIG. 6) instead of a carrier in a conventional system. The communication system or apparatus according to the present invention modulates the chip rate with the polarity of the wavelet (ie, +1 or −1). They also modulate the polarity of the chip sequence at the bit rate. Thus, in contrast to a conventional DPSK system, in a communication system and apparatus according to the present invention, the bit time (see signal 563 shown in FIG. 6) includes a wavelet encoding sequence.

  Note that further functions are performed by the circuitry associated with the receiver 19. As known to those skilled in the art who understand the advantages of the description of the present invention, for example, such a circuit restores the data bits, restores the timing of the chip sequence, and reduces the integration time of the integrator 918 depending on the signal quality. Make fine adjustments.

  Note that in the exemplary embodiment described above, each data bit is associated with a length N spreading code. That is, barber codes having lengths N = 2, 3, 4, 5, 7, 11, and 13 can be used. Therefore, N chips can be associated with one data bit. As an example, “1” can be transmitted by using the length 7 Barker code (see Table 2 above) and the sequence 111-1-11-1. Similarly, by using the sequence 1-1-111-11, "0" can be transmitted (that is, a sequence obtained by multiplying each digit of the previous sequence by -1).

  In other embodiments according to the invention, codes longer than those required for encoding one bit can be used. This provides several advantages. First, the spectrum of the resulting signal becomes more similar to white noise (ie, the advantage of spectrum “whiteness”).

  Second, channelization can be performed using such codes. Longer codes have a relatively large number of nearly orthogonal family members. By using such family members, it is possible to represent various symbols (that is, bit groups) and perform more effective channelization.

  As an example, a PN sequence generated by TIA-95 code division multiple access (CDMA) can be used. Such a sequence is 32,768 chips long. By multiplying the channel and the symbol (eg, using an exclusive-or operation) with a PN sequence (at a chipping rate) using a Hadamard code or a Walsh code (ie, 1111111100000000, as those skilled in the art will understand, It can also be defined (with a repeating sequence such as 1111000011110000, 1100110011001100, etc.). Thus, a chip group uniquely defines a symbol or channel. With such a technique, a signal having a relatively smooth spectrum can be obtained using the code lengths 32,767.

  In addition to channeling using relatively long codes, other techniques such as time division multiplexing and space division multiplexing (separate links using directional antenna techniques) if desired It can also be used. Such techniques are within the knowledge of those skilled in the art who understand the advantages of the description of the invention.

  As described above, in addition to encoding transmission data in an embodiment according to the present invention, error correction coding (ECC) can be performed if desired. For example, if desired, the ECC is adapted to the data input 44 shown in FIG. Many of such codes exist in the art and can be adapted to the communication system and apparatus according to the present invention, as those skilled in the art will appreciate the benefits of the description of the present invention. Such codes include BCH codes, Reed-Solomon codes, and Hamming codes.

  As described above, the carrier signal (for example, the carrier signal 21 shown in FIG. 2) can constitute a sine wave carrier signal or a non-sine wave carrier signal. FIG. 21 shows some examples of signal waveforms corresponding to non-sinusoidal carrier signals. FIG. 21 includes a repeated pattern “1010” of the chip 2022. Signal 2021 corresponds to the “1010” repeating pattern of the chip. As shown in FIG. 21, the signal 2021 has a gap 2023 of any length between its segments (of course having the parameters of the signal 2022).

  Another aspect of the invention relates to a plurality of separately modulated harmonic signals (eg, harmonics of any frequency such as a clock frequency). That is, in the communication apparatus according to the present invention, various harmonic signals can be modulated with either the same data stream or different (or set) different data streams. Thus, the effective data rate constitutes the sum of all data rates that modulate the harmonic signal.

  Also, if desired, each harmonic signal can be selectively enabled or turned on. In other words, the harmonic signals can be configured separately. In some configurations, the harmonic signals are not turned on or enabled simultaneously. In short, if desired, one harmonic signal or frequency is expected as a function of time from one harmonic signal or frequency.

  Configuring by selectively turning on the harmonic signal has a tremendous advantage in the communication device or system. A communication device or system can operate in the presence of multipath interference without the use of coding. That is, such a device or system operates in an environment where there is a multipath effect without encoding the signal that modulates each harmonic signal (as done in the above embodiments). Of course, please note. If desired, encoding can still be used, but this need not be done to suppress the effects of multipath interference.

  In order to suppress the effects of multipath interference, a communication device or system according to the present invention transmits an impulse of any harmonic frequency or channel and attenuates the multipath echo of that channel before transmitting the next. Wait for you to do. For example, assume that multipath interference in any environment has a delay spread of 25 ns. Thus, since the multipath is attenuated, the echo before receiving the next impulse or signal (the product of the harmonic signal and the signal chip or bit with one data bit) is about 20 ns time.

  By using multiple harmonic signals (ie, two or more harmonics) or frequencies, multiple data bits can be transmitted. That is, if the first data bit related to the frequency of the first harmonic signal is transmitted and the second data bit related to the frequency of the second harmonic signal is transmitted and the Nth harmonic signal is used, This is repeated until the next data bit (ie, data bit N) is transmitted. This cycle can be repeated if desired.

  The delay between subsequent transmissions using any harmonic signal can attenuate the multipath echo so that echoes from one transmission do not interfere with subsequent transmissions using that harmonic signal. In short, it takes advantage of the fact that there are a sufficient number of frequency and time combinations before the next transmission with any harmonic frequency. Many multipath echoes exist at attenuated frequencies. Further, by sufficiently separating the transmission frequencies, it is possible to reduce interference from multipath echoes of one harmonic frequency in transmission related to another harmonic frequency.

  FIG. 22 shows an exemplary embodiment of a transmitter 2200 according to the present invention that uses separately modulated harmonic signals. Note that the dashed lines in FIG. 22 indicate that a circuit operating at a relatively low frequency is separated from other circuits operating at a relatively high frequency. If desired, lower frequency circuitry can be included in one IC, and higher frequency circuitry can be included in another IC.

The reference clock 41 generates a signal using a desired frequency. For example, a sine wave is generated at a frequency f _osc . The reference clock 41 can be implemented in a variety of ways by using various techniques included in the knowledge of those skilled in the art who understand the benefits of the present description.

The reference clock 41 is connected to the harmonic generator 2220. Based on the clock signal, received from the reference clock 41, the harmonic generator 2220 generates the m th harmonic signal of the frequency of the clock reference 41. For example, an arbitrary a clock frequency f _osc , that is, the second harmonic signal output from the harmonic generator 2220 has a frequency of 2 · _fosc , and the mth harmonic signal generally has a frequency of m · _fosc. The same applies to the following.

  Note that m can be varied while transmitter 2200 is operating, if desired. That is, m is changed per data bit or is changed on a chip-by-chip basis. By changing m, a desired harmonic signal having an arbitrary frequency can be generated. Accordingly, if m = 3 is used, the third harmonic can be generated, and if m = 9, the ninth harmonic can be generated, and so on.

  As is well known to those skilled in the art who understand the advantages of the present description, similar to the harmonic generator 49 described above, the harmonic generator 2220 can be implemented in various ways. As an example, a frequency synthesizer can be used. By changing the control signal (eg, control voltage) of the frequency synthesizer, the output frequency of the frequency synthesizer can be changed. Therefore, if desired, a desired harmonic can be generated by applying a level of the control signal corresponding to the desired value of m.

  The harmonic generator 2220 generates a harmonic signal in synchronization with the reference clock. Various circuits and techniques can be used to synchronize one or more harmonics to the reference clock. Such circuitry and techniques are within the knowledge of those skilled in the art who understand the advantages of the description of the invention described above.

  The transmitter 2200 also includes a signal shaping circuit 2218 and a mixer 2202. If desired, the data shaping circuit 2218 and the mixer 2202 can be used to shape (or filter) the data signal 2206 as described in detail below. In the embodiment using this option, the output signal 2208 constituting the data pulse shaped by the mixer 2202 is generated.

  The transmitter 2200 also includes a mixer 2204 and an antenna 48. Output signal 2208 is provided to one input of mixer 2204. The output signal 2210 of the harmonic generator 2220 is supplied to the other input of the mixer 2204. The output signal of the mixer 2204 constitutes a modulated RF signal 2212. The antenna 48 receives the modulated RF signal 2212 from the mixer 2204 and propagates it to the transmission medium.

  Note that by changing the value of m, the transmitter 2200 can be made to a heterodyne operating frequency (shaped data pulse) of the output signal 2208 of the mixer 2202 to another RF frequency. That is, by changing the value of m to a function of time, as described above, the output frequency of the transmitter 2200 is hopped to various frequencies as a function of time. The value of m can be varied in various ways, as is well known to those skilled in the art who understand the advantages of the present description. For example, if desired, various functions of transmitter 2200 can be controlled, such as selecting the value of m using a controller (not explicitly shown).

  As is well known to those skilled in the art of understanding the description of the invention, integer or non-integer (eg, fractional) values of m can be used if desired. Thus, in general, the operating frequency of the output signal 2208 of the mixer 2202 can be derived if desired by using an integer or non-integer value of m. In other words, the operating frequency of the output signal 2208 of the mixer 2202 need not constitute (but can constitute) an integer number of harmonics of the clock signal. Rather, it relates to the clock frequency in any desired or arbitrary manner. For example, the clock frequency can constitute a fraction of the operating frequency of the output signal 2208. Such operating frequencies can also be generated using a frequency synthesizer, such as a fractional M synthesizer, as is well known to those skilled in the art who understand the advantages of the present description.

  Also, if desired, the intelligence signal or information signal can be modulated in various ways to generate the data signal 2206. As described above, as can be understood from the present technology as described above, BPSK modulation, quadrature amplitude modulation (QAM), QPSK modulation, or the like can be applied. As is well known to those skilled in the art who understand the advantages of the description of the invention, the choice of modulation method depends on the design and performance specifications of the particular implementation.

  FIG. 23 shows an exemplary embodiment of a receiver 2300 according to the present invention for receiving separately modulated harmonic signals. The receiver 2300 includes an antenna 58, a mixer 2314, a mixer 2316, an integrator / sampler (integrator / sample and hold) 2303, a controller 2306, a baseband template generator 2312, a phase locked loop (PLL) 2319, a harmonic Wave generator 2220.

  The antenna 58 receives the RF signal and supplies it to one input of the mixer 2314. The output signal of mixer 2316 constitutes the second input of mixer 2314. Baseband template generator 2312 generates a template signal that constitutes one input of mixer 2316. The output of harmonic generator 2220 constitutes the second input of mixer 2316.

  The output of the baseband template generator 2312 matches the output of the signal shaping circuit 2218 of the transmitter 2200 (see FIG. 22). Baseband template generator 2312 generates its output under the control of PLL 2319. Using the feedback in the receiver 2300, the PLL 2319 generates a reference signal 2322 that is a first output signal and supplies the reference signal 2322 to the baseband template generator 2312.

When receiver 2300 is synchronized with the desired RF signal, reference signal 2322 is configured similarly to the reference signal used in the corresponding transmitter of the RF signal. For example, referring to FIGS. 22 and 23, when the receiver 2300 synchronizes signal transmission by the transmitter 220, the reference signal 2322 generates the same signal as the reference signal generated by the clock reference 41 (see FIG. 22). That, PLL2319 is to have a frequency _{f osc,} and generates a reference signal 2322.

The PLL 2319 generates a second output signal 2328 having a frequency f _osc that is supplied to the harmonic generator 2220. Harmonic generator 2220 operates in conjunction with transmitter 2200 shown in FIG. 22 as described above. Therefore, the harmonic generator 2220, a harmonic signal, and supplies to the mixer 2316 with a frequency m _{· f osc.}

  As described above, the output of mixer 2316 is supplied to one input of mixer 2314. Receiver 2300 uses the output of mixer 2314 to control the feedback loop including PLL 2319 so that the output of mixer 2316 matches the RF signal received from antenna 58. The control loop includes an integrator / sampler 2303, a controller 2306, and a PLL 2319.

  The output of the mixer 2314 provides one of the two voltage levels as its output, depending on the data value received by the receiver 2300 that is supplied to the input of the integrator / sampler 2303. For example, if receiver 2300 receives binary zeros, the output of integrator / sampler 2303 constitutes a negative voltage. Conversely, when receiver 2300 receives binary 1, integrator / sampler 2303 provides a positive voltage as its output.

  The output of integrator / sampler 2303 is supplied to the input of controller 2306. The controller 2306 generates a data value according to the voltage level received from the integrator / sampler 2303. For example, in response to a positive voltage present at the output of the integrator / sampler 2303, the controller 2306 can generate a binary 1 bit having a desired digital level.

  Note that the controller 2306 can perform filtering, shaping, etc. of the data signal if desired. Controller 2306 also provides feedback control signal 2325 to PLL 2319, thus affecting the frequency of the signal generated by PLL 2319. In addition, the controller 2306 determines the value of m and supplies the value to the harmonic generator 2220.

  As described above, the harmonic generator 2220 generates the m-th harmonic of the output signal 2328 of the PLL 2319 as its output. Note that the sequence of values of m is determined as a function of receiver and transmitter time. During operation, the receiver and transmitter use various values of m according to a predetermined sequence.

  Note that the feedback loop can be implemented in receiver 2300 in various ways, if desired. As will be appreciated by those skilled in the art who understand the advantages of the description of the present invention, the choice of implementation depends on a number of factors such as design and performance specifications and characteristics. As known to those skilled in the art who understand the advantages of the description of the present invention, the feedback loop uses the baseband template generator 2312 to configure the synchronization mechanism to transmit the signal (ie, template receiver or matching template receiver). Receive.

  As described above, various frequency channels can be used in the communication apparatus and system according to the present invention by changing the value of m. Also, changing these values as a function of time by changing the value of m as a function of time. Accordingly, the channel frequency plan and channel timing plan of the communication apparatus and system according to the present invention can be specified. By varying the frequency and channel timing plan, a wide variety of communication devices and systems can be designed and implemented if desired.

  Table 3 below shows an example of the channel frequency and timing plan in the illustrative embodiment of the communication apparatus or system according to the present invention.

表３の例は、６つのチャネルを用いる通信装置またはシステムに対応する。また、装置あるいはシステムは、それぞれ８ｎｓの持続時間の６つの時間スロットを用る。従って、時間スロットは、４８ｎｓのインターバルで繰り返す。用いられる高調波は、第２８次高調波から第３８次高調波の範囲である。別の言い方では、ｍは、２８から３８の範囲である。１２５ＭＨｚのクロック基準周波数を用いると、チャネルは周波数３．５０ＧＨｚ〜４．７５ＧＨｚの範囲となる。

The example in Table 3 corresponds to a communication device or system that uses six channels. The device or system also uses 6 time slots, each with a duration of 8 ns. Therefore, the time slot repeats at an interval of 48 ns. The harmonics used range from the 28th harmonic to the 38th harmonic. In other words, m ranges from 28 to 38. With a clock reference frequency of 125 MHz, the channel is in the frequency range of 3.50 GHz to 4.75 GHz.

  That is, at time t = 0, m = 30 corresponds to a harmonic frequency of 3.75 GHz. This frequency corresponds to channel 2. In the latter 8 nanoseconds, t = 8 ns, m = 34 corresponds to a frequency of 4.25 GHz, which corresponds to channel 4. The same applies hereinafter. The frequencies shown in the second column of Table 3 indicate the frequencies of the on or enable harmonic signal (ie, the transmitter has modulated and transmitted).

  Note that the channels and their corresponding frequencies can be arranged in various ways, if desired. In the embodiment corresponding to Table 3, it is also possible to select the channel frequency corresponding to the time slot as far as possible from the neighboring time slots. Referring to Table 3, note that the neighboring channels are separated by the time of at least two time slots (ie, 16 ns). By selecting the channel, frequency, and timing plan in this way, interference between channels is reduced or minimized. Increase or maximize channel separation to help attenuate multipath interference.

  However, if desired, it has many other channels, frequencies, time slots, harmonics, rather than using the channel plan of Table 3, as is well known to those skilled in the art who understand the advantages of the description of the invention. Note that a wide variety of other devices or systems may be used. Depending on the desired system performance and design specifications, channels, frequencies and timing plans can be used to improve multipath blocking performance, channelize and adapt multiple users within a communication system.

  Depending on any arbitrary channel frequency and timing plan, various modulation methods can be used if desired, as is well known to those skilled in the art who understand the advantages of the description of the invention. Examples of modulation methods include BPSK, QPSK, 8-QAM, and 16-QAM. Selecting the channel and the type of modulation technique used affects the overall data rate of the communication system or device.

  Table 4 below shows examples of channels used and the approximate data rate that can be achieved for throughput (megabits per second) for various modulation techniques.

表４に示すような６つのチャネルを用いるよりもむしろ、所望ならば、もっと少ない数のチャネルを用いることができることに留意されたい。本発明の説明の利点を理解する当業者が周知のように、チャネル数と用いる変調技術との選択は、システム性能および設計仕様や考え方等の要因による。

Note that a smaller number of channels can be used if desired, rather than using six channels as shown in Table 4. As is well known to those skilled in the art who understand the advantages of the description of the present invention, the choice of the number of channels and the modulation technique used depends on factors such as system performance and design specifications and concepts.

  Table 4 also corresponds to a UWB system with a bandwidth of about 500 MHz per channel. If desired, the idea of the present invention can be applied to various UWB systems with other bandwidths. A bandwidth of 500 MHz corresponds to the minimum bandwidth defined as UWB in US Federal Communications Commission Enforcement Rules, 47 Federal Enforcement Rules Part 15.

  Note further that Table 4 corresponds to a system with a pulse repetition rate of about 20.83 MHz. This pulse repetition rate corresponds to a pulse of 8 nanoseconds that is transmitted at a 1/48 ns (about 20.83 MHz) pulse repetition rate every 48 nanoseconds (one cycle is 125 MHz reference).

The last row of Table 4 labeled “Est. E _b / N ₀ (dB)” represents the estimated or approximate energy used for each transmitted bit in which noise is present. Referring to Table 4, the described modulation method shows that the BPSK modulation has the lowest amount of energy per bit relative to the noise density ratio (7 dB for a bit error rate of about 0.1%), but the total data throughput is It is also the lowest. Conversely, 16-QAM has the highest energy per bit for the noise density ratio (16 dB for a bit error rate of about 0.1%), but the highest total data throughput (about 8 hours than BPSK). high). In general, the more complex the modulation method, the higher the energy level used to transmit bits with a specified bit error rate where noise is present.

  From the information in Table 4, it is possible to design and implement a communication device or system that meets IEEE 802.15.3a proposed by the draft standard. The proposed draft specifies a data rate of about 110 megabits per second, about 200 megabits per second, and about 480 megabits per second. The cells highlighted in bold in Table 4 indicate a combination of the modulation method and the number of channels used so that such a device or system can be implemented in a flexible manner.

  This flexibility is desirable when the power spectral density is limited so that the total transmit power is proportional to the total bandwidth used (ie, the number of channels used in Table 4) due to the specified regulatory transmission limits. Because it becomes. Therefore, transmission of 125 Mb / s can be performed by using three channels by using QPSK or by using six channels by using BPSK. If six channels are used, the total radiated power is twice the total radiated power of the three channels in the extended range communication. Thus, in some systems, bandwidth can be exchanged for range at any or desired data rate.

One aspect of the apparatus or system according to the present invention relates to its extensibility. That is, by using the _f osc 125 MHz in the next center frequency may be a frequency to design a channel of the plurality of 500MHz width in the range of 3.1～5.2GHz.
f ₁ = 28 f _osc = 3.500 GHz
f ₂ = 29 f _osc = 3.625 GHz
f ₃ = 30 f _osc = 3.750 GHz
...
And f ₁₃ = 40 f _osc = 5.000 GHz.

In creating the rules, the FCC has limited UWB emissions up to -41.3 dBm per MHz. For each channel, the power can be determined from:
P _c = -41.3 + 10 log (2.374 f _osc )
Or -16.6 dBm per channel
Thus, the total radiated power can be increased by increasing the number of channels to obtain a higher data rate. For example, two channels supply power that is 3 dB larger than one channel. As another example, four channels supply power that is 6 dB greater than one channel. The same applies hereinafter. By using multiple channels, if one modulation method is used (eg, not switching from BPSK to QPSK, etc.), the total radiated power is increased at the same rate as increasing the total data rate or data throughput.

  As a result, the bandwidth can be exchanged for the communication range while the communication range remains approximately constant with increasing data rate or data throughput as described above. Thus, the device or system according to the present invention provides the desired extensibility features such that increasing the data rate or data throughput does not reduce the communication range. That is, by increasing the number of channels, communication with higher data throughput can be achieved in a desired range, so that the total radiated power increases.

  It should be noted that the above examples having specific system parameters, such as frequency and frequency range, constitute an illustrative embodiment of the idea of the invention. As is well known to those skilled in the art who understand the advantages of the present description, various other system parameters (eg, frequency and frequency range) may be used, if desired, depending on various factors such as the desired design and performance specifications. You can also

As an example, two channels in the 3.1-5.1 GHz band can be used with the next center frequency at f _osc = 232 MHz (ie, the channel is wider than 500 MHz).
f ₁ = 16 f _osc = 3.712 GHz
f ₂ = 20 f _osc = 4.640 GHz
Or m = 16, 20 respectively
P _c = -41.3 + 10 log (2.374 f _osc )
Or -13.9 dBm.

  By convention, the FCC allows UWB radiation in the frequency band or range of 3.1 to 10.6 GHz. The exemplary channel plan described is in compliance with FCC rules, but coexisting communications with UNII bands are also possible. Therefore, when it is desired to avoid possible interference with communication in the UNI band, an example of a desired frequency range is configured in the range of 3.1 to about 5.2 GHz.

By creating the FCC rule referred to above, one mask is designated as a bandwidth defined by −10 dB points. In the above example, 10 dB point occurs at _2f osc = 464MHz, -20dB point occurs at _2.62f osc = 607.84MHz. Therefore, an apparatus or system according to an illustrative embodiment based on this example meets the FCC specification of −10 dB at 3.1 GHz. Note that in this example, the two center frequencies correspond to a bandwidth of 928 MHz, and the two channels are here adapted to the desired frequency range of 3.1-5.2 GHz.

  Of course, other embodiments according to the present invention can be implemented using a wide range of parameters such as frequency plan, modulation method, etc., as is well known to those skilled in the art who understand the advantages of the description of the present invention. In fact, as described above, other frequency synthesis methods in which the value of m is not an integer can be used.

  Another aspect of the present invention relates to signal shaping in a communication device. That is, the signal shaping circuit 2218 shown in FIG. 22 performs a method of shaping, processing, or filtering the output signal 2202A of the reference clock 41 to generate the shaped output signal 2204A. The shaped output signal 2204A is supplied to one input of the mixer 2202 as described above.

  The signal shaping circuit 2218 affects the spectrum of the output signal of the mixer 2204, but basically constitutes the transmission signal of the transmitter 2200. That is, rather than mixing the signal 2204 A shaped using the signal shaping circuit 2218 with the data signal 2206, the data signal 2206 can simply be supplied to the mixer 2204. As a result of bypassing or not using the signal shaping circuit 2218, the spectrum of the transmitted signal includes relatively high sidelobe levels. These sidelobe levels are compatible with the desired mask, such as a mask defined by the FCC or another regulatory agency.

  By using the signal shaping circuit 2218, the side lobe of the transmission signal can be reduced or lowered. As a result, the spectrum of the transmitted signal can more easily meet the more stringent spectrum emission or mask requirements. Note that a similar signal shaping function applies to the receiver that receives and processes the signal transmitted by transmitter 2200.

  Referring to FIG. 23, the receiver 2300 constitutes a matched template or matched filter receiver. Baseband template generator 2312 forms the same or similar signal shaping functionality as signal shaping circuit 2218 performs at transmitter 2200. In other words, as described above, the output of the baseband template generator 2312 matches the output of the signal shaping circuit 2218 of the transmitter 2200.

  Note that the signal shaping circuit 2218 (and the corresponding signal shaping of the receiver 2300) can form virtually any desired signal shaping, processing, or filtering function, if desired. For purposes of explanation, DC shaping components (such as DC voltage) can be added by signal shaping circuit 2218, as is well known to those skilled in the art who understand the advantages of the description of the present invention (eg, full wave of the input signal). A magnitude function or the like (by rectification) can be formed.

  Also, if desired, various functions can be combined. For example, a DC offset can be added to the magnitude function. In general, by configuring the transfer function of the signal shaping circuit 2218, a wide variety of signal shaping functions or a combination of functions can be applied. Selective use of the function is a factor in various designs and performance, such as desired spectral characteristics and / or desired levels deviating from band energy, as is well known to those skilled in the art who understand the benefits of the description of the invention. based on.

  Rather than shaping the signal using analog circuitry, digital or mixed mode circuitry can be used if desired. For example, the signal samples can be stored in a memory such as a read only memory (ROM). Based on the input signal to the signal shaping circuit 2218, various addresses in the memory can be accessed using a counter, and a desired signal can be generated by the output of the signal shaping circuit 2218.

  By using an appropriate transfer function of the signal shaping circuit 2218, the spectrum of the data signal 2206 can be smoothed. In other words, the high frequency data of the baseband data signal 2206 can be reduced. As described above, the data signal 2206 generally has a pulse shape (eg, a rectangle or a pulse train). For example, assume that the signal shaping circuit 2218 applies the magnitude function to the output signal 2202A of the reference clock 41.

  The output signal 2204A of the signal shaping circuit 2218 constitutes a rectified cosine signal whose spectrum contains less high frequency data than the spectrum of the data signal 2206. Accordingly, the output signal 2212 of the mixer 2204, i.e., the transmitted signal, has a lower level sidelobe.

  In the above example, it should be noted that the magnitude function can be implemented without the use of analog filter components, as is well known to those skilled in the art who understand the advantages of the description of the invention. Thus, if desired, the magnitude function can be implemented in an IC that primarily includes digital circuitry. This further forms processing and manufacturing flexibility, providing higher reliability and lowering costs.

  FIG. 24 and FIG. 25 show sample waveforms of an example of the magnitude function implemented in the illustrative embodiment of the pulse shaping circuit 2218 according to the present invention. That is, FIG. 24 shows one cycle of output signal 2208 (assuming rectangular data signal 2206) of mixer 2202 shown in FIG. That is, the reference clock 41 generates a cosine signal that is supplied to the signal shaping circuit 2218. The signal shaping circuit 2218 processes the signal to generate the magnitude, and supplies the obtained signal (signal 2204A) to the mixer 2202. The mixer 2202 mixes the signal 2204A with the input data signal 2206 to generate an output signal 2208.

FIG. 25 shows a Fourier transform of the signal shown in FIG. In other words, FIGS. 24 and 25 represent the time and frequency domain of the output signal 2208 of the mixer 2202, respectively. Therefore, the waveform shown in FIG. 24 represents a time signal.
s (t) = cos (2πf _osct )
The spectrum shown in FIG. 25 indicates the spectrum s (t) or S (f).

この例示の実施の形態では、信号整形回路２２１８により、マグニチュード機能を実施する。入力データ信号２２０６で乗算した余弦関数の大きさ（すなわち、信号整形回路２２１８の出力信号２２０４Ａ）は、図２４に示すように（しかしながら、図２４は１サイクルの信号２２０８を示していることに留意されたい）、ミキサ２２０２の出力信号２２０８を生成する。上述のように、図２５は、図２４に示す信号のフーリエ変換を示す。効果的には、このような実施の例では、入力チップを余弦関数により重み付けする。

In this exemplary embodiment, the signal shaping circuit 2218 implements the magnitude function. Note that the magnitude of the cosine function multiplied by the input data signal 2206 (ie, the output signal 2204A of the signal shaping circuit 2218) is as shown in FIG. 24 (however, FIG. 24 shows the signal 2208 in one cycle. The output signal 2208 of the mixer 2202 is generated. As mentioned above, FIG. 25 shows the Fourier transform of the signal shown in FIG. Effectively, in such an embodiment, the input chip is weighted with a cosine function.

Note that the maximum chip rate constitutes twice the frequency of the reference clock, ie 2 f _osc . However, it is also possible to transmit a coarse chip at the next rate.

ここで、

here,

また、
Ｐ_ｃ＝−４１．３＋１０ｌｏｇ（２．３７４ｆ_ｏｓｃ）
−１０ｄＢポイントと−２０ｄＢポイントはそれぞれ、２ｆ_ｏｓｃと２．６２ｆ_ｏｓｃとを構成する。３．１ＧＨｚを越える（ＦＣＣ規定ののマスク端）最も近い周波数を書き込むこともできる。信号レベルは−２０ｄＢ（またはこれ未満）、ｆ_１は次の通りである。
ｆ_１＝（ｍ−２．６２）・ｆ_ｏｓｃ
ここで、ｍは整数である。

Also,
P _c = -41.3 + 10 log (2.374 f _osc )
Each -10dB point and -20dB _point, constitutes a _{2f osc} and _{2.62f osc.} It is also possible to write the closest frequency exceeding 3.1 GHz (FCC prescribed mask edge). The signal level is −20 dB (or less), and f ₁ is as follows.
f ₁ = (m−2.62) · _fosc
Here, m is an integer.

It should be noted that the baseband signal is composed of the signal shown in FIG. 24 and the spectrum associated therewith shown in FIG. That is, the signal spectrum shown in FIG. 24 is centered at approximately zero frequency or DC. As can be seen from the transmitter of FIG. 22, the baseband signal can also be heterodyned to align to a relatively high frequency (RF frequency). That is, using a mixer 2204, an output signal 2208 of (by mixing with the signal 2210) mixers 2202 and heterodyne and aligning approximately frequency m _{· f osc.}

  The spectrum of the signal 2208 is moved to the frequency band by heterodyne processing. The frequency band can constitute a desired frequency band such as a band defined by a supervisory organization (for example, FCC) or any other specified, designated or designed frequency band. By using the inventive concepts described herein, the RF device or system can be adapted so that the shifted spectrum matches or matches the desired or specified mask, eg, the mask specified in the FCC rule creation. Can be designed.

  FIG. 26 shows sample waveforms in an exemplary embodiment of a transmitter according to the present invention, such as transmitter 2200 of FIG. A waveform 2605 corresponds to the output signal 2202A of the reference clock 41. A waveform 2610 shows the shaped output signal 2204A. That is, the output of the signal shaping circuit 2218. In this particular embodiment, shaped output signal 2204A constitutes the magnitude of signal 2202A. However, as described above, if desired, the signal shaping circuit 2218 can be configured to perform virtually any signal shaping or transfer function.

  Waveform 2615 shows the input data signal 2206 (see FIG. 22). Waveform 2620 shows output signal 2208 as a function of time. That is, the output signal of the mixer 2202. Note that, as described above, waveform 2620 corresponds to the product of waveform 2610 and waveform 2615 (mixed). As described above, waveform 2620 corresponds to a baseband signal (ie, a signal that is centered at approximately zero frequency, ie, DC).

  A waveform 2625 shows the output signal 2210 of the harmonic generator 2220. Note that waveform 2625 corresponds to a particular value of m. As described above, the value of m varies as a function of time. Accordingly, the frequency of waveform 2625 also varies as a function of time (proportional to the value of m).

  Waveform 2630 shows the output signal of mixer 2204, which constitutes modulated RF signal 2212. Note that the mixer 2204 mixes the output signal 2208 (baseband signal) with the output signal 2210 (RF signal) of the harmonic generator 2220 to produce a modulated RF signal 2212.

In-phase and quadrature channels can be generated as part of the heterodyne method, as is well known to those skilled in the art who understand the advantages of the present description. That is, the in-phase or I channel can be generated by mixing the output signal 2208 or pulse with a cosine signal. Therefore,
s _I (t) = [cos (2πf _osc t)] · cos (2πf ₀ t)
and

ここで
ｆ_０＝ｍ・ｆ_ｏｓｃ
図２７は、例示のＩチャネルパルス（移動またはヘテロダインした信号を生成したもの）を時間の関数として示す。図２８は、図２７の信号スペクトラムの大きさを示す。ヘテロダインによりベースバンド信号のスペクトラムを移動して、およそ相対的に高周波数（約４ＧＨｚ）に調心したことに留意されたい。

Where f ₀ = m · f _osc
FIG. 27 shows an exemplary I-channel pulse (which generated a moving or heterodyned signal) as a function of time. FIG. 28 shows the magnitude of the signal spectrum of FIG. Note that the spectrum of the baseband signal was moved by the heterodyne and centered at a relatively high frequency (approximately 4 GHz).

Conversely, the output signal 2208, or pulse, can be mixed with a sinusoid to produce a quadrature or Q channel. Therefore,
s _Q (t) = [cos (2πf _osc t)] · sin (2πf ₀ t)
and

図２９は、例示のＱチャネルパルス（移動またはヘテロダインした信号を生成したもの）を時間の関数として示す。図３０は、図２９の信号スペクトラムの大きさを示す。ヘテロダインにより、スペクトラムを移動して、およそ相対的に高周波数（約４ＧＨｚ）に調心したことに留意されたい。

FIG. 29 shows an exemplary Q channel pulse (which generated a moving or heterodyned signal) as a function of time. FIG. 30 shows the magnitude of the signal spectrum of FIG. Note that due to the heterodyne, the spectrum has been shifted to approximately the higher frequency (approximately 4 GHz).

Note further that the S (f) formula for the I and Q channels is the same as the spectrum magnitudes shown in FIGS. As is well known to those skilled in the art who understand the advantages of the description of the invention, the phase of the spectrum is different for the I and Q channels. (However, as described above, FIG. 28 and FIG. 30 show the magnitude of each spectrum, so the phase difference is not shown.)
As described above, by using the signal shaping circuit 2218, the size of the side lobe (see FIG. 22) shown in the output spectrum or profile of the transmitter 2200 can be reduced. FIG. 31 and FIG. 32 show examples in which how the pulse is shaped affects the size of the side lobe.

  FIG. 31 shows two signals as a function of time corresponding to a cosine shaped pulse and an unshaped pulse in an illustrative embodiment according to the present invention. Signal 3105 corresponds to the cosine shaping pulse of output signal 2212 (see FIG. 22). In other words, using the signal shaping circuit 2218 corresponds to a state in which the magnitude function is implemented. On the other hand, signal 3110 corresponds to a state where no signal shaping is applied to signal 2202A. That is, in the latter case, the signal 2204A constitutes a rectangular pulse, that is, a DC level.

  FIG. 32 shows a spectrum obtained by using the signal shaping shown in FIG. Spectrum 3205 corresponds to a cosine weighted pulse (shown as signal 3105 in FIG. 31). The spectrum 3210 corresponds to a state where no signal shaping is applied. Note that the side lobes of spectrum 3205 are smaller in size than the side lobes of spectrum 3210. The spectrum mask 3215 indicates a desired or designated mask such as a mask defined by the FCC.

  As described above, virtually any signal shaping can be applied via the signal shaping circuit 2218. FIGS. 33 and 34 further show examples of how shaping the pulse affects the sidelobe size.

  FIG. 33 shows, as a function of time, two signals corresponding to a Gaussian shaped pulse and an unshaped pulse in an illustrative embodiment according to the present invention. The signal 3105 is a Gaussian shaped pulse of the output signal 2212 (see FIG. 22), in other words, at its output,

により説明したように、信号整形回路２２１８がガウス整形パルス（または整形パルスに近似しているもの）を生成する状態に対応する。図３１と同様に、出力信号３１１０は、信号整形を信号２２０２Ａに全く適用しない状態に対応する。

This corresponds to the state in which the signal shaping circuit 2218 generates a Gaussian shaping pulse (or an approximation to the shaping pulse) as described above. Similar to FIG. 31, output signal 3110 corresponds to a state where no signal shaping is applied to signal 2202A.

  FIG. 34 shows a spectrum obtained using the signal shaping shown in FIG. Spectrum 3405 corresponds to a Gaussian weighted pulse (shown as signal 3305 in FIG. 33). A spectrum 3410 corresponds to a state where no signal shaping is applied. Note that, as in FIG. 32, the side lobes of spectrum 3405 are smaller in size than the side lobes of spectrum 3410. Similar to the spectrum mask 3215 shown in FIG. 32, the spectrum mask 3415 represents a desired or designated mask such as a mask defined by the FCC.

  Note that selecting the signal shaping function implemented or applied by the signal shaping circuit 2218 does not affect the characteristics of the main lobe in the spectrum of the resulting output signal of the transmitter. That is, even if a certain signal shaping method (for example, the above-described technique) tends to reduce the size of the spectrum side lobe, the main lobe characteristic tends to remain relatively unchanged. As an example, it should be noted that in FIG. 32, the main lobe of spectrum 3205 is generally the same shape and size as the main lobe of spectrum 3210.

  Depending on the signal shaping performed or applied by the signal shaping circuit 2218, the magnitude of the side lobe (that is, the signal 2212 in FIG. 22) of the spectrum of the signal obtained from the output of the mixer 2202 may be too large. That is, the side lobe of the spectrum of the obtained signal exceeds the limit defined by the specific mask. In such a case, the signal 2212 may be filtered before being applied to the antenna 48. Filter transfer functions (ie, filtering characteristics) can be configured or designed to remove or reduce energy at a certain frequency or within a certain frequency band. This reduces the size of the sidelobe that would otherwise not meet the constraints of a particular mask.

(System and method for generating and receiving UWB signals based on sustained waveforms)
The present invention discloses a system and method for generating and receiving ultra-wideband signals based on sustained waveforms. The basic part of the technology was mainly developed around a broadband impulse type signal with one or more cycles in the pulse type waveform. Typically, these systems are primarily based on spread spectrum pulse shapes, rather than code or modulation. However, typically, the code is used to shape the spectrum and otherwise smooth the comb. Also, because these pulses are often generated by a trigger pulse generator or edge source, they are not a direct result of the carrier and typically do not result in a carrier-based signal.

  However, these pulses are usually fixed at the center frequency, bandwidth, and pulse shape, and it is difficult to control this shape due to stray inductance and capacitance at microwave frequencies that vary within acceptable limits. This is because the shape is generated by directly using highly affected analog components. Therefore, these pulse technologies are not suitable for the precise frequency control required by legal agencies and can easily switch between such pulses as required by certain system requirements such as multiple channel separations. Therefore, it cannot be operated within a plurality of frequency bands.

  Thus, there is a need to improve the signal generation technology and associated receiver technology to meet the market needs for specific UWB applications.

  The present invention described herein is a system and method for generating and receiving UWB signals that use continuous or generally continuous waveforms. The advantages of this method are many and will become apparent from the detailed description of various embodiments. These advantages include accurate spectrum control as a result of linking spectrum generation to a reference oscillator designed to have an accurate frequency, such as a crystal oscillator. In addition, accurate spectrum shaping is obtained by using a plurality of carriers, and each part of the spectrum is controlled.

  Another advantage of the present invention is that as a result of generating a plurality of frequencies separated from the signal, a frequency separated from a plurality of user subband channels with high separability is generated. Each occupies part of the total bandwidth. The frequency separated from these signals also includes time coding, enabling code channels within the sub-band, maximizing spatial spectrum reuse and obtaining maximum spatial spectrum efficiency.

  A further advantage of the present invention is that multiple frequencies separated from the signal can be phase or amplitude encoded for multi-user operation or multi-frequency modulation.

  An advantage of the present invention is further that the spreading bandwidth is set or adjusted to provide the best match to the environment to obtain maximum signal gain in multipath. A further advantage is that multiple subbands are used to efficiently collect signal energy for maximum range performance or to improve fast data rate capability.

  It also increases fast data rate capability by using multiple frequencies separated from these signals to separately modulate the signals to allow parallel data transmission. This modulation can also be performed using a spatially efficient modulation type to obtain maximum spatial spectrum efficiency.

  With a plurality of frequencies separated from the signal, a signal can be generated and received in a wide range. Some of the multiple frequency architectures are unique, but because they can efficiently generate and receive wideband pulse signals, they can also communicate with traditional pulse design architectures.

  In one embodiment, the multiple frequencies are generated separately. In another, multiple frequencies are generated using a common modulo period. In another, the plurality of frequencies are harmonics of a common base frequency.

  In one embodiment, multiple frequencies are generated using multiple phase locked loops connected to a common reference. These multiple phase-locked loops can utilize separate frequency dividers, can utilize comb samplers, and can switch multiple dividers off when not needed and multiple By sharing the division function among a plurality of carriers using the conversion processing, the deterministic division characteristic of the frequency divider can be used together with the low power consumption and the comb sampler used efficiently.

  In one embodiment, the multiple frequencies are further controlled in the spectrum by filtering. In another embodiment, multiple frequencies are further controlled in the spectrum by a time domain window function. The time domain window function takes the form described by the DSP. Rectangle, triangle, Hamming, Hanning, Kaiser, Dorf Chebyshev, Gauss, etc. This time domain window function is applied at a common modulo period or at a common base frequency.

  In one embodiment, additional frequencies are generated by generating a first set of frequencies and then mixing the first set of frequencies to generate sum and difference frequencies.

  Embodiments further allow full Nyquist sampling of wideband inputs by dividing the band into low frequency components that can be sampled with a practical sampler by multiple frequencies.

  In another embodiment, the receiver uses generally the same sampling process by performing code rake processing and improving the signal to noise ratio by the receiver, but also adjusts the received signal weighting factor. If possible, deviate from the transmitted code in a multipath environment and make noise from the gain signal. These weightings by performing channel measurements and sample time matching, or by measuring performance (SNR, etc.) to obtain each change in the best weight sample weight selected for reception Factors can be adjusted to the channel pattern.

  A further advantage of the disclosed architecture is to efficiently generate these signals with minimal complexity and cost. This architecture enables consumer and commercial applications that maximize integration on semiconductor chips, benefit people's lives, and stimulate the economy.

  Further benefits and advantages of the present invention will become apparent from the detailed description of the preferred embodiment disclosed herein.

  The present invention will be described in more detail with reference to FIG. FIG. 35 is an exemplary block diagram of a transmitter according to the present invention. The transmitter includes a plurality of sine wave generators 3502 that generate sine wave signals 3504, also referred to as subcarrier signals or carrier signals 3504, an integration node 3506 that accumulates the sine wave signals 3504, and a modulation that modulates the sine wave function with a data signal 3510. 3508. The obtained modulated signal is transmitted to the antenna 3512 and transmitted. In one embodiment, sine wave signals 3504 need not be synchronized with each other or with data 3510. However, in another embodiment, it can be synchronized with a sinusoidal signal.

  FIG. 36 is a diagram illustrating an exemplary spectrum of a signal generated by a transmitter according to the present invention. The spectrum shown in FIG. 36 represents an unmodulated signal including comb 3604, one comb for each sinusoidal signal 3504. When modulating, the spectrum of each sine wave signal is spread by modulation, occupying the gaps between the combs 3604, and a relatively flat spectrum occupying the range of a plurality of sine wave signals 3504. .

In one embodiment, the sine wave signal is synchronized with the common reference signal and becomes a set of sine wave signals. This is at the harmonic frequency of the common reference signal, and can be specified by each harmonic number 3602. For example, if the carrier is synchronized with multiple 100 MHz, the 3.1-5.15 GHz range will be filled with multiple 32-50 carrier operating at 100 MHz reference harmonic frequency, so the specified This creates a signal for use in an ultra wideband system that fills the available bandwidth up to the power limit across the entire bandwidth indicated by the mask. (Note that the current UWB band has expanded to 10 GHz, but avoiding interference from the NII band by interrupting at 5.15 GHz.)
FIG. 37 is an exemplary block diagram of a receiver according to the present invention. The receiver comprises a plurality of parallel channels. Each channel tracks a separate sinusoidal component of the transmitted signal separately. After taking the correspondence, the output of each channel is integrated. The resulting integrated signal is then filtered, amplified and demodulated. Referring to FIG. 37, a signal 3712 is received using an antenna 3512 and the signal 3712 is connected to a mixer stage 3702 that correlates with a sine wave signal 3502 and generates a correspondence output. The correspondence output is connected to the detection path and the tracking path. If necessary, the detection path signal is integrated 3703 with signals from other channels, filtered 3704 (integrated), and amplified 3706. Next, the signal 3712 is demodulated by the demodulator 3708 based on the original modulation to obtain the data signal 3510. In the case of flip modulation, for example, demodulator 3708 can be a comparator.

  Also, the correspondence output is supplied to the tracking path. This is to filter this output and supply it to 3710, the frequency control of oscillator 3502, to maintain signal tracking. If the modulation used by the transmitter results in a carrierless signal, it is necessary to use a technique such as a Costas loop or decision feedback 3714 as shown in FIG.

  FIG. 38 shows a tracking loop used in FIG. The tracking loop tracks one carrier from the transmitter as shown in FIG.

  FIG. 39 is an exemplary block diagram of a transmitter using multiple synchronous sine wave generators 3502 according to the present invention. In FIG. 39, clock 3902 drives comb generator 3904 to generate a signal with sufficient harmonics in the desired band. A typical waveform of the comb generator 3904 is a rectangular wave, a triangular wave, or an impulse train. Each harmonic signal is filtered by a bandpass filter 3906 to produce a sinusoidal signal having approximately one frequency for each filter output. Next, the sine wave signal is integrated 3506 to generate a composite signal. Next, the composite signal is modulated 3508 and transmitted.

  The clock 3902 also drives the data path to modulate 3508, 3910, transmit 3512, and synchronize with the clock 3902. The code 3908 is also used with data to generate an encoded data signal 3914 to further spread the spectrum, improve multiple user access, and achieve other benefits of encoding. For example, the system can be designed with a reference frequency of 100 MHz to generate a plurality of sinusoidal signals from 3.1 GHz to 5.0 GHz. By clocking the data at 100 MHz / 11, that is, 9.0909 Mbps, the data can be transmitted using the Barker 11 length code. Each data bit value (1 or -1) is used to modulate the polarity of the corresponding 11 chip code sequence at a chip rate of 100 Mcps (megachips per second).

  FIG. 40 is an exemplary block diagram of a transmitter according to the present invention in which each sinusoidal signal is modulated and synthesized separately. FIG. 40 also shows details of a splitter 4002 that divides the comb signal into a plurality of paths. The transmitter of FIG. 40 is combined with the features shown in FIG. 39 to achieve encoding and synchronization. The transmitter of FIG. 40 can also use a complex modulation method such as IQ modulation that modulates each sinusoidal signal separately.

  FIG. 41 is an exemplary block diagram of a transmitter according to the present invention. This produces a plurality of sine wave signals by mixing two sine wave signals to produce an accumulated and difference frequency. Referring to FIG. 41, the 3.5 GHz 3502A oscillator and the 500 MHz 3502B oscillator may or may not be synchronized if desired in a particular system. Two oscillator signals are supplied to the mixer 4102. When the mixer 4102 is a balanced mixer, the integrated and difference frequencies are generated at the output, and the input frequency is canceled at the output. If the mixer is not a balanced mixer, there is an input signal. Further, when a certain oscillator frequency is desired as an output, a balance mixer can be used to add the desired oscillator frequency to the output of the mixer using an integrating circuit or a combiner. Next, the output of mixer 4102 is modulated 3508 by data or encoded data, often referred to as chip 3914.

  FIG. 42 is an exemplary block diagram of a transmitter according to the present invention in which multiple sinusoidal signals are generated by separate phase locked oscillators. Referring to FIG. 42, the reference oscillator 3902 drives a plurality of phase locked oscillators 4202 and a data path. Each phase-locked oscillator 4202 includes a phase detector having a phase-locked loop structure, a filter, and a frequency-controllable oscillator. Each oscillator output is configured to operate at a plurality of different reference frequencies by using different division ratios in the phase locked loop circuit. The phase locked loop architecture allows the relative amplitude and phase of each sine wave generator to be accurately controlled. In the data path, the data 3510 is clocked 4204 in synchronization with the reference oscillator 3902 before modulating the composite sine wave signal 3508.

  FIG. 43 is an exemplary block diagram of the transmitter shown in FIG. 42, showing further details and features. The system of FIG. 43 illustrates using a comb generator 3904 to drive the phase locked loop 4202 and includes a code sequence in the data path.

  FIG. 44 shows an exemplary phase locked loop 4202 used in the transmitter of FIG. The phase locked loop 4202 includes a reference oscillator 3902, a phase detector 4402, a loop filter 4404, a sine wave generator 3502, and a frequency divider 4406. In the transmitters of FIGS. 42 and 43, each loop shares a common reference oscillator 3902. Each phase-locked loop 4202 uses different frequency division numbers for the frequency divider 4406 to generate signals of different harmonic frequencies of the reference oscillator 3902.

  FIG. 45 shows another phase locked loop 4202 that uses a sampling detector 4502. By eliminating the need for frequency divider 4406 circuitry, sampling detector 4502 can simplify the system in some cases. Since sampling detector 4502 samples the oscillator signal at some point in time using a clock edge or wideband impulse, sampling detector 4502 can generate synchronization with any integer multiple of the reference oscillator 3902 frequency. To use the sampling detector 4502 in the phase locked loop 4202 of the transmitter of FIG. 43, the oscillator is designed to operate at a narrow range of frequencies so that it does not lock to the wrong harmonic multiple.

  FIG. 46 is an exemplary block diagram of a sampling phase locked loop 4202 including a switching counter. In a system using many phase locked loops 4202, the accumulated power consumption becomes very large. The phase locked loop 4202 of FIG. 46 saves power by configuring two phase detectors and two feedback loops. First, the first signal path is enabled by closing the input switch 4610 and the output switch 4616. In the first signal path, the frequency divider 4406 feeds the first phase detector 4602, the first filter 4604 initializes the phase locked loop 4202, and the controlled oscillator 3502 is the correct harmonic of the reference oscillator 3902. Ensure synchronization with wave frequency. After the initial synchronization, the second phase detector 4606, which is the sampling detector 4502, is enabled via the error signal combiner 4614. A second phase detector 4606 is used to maintain synchronization and ensure frequency stability. When the second phase detector 4606 is enabled, the power of the first phase detector 4602 and the frequency divider 4406 decreases or switches to a low power mode.

  FIG. 47 is an exemplary block diagram of a system including a multiplex counter. Multiplexing the phase detector 4602 associated with one counter 4406 in several phase locked loops 4202 further reduces power and components in the system of FIG. At the time of initialization, the switch 1a and the switch 1b are closed, and the counter 4406, the phase detector 4602, and the filter 4604 are connected to the oscillator I3502-1. The resulting phase locked loop then operates to synchronize the oscillator 3502-1 to the selected harmonic of the reference oscillator 3902. Once synchronized, loop control is transferred to sampling phase detector 4606-1. In one embodiment, this transition is performed by deactivating the multiple phase detector 4602 and connecting it to the sampling phase detector 4606-1. In another embodiment, a combiner 4614-1 is used and the sampling phase detector 4606-1 is left connected from the beginning, but if the signals from the two phase detectors do not match, the multiple phase detector 4602 Multiple phase detector 4602 is more authoritative than sampling phase detector 4606-1 so that the output of sampling phase detector 4606-1 can be exceeded. The combiner comprises a register network or other component to ensure greater authority than the multiple phase detector 4602. Therefore, all that is necessary is to release the multiple phase detector 4606-1 using the switch 1 b and shift the control of the phase locked loop 4202.

  Once synchronized with the first oscillator 3502-1, the next oscillator 3502-2 and associated components 4606-2, 4608-2, 4614-2, switch 2a, switch 2b perform this process. Are repeated in order, and the division value N is changed to match each oscillator.

  As a fail-safe reliable measure, once all the oscillators are synchronized, the multiple phase detector 4602 is periodically connected to each oscillator 3502-1 and 3502-2 to verify the correct frequency and the frequency is incorrect. If so, resynchronize the oscillator. The correct frequency can be verified by a stable DC offset at the output of the multiple phase detector 4602. When there is a frequency error, the output of the phase detector 4602 becomes a complete AC signal.

  FIG. 48A is a Matlab code segment used for a simulation that integrates seven sine wave signals to generate a wideband pulse train. FIG. 48B is a graph showing a composite waveform generated using the seven sine wave signals of FIG. 48A. Referring to FIG. 48A, this figure shows that a sinusoidal signal for multiple harmonics can be synthesized to generate a wideband pulse waveform. Although only one modulo of the waveform is shown in the graph, since this waveform is repeated at the fundamental frequency, the repeated waveform becomes a broadband pulse train. Accordingly, a plurality of sinusoidal signals based on transmitters or receivers can be designed to transmit and receive pulse signals, thus enabling communication with a pulsed ultra-wideband system. Furthermore, other waveforms can be generated by changing the relative phase and amplitude of the sinusoidal signal.

  FIG. 49A is a Matlab code segment used in a simulation for accumulating seven sine wave signals to generate a band-limited pulse train. FIG. 49B is a graph showing a composite waveform generated using the seven sine wave signals of FIG. 49A. FIG. 49A shows an approximate simulation of the 15th to 22nd harmonic frequencies. A 3.1 GHz to 5.0 GHz band with the same relative bandwidth specified by the possible UWB bands. Accordingly, the band-limited pulse in the 3.1 to 5.0 band has a waveform similar to that shown in FIG. 49B.

  FIG. 50 is a block diagram illustrating a transmitter configured as shown in FIG. 43 to obtain two sine waves in a composite sine wave signal path. This data is modulated with the Barker 7 code.

  FIG. 51 is a block diagram illustrating a system that is equivalent or compatible with that shown in FIG. 50 but uses 14 sine wave generators 3906 to generate the encoded pulse train. The 14 carriers of FIG. 51 are each adjusted in phase and amplitude using a weighting function 5102 to produce a waveform sufficiently similar to that produced by the system of FIG. 50 that is compatible with the two systems. That is, the signal transmission by the system of FIG. 51 can be received by a receiver designed so that the signal transmission can be received by the system of FIG.

  FIG. 52A shows an exemplary protocol in which the transmitter sends a known pattern to synchronize the receiver, and then tunes the weights before sending the known pattern to send the header and data. FIG. 52B shows that the transmitter alternately sends a known pattern to synchronize the receiver, then tunes the weights before sending the known pattern to send the header and data, and repeats this pattern several times. An exemplary protocol is shown.

  53, 54, and 55A-E illustrate signal repetition in a multipath environment.

  FIG. 53 shows the received signal including many reflections from the impulse transmitter. A typical impulse receiver can only receive energy in the vicinity of one sample due to the bandwidth of the receiver. This is because only one lobe 5302 surrounds the sample time shown in FIG. However, the total energy 5304 available for reception is the sum of the energy under all signals. Therefore, an improved receiver can receive all energy.

  FIG. 54 is a plot of signal strength versus distance for a typical UWB system with noise. In a fairly typical environment, the total energy described in FIG. 53 decays by 1 / r squared 5402. It is basically a free space path loss. However, due to the scattering of energy and the fact that the total energy as shown in FIG. 53 cannot be received, in a scattering environment such as an indoor environment, the UWB sampling receiver has an energy of 1 / r with respect to the distance. It attenuates 5404 by the third power. The ratio of the two curves 5406 shows the signal gain achievable from an advanced receiver design.

  FIG. 55A shows an exemplary channel model for a UWB environment. Each arrow indicates the time delay, reflection amplitude and polarity in this environment. The simulation reception signal can be constructed by integrating the number of copies of the transmission waveform according to the delay, polarity, and amplitude of the channel model.

  The use of a narrowband element for receiving a wideband signal will be described with reference to FIGS. A wideband signal can also be received using a plurality of narrowband channels. FIG. 55A shows an impulse channel model. FIG. 55B shows a first integration interval relative to FIG. 55A, and FIG. 55C shows a narrowband pulse signal associated with the integration interval of FIG. 55B. FIG. 55D shows a second integration interval relative to FIG. 55A. FIG. 55E shows a broadband pulse signal associated with the integration interval of FIG. 55D.

  55B and 55C show the integration intervals associated with the narrowband pulses. A receiver based on narrowband pulses can in some cases integrate the energy of a significant portion of the total received signal that is highly reflective. With the wideband pulses and associated integration intervals shown in FIGS. 55D and 55E, only a small portion of the total received energy can be received. In fact, one arbitrary narrowband channel adds and cancels reflections and accumulates reflections. Typically, the final result adds more than offsets the energy. However, a set of parallel narrowband channels tuned to different frequencies can obtain most of the available signals.

  FIG. 56 is a block diagram of a multi-carrier wavelet, pulse, or coded wave train signal generator. The system of FIG. 56 can be used to generate a transmitter signal or a receiver template signal. The operating principle shown in FIGS. 49A, 49B, 50A, and 50B can be implemented using the system of FIG. Although four channels are shown, any number of channels can be used and the present invention is considered to include any practical number of channels. In general, this disclosure refers to a plurality of channels if not apparent from the context, and any number of channels may be applied in the present invention.

  Referring to FIG. 56, a reference oscillator 3902 drives a number of sine wave generators 3502, ie carrier generators 3502. Depending on the controller 5612, each may be separately phase modulated 5602, 5608 and / or separately amplitude modulated 5604, 5610. Next, the obtained carriers are integrated 3506 to generate a composite signal 5616 output. Also, the reference 3902 is divided 5606 into frequencies to generate a time interval 5614 that represents one or more modulo repetition cycles of the composite sinusoidal signal waveform 5616.

  FIG. 57 shows a block diagram of a transmitter 5700 that uses the system of FIG. Referring to FIG. 57, data is received and synthesized with code 5702. The resulting signal is used to modulate composite sine wave signal 5616. Next, a modulated composite sine wave signal 3912 is transmitted.

  FIG. 58 is a block diagram of a receiver using the system of FIG. Referring to FIG. 58, composite sine wave signal 5616 is modulated with code 5702. The resulting encoded composite sinusoidal signal is used as an interrelated receiver template. In the reception path, the signal is received by the antenna 3512, amplified 5802 if necessary, supplied to the mixer 5804, and multiplied by the encoded composite sine wave signal 5812. The mixer 5804 output is filtered 5806 and provided to a demodulator 5808 to detect data and generate a data signal 3510.

  FIG. 59 further shows details and features relating to the receiver of FIG. Referring to FIG. 59, reference oscillator 3902 drives comb generator 3904 and a number of phase locked loop oscillators 3502 are driven as sine wave generators 3502.

  In FIG. 59, the controller 5908 receives signal quality information 5906 in the form of signal level, noise level, and signal-to-noise ratio. The signal quality information 5906 is used to adjust the phase 5602 and amplitude 5604 of the signal from the sine wave generator 3502 to improve the correspondence match 5804, thus improving the received signal quality 5906.

  By receiving the signal and measuring the signal quality 5609, phase and amplitude adjustments 5602, 5604 can be made applicable. This is done by making incremental adjustments in one parameter such as phase 5602 or amplitude 5604 of one sinusoidal signal 3502 and measuring the change in signal quality 5609. If the quality 5609 improves, the incremental adjustment is further increased and the change in signal quality is measured again. Adjust the time-specific parameters (phase or amplitude from unadjusted carrier) and continue this process until the signal quality does not improve. Once all the parameters for the full sine wave generator 3502 have been adjusted, the process begins again.

  It is also possible to adjust the phase and amplitude in synchronization with a known transmitter signal. The transmitter can transmit one carrier in a known sequence time according to a well-known protocol to optimize the phase and gain parameters at high speed. Using this protocol, the data rate can be increased after first ensuring reliable communication over a partially optimal channel at a slow data rate and then establishing the optimal phase and amplitude parameters.

  The multi-carrier receiver shown in FIGS. 58 and / or 59 can receive signals generated by an impulse transmitter, another multi-carrier transmitter of FIG. 57, or another multi-carrier transmitter with a different number of carriers. Designed with the architecture of FIGS. 58 and 59 for the same reason that the systems of FIGS. 51 and 52 are configured to be compatible, the carrier is adjusted in phase and amplitude to match a wide range of waveforms. Compatibility with other systems can be obtained. The specific phase and amplitude can be fixed by design or determined as applicable by the receiver reception time.

(Receiver tracking)
The receivers of FIGS. 58 and 59 can track the received signal by filtering the correlator output 5804, feeding back the filtered correlator signal, and controlling the frequency of the reference oscillator 3902. In FIG. 59, the filtered correlator signal is further processed by a controller to generate a frequency control signal 5910. In addition, the phase of the reference oscillator 3902 can be controlled. Further, phase control 5602 of each harmonic can be controlled. To control the phase of each carrier phase control 5602 to keep the waveform 5812 constant, each phase control 5602 is shifted by the same amount of time. Therefore, each phase control is shifted by a different amount depending on the frequency. Higher frequencies shift at a higher rate per second than lower frequencies. The angular velocity of the shift is directly proportional to the frequency.

  A general description of the tracking system can be found in US Pat. No. 6,556,621, US serial numbers 10/356, 995, US serial numbers 09 / 711,026. The entire contents are incorporated herein by reference for multiple purposes.

  The present invention can include combined carrier modulation and channelization. Referring to FIG. 36, a system based on multiple sinusoidal signals further has the advantage of enabling novel techniques for modulation and channelization. A transmitted waveform can be generated between certain minimum and maximum frequencies with certain spectral characteristics including a flattened energy spectrum as characterized by the FCC mask.

  By combining various carriers having different frequencies, basically the same spectrum characteristics of the transmission waveform can be obtained. For example, the present invention can use 20 carriers at 100 MHz intervals (eg, 3.2 GHz, 3.3 GHz, 3.4 GHz, etc.). In the present invention, ten carriers can be used at 200 MHz intervals (for example, 3.2 GHz, 3.4 GHz, 3.6 GHz, etc.). In the present invention, four carriers can be used at intervals of 500 MHz (for example, 3.2 GHz, 3.7 GHz, 4.2 GHz, etc.).

  Channelization can be performed based on carrier characteristics that are combined to obtain an arbitrary transmit waveform as specified by one or more codes. Different codes, combinations of code subset codes, and multiple subsets of codes can be applied to each successive transmit waveform or group of consecutive transmit waveforms.

  The present invention can use a frequency selectable code. A channel can be defined by a combination of specific carrier frequencies synthesized into the transmit waveform (eg, N possible carrier frequencies used by the system, channel 1 is defined by carrier frequencies 1, 3, 7, 13, 18) , Channel 2 is defined by carriers 2, 7, 10, 15, 17, etc.).

The present invention can use a phase code. A channel can be defined by a combination of phase relationships between carriers synthesized in the transmit waveform. In particular, for a certain reference time, each carrier used may have a phase relationship defined with respect to other carriers. For example, at time t ₀ , the three carriers 100 MHz, 200 MHz, and 300 MHz may have a phase relationship of 0 °. After a certain period T, the three carriers may again have a 0 ° phase relationship. Two full cycles are performed on the first carrier, three full cycles are performed on the second carrier, and six full cycles are performed on the third carrier. Similarly, at time t ₀ , the second and / or third carrier may have an N ° phase relationship with the first carrier and / or each other. N can be 180 or some other number.

  The present invention can use an amplitude code. A channel can be defined by a combination of specific carrier amplitudes combined with the transmit waveform (eg, N possible carrier amplitudes used by the system, channel 1 is defined by carrier amplitudes 1, 3, 7, 13, 18 Channel 2 is defined by carrier amplitudes 1, 7, 10, 15, 17, etc.). An amplitude code can be used to define a reverse carrier and / or carrier to be blank (eg, zero amplitude).

  The present invention can use a combination of codes. Encoding can be used to define channels based on a combination of frequency, phase relationship, and amplitude. Three or more carriers of the same frequency but different phase relationship and / or amplitude can be defined.

  Channelization can be defined based on carrier characteristics combined to obtain a sequence of transmit waveforms, as specified by one or more codes. The combined carriers and carrier characteristics to generate individual transmit waveforms can also be based on one or more codes or can be the same for each transmit waveform.

  The present invention can use amplitude encoding. The sequence of N transmit waveforms can also be controlled by a code. This code defines the relative amplitude of each of the N transmit waveforms. An amplitude code can be used to define an inverse waveform and / or blanking the waveform (eg, zero amplitude). The sequence of N transmission waveforms can be controlled by a code, for example, by a Barker 11 code. This code defines whether each transmit waveform of the N transmit waveforms is reversed.

  The present invention can use time coding. The sequence of N transmit waveforms can also be controlled by code (ie, time hopping code). This code defines the relative timing of each transmission waveform of the N transmission waveforms.

  The transmit waveform code and transmit waveform sequence code identified above may include one or more fixed or non-fixed (ie, delta) value ranges and / or discrete value layouts, as described in some of the TDC encoded patent applications. It matches. Such codes may be algebraic codes (secondary congruential, hyperbolic congruential, etc.), pseudo-random codes or design codes such as Kumar codes (eg, Design II or III), or other Barker codes, Walsh It can be any one of many well-known codes such as a code, a scissor code, a gold code, and the like. This is used to obtain the corresponding correspondence and / or spectrum characteristics of any designed code.

  A code can be selected based on operating environment characteristics such as determined (eg, measured) multipath characteristics, interference characteristics, noise characteristics, and the like. For example, a code rake approach can be used. Similarly, one or more characteristics (eg, frequency, phase) of one or more combined carriers may be altered and transmitted by an interference rejection / avoidance approach to mitigate the effects of one or more interference signals. Generate a waveform.

  The code can be used to avoid certain carrier frequencies and / or phase relationships without determining operating environment characteristics (ie, real time). A code defining a waveform sequence can be used in combination with a delay code.

  By changing at least one carrier characteristic (eg, amplitude, phase, frequency), modulation can be performed on a single carrier basis. By filtering the received waveform and extracting each carrier, the data present on the carrier can be demodulated. Whether or not a carrier is present, it can be used to modulate data.

  By shifting the waveform to be transmitted with time (for example, waveform position modulation), modulation can be performed on a single transmission waveform basis. By modulating the transmitted waveform (eg, inverse vs. non-inverted or discrete vs. reference amplitude), modulation can be performed on a single transmit waveform basis. By determining the presence (ie, no waveform is present), modulation can be performed on a single transmitted waveform basis. By combining the above (for example, QFTM type modulation), modulation can be performed on the basis of one transmission waveform.

  By shifting the waveform transmission sequence with time (waveform sequence position modulation), modulation can be performed on a single transmission waveform sequence basis. By modulating the transmitted waveform sequence (e.g., reverse vs. non-reverse, or discrete vs. reference amplitude), modulation can be performed on a single transmit waveform sequence basis. By determining the presence (ie, there is no waveform sequence, such as no waveform sequence or a different waveform sequence), modulation can be performed on a single transmit waveform sequence basis. By combining the above (for example, QFTM type modulation), modulation can be performed based on one transmission waveform sequence.

  The present invention can be implemented in local area networks, personal area networks, routers, switches, nodes and / or devices such as laptop computers, personal digital assistants, video displays and the like. For example, a router may include a transmitter that implements the present invention and a receiver that implements the present invention. Both of these support the circuit.

  As will be appreciated by those skilled in the art who understand the advantages of the description of the invention, the apparatus and method according to the invention are flexible and can be implemented in a wide range. A wide variety of semiconductor materials and technologies can be used to design, implement and manufacture communication devices and systems according to the present invention. For example, if desired, silicon, thin film technology, silicon on insulator (SOI), silicon germanium (SiGe), gallium arsenide (GaAs) can be used.

  If desired, an n-type metal oxide semiconductor (NMOS), a p-type metal oxide semiconductor (PMOS), a complementary metal oxide semiconductor (CMOS), a bipolar junction transistor (BJT), a combination of BJT and a CMOS circuit Such systems and devices can be implemented using (BiCMOS), heterojunction transistors, and the like. As is well known to those skilled in the art who understand the advantages of the description of the present invention, the choice of semiconductor material, technology, and design method depends on the design and performance specifications of the particular application.

  It should be noted that the communication system and apparatus according to the present invention can be manufactured with high yield, high reliability, and low cost by utilizing standard semiconductor devices and manufacturing techniques. For example, such a system and device can be fabricated using a standard mixed signal CMOS process. This flexibility enables the manufacture and sale of consumer high-speed data communication products, professional products, health care products, industrial supplies, scientific equipment, military equipment, etc. using the communication system and apparatus of the present invention.

  Although the above description of the communication system and apparatus relates to wireless communication, the disclosed inventive concepts may be used in other situations, as is well known to those skilled in the art who understand the advantages of the description of the invention. You can also. For example, if desired, communication systems and devices using high-speed data communication land lines (ie, cables, optical fibers, indoor wiring, coaxial lines, parallel two lines, telephone lines, cable TV lines, etc.) can be implemented.

  In other words, the transmitter and receiver can be connected together via a transmission line, such as a wire line or optical fiber, by striking the antenna (and any associated circuitry). Such a system provides the same or similar advantages as a wireless counterpart. That is, a UWB signal can coexist with other signals on the same transmission medium.

The spectrums (for example, FIGS. 13 to 16) shown in various figures represent transmission and emission spectra. Since radio wave propagation in free space does not exhibit frequency dependence, the electric field strength PSD of the receiver is the same as the transmission PSD, and the signal is attenuated by 1 / (4πr ² ). As described above, when a signal is received by a constant aperture type antenna, an example of a constant aperture antenna in which the received spectrum is the same as the transmission spectrum is a broadband horn or a broadband parabola whose gain increases with the square of the frequency. is there.

On the other hand, when a signal is received by a constant gain type antenna, the received spectrum has a load 1 / f ² characteristic. An example of a suitable constant gain antenna is a wide dipole whose gain is essentially flat with frequency. In a non-free space environment, it exhibits frequency dependence to some extent. However, these effects are basically the same whether or not a constant aperture or constant gain antenna is used.

  Referring to the figure, the various blocks shown (eg, FIG. 3 or FIG. 5) primarily illustrate conceptual functions and signal flow. In order to implement an actual circuit, it may or may not include separately identifiable hardware of various functional blocks. For example, if desired, the functionality of the various blocks can be combined into one circuit block. Also, if desired, the functionality of one block can be implemented in several circuit blocks. The choice in implementing the circuit will depend on various factors such as the specific design and performance specifications for performing any implementation, as will be appreciated by those skilled in the art having read the present disclosure.

  In addition to those described herein, other variations and modifications of the present invention will become apparent to those skilled in the art who understand the advantages of the description of the invention. Accordingly, this description teaches those skilled in the art how to practice the present invention and should be construed as illustrative only.

  The forms of the invention shown and described are to be understood as the presently preferred embodiments. Without departing from the scope of the invention described in this document, those skilled in the art can make various changes in shape, size and configuration. For example, those skilled in the art can substitute equivalent elements for those shown and described herein. Also, those skilled in the art who understand the advantages of the description of the invention can use certain features of the invention separately from other features without departing from the scope of the invention.

The accompanying drawings are merely illustrative of exemplary embodiments of the invention and are not to be construed as limiting the scope of the invention. The novel idea of the disclosure is suitable for other equally effective embodiments. In the drawings, the same reference numerals in two or more drawings indicate the same or equivalent functions, components, or blocks.
FIG. 1 shows several power spectral density (PSD) profiles in various embodiments according to the present invention. FIG. 2 shows exemplary signal waveforms corresponding to a high speed data communication UWB device. FIG. 3 illustrates an exemplary embodiment of a high speed data communication UWB transmitter according to the present invention. FIG. 4 shows exemplary waveforms corresponding to a high speed data communication UWB transmitter according to the present invention. FIG. 5 illustrates an exemplary embodiment of a high speed data communication UWB receiver according to the present invention. FIG. 6 shows exemplary waveforms corresponding to a high speed data communication UWB receiver according to the present invention. FIG. 7 shows the timing relationship of various signals in the high-speed data communication UWB transmitter according to the present invention. FIG. 8 illustrates an exemplary desired or predetermined PSD profile corresponding to two modes of operation for illustrative purposes according to the present invention. FIG. 9 shows a PSD profile for an exemplary embodiment of the invention using higher order harmonics. FIG. 10 shows an illustrative PSD profile in an exemplary embodiment according to the present invention. FIG. 11A shows an exemplary output signal for one cycle of a transmitter in a UWB communication device according to the present invention. FIG. 11B shows another example output signal of one cycle of the transmitter in the UWB communication device according to the present invention. FIG. 12 illustrates the timing relationship between several signals in an exemplary embodiment according to the present invention. FIG. 13 shows several PSD profiles of an illustrative embodiment according to the present invention. FIG. 14 shows some PSD profiles of another exemplary embodiment according to the present invention. FIG. 15 shows a PSD profile of another embodiment according to the present invention for illustrative purposes. FIG. 16 shows a PSD profile of another exemplary embodiment of a communication system or apparatus according to the present invention. FIG. 17 shows an exemplary embodiment according to the present invention of a communication system incorporating mode switching. FIG. 18 shows an illustrative chipping sequence used in the communication system and apparatus according to the present invention. FIG. 19 shows an exemplary embodiment 19 of a differential receiver according to the present invention. FIG. 20 shows a set of offset quadrature phase shift modulation (OQPSK) UWB signals in an exemplary embodiment according to the present invention. FIG. 21 shows a set of chipping signal waveforms in an exemplary embodiment according to the present invention. FIG. 22 shows an exemplary embodiment of a transmitter according to the present invention using separately modulated harmonic signals. FIG. 23 shows an exemplary embodiment of a receiver according to the invention for receiving separately modulated harmonic signals. FIG. 24 shows a sample waveform in the illustrative embodiment according to the present invention. FIG. 25 shows a Fourier transform of the signal shown in FIG. FIG. 26 shows a sample waveform in an exemplary embodiment of a transmitter according to the present invention. FIG. 27 shows an exemplary in-phase channel pulse as a function of time in an illustrative embodiment according to the present invention. FIG. 28 shows the magnitude of the signal spectrum of FIG. FIG. 29 shows an exemplary quadrature channel pulse as a function of time in an illustrative embodiment according to the present invention. FIG. 30 shows the magnitude of the signal spectrum of FIG. FIG. 31 shows two signals as a function of time in an illustrative embodiment according to the invention. FIG. 32 shows the spectrum obtained from using the signal shaping shown in FIG. FIG. 33 shows two signals as a function of time in another illustrative embodiment according to the present invention. FIG. 34 shows the spectrum obtained from using the signal shaping shown in FIG. FIG. 35 is a schematic diagram showing a transmitter representing an embodiment of the present invention. FIG. 36 shows a spectrum of a signal generated by the transmitter representing the embodiment of the present invention. FIG. 37 is a schematic diagram showing a receiver representing an embodiment of the present invention. FIG. 38 shows a high-level schematic diagram of a receiver that represents an embodiment of the present invention. FIG. 39 is a schematic diagram showing a transmitter representing an embodiment of the present invention. FIG. 40 is a diagram showing a transmitter representing an embodiment of the present invention. FIG. 41 is a diagram showing a transmitter representing an embodiment of the present invention. FIG. 42 is a schematic diagram showing a transmitter representing an embodiment of the present invention. FIG. 43 is a schematic diagram showing a transmitter representing an embodiment of the present invention. FIG. 44 is a diagram showing a phase locked loop representing the embodiment of the present invention. FIG. 45 is a diagram showing a phase-locked loop representing the embodiment of the present invention. FIG. 46 is a diagram showing a sampling phase-locked loop representing the embodiment of the present invention. FIG. 47 is a schematic diagram showing a system including a multiplex counter representing the embodiment of the present invention. FIG. 48A shows a Matlab code segment using an integration simulation of seven sine wave signals to generate a wideband pulse train * representing an embodiment of the present invention. FIG. 48B shows a graph representing a composite waveform generated using the seven sine wave signals of FIG. 48A. FIG. 49A shows a Matlab code segment using an integration simulation of seven sine wave signals to generate a band limited pulse train representing an embodiment of the present invention. FIG. 49B is a graph showing a composite waveform generated using the seven sine wave signals of FIG. 49A. FIG. 50 is a schematic diagram showing a transmitter representing an embodiment of the present invention. FIG. 51 is a schematic diagram showing a transmitter representing an embodiment of the present invention. FIG. 52A shows an exemplary protocol representing an embodiment of the present invention. FIG. 52B shows an exemplary protocol representing an embodiment of the present invention. FIG. 53 shows a received signal from an impulse transmitter that includes many reflections, representing an embodiment of the present invention. FIG. 54 is a plot showing signal strength versus distance for a noisy UWB system that represents an embodiment of the present invention. FIG. 55A shows an impulse channel model representing an embodiment of the present invention. FIG. 55B shows a first integration interval relative to FIG. 55A. FIG. 55C shows a narrowband pulse signal associated with the integration interval of FIG. 55B. FIG. 55D shows a second integration interval relative to FIG. 55A. FIG. 55E shows a broadband pulse signal associated with the integration interval of FIG. 55D. FIG. 56 is a schematic diagram showing a multicarrier wavelet pulse or an encoded wave train signal generator representing the embodiment of the present invention. FIG. 57 is a schematic diagram illustrating a transmitter using the generator of FIG. FIG. 58 is a schematic diagram showing a receiver representing an embodiment of the present invention. FIG. 59 is a schematic diagram showing a receiver representing an embodiment of the present invention.

Claims (17)

Below we generate several generally continuous waveforms
Accumulate multiple continuous waveforms,
Modulate multiple accumulated waveforms to generate a spread spectrum signal,
A method of transmitting a spread spectrum signal, including transmitting a spread spectrum signal. The method of claim 1, further comprising synchronizing each of the plurality of generally sustained waveforms to a common reference using one of the plurality of phase locked loops before generation and after integration. The method of claim 1, wherein modulating the accumulated plurality of generally sustained waveforms includes modulating the accumulated plurality of generally sustained waveforms with the encoded data to generate a spread spectrum signal. The method of claim 1, wherein the frequency spread spectrum signal comprises an ultra wideband signal. A comb generator with a spread spectrum transmitter,
An integration node connected to the comb generator;
A modulator connected to the integrating node;
Spread spectrum signal output connected to the modulator,
A data encoder connected to the modulator;
Data input connected to the data encoder,
A code source connected to the data encoder,
A device comprising a clock connected to a data encoder and a comb generator. 6. The apparatus of claim 5, further comprising a plurality of harmonic related phase locked loop circuits connected to the integrating node, each harmonic related phase locked loop circuit connected to a comb generator. An ultra-wideband router comprising the device according to claim 5. Below we generate several generally continuous waveforms
Accumulate multiple continuous waveforms,
Modulate multiple accumulated continuous waveforms with code,
Mixing multiple modulated and generally continuous waveforms with a spread spectrum signal to generate a baseband signal,
Integrate the baseband signal
A method of receiving a spread spectrum signal, including detecting data. 9. The method of claim 8, further comprising synchronizing each of the plurality of generally sustained waveforms to a common reference using one of the plurality of phase locked loops before generation and after integration. 9. The method of claim 8, further comprising adaptably adjusting the phase of each of the plurality of generally continuous waveforms, wherein the plurality of generally continuous waveforms are associated as harmonics. The method of claim 10, further comprising adaptably adjusting an amplitude of each of the plurality of generally continuous waveforms, wherein the plurality of generally continuous waveforms are associated as harmonics. 9. The method of claim 8, further comprising adaptably adjusting the amplitude of each of the plurality of generally continuous waveforms, wherein the plurality of generally continuous waveforms are associated as harmonics. The method of claim 8, wherein the frequency spread spectrum signal comprises an ultra wideband signal. Below the comb generator,
An integration node connected to the comb generator;
A modulator connected to the integrating node;
A code source circuit connected to the modulator;
A clock connected to the code source circuit and the comb generator;
A multiplier connected to the modulator;
Spread spectrum signal input connected to the multiplier,
An integrating circuit connected to the multiplier;
An apparatus comprising a spread spectrum signal receiver with a data output connected to an integrating circuit. 15. The apparatus of claim 14, further comprising a plurality of harmonic related phase locked loop circuits connected to the integrating node, each harmonic related phase locked loop circuit connected to a comb generator. The apparatus of claim 14, wherein the multiplier includes a mixer. An ultra-wideband router comprising the device according to claim 14.

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