JP2006246323A - Direct conversion receiver - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a direct conversion receiver capable of canceling disturbing waves with a relatively small circuit scale, without using a frequency synthesizer. <P>SOLUTION: Cancel signal generating sections 21 and 24 extract disturbing waves component from the demodulation output of an orthogonal demodulator 15 for demodulating a received signal and generate opposite-phase outputs. The outputs are added as cancel signals to the demodulation output. The frequencies and amplitudes of the cancel signals of the cancel signal generation sections 21 and 24 are controlled by a control section 30. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a direct conversion receiver used in a wireless communication device such as a mobile phone terminal.

  In recent years, it has become common to use a direct conversion system that can realize a receiver with a small number of parts as compared with a conventional superheterodyne system or the like for a receiver of a mobile phone terminal.

  FIG. 9 shows a general block configuration of a direct conversion receiver of a system that performs transmission and reception at the same time, such as a CDMA system. A signal received by the antenna 11 is amplified by a low noise amplifier (LNA) 13 via a duplexer 12, and a frequency outside the reception band is suppressed by an RF band pass filter 14. The frequency is directly converted to the orthogonal I-phase and Q-phase components of the frequency band. From the I-phase and Q-phase components after conversion to baseband, desired channel signals are selected by low-pass filters 32 and 33, respectively, and A / D conversion at the subsequent stage is performed by variable gain amplifiers 34 and 35, respectively. Amplified or attenuated to a signal level appropriate for the instrument (not shown). The low-pass filters 32 and 33 are normally configured by active filters.

  On the other hand, mobile phone terminals are provided with specifications regarding resistance to interference waves. Many of the test conditions of these specifications are very severe. The desired wave is set to a very weak level close to the reception sensitivity point, while the disturbing wave is set to a level several tens dB higher than the desired wave. When the frequency of the jamming wave is far from the desired wave, the jamming wave is suppressed by a duplexer or an RF bandpass filter, so that it does not become a big problem. However, under the condition that the interference wave is close to the desired wave, it cannot be expected to be suppressed by the duplexer or the RF band pass filter. Therefore, the interference wave is amplified to a very large level by the LNA and the quadrature demodulator before reaching the input terminals of the low-pass filters 32 and 33 in the baseband portion, and saturates the circuit of the low-pass filter. . The saturated circuit can no longer exhibit the desired transfer function, the interference wave is not sufficiently suppressed, and the subsequent variable gain amplifiers 34 and 35 are saturated. As a result, the bit error rate is degraded.

In order to avoid the above problems, the technique described in Patent Document 1 employs the following method. FIG. 10 shows a direct conversion receiver disclosed in Patent Document 1. In FIG. 10, the same components as those shown in FIG. Of the signal frequency-converted to the baseband frequency band, only the interference wave is detected by the detection means including the high-pass filter 41 and the detection circuit 42, and the control unit 40 determines the synthesizer 44 and the variable attenuation based on the detection signal. The interfering wave canceling signal is generated by controlling the unit 45, and the canceling signal is combined with the received signal in the RF band, so that only the interfering wave is reduced. This operation is performed autonomously.
JP 2001-102942 A

  However, the use of the frequency synthesizer as the cancellation signal generating means has the following disadvantages. A frequency synthesizer is composed of a voltage-controlled oscillator (VCO), frequency divider, phase comparator, charge pump, etc., and its circuit scale is large. hard. In particular, an on-chip inductor used for a VCO occupies a large area. Further, in the situation where the interference wave is close to the desired wave, there is a concern that the cancellation signal generating VCO and the desired signal demodulating VCO may be combined to cause a serious influence on reception characteristics such as phase noise degradation. Is done. Furthermore, the frequency steps that can be locked by a frequency synthesizer configured using a phase-locked loop (PLL) are limited by discrete comparison frequencies. In order to cancel the interfering wave with certainty, it is important that the deviation between the frequency of the interfering wave and the frequency of the canceling signal is small. Therefore, the method described in Patent Document 1 may not cancel the interfering wave well. .

  The present invention has been made in such a background, and an object thereof is to provide a direct conversion receiver capable of satisfactorily canceling an interference wave with a relatively small circuit scale without using a frequency synthesizer. is there.

  A direct conversion receiver according to the present invention includes quadrature demodulation means for demodulating a received signal, and cancellation signal generation means for extracting an interference wave component from the demodulated output of the quadrature demodulation means and outputting it in reverse phase. The cancellation signal output from the means is added to the demodulated output to cancel the interference wave component in the received signal. That is, the interference wave component is extracted from the reception signal down-converted to the baseband frequency band, and added to the original signal in the opposite phase, thereby canceling the interference wave component in the reception signal.

  The cancellation signal can be added by providing a buffer for receiving the demodulated output, and adding the output of the buffer and the cancellation signal with an adder. Alternatively, a low-pass filter composed of a plurality of stages of operation units for selectively outputting a desired channel signal from the demodulated output may be provided, and the first-stage operation unit of the low-pass filter may be used.

  The cancellation signal generating means includes, for example, a fourth-order high-pass filter, and further includes control means for generating a cutoff frequency control signal for controlling the cutoff frequency so that the cutoff frequency becomes equal to the interference wave frequency.

  The cancellation signal generating means further includes a variable gain amplifier that adjusts the amplitude of the output of the fourth-order high-pass filter, and the control means is configured so that the amplitude of the cancellation signal is equal to the signal amplitude of the cutoff frequency. Preferably, a gain control signal for controlling the gain of the variable gain amplifier is generated. The high-pass filter can be composed of, for example, a MOSFET-C filter that can adjust channel resistance by changing the gate bias of a MOS transistor.

  According to the present invention, in the direct conversion receiver, the interference wave component is extracted from the received signal and added to the original signal in the opposite phase, so that the interference wave component in the received signal is canceled out. It is possible to obtain a receiver having excellent resistance to interference waves under conditions where the received power of waves is weak.

  Further, since it is not necessary to up-convert the signal that cancels the interference wave to the RF frequency, a circuit that occupies a large area such as a VCO is not required. Therefore, it can be realized with a relatively small circuit scale. Further, it is possible to avoid an adverse effect on reception characteristics due to having an extra VCO.

  Furthermore, since the cancellation signal generated by the cancellation signal generation means is locked to the frequency of the interference wave, it is possible to cancel the interference wave with certainty.

  Hereinafter, preferred embodiments of the present invention will be described in detail.

  FIG. 1 is a block diagram showing a configuration of a direct conversion receiver according to an embodiment of the present invention. A signal received by the antenna 11 is amplified by a low noise amplifier (LNA) 13 via a duplexer 12, and after an RF band pass filter 14 suppresses a frequency outside the reception band, an orthogonal demodulator (orthogonal demodulation means) 15, the frequency is directly converted (down-converted) into the I-phase and Q-phase components of the baseband frequency band. The quadrature demodulator 15 has a pair of mixers. Both mixers receive the oscillator output directly from the local oscillator 17 and through the 90 ° phase shifter 16, respectively. The cancellation signal generation unit (cancellation signal generation means) 21 extracts a disturbing wave component from a part of the signal converted to baseband by a high-pass filter (described later), changes its amplitude and phase, The original signal through 22 and 23 is combined. At this time, if the amplitude of the cancellation signal is the same as that of the original signal, and the phase of the interference wave is opposite, the interference wave is canceled and the interference wave level input to the low-pass filters 32 and 33 at the subsequent stage Is reduced. Therefore, the circuit blocks after the low-pass filters 32 and 33 can operate normally, and good reception characteristics can be obtained. The detection unit (detection unit) 31 detects the phase and level of the combined interference wave and sends it to the control unit (control unit) 30. The control unit 30 controls the phases and amplitudes of the cancellation signal generation units 21 and 24 so that the combined interference wave level is minimized.

  In an actual implementation, the buffers 22 and 23 in FIG. 1 may be unnecessary. This can be realized, for example, by synthesizing the interference canceling signal in the first stage circuit block of the low-pass filter.

  An example of a method for inputting a cancellation signal to the low-pass filter first-stage amplifier circuit block will be described. FIG. 2 is an example in which the low-pass filter 32 (33) is configured by an RC filter using operational amplifiers (operation units) OP1 to OP5. The output of the first stage operational amplifier OP1 is supplied to the detection unit 31. FIG. 3 shows a more specific circuit configuration example of the first-stage operational amplifier OP1 among the operational amplifiers OP1 to OP5 constituting the low-pass filter 32 (33). The input terminal pair (Comp-Comp_X) in the circuit of FIG. 3 is a terminal pair that inputs a cancellation signal from the cancellation signal generation unit 24, and an alternating current corresponding to the cancellation signal is input from the input terminal pair IN-INX. Subtract from signal.

  FIG. 4 shows a configuration example of the cancellation signal generation unit 21 (24). The cancellation signal generation unit 21 (24) includes a fourth-order high-pass filter 211 and a variable gain amplifier 212. The high-pass filter 211 of this example has a fourth order Butterworth characteristic. The fourth-order Butterworth filter has a feature that the phase rotates 180 degrees at the cutoff frequency. The cutoff frequency of the high-pass filter 211 is controlled to be equal to the interference wave frequency by the cutoff frequency control signal Fc. The gain of the variable gain amplifier 212 is controlled by the gain control signal Gc from the control unit 30 so that the amplitude of the cancellation signal becomes equal to the signal amplitude of the cutoff frequency. With such a configuration, the output signal of the cancellation signal generation unit 21 (24) can obtain a signal having the same amplitude and an opposite phase as the original signal.

  Note that the fourth-order high-pass filter does not necessarily have Butterworth characteristics. A transfer function such as Chebyshev also has a characteristic that the phase rotation is 180 degrees near the cutoff frequency. If the control unit 30 is configured to control the cutoff frequency of the high-pass filter so that the combined interference wave level is minimized, the cutoff frequency of the high-pass filter and the frequency of phase rotation = 180 degrees And do not necessarily match. The gain of the variable gain amplifier is controlled by a gain control signal Gc from the control unit 30.

  A method for controlling the cutoff frequency of the high-pass filter 211 of the cancellation signal generation unit 21 will be described with reference to FIG. The same applies to the cancellation signal generation unit 24. In this example, the high-pass filter 211 is composed of a MOTFET-C filter, and the channel resistance of the MOS transistor can be changed by controlling the gate bias of the MOS transistor operating in the triode region. The frequency adjustment capital 300 is configured as a so-called DLL (Delay Locked Loop) circuit as a part of the control unit 30, and passes the phase of the output signal from the reference filter 301 having a function as a delay circuit and the reference filter 301. It is configured to lock when the phase of the signal that is not shifted is shifted by a predetermined value (for example, 90 degrees). An output signal from the reference filter 301 and a signal that does not pass through the reference filter 301 are multiplied in the multiplier 302, and the multiplier 302 outputs a signal composed of a signal component twice the interference wave and a DC component, This is supplied to the loop filter 303. The loop filter 303 extracts only the DC component and supplies this to the charge pump 304 as a control signal. Then, until the phase difference between the output signal of the reference filter and the signal that does not pass through the reference filter reaches a predetermined value (for example, 90 degrees), the voltage is charged in the charge pump 304, and the reference filter 301 is To generate a gate voltage at a target level. This gate voltage is also supplied as a cut-off frequency control signal Fc to the high-pass filter 211 that is the adjusted filter, and the cut-off frequency of the high-pass filter 211 is controlled.

  When the method shown here is used, since the phase is controlled autonomously, the detection unit 31 shown in FIG. 1 does not need to detect the phase, which is useful for reducing the circuit scale. Of course, the phase may be detected by a detection unit such as the detection unit 31 and the control unit may generate a control signal using a control program.

  In the case of the example shown here, the cutoff frequency of the high-pass filter is locked to a frequency where a strong interference wave exists, but if there is no interference wave, it sticks to the upper limit or the lower limit of the cutoff frequency adjustment range. . The adjustment range of the cut-off frequency of the high-pass filter 211 needs to be designed so as not to affect the desired signal band even when the cut-off frequency sticks to the lower limit of the adjustment range. In the case of the W-CDMA system, since the bandwidth of the desired channel baseband signal is 1.92 MHz, the circuit constant can be selected so that the lower limit of the cutoff frequency adjustment range of the high-pass filter is about 8 MHz, for example. desirable.

  Next, a gain control method of the variable gain amplifier 212 of the cancellation signal generation unit 21 will be described with reference to FIG. The same applies to the cancellation signal generation unit 24. The detection unit 31 includes a high-pass filter 311 and a detector 312. The high pass filter 311 plays a role of removing a desired signal from the combined signal. The detector 312 obtains a detection signal by a technique of detecting an AC signal using, for example, a diode and smoothing it with a low-pass filter. This detection signal is negatively fed back as a gain control signal Gc to the variable gain amplifier 212 of the cancellation signal generation unit 21 via the feedback circuit 306, thereby forming an automatic gain control loop. The feedback circuit 306 constitutes a part of the control unit 30. The operation of the automatic gain control loop converges so that the interference wave is most suppressed.

  Although an analog automatic gain control method is shown here, a configuration in which gain control is digitally performed using an A / D converter, a memory, a processor, or the like may be employed.

  FIG. 7 shows the amplitude frequency characteristic at the output terminal of the first stage operational amplifier OP1 of the low-pass filter 32 (33). The frequency characteristic in which the frequency of the interference wave (in this example, 11.3 MHz) is suppressed is exhibited, and the interference wave level input to the amplifier at the subsequent stage is reduced. In the example of FIG. 7, 17.6 dB is suppressed at 11.3 MHz, which corresponds to about 1/8 in terms of amplitude.

  FIG. 8 shows the amplitude frequency characteristics at the output of the low-pass filter 32 (33). Similar to FIG. 7, the frequency characteristic in which the frequency in which the interference wave exists is suppressed is exhibited. At first glance, the frequency characteristic of FIG. 8 seems to be the same as the transfer function in which the order of the low-pass filter is increased and the zero point is arranged near 11 MHz. However, when a similar transfer function is realized by a low-pass filter, when attention is paid to the frequency response at the output of each operational amplifier constituting the filter, the notch near the interference wave frequency is a relatively second half of the filter circuit (for example, the fourth stage). And thereafter). This means that the operational amplifier in the first half of the filter (for example, the first stage to the third stage) must handle a large level of disturbing waves, and as a result, the filter circuit cannot operate normally. In the present embodiment, since the frequency response as shown in FIG. 7 is realized in the first stage of the filter, the entire circuit of the low-pass filter can operate normally.

  The preferred embodiments of the present invention have been described above, but various modifications and changes other than those mentioned above can be made.

It is a block diagram which shows the structure of the direct conversion receiver which concerns on embodiment of this invention. It is a circuit diagram which shows the example which comprised the low-pass filter shown in FIG. 1 with RC filter using operational amplifier OP1-OP5. FIG. 3 is a circuit diagram illustrating a more specific configuration example of an operational amplifier OP1 at the first stage of the low-pass filter illustrated in FIG. It is a block diagram which shows the structural example of the cancellation signal production | generation means shown in FIG. FIG. 5 is a diagram illustrating a circuit configuration for controlling a cutoff frequency of a high-pass filter of the cancellation signal generation unit illustrated in FIG. 4. FIG. 5 is a diagram illustrating a circuit configuration that performs gain control of a variable gain amplifier of the cancellation signal generation unit illustrated in FIG. 4. It is a graph which shows the amplitude frequency characteristic in the output terminal of the first stage operational amplifier of the low-pass filter shown in FIG. It is a graph which shows the amplitude frequency characteristic in the output of the low-pass filter shown in FIG. It is a block diagram which shows the general structure of the direct conversion receiver of the system which performs transmission / reception simultaneously, such as a CDMA system. It is a figure which shows the direct conversion receiver of patent document 1. FIG.

Explanation of symbols

DESCRIPTION OF SYMBOLS 11 ... Antenna, 12 ... Duplexer, 14 ... Band pass filter, 15 ... Quadrature demodulator, 16 ... Phase shifter, 17 ... Local oscillator, 21 ... Cancellation signal generation part, 22, 23 ... Buffer, 24 ... Cancellation signal generation part , 30: control unit, 31: detection unit, 32, 33 ... low-pass filter, 34, 35 ... variable gain amplifier, 40 ... control unit, 41 ... high-pass filter, 42 ... detection circuit, 44 ... synthesizer, 45 DESCRIPTION OF SYMBOLS ... Variable attenuator, 211 ... High pass filter, 212 ... Variable gain amplifier, 300 ... Frequency adjustment capital, 301 ... Reference filter, 302 ... Multiplier, 303 ... Loop filter, 304 ... Charge pump, 306 ... Feedback circuit, 311 ... high-pass filter, 312 ... detector, Fc ... cutoff frequency control signal, Gc ... gain control signal

Claims (7)

  A direct conversion receiver that directly converts the received signal in the RF frequency band to the baseband frequency band, extracts the interference wave component from the received signal down-converted to the baseband frequency band, and reverses the phase to the original signal. A direct conversion receiver that cancels out interference wave components in a received signal by adding. A direct conversion receiver that directly converts a received signal in an RF frequency band to a baseband frequency band,
Orthogonal demodulation means for demodulating the received signal;
A cancellation signal generating means for extracting an interference wave component from the demodulated output of the orthogonal demodulating means and outputting it in an opposite phase;
A direct conversion receiver characterized in that an interference signal component in a received signal is canceled by adding the cancellation signal output from the cancellation signal generating means to the demodulated output.   3. The direct conversion receiver according to claim 2, further comprising: a buffer that receives the demodulated output; and an adder that adds the output of the buffer and the cancellation signal.   A low-pass filter including a plurality of stages of operation units for selectively outputting a desired channel signal from the demodulated output is provided, and the cancellation signal is added in an operation unit of the first stage of the low-pass filter. Item 3. The receiver according to Item 2.   The cancellation signal generating means includes a fourth-order high-pass filter, and further includes control means for generating a cutoff frequency control signal for controlling the cutoff frequency so that the cutoff frequency is equal to the interference wave frequency. The direct conversion receiver according to claim 2.   The cancellation signal generation means includes a variable gain amplifier that adjusts the amplitude of the output of the fourth-order high-pass filter, and the control means adjusts the variable so that the amplitude of the cancellation signal is equal to the signal amplitude of the cutoff frequency. 6. The direct conversion receiver according to claim 5, wherein a gain control signal for controlling a gain of the gain amplifier is generated.   7. The direct conversion receiver according to claim 5, wherein the high-pass filter includes a MOSFET-C filter capable of adjusting a channel resistance by changing a gate bias of a MOS transistor.

Images