JP2005354788A - Inverter control device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To obtain an inverter control device that can be reduced in the size of a backup charging circuit, is less in circuit loss and can inexpensively be provided. <P>SOLUTION: The inverter control device comprises: a converter 4 that converts an alternating current to a direct current; an inverter 8 that converts a direct current to an alternating current; a plurality of main circuit capacitors 6, 7 connected in series between DC bus bars that connect the converter and the inverter; and a plurality of power supply devices 11, 12 connected between each terminal of the main circuit capacitor, respectively. The power supply device has a total output voltage that exceeds a voltage of the converter, and an output limit characteristic equal to that of the other power supply device, and is provided with a switching means 50 that switches output limit levels. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to an inverter control device, and more particularly to an improvement of an inverter control device in which capacitors of a DC smoothing circuit are connected in series.

A switch that controls the turning on of power between the converter's AC input terminal and AC power supply by connecting a capacitor to the DC part of the converter that converts AC to DC and the inverter device that outputs AC power of arbitrary waveform from the obtained DC In order to prevent an inrush current by charging a capacitor with a switching power supply in advance before turning on the switch so that an excessive current flows through the contact and converter of the switch when the switch is turned on and does not cause a failure. A method has been proposed (see, for example, Patent Document 1).
In addition, when the capacitor is composed of a series connection of capacitors and a DC voltage is applied to both ends of the capacitor, the shared voltage of each capacitor becomes unbalanced and exceeds the rated voltage due to the deviation of the leakage current of each capacitor. Can cause damage to the capacitor. On the other hand, a circuit has been proposed in which the influence of leakage current is corrected by a transistor and the voltage between terminals of each capacitor is kept uniform (see, for example, Patent Document 2).

JP 2001-261245 A (first page, FIG. 1) Japanese Patent Laid-Open No. 10-295081 (first page, FIG. 1)

The inverter device that precharges the capacitor with the switching power supply described in Patent Document 1 has advantages in that it is more energy efficient and can be reduced in size as compared with a conventional charging method in which a resistor and an electromagnetic switch are combined. However, when combined with an inverter device that uses an external resistor connected in parallel for leakage current correction of the capacitor series body, a large amount of power is consumed by the external resistor at all times, so it is necessary to expand the capacity of the switching power supply to maintain the charging speed At the same time, there is a problem that the energy efficiency improved by the switching power supply becomes less effective due to the influence of the external resistance.
Further, in the example illustrated in Patent Document 2, from the viewpoint of the availability and reliability of parts, generally, as shown in FIG. A small balance resistor is connected to the offset to correct the deviation caused by the leakage current.

  The present invention has been made to solve the above-described problems of the prior art, and it is possible to reduce the size of the precharge circuit, to reduce the circuit loss, and to obtain an inverter control device that can be provided at low cost. It is the purpose.

  An inverter control device according to the present invention includes a converter that converts alternating current to direct current, an inverter that converts direct current to alternating current, a plurality of main circuit capacitors connected in series between the converter and the direct current bus connecting the inverter, In the inverter control device comprising a plurality of power supply devices connected between the terminals of the main circuit capacitor, the power supply device has a total output voltage exceeding the voltage of the converter and has an output limiting characteristic equivalent to each other. And switching means for switching the output restriction level.

  In the present invention, as a plurality of power supply devices respectively connected between the terminals of the main circuit capacitor, the total output voltage exceeds the voltage of the converter, has the same output limiting characteristics, and has an output limiting level. By providing switching means for switching, it is possible to reduce the size of the precharge circuit, reduce the circuit loss, and obtain an inverter control device that can be manufactured at low cost.

Embodiment 1 FIG.
1 to 5 illustrate an inverter control apparatus according to Embodiment 1 of the present invention. FIG. 1 is an overall configuration diagram of the inverter control apparatus, and FIG. 2 is a specific configuration of the flyback regulator shown in FIG. FIG. 3 is an internal block diagram showing details of the power supply control circuit shown in FIG. 2, and FIG. 4 is an output current of the power supply device from when the charging circuit shown in FIG. FIG. 5 is a timing chart for explaining the operation of the flyback regulator shown in FIGS. 1 to 3. Note that the same reference numerals denote the same or corresponding parts throughout the drawings. In the figure, 1 is a three-phase AC power source, 2 is a power-on switch such as a no-fuse breaker, 3 is an electromagnetic contactor, 4 is a converter that converts a three-phase AC voltage into DC, and is composed of a diode converter or a transistor converter. .

  Reference numeral 5 denotes a DC smoothing capacitor, which is constituted by a series connection body of two main circuit capacitors 6 and 7 in the first embodiment. 8 is an inverter that converts a DC voltage into a voltage having an arbitrary waveform, and 9 is an electric motor. Reference numeral 10 denotes a flyback regulator as a switching power supply, which includes a first power supply device 11 connected to the main circuit capacitor 6 and a second power supply device 12 connected to the main circuit capacitor 7. In the first embodiment, as shown in FIG. 2, the first and second power supply devices 11 and 12 use windings 25 and 26 and rectifier diodes 27 and 28 of the multi-output flyback regulator 10, respectively. It is configured.

  Reference numeral 13 denotes a charge completion detection circuit for the DC smoothing capacitor 5, 14 receives a signal from the charge completion detection circuit 13, and outputs an output limit level switching command for the flyback regulator 10 via a signal line 14a. 3 is a control circuit for supplying 3 through a signal line 14b, 15 and 16 are respectively a diode module and a smoothing capacitor for rectifying AC power and supplying DC power to the flyback regulator 10, 17 is a power supply control circuit for the flyback regulator 10, 18 Is a current detection resistor that detects a current flowing through the primary winding 23 of the flyback regulator 10 and the switching element 20 that is turned on by inputting a high signal, and 19 is a snubber circuit that protects the switching element 20 from being applied with an excessive voltage. It is. Reference numeral 24 denotes a winding of the high frequency transformer, 22 denotes a rectifier diode that rectifies an AC voltage generated between the terminals of the winding 24, and 21 denotes a smoothing capacitor. The voltage obtained by rectifying the windings 24, 25, and 26 is supplied to the power supply control circuit 17 as a voltage feedback signal of the main circuit capacitors 6 and 7 by utilizing the fact that the ratio of the windings follows.

  As shown in FIG. 3, the power supply control circuit 17 of the flyback regulator 10 amplifies the voltage obtained by subtracting the rectified voltage of the winding 24 from the DC reference voltage source 31 that sets the charging start voltage to an appropriate level, and corrects the phase. Output circuit 29, a comparator 32 that compares the output voltage of the feedback circuit 29 with the magnitude of the triangular wave transmitter 30, and an output of the current detection resistor 18 that is input to a voltage dividing circuit that includes resistors 35, 37, and 38. A comparator 34 that compares the divided voltage of the voltage with a DC reference voltage source 39 that sets an output restriction level; an RS flip-flop 40 that generates a PWM pulse signal according to the outputs of the comparators 32 and 34; and an output restriction And a switch 36 for switching the level.

  The comparator 34 outputs a cut-off signal for the switching element 20 when the current of the switching element 20 reaches the output limit level, and prevents further current from flowing. When the switch 36 for switching the output restriction level is opened, the voltage of the current detection resistor 18 is transmitted to the comparator 34 at a higher level, so that the switching element 20 is cut off at a lower current level, and as a result. The output limit level is lowered. Conversely, when the switch 36 is turned on, the output restriction level increases. Reference numeral 50 denotes switching means for switching the output restriction levels of the power supply apparatuses 11 and 12, and includes the control circuit 14, the signal line 14a, resistors 35, 37, and 38, a switch 36, and the like.

Next, an outline of the operation of the first embodiment configured as described above will be described.
The electromagnetic contactor 3 is shut off before the power-on switch 2 is turned on. When the power-on switch 2 is turned on, the voltage of the AC power source 1 is converted into DC by the diode module 15 and the smoothing capacitor 16 and supplied to the first power source device 11 and the second power source device 12 to start the operation of the charging circuit. To do. At the start of operation, no charges are accumulated in the main circuit capacitors 6 and 7, and the voltage between the terminals is zero. Accordingly, the output voltages of the first and second power supply devices 11 and 12 are zero, and the output currents of the first and second power supply devices 11 and 12 at this time are represented by the curve 41 shown by the solid line in FIG. The main circuit capacitors 6 and 7 are charged with the current corresponding to the point a, and the voltage between the terminals of the main circuit capacitors 6 and 7 increases. In the process, the charging currents of the first and second power supply devices 11 and 12 move from the point a of the curve 41 toward the point b.

  When the charge completion detection circuit 13 detects that the total value of the voltage across the terminals of the main circuit capacitors 6 and 7 exceeds the rectified voltage of the AC power supply 1, the control circuit 14 turns the electromagnetic contactor 3 through the signal line 14a. The converter 4 and the inverter 8 become operable. Before the electromagnetic contactor 3 is turned on, the total value of the voltages across the terminals of the main circuit capacitors 6 and 7 exceeds the control voltage of the converter 4, but after the electromagnetic contactor 3 is turned on, the first and second power supply devices 11, Compared with the output power of 12, the power handled by the converter 4 and the inverter 8 is overwhelmingly large, so the total value of the voltages across the terminals of the main circuit capacitors 6 and 7 is reduced to the control voltage of the converter 4. At this time, by arranging the output limiting characteristics of the first and second power supply devices 11 and 12, the DC bus voltage is evenly distributed to the main circuit capacitors 6 and 7, which is excessive to a specific capacitor. It is possible to prevent an excessive voltage from being applied.

  After starting, a current corresponding to half of the control voltage of the converter 4 flows continuously from the first and second power supply devices 11 and 12, but the curve 41 in FIG. 4 indicates the charging of the main circuit capacitors 6 and 7. Since the time is determined, the output restriction level should normally be set as high as possible. However, in such a case, after the converter 4 and the inverter 8 are started, a large current continues to flow through the first and second power supply devices 11 and 12, and the charging circuit gradually generates heat. In order to prevent this, it is necessary to provide a heat dissipation mechanism necessary and sufficient to continuously output the charging current. However, since the current required to equalize the voltages of the main circuit capacitors 6 and 7 after starting is about the leakage current of the capacitors and smaller than the charging current, the curve 42 shown by the broken line in FIG. By significantly reducing the output restriction level, the output current after starting can be reduced, the heat dissipation mechanism can be greatly simplified, and the circuit can be miniaturized.

  In FIG. 4, the charging current changes from a to b in the curve 41 during the preliminary charging, reaches the charging end voltage Ve, and when the switch 36 is opened, the charging current changes from b to e, f. When the magnetic contactor 3 is turned on and the converter 4 and the inverter 8 become operable, the charging current changes in the range from c to d of the curve 42. It shows that

Hereinafter, the flyback regulator 10 and its power supply control circuit 17 according to the first embodiment will be described in more detail with reference to FIGS. 2 and 3.
When the power-on switch 2 is turned on and power is supplied to the charging circuit, the voltages of the main circuit capacitors 6 and 7 are zero, and the charging completion detection circuit 13 indicates that the charging is not completed. This is transmitted to the control circuit 14. In response to this, the switch 36 is turned on via the signal line 14a, and the charging circuit starts operating in a state where the output restriction level is high. Since the winding voltage of the high-frequency transformer is proportional to the winding ratio, the number of turns of the windings 24, 25, and 26 is N24, N25, and N26, respectively, and the charging voltages of the capacitors 6, 7, and 21 are V6 and V7, respectively. , V21, the voltage ratio between terminals of each capacitor during charging is roughly as shown in (Expression 1).

  V6: V7: V21≈N25: N26: N24 (Formula 1)

  Accordingly, the voltage of the smoothing capacitor 21 is proportional to the voltage between the terminals of the main circuit capacitors 6 and 7 and is almost zero. This voltage is input to the feedback circuit of the power supply control circuit 17 of the flyback regulator 10 and subtracted from the DC reference voltage source 31 that outputs a voltage corresponding to the charging end voltage Ve shown in FIG. The signal is shaped to an appropriate size and frequency response by an amplifier or the like and input to the plus terminal of the comparator 32.

  Since the rectified voltage of the winding 24 is significantly smaller than the DC reference voltage source 31 at the beginning of charging, a positive voltage is input to the feedback circuit 29 and also to the output of the feedback circuit 29 as shown by a waveform 44 in FIG. A large positive voltage is output. On the other hand, a signal of the triangular wave oscillator 30 that generates a carrier signal for PWM modulation as shown by a waveform 43 in FIG. As a result, a waveform as shown in FIG.

Since in the T _on period R input of the RS flip-flop 40 is low, the output of the RS flip-flop 40 as shown in FIG. 5 (5) is a _{T on} period in high. During this time, the switching element 20 is turned on, and the rectified voltage Vc is applied to the winding 23 of the high-frequency transformer. At this time, voltages are generated in the other windings 24, 25, and 26 of the high-frequency transformer in a direction in which a reverse bias voltage is applied to the rectifier diodes 22, 27, and 28 connected to the windings. The diode is turned off. Therefore, current flows only in the winding 23 of the high-frequency transformer 33, the inductance of the winding 23 When _{L 23,} current I (t) is Equation 2 that flows through the winding 23, and as shown in FIG. 5 (8) Increasing over time.

但し、
Ｖｃ：巻線２３に印加される整流電圧。
Ｌ_２３：巻線２３のインダクタンス。
ｔ：時間。

However,
Vc: Rectified voltage applied to the winding 23.
L ₂₃ : Inductance of the winding 23.
t: time.

This current generates a voltage across the terminals of the current detection resistor 18, the voltage is divided by the resistors 35,37,38 constituting the voltage dividing circuit, set at a voltage whose level corresponds to the output limit level I _CLM When the level of the DC reference voltage source 39 is reached, the output of the comparator 34 becomes high, the output of the RS flip-flop 40 becomes low, and the switching element 20 is cut off. For this reason, a counter electromotive force is generated in the high-frequency transformer, and the switching element is conducted in each winding to generate a voltage having a polarity opposite to that of the _Ton period. As a result, the rectifier diodes 22, 27, and 28 are turned on, and the main circuit capacitors 6 and 7 and the smoothing capacitor 21 are charged. During operation of the flyback regulator 10, the voltage appearing between the terminals of the windings 25 and 26 is proportional to the number of turns. Therefore, by arranging the number of turns N25 and N26 of the windings 25 and 26, the main circuit capacitors 6 and 7 are arranged. The voltage between terminals can be equalized.

After the charging is completed and the electromagnetic contactor 3 is turned on, the voltage between the terminals of the main circuit capacitors 6 and 7 is pulled down by the control voltage of the converter 4 and is supplied from the windings 24, 25 and 26 at this time. The total power P ₂₃ is expressed by (Equation 3) when the circuit loss is ignored.

但し、
Ｔ_ｏｎ：スイッチング素子２０の導通期間。
Ｔ：ＰＷＭ信号のキャリア周期。
Ｉ_ＣＬＭ：巻線２３の電流制限レベル（出力制限レベル）。

However,
T _on : The conduction period of the switching element 20.
T: Carrier period of the PWM signal.
I _CLM : Current limit level of winding 23 (output limit level).

As shown in (Equation 3) above, the charging power is proportional to the square of the current limit level _ICLM of the winding 23. Therefore, at the time of start-up, the switch 36 is turned on to increase the current limit level of the winding 23 so that charging can be performed at high speed. After the start-up, the electromagnetic contactor 3 is turned on via the signal line 14b to charge the charging circuit. 4 is pulled down by the control voltage of the converter 4 and the switch 36 is opened, the current limit level is reduced as shown by the curve 42 in FIG. The charging circuit can be reduced in size.

  As described above, in the first embodiment of the present invention, the total value of the output voltages of the first and second power supply devices 11 and 12 is set to be equal to or higher than the control voltage of the converter 4, and at the same time, the output limiting function is provided. Thus, the voltage across the terminals of the main circuit capacitors 6 and 7 can be equalized. It is possible to charge the main circuit capacitors 6 and 7 in a short time by increasing the output limit level of each power supply device 11 and 12 at the start, and the output limit of the flyback regulator 10 as a switching power supply after the start is leaked from the capacitor. By reducing the level to a level necessary and sufficient to compensate for the current component, heat generation due to overload of the first and second power supply devices 11 and 12 when the voltage of the converter 4 is reduced, and the main circuit capacitors 6 and 7 connected in series Unnecessary charging current of the power supply device due to voltage mismatch between the terminals of the main circuit capacitors 6 and 7 due to capacitance mismatch and charging / discharging current flowing through the DC smoothing capacitor 5 during operation of the inverter 8 is limited. Heat generation can be suppressed, and the power supply device can be downsized.

  In addition, since the first and second power supply devices 11 and 12 are switching power supplies having an output limiting function, a resistance is connected in series with the output of the power supply, and the output is directly limited by the resistance. Energy efficiency can be improved and downsized. Further, since the output limit level is changed by switching the current limit value of the primary side winding 23, the switching means 50 can be constituted by a low-power circuit, so that the size and cost can be reduced.

Embodiment 2. FIG.
FIG. 6 is an internal block diagram showing details of the power supply control circuit of the inverter control apparatus according to Embodiment 2 of the present invention. Other configurations are the same as those illustrated and described in FIGS. 1 and 2 according to the first embodiment, and therefore, differences from the first embodiment will be mainly described below. In the figure, reference numeral 30 denotes a triangular wave oscillator that generates a carrier signal for PWM modulation. In the second embodiment, when the charge completion detection circuit 13 detects the completion of charge and transmits this to the control circuit 14, the control circuit 14 is configured such that the frequency of the carrier signal is reduced via a signal line 14a. In the second embodiment, the switching means 50 for switching the output restriction level is configured using the control circuit 14, the signal line 14a, the triangular wave transmitter 30, and the like. Other reference numerals have the same functions as those in FIG.

  Next, the operation of the second embodiment configured as described above will be described. Since the flyback regulator 10 of FIG. 1 is configured as shown in FIG. 2 as in the first embodiment, the characteristics thereof are the same as those of the first embodiment and follow the above (Equation 3). As shown in (Equation 3), the total power output from the high-frequency transformer 33 is inversely proportional to the period which is the reciprocal of the frequency of the carrier signal. Therefore, by reducing the carrier frequency after completion of charging, the same effect as in the first embodiment can be obtained.

  As described above, according to the second embodiment of the present invention, the first and second power supply devices 11 and 12 for precharging the main circuit capacitors 6 and 7 connected in series have the switching power supply having an output limiting function. As a result, it is possible to improve energy efficiency and reduce the size as compared with the case where a resistor is connected in series with the output of the power source and the output is directly limited by the resistor. In addition, since the output limit level is changed by switching the switching frequency, the switching function can be configured with a low-power circuit, so that the size and cost can be reduced.

  In the above description of the embodiment, the example in which the DC smoothing capacitor 5 includes the two main circuit capacitors 6 and 7 and the flyback regulator 10 is used as a switching power supply has been described. is not. Further, the switching means 50 for switching the output restriction level is not limited to the configuration of the above embodiment.

BRIEF DESCRIPTION OF THE DRAWINGS It is a whole block diagram of the inverter control apparatus concerning Embodiment 1 of this invention. FIG. 2 is a block diagram specifically showing the configuration of the flyback regulator shown in FIG. 1. FIG. 3 is an internal block diagram showing details of a power supply control circuit shown in FIG. 2. It is a characteristic view which shows the change of the output current of a power supply device from the time the charging circuit shown in FIG. It is a timing chart explaining operation | movement of the flyback regulator shown in FIGS. It is an internal block diagram which shows the detail of the power supply control circuit which is the principal part of the inverter control apparatus by Embodiment 2 of this invention.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 Three-phase AC power supply, 2 Power-on switch, 3 Magnetic contactor, 4 Converter, 5 DC smoothing capacitor, 6, 7 Main circuit capacitor, 8 Inverter, 9 Electric motor, 10 Flyback regulator (switching power supply), 11 1st Power supply device, 12 second power supply device, 13 charge completion detection circuit, 14 control circuit, 14a, 14b signal line, 15 diode module, 16 smoothing capacitor, 17 power supply control circuit, 18 current detection resistor, 19 snubber circuit, 20 switching elements, 21 smoothing capacitors, 22 rectifier diodes, 23 primary windings, 24, 25, 26 windings, 27, 28 rectifier diodes, 29 feedback circuits, 30 triangular wave oscillators, 31 DC reference voltage sources, 32, 34 Compare Motor, 33 high-frequency transformers, 35,37,38 resistance, 36 switch, 39 a DC reference voltage source, 40 RS flip-flop, 50 switching means.

Claims (4)

  A converter for converting alternating current into direct current, an inverter for converting direct current converted by the converter into alternating current, a plurality of main circuit capacitors connected in series between the converter and a direct current bus connecting the inverter, and these main circuits In the inverter control device comprising a plurality of power supply devices connected between the respective terminals of the capacitor, the power supply device has an output limiting characteristic equivalent to each other while the total output voltage exceeds the voltage of the converter. And an inverter control device comprising switching means for switching an output restriction level of the power supply device.   The power supply apparatus comprises a switching power supply using a transformer, and the output is limited by configuring so that the maximum value of the primary side winding current of the transformer does not exceed a certain value. The inverter control device according to Item 1.   The inverter control device according to claim 2, wherein the switching means switches the output restriction level by switching the maximum value of the primary side winding current of the transformer.   The inverter control device according to claim 2, wherein the switching means switches the output restriction level by switching a switching frequency of the switching power supply.

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