JP2005341131A - Mimo wireless signal transmission system and method therefor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a MIMO wireless signal transmission system and a method for more effectively exhibiting error capability of soft decision error correction decoding, by multiplying an MLD determination signal for each transmitted signal t<SB>j</SB>by a magnitude of a transfer coefficient h<SB>ij</SB>through which each signal t<SB>j</SB>passes, that is, a likelihood weighting factor with SNR of each transmitted signal t<SB>j</SB>reflected. <P>SOLUTION: A wireless signal transmitter 1 has transmission antennas 1-6-1 to 1-6-N and transmits an error corrected and encoded wireless signal T by means of the same frequency. A wireless signal receiver 2 has reception antennas 2-1-1 to 2-1-M, and has a transfer response matrix estimation means 2-3 for a transfer response matrix H, an MLD means 2-4 for generating a replica H×(^T<SB>s</SB>) of a received signal R for determining (=T) which minimizes ¾R-H×(^T<SB>s</SB>)¾ to be an original transmitted signal, and soft decision error correcting and encoding means 2-8-1 to 2-8-N for soft decision error correcting and encoding (=T), and uses a predetermined value as the likelihood weighting factor. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a radio signal transmission system and method for configuring a multiple-input multiple-output (MIMO) channel using a plurality of transmitting antennas and a plurality of receiving antennas in a radio communication system.

In a wireless communication system, there is provided a MIMO wireless transmission system capable of improving transmission capacity without expanding a use frequency bandwidth by spatially multiplexing a plurality of signals at the same frequency using a plurality of transmitting antennas and a plurality of receiving antennas. As a wireless transmission system that dramatically improves frequency utilization efficiency, it is attracting attention.
In such a MIMO wireless transmission system, since a plurality of different transmission signals overlap each other in the spatial multiplexing signal received by each receiving antenna, a space for determining a specific transmission signal from such a spatial multiplexing signal on the reception side. A multiple signal determination method is required.

  As such a spatial multiplexing signal determination method, a reception signal replica at each reception antenna is generated from a transmission channel candidate point and an estimated MIMO channel, and the difference between the reception signal actually received by each antenna and the Euclidean distance between each reception signal replica is generated. It is known that the best transmission quality can be realized theoretically by using MLD (Maximum Likelihood Detection) that estimates a transmission signal candidate point that minimizes the sum of squares of 2 as the original transmission signal.

It is assumed that the wireless signal transmission apparatus includes N transmission antennas, the wireless signal transmitted from each transmission antenna j is t _j, and a set of these is a transmission signal vector T = (t ₁ , t ₂ ,. ., T _N ) _Let ^t be. However, ^t means a transposed vector.
Further, it is assumed that the radio signal receiving apparatus is provided with M receiving antennas, the received signals received by each receiving antenna i are denoted by r _i, and these sets are defined as received signal vectors R = (r ₁ , r ₂ ,. .., r _M ) ^t .
In addition, a transmission coefficient between the transmission antenna j and the reception antenna i is set as h _ij , and a matrix having h _ij as i row and j column component elements is set as a transfer coefficient matrix H. The relationship between T, R, and H is expressed as the following formula (1).

Here, N is a noise signal vector composed of a noise signal associated with each received signal, and N = (n ₁ , n ₂ ,..., _{N N} ) ^t . MLD is intended to be estimated transfer coefficient matrix H on the receiving side, generating all transmission signal candidate points from the transfer coefficient matrix H estimated as (^ T _s), a replica H × reception signal (^ T _s) To do.

The replica H × (^ _{T s)} and the Euclidean distance difference between the received signal R | determined that the minimizing ^(~ T) original transmission signal _{| R-H × (^ T} s). That is, the determination is made as in the following equation (2).

However, when a soft decision error correction code is applied to a MIMO wireless transmission system to which such MLD is applied, a signal determined by MLD becomes a fixed transmission signal candidate point (^ T _s ), that is, a hard decision value. There is a problem that the soft decision gain of the soft decision error correcting code cannot be obtained sufficiently.
To solve this, MLD judgment signal ^{(~ T),} a technique for multiplying the likelihood weighting coefficients to weight that reflects the reliability of the MLD determination result is considered. As such a likelihood weighting factor, the second smallest value d ₂ and the first smallest value of the Euclidean distance difference | R−H × (^ T _s ) | between the received signal received by each antenna and each received signal replica. squared difference _d ² 2 _-d ^{1 2} method using the d ₁ have been conventionally proposed (non-Patent Document 1).

FIG. 10 shows the likelihood weighting factor of the present technique illustrated in the received signal complex space (r ₁ , r ₂ ). For the sake of simplicity, consider a MIMO radio signal transmission system in which the number of transmission antennas is two and the number of reception antennas is two, and BPSK (Binary Phase Shift Keying) modulation is used as modulation / demodulation of a transmission signal. Four combinations (t ₁ , t ₂ ) = (1, 1), (1, −1), (−1, 1), (−1, BPSK modulated transmission signals t ₁ , t ₂ -1), and there are replica signals h ₁ + h ₂ , h ₁ -h ₂ , -h ₁ + h ₂ , and -h ₁ -h ₂ for each. The grid points ● in FIG. 10 correspond to this.

The vectors h ₁ and h ₂ are sets of transmission coefficients of the wireless transmission path through which the transmission signals t ₁ and t ₂ pass, respectively, and h ₁ = (h ₁₁ , h ₂₁ ) ^t , h ₂ = (h ₁₂ , H ₂₂ ) ^t . When the received signal (r ₁ , r ₂ ) is received at the x point (FIG. 10), the closest replica signal from the x point is h ₁ -h ₂ and the second closest replica signal is -h ₁ -h. _{2, and} likelihood weighting coefficient d ₂ ² -d ₁ ² of the present approach is the squared differences of the distance between two points of these replica signal and a received signal.
If the difference between d ₂ and d ₁ is large, d ₁ is remarkably small in Euclidean distance | R−H × (^ T _s ) | in comparison with other transmission signal candidate points (^ T _s ), and d ₁ is minimum. since the reliability of the result of a transmission signal ^(~ T) are considered to be high, likelihood weighting coefficient _d ^{2 2} _-d ^{1 2} of the present approach reflects the reliability of the determination signal of MLD ^(~ T) It is thought that.

Therefore, by multiplying such a likelihood weighting factor to each MLD determination signals ^{(~ T),} the input value to the soft decision error correction decoder is a soft decision value for the reliability reflecting the MLD determination signal . Therefore, according to this method, the error rate after error correction decoding can be significantly improved as compared with the case where no likelihood weighting coefficient is multiplied.
On the other hand, when there is a difference in the size of the transmission coefficient h _ij of the wireless transmission path through which each transmission signal t _j passes, it is the total reception signal power of each transmission signal t _j received by all reception antennas.

Since there is a difference for each _j , there is a difference in the SNR of each transmission signal t _j .
However, since all the same likelihood weighting factor present likelihood weighting coefficient d _{² 2} -d _{1 2} for each transmit signal t _j simultaneously transmitted, the difference between the SNR of each transmitted signal t _j transmitted simultaneously Cannot be reflected and the error correction capability of soft decision error correction decoding cannot be maximized.
Hori, Mizoguchi, Morikura, “Viterbi decoding soft decision method in OFDM system using SDM”, IEICE General Conference, B-5-252, March. 2001.

The present invention has been made in view of these circumstances, transfer the objects, in a MIMO wireless transmission system according to the MLD, the relative MLD determination signal for each transmission signal t _j, through which the respective transmit signal t _j A MIMO radio signal transmission system that more effectively exhibits the error capability of soft decision error correction decoding by multiplying the likelihood weighting factor reflecting the magnitude of the coefficient h _ij , that is, the SNR of each transmission signal t _j And providing a method.

The MIMO radio signal transmission system according to claim 1 includes N (N is an integer of 2 or more) transmission antennas, and a radio signal T subjected to error correction coding using the same frequency from the transmission antennas. (T = (t ₁ , t ₂ ,..., T _N ) means ^t . Also, ^t means a transposed vector. Also, _i in t _i means a transmitting antenna number.) A wireless signal transmission apparatus and M (M is an integer of 1 or more) reception antennas are provided, and N × M wireless transmission paths that are combinations of the N transmission antennas and the M reception antennas are transmitted. A transfer response matrix H (H (i, j) component h _ij represents a response between the transmitting antenna j and the receiving antenna i. J is an integer from 1 to N, and i is 1) And an estimation means of M or less, and the estimation means A constant has been transmitted response matrix H, transmission signal candidate points (^ T _s) to generate a replica H × reception signal (^ T _s) from the replica H × (^ T _s) and the M each receive a Euclidean distance difference | R− with the received signal R of the antenna (R = (r ₁ , r ₂ ,..., R _M ) ^t . Also, _j of r _j means the receiving antenna number). H × _{(^ T} s) | a to minimize the determined MLD unit and ^(~ T) original transmission signal, soft-decision error correction is determined by the MLD means ^(~ T) for soft-decision error correction decoding In a MIMO radio signal transmission system comprising a radio signal receiving apparatus having decoding means, the radio signal receiving apparatus transmits a transmission signal t _k (k is 1 or more, N) as soft decision information used for the soft decision error correction decoding means. As a likelihood weighting factor for Left pseudo-inverse of the transfer response matrix ^{H (H H H) -1 H} H ((H H H) -1 H H s-row t column component (s is 1 or more N an integer, t is 1 to M integer.) after taking the sum in the column direction of the square of the absolute value of the k-th row element components h'and _st) and is reciprocal square root values of

It is characterized by using.

The radio signal transmission system according to claim 2, wherein the soft decision as the likelihood weighting factor for the transmit signal t _k used in the error correction decoding means, wherein in the A and the total transmitted signal candidate points (^ T _s) Of the Euclidean distance difference | R−H × (^ T) |, the product of the square root √ (d ₂ ² −d ₁ ² ) of the square difference of the second smallest distance d ₂ and the minimum distance d ₁ A × √ ( d ₂ ² −d ₁ ² ) is used.

The radio signal transmission system according to claim 3, wherein the soft decision as the likelihood weighting factor for the transmit signal t _k used in the error correction decoding means, wherein in the A and the total transmitted signal candidate points (^ T _s) Euclidean distance difference | R-H × (^ T ) | of, is characterized by using the second a product a × _d 2 / _{d 1} of a small distance _{d 2} and the ratio of the minimum distance _{d 1.}

Moreover, MIMO radio signal transmission system according to claim 4, wherein the transmit signal T OFDM signal, as the likelihood weighting factor for the sub-carrier component of the transmission signal t _k to be used for the soft-decision error correction decoding means, sub The above-mentioned A of the carrier component f is used.

Moreover, MIMO radio signal transmission system according to claim 5, wherein the transmit signal T OFDM signal, as the likelihood weighting factor for the sub-carrier component of the transmission signal t _k to be used for the soft-decision error correction decoding means, sub The A × √ (d ₂ ² −d ₁ ² ) of the carrier component f is used.

Moreover, MIMO radio signal transmission system according to claim 6, wherein the transmit signal T OFDM signal, as the likelihood weighting factor for the sub-carrier component of the transmission signal t _k to be used for the soft-decision error correction decoding means, sub The A × d ₂ / d ₁ of the carrier component f is used.

Further, the MIMO radio signal transmission method according to claim 7 performs error correction coding using the same frequency transmitted from a radio signal transmission device having N (N is an integer of 2 or more) transmission antennas. The applied radio signal T (T = (t ₁ , t ₂ ,..., T _N ) means ^t , ^t means a transposed vector, and _i in t _i means a transmitting antenna number. .) In a radio signal receiving apparatus having M (M is an integer of 1 or more) receiving antennas and having a transfer response matrix estimating means, likelihood weighting coefficient calculating means, MLD means, and soft decision error correcting decoding means In the receiving MIMO radio signal transmission method, the transfer response matrix H (the (i, j) component h _ij of H) means the transfer response between the transmission antenna j and the reception antenna i by the transfer response matrix estimation means. Is 1 or more N Under integer, i is estimating a an integer of 1 to M.),
The likelihood weighting coefficient calculation means uses the transmission response matrix H estimated by the transmission response matrix estimation means, and uses the likelihood weighting coefficient for N transmission signals t _j.

Calculating a, a step of storing the likelihood weighting factor likelihood weighting factor A calculated by the calculation means, by the MLD unit determines the N transmission signals t _j, likelihood for the determined t _j Calculating the degree L _j , the likelihood calculated by the MLD means, the value L _j × A obtained by multiplying the accumulated A as a soft decision value, and the soft decision error correction decoding means And a step of correcting and decoding the soft decision value obtained by the MLD means.

Further, in the MIMO radio signal transmission method according to claim 8, the MLD means uses 2 of the Euclidean distance differences | R−H × (^ T) | at the A and all transmission signal candidate points (^ T). Calculating the product A × √ (d ₂ ² −d ₁ ² ) of the second smallest distance d ₂ , the minimum distance d ₁ and the square root of the square difference √ (d ₂ ² −d ₁ ² ), A × √ (d ₂ ² −d ₁ ² ) is used as the likelihood weighting coefficient.

Further, in the MIMO radio signal transmission method according to claim 9, the MLD means uses 2 of Euclidean distance differences | R−H × (^ T) | between the A and all transmission signal candidate points (^ T). further comprising the step of computing the product a × _d 2 / _{d 1} of a small distance _{d 2} and the ratio of the minimum distance _{d 1} to the second, as the likelihood weighting factor, the use of a × _d 2 / _{d 1} Features.

The MIMO radio signal transmission method according to claim 10 is used in the step of correcting and decoding the soft decision value obtained by the MLD means by the soft decision error correction decoding means, wherein the transmission signal T is an OFDM signal. A is used as a likelihood weighting factor for a subcarrier component of a transmission signal t _k (k is an integer of 1 to N).

The MIMO radio signal transmission method according to claim 11 is used in the step of correcting and decoding the soft decision value obtained by the MLD means by the soft decision error correction decoding means, wherein the transmission signal T is an OFDM signal. A × √ (d ₂ ² −d ₁ ² ) is used as a likelihood weighting factor for a subcarrier component of a transmission signal t _k to be transmitted (k is an integer of 1 to N).

The MIMO radio signal transmission method according to claim 12 is used in the step of correcting and decoding the soft decision value obtained by the MLD means by the soft decision error correction decoding means, wherein the transmission signal T is an OFDM signal. A × d ₂ / d ₁ is used as a likelihood weighting factor for a subcarrier component of a transmission signal t _k (k is an integer of 1 to N).

  According to the first aspect of the present invention, the likelihood weighting factor reflecting the SNR of each transmission signal

By multiplying, the error capability of soft decision error correction decoding can be effectively exhibited.

According to the second aspect of the present invention, by multiplying the likelihood weighting coefficient A × √ (d ₂ ² −d ₁ ² ) reflecting the SNR of each transmission signal, soft decision error correction decoding is performed. The error ability can be exhibited more effectively.

According to the third aspect of the invention, the error capability of the soft decision error correction decoding is further improved by multiplying the likelihood weighting factor A × d ₂ / d ₁ reflecting the SNR of each transmission signal. Can be demonstrated.

  Moreover, according to the invention of Claims 4-6, the error correction capability of soft decision error correction decoding can be more effectively exhibited in a frequency selective fading environment.

Hereinafter, a MIMO radio signal transmission system according to an embodiment of the present invention will be described in detail with reference to the drawings.
FIG. 1 is a block diagram of a radio signal transmitting apparatus 1 and a radio signal receiving apparatus 2 constituting a MIMO radio signal transmission system according to the first embodiment of the present invention.
N (N is an integer of 2 or more) independent data signal sequences # 1, # 2,..., #N are input to the wireless signal transmission device 1. Note that the data signal sequences # 1, # 2,..., #N may be configured such that N data signal sequences independent from the outside such as a PC are supplied. May be configured such that one data signal sequence is supplied to the wireless device and the wireless device decomposes the data signal sequence into N data signal sequences by serial-parallel conversion. Further, the wireless device itself may generate N independent data signals.

The radio signal transmission apparatus 1 performs error correction coding on each signal sequence # 1, # 2,..., #N with respect to the data signal sequences # 1, # 2,. N error correction coding units 1-1-1 to 1-1-N are provided. Further, the output signals of the N error correction coding units 1-1-1 to 1-1-N are modulated into N radio signals T = (t ₁ , t ₂ ,..., T _N ) ^t . It has N modulation means 1-2-1 to 1-2N.
Further, the radio signal receiving apparatus 2 includes path transfer response matrix estimation pilot signal generating means 1-3 for generating a path transfer response matrix estimation pilot signal for estimating the transfer response matrix H. Further, N signal multiplexing means 1-4-1 to 1-4-N for adding a path transfer response matrix estimation pilot signal to the radio signal T are provided.

Also, N frequency converting means 1-5-1 to 1-5-N for converting the output signals of the N signal multiplexing means 1-4-1 to 1-4-N into radio frequencies are provided. In addition, N transmission antennas 1-6-1 to 1-6 -N that transmit the output signals of the N frequency conversion units to the space are provided.
The error correction coding means 1-1-1 to 1-1-N generate data signal sequences # 1, # 2,..., #N from one data signal sequence by serial-parallel conversion. In such a case, a configuration in which only one is placed before the serial-parallel conversion is also conceivable.

On the other hand, the radio signal receiving apparatus 2 includes M reception antennas 2-1-1 to 2-1 -M. In addition, M frequency conversion units 2-2-1 to 2-2M for frequency-converting signals received by the reception antennas 2-1 to 2-1-M to the baseband are provided.
Further, there is provided a transfer response matrix estimation means 2-3 for estimating the path transfer response matrix H from the path transfer response matrix estimation pilot signals received by the M reception antennas 2-1-1 to 2-1 -M. Also, using the transfer response matrix H estimated by the transfer response matrix estimation means 2-3, M received signals R = (r ₁ ) received by the M receiving antennas 2-1-1-2-1 -M. _{, r} 2, ···, from r ^{M) t,} the original N transmission signals by ^{_{MLD (~ T) = ((}} ~ t 1), (~ t 2), ···, (~ t N) ) MLD means 2-4 for determining ^t .

  Also, N likelihood weighting factors are determined from the transfer response matrix H estimated by the transfer response matrix estimation means 2-3.

Likelihood weighting coefficient calculating means 2-6.
Moreover, N sets of transmission signals determined in MLD unit _{^{2-4 (~ T) = ((}} ~ t 1), (~ t 2), ···, (~ t N)) the likelihood for ^t N likelihood calculating means 2-5-1 to 2-5-N to be calculated are included. Further, the N likelihood weighting coefficients A calculated by the likelihood weighting coefficient calculating means 2-6 are added to the likelihoods calculated by the N likelihood calculating means 2-5-1 to 2-5-N. N likelihood weighting coefficient multiplying means 2-7-1 to 2-7 -N each multiplying each other are provided.
Also, soft decision error correction decoding means 2-8-1 to 2-8-N for performing soft decision error correction on the N output signals of the likelihood weight coefficient multiplication means 2-7-1 to 2-7-N, respectively. Have

When the error correction coding means in the radio signal transmission apparatus 1 is composed of one piece, the error correction decoding means in the radio signal reception apparatus 2 is also N pieces from the N likelihood weight coefficient multiplication means 2-6. There is a case where only one output signal is formed in the subsequent stage after the output signal is converted into one signal string by parallel-serial conversion.
The radio signal transmission device 1 uses the error correction coding means 1-1-1 to 1-1-N to convert N transmission signals T = (t ₁ , t ₂ ,..., T _N ) ^t into error correction codes. Then, prior to the transmission signal T = (t ₁ , t ₂ ,..., T _N ) ^t subjected to N error correction coding, the radio signal receiving apparatus 2 sets the path transfer response matrix H The radio signal receiving apparatus 2 for estimation generates a transmission response matrix estimation pilot signal having a known signal pattern by the transmission response matrix estimation pilot signal generating means 1-3. Then, the transmission signal T = (t ₁ , t ₂ ,..., T _N ) ^t subjected to N error correction coding by N signal multiplexing means 1-4-1 to 1-4-N. A pilot signal for estimating a transfer response matrix is added before.

The transmission response matrix estimation pilot signal may be any pilot signal pattern as long as the wireless signal receiving apparatus 2 can estimate the transmission response matrix H. For example, a mode in which the same pilot signal pattern is transmitted from each transmission antenna in a time division manner is conceivable. This signal pattern is shown in FIG.
In FIG. 2, for example, the same pilot signal pattern is transmitted in order from the transmission antenna 1-6-1, and other transmission antennas are not transmitting while transmitting a pilot signal from a certain transmission antenna. For example, when a pilot signal is transmitted from the transmission antenna 1-6-j (j is an integer of 1 or more and N or less), the reception antenna 2-1-i (i is 1 or more and M or less) of the radio signal reception device 2. ), Only the pilot signal via the reception antenna 2-1-i is received from the transmission antenna 1-6-j. Therefore, if the radio signal reception device 2 knows the pilot signal pattern, the transmission antenna 1 The transfer response _hij between -6-j and the receiving antenna 2-1-i can be estimated. Hereinafter, if a similar operation is performed for each pilot antenna sequentially transmitted from each transmission antenna, the transmission responses h _ij of all combinations of N transmission antennas and M reception antennas, that is, transmission responses, are performed. The matrix H can be estimated.
The operation for estimating the transfer response matrix H in the radio signal receiving apparatus 2 described above is performed by the transfer response matrix estimating means 2-3 in FIG.

N multiplexing means 1-4-1 to 1-4-N transmit responses to transmission signals T = (t ₁ , t ₂ ,..., T _N ) ^t subjected to N error correction codings. The N signals to which the matrix estimation pilot signal is added are frequency-converted to a radio frequency band by N frequency converting means 1-5-1 to 1-5-N, and N transmitting antennas 1-6-1 are transmitted. -1-6-N is transmitted to the space.
The radio signal receiving apparatus 2 converts N transmission signals transmitted from N transmission antennas 1-6-1 to 1-6-N into M reception antennas 2-1 to 2-1-M. After receiving M received signals, the frequency conversion means 2-2-1 to 2-1-M converts the frequency to the baseband.

MLD unit 2-4, using the transmission response matrix H estimated by transmission response matrix estimation unit 2-3 from M received signal, based of the N transmission signals _{^{(~ T) = ((~}} t 1) , judges ^{_{(~ t 2), ···,}} (~ t N)) t. On the other hand, the likelihood weighting coefficient calculating means 2-6 calculates N likelihood weighting coefficients A using the transfer response matrix H estimated by the transfer response matrix estimating means 2-3.
N number of likelihood calculating means 2-5-1～2-5-N is, the N transmission signals MLD means determines _{^{(~ T) = ((~}} t 1), (~ t 2), ·· · ^to calculate each likelihood respect _(~ ^{t N))} t. In this case, when the multi-level number of a modulation scheme and m are each transmitted signal calculation ^{(~ t} _j) m likelihoods for each, m × N number of likelihood in total is calculated.

Likelihood weight coefficient multiplication means 2-7-1 to 2-7 -N calculate the likelihood of m × N likelihoods calculated by N likelihood calculation means 2-5-1 to 2-5 -N. each of N likelihood weighting coefficient a Doomomi coefficient calculation unit 2-6 is calculated by multiplying the each other correspond to each transmitted signal ^{(~ t} _j), N pieces of soft decision error correction decoding means 2-8 Input to -1 to 2-8-N.
N soft decision error correction decoding means 2-8-1 to 2-8 -N perform soft decision error correction on each of the N input signal sequences inputted, and finally the original N pieces data signal sequence ^{# (~} 1), # (1-2), playing ..., ^# a ^(~ N).

In order to clarify the physical meaning of the likelihood weighting factor A of the present embodiment, FIG. 3 shows the likelihood weighting factor of the present embodiment and the arrangement of replica signals in the received signal complex space (r ₁ , r ₂ ). An illustration of the relationship is shown. As in FIG. 10, for simplicity of explanation, a MIMO radio signal transmission system in which the number of transmission antennas is two, the number of reception antennas is two, and BPSK modulation is used as modulation / demodulation of a transmission signal will be described. In the following description, the same holds true for multilevel modulation such as QPSK, 16QAM, etc., in which the modulation scheme of the transmission signal is orthogonal to each other.

Similar to FIG. 10, four combinations (t ₁ , t ₂ ) = (1, 1), (1, −1), (−1, 1) with respect to the transmission signals t ₁ , t _{2 of} BPSK modulation. ), (−1, −1), and there are replica signals h ₁ + h ₂ , h ₁ −h ₂ , −h ₁ + h ₂ , and −h ₁ −h ₂ for each. The grid points in FIG. 3 correspond to this.
As in FIG. 10, the vectors h ₁ and h ₂ are sets of transmission coefficients of wireless transmission paths through which the transmission signals t ₁ and t ₂ respectively pass, and h ₁ = (h ₁₁ , h ₂₁ ) ^t , h ₂ = (h ₁₂ , h ₂₂ ) ^t . Also, e ₁ and e ₂ are unit vectors of size 1 that are orthogonal to h ₂ and h ₁ , respectively.

The reliability of the transmission signal t ₁ determined by the MLD means 2-4 is the distance between replica signals having the same transmission signal t ₂ but different transmission signals t ₁ , that is, (t ₁ , t ₂ ) = (1, 1). And (t ₁ , t ₂ ) = (− 1,1) (or (t ₁ , t ₂ ) = (1, −1) and (t ₁ , t ₂ ) = (− 1, −1)) This is the Euclidean distance difference between the corresponding replica signals h ₁ + h ₂ and -h ₁ + h ₂ . Although this Euclidean distance difference depends on the angle between h ₁ and h ₂ and the magnitude of h ₂ , the minimum value is equal to twice the inner product value <e ₁ | h ₁ > of h ₁ and e ₁ .

Therefore, the inner product value <e ₁ | h ₁ > is considered to reflect the reliability of the transmission signal t ₁ determined by the MLD. Similarly, the inner product value <e ₂ | h ₂ > is considered to reflect the reliability of the transmission signal t ₂ determined by the MLD means 2-4.
Note that the likelihood weighting coefficient A in the present embodiment is the inner product value <e _t | h _t > itself due to the property of the left pseudo inverse matrix. Therefore, the likelihood weighting coefficient A in the present embodiment reflects the reliability of each transmission signal determined by MLD from the aspect of the transmission coefficient of the wireless transmission path through which each transmission signal passes, and the soft decision error for the MLD determination signal The error correction effect of the correction decoding can be expected as compared with the case where the conventional weight coefficient is used.

Next, a MIMO radio signal transmission method according to the first embodiment of the present invention will be described.
FIG. 4 is a flowchart showing the operation of the radio signal receiving apparatus 2 according to the first embodiment. Signals received by the receiving antennas 2-1-1 to 2-1 -M of the radio signal receiver 2 are transmitted through the frequency conversion means 2-2-1 to 2-2 -M and the transfer response matrix estimation means 2. -3.
In the transfer response matrix estimation means 2-3, the transfer response matrix H is estimated (step S11).
Next, in the likelihood weighting factor calculation unit 2-6, the N transmission signals t _k transmitted from the radio signal transmission device 1 using the transmission response matrix H estimated by the transmission response matrix estimation unit 2-3. A likelihood weighting factor A is calculated for (step S12).
Next, the likelihood weighting coefficient A calculated by the likelihood weighting coefficient calculating means 2-6 is accumulated (step S13).

Next, it is determined whether or not the radio signal receiver 2 has received the transmission signal T transmitted from the radio signal transmitter 1 (step S14). When the transmission signal T is received, “YES” is determined in the step S14, and the process proceeds to the step S15. On the other hand, when the transmission signal T is not received, the process of step S14 is repeated until the transmission signal T is received.
In step S14, when receiving a transmission signal T, the likelihood calculating unit 2-5-1~2-5-N, N-number of transmission signals ^(~ _{t k)} is determined, it is determined ^(~ Likelihood L _k is calculated for t _k ) (step S15).
Then, the likelihood _{L k} calculated for the N transmit signals ^(~ _{t k),} the stored A is multiplied by the value of _{L k} × A is determined as a soft decision value (step S16) .
Next, in the soft decision error correction decoding means 2-8-1 to 2-8 -N, soft decision error correction decoding is performed using the value of L _k × A and transmitted from the radio signal transmission apparatus 1. The transmitted signal T is reproduced (step S17).
Next, it is determined whether or not reception of the transmission signal T has been completed (step S18). When the reception of the transmission signal T is finished, the process is finished. On the other hand, if the reception of the transmission signal T has not ended, the process proceeds to step S19.
In step S19, it is determined whether or not the transfer response matrix can be estimated. If the transfer response matrix can be estimated, the process proceeds to step S12. On the other hand, if the transfer response matrix cannot be estimated, the process proceeds to step S14.

Next, a MIMO radio signal transmission system according to the second embodiment of the present invention will be described.
FIG. 5 is a block diagram of the radio signal transmitting apparatus 1 and the radio signal receiving apparatus 4 constituting the MIMO radio signal transmission system according to the second embodiment of the present invention.
The wireless signal transmission device 1 has the same configuration as the wireless signal transmission device 1 of FIG. The radio signal receiving apparatus 4 has the same configuration except for the MLD means 2-4 and the likelihood weight coefficient calculating means 2-6 of the radio signal receiving apparatus 2 of FIG. MLD unit 4-4 in the radio signal receiving apparatus 4 with a small Euclidean distance _{d 2} is the second, to output a smaller Euclidean distance _{d 1} to the likelihood weighting factor calculation unit 4-6 to the first.
Likelihood weight coefficient calculation means 4-6 in the radio signal receiving apparatus 4 is a likelihood weight from the transfer response matrix H estimated by the transfer response matrix estimation means 2-3 and the outputs d ₂ and d ₁ from the MLD means 4-4. The coefficient A × √ (d ₂ ² −d ₁ ² ) is calculated and output to the likelihood weight coefficient multiplication means 2-7-1 to 2-7 -N.

The likelihood weighting factor of the present embodiment is obtained by multiplying the conventional likelihood weighting factor (d ₂ ² -d ₁ ² ) and the weighting factor A used in the first embodiment by matching the dimensions, and A synergistic effect of the likelihood weighting factor can be expected.

Next, a MIMO radio signal transmission method according to the second embodiment of the present invention will be described.
FIG. 6 is a flowchart showing the operation of the radio signal receiving apparatus 4 according to the second embodiment.
The processes of steps S25, S26, and S27 are different from those of the radio signal receiving apparatus 2 according to the first embodiment. That is, in step S25, the likelihood calculating unit 1-5-1-1-5-N, it is determined N transmission signals ^(~ _{t k),} along with the likelihood _{L k} is calculated for it is determined The minimum Euclidean distance d ₁ and the second smallest Euclidean distance d ₂ are calculated for the transmission signal T.
In step S26, the likelihood _{L k} calculated for N transmission signals _{t k,} in addition to further √ of accumulated _{^{A (d 2 2 -d 1 2}} ) are multiplied, √ (d _{The value of 2} ² −d ₁ ² ) × A × L _k is obtained as the soft decision value.
In step S27, the soft decision error correction decoding means 2-8-1 to 2-8-N uses the value of √ (d ₂ ² -d ₁ ² ) × A × L _k to correct the soft decision error. Decoding is performed, and the transmission signal T transmitted from the wireless signal transmission device 1 is reproduced.
The processing of the other steps (steps S21 to S24, S28, S29) is the same as the processing (FIG. 4) in the first embodiment, and the description thereof is omitted.

Next, a MIMO radio signal transmission system according to the third embodiment of the present invention will be described.
The configuration of the MIMO radio signal transmission system according to the third embodiment of the present invention is the same as the configuration of the MIMO radio signal transmission system according to the second embodiment shown in FIG.
The difference from the MIMO radio signal transmission system according to the second embodiment is that the likelihood weighting coefficient calculating means 4-6 in the radio signal receiving apparatus 4 uses the transfer response matrix H and MLD means estimated by the transfer response matrix estimating means 2-3. The likelihood weighting factor A × d ₂ / d ₁ is calculated instead of the likelihood weighting factor A × √ (d ₂ ² −d ₁ ² ) from the outputs d ₂ and d ₁ from 4-4. The configuration other than the likelihood weight coefficient calculation unit 4-6 is the same as that of the second embodiment.
The likelihood weighting factor of the third embodiment uses a ratio as an amount that reflects the difference between d ₂ and d ₁ , and a further effect can be expected than the likelihood weighting factor of the second embodiment.

Next, a MIMO radio signal transmission method according to the third embodiment of the present invention will be described.
FIG. 7 is a flowchart showing the operation of the wireless signal receiving device 4 according to the third embodiment.
Steps S36 and S37 are different from the processing according to the second embodiment. That is, in step S36, and the likelihood _{L k} calculated for N transmission signals _{t k,} the accumulated ^{_{A √ (d 2 2 -d 1}} 2) rather _than, d 2 / _{d 1} is multiplied Then, the value of d ₂ / d ₁ × A × L _k is obtained as the soft decision value.
Further, in step S37, soft decision error correction decoding means 2-8-1 to 2-8-N performs soft decision error correction decoding using the value d ₂ / d ₁ × A × L _k. The transmission signal T transmitted from the wireless signal transmission device 1 is reproduced.
The processing of other steps (steps S31 to S35, S38, S39) is the same as the processing (FIG. 6) in the second embodiment, and the description thereof is omitted.

Next, a MIMO radio signal transmission system according to the fourth embodiment of the present invention will be described.
FIG. 8 is a block diagram of a radio signal transmitting apparatus 5 and a radio signal receiving apparatus 6 constituting a MIMO radio signal transmission system according to the fourth embodiment of the present invention.
In this embodiment, the MIMO radio signal system according to the first embodiment is applied to an OFDM (Orthogonal Frequency Division Multiplexing) scheme.
The radio signal transmission device 5 includes N IFFT (Inverse-Fast-Fourier Transform) means 5- following the N modulation means 1-2-1 to 1-2-N in the radio signal transmission apparatus 1 of FIG. 3-1 to 5-3 -N, and output signals T = (t ₁ , t ₂ ,..., T _N ) ^t of the modulation means 1-2-1-1-2 -N. OFDM modulation is performed by IFFT means 5-3-1 to 5-3-N.

  On the other hand, the radio signal receiving apparatus 6 includes M FFT (Fast-Fourier Transform) means 6-4-1 to 6-4-N in the preceding stage of the MLD means 2-4 in the radio signal receiving apparatus 2 of FIG. The transmission response matrix estimation means 2-3, the MLD means 2-4, the N likelihood calculation means 2-5-1 to 2-5-N, the likelihood weight coefficient calculation means 2-6, and the N pieces Likelihood weight coefficient multiplication means 2-7-1 to 2-7 -N are the transfer response matrix estimation means 2-3 and MLD means 2-4 in the radio signal receiving apparatus 2 (FIG. 1) of the first embodiment. N likelihood calculating means 2-5-1 to 2-5-N, likelihood weighting coefficient calculating means 2-6 and N likelihood weighting coefficient multiplying means 2-7-1 to 2-7-N Each processing is performed independently for each subcarrier component.

Ie, N number of transmission signal by MLD unit 2-4 for each sub-carrier component (~ T) = ((~ t 1), (~ t 2), ···, (~ t N)) is t determined is, N pieces of transmission signals for each subcarrier by the likelihood calculation unit 2-5-1~2-5-N (~ T) = ((~ t 1), (~ t 2), ···, (~ t N)) m × N number of likelihood with respect to t is calculated, each of N likelihood weighting coefficient a is calculated by the likelihood weighting coefficient calculation unit 2-6 for each sub-carrier component, the N likelihood weighting factor multiplier m × N number of likelihood for each subcarrier component by 2-7-1～2-7-N and N likelihood weighting factors (~ T) = ((~ t 1), (~ t 2 ), ···, are multiplied by each other (~ t N)) which t corresponds to the respective transmission signals (~ t j). The output signals of the respective likelihood weight coefficient multiplying means 2-7-j are outputted by the number of subcarriers, but all the subcarrier components are inputted to the same soft decision error correction decoding means 2-8-j. Soft decision error correction decoding means 2-8-j performs error correction on the input signals of all the sub-carrier components, the original data signal sequence # (~ j) are reproduced.
Error correction is similarly performed in N soft decision error correction decoding units 2-8-1 to 2-8 -N, and finally N original data signal sequences # (1 to 1), # (2 to 2 ), ···, # (~ N ) is played.

Order all subcarrier component of the output signal of likelihood weighting coefficient multiplier unit 2-7-j input to the soft-decision error correction decoding means 2-8-j is the order of the original data signal sequence # (~ ^j) Correspond. If an interleaver is performed in the wireless signal transmission device 5, a deinterleaver corresponding to the interleaver is performed in the wireless signal reception device 6.
In the present embodiment, the OFDM scheme is further applied to the MIMO radio signal transmission system according to the first embodiment, so that the first embodiment can be compared with the first embodiment in a frequency selective fading environment as compared with the first embodiment. The soft decision gain can be further improved by using the likelihood weighting coefficient A.

Next, a MIMO radio signal transmission method according to the fourth embodiment of the present invention will be described.
The flowchart showing the MIMO radio signal transmission method according to the fourth embodiment is the same as the flowchart (FIG. 4) showing the MIMO radio signal transmission method according to the first embodiment.
However, in the flowchart of FIG. 4, the processing in steps S11 to S16 and S19 is different in that the processing is performed in units of subcarriers, and the processing in steps S17 and S18 is different in that the processing is performed in all subcarriers.

Next, a MIMO radio signal transmission system according to the fifth embodiment of the present invention will be described. The configurations of the radio signal transmitting apparatus and the radio signal receiving apparatus constituting the MIMO radio signal transmission system according to the present embodiment are the same as the configuration of the MIMO radio signal transmission system according to the fourth embodiment (FIG. 8). The MIMO radio signal transmission system according to the fifth embodiment is an application of the OFDM scheme to the MIMO radio signal transmission system according to the second embodiment.
In FIG. 8, IFFT means 5-3-1 to 5-3 -N and FFT means 6-4-1 to 6-4 -N are arranged in the wireless signal transmitter 5 and the wireless signal receiver 6, respectively. Yes. The processes from the error correction coding means 1-1-1 to 1-1-N to the likelihood weight coefficient multiplying means 2-7-1 to 2-7-N are the MIMO radio signal transmission system according to the second embodiment. The same processing as the processing from the error correction coding means 1-1-1 to 1-1-N to the likelihood weight coefficient multiplying means 2-7-1 to 2-7-N in FIG. To do. The processes of the soft decision error correction decoding means 2-8-1 to 2-8 -N after the likelihood weight multiplication means 2-7-1 to 2-7 -N are the same as those in the fourth embodiment (FIG. 8). This is the same as the decision error correction decoding means 2-8-1 to 2-8-N.

In the MIMO radio signal transmission system according to the fifth embodiment, by applying the OFDM scheme to the MIMO radio signal transmission system according to the second embodiment, the likelihood weighting coefficient A × √ in a frequency selective fading environment. The soft decision gain can be further improved by using (d ₂ ² -d ₁ ² ).

Next, a MIMO radio signal transmission method according to the fifth embodiment of the present invention will be described.
The flowchart showing the MIMO radio signal transmission method according to the fifth embodiment is the same as the flowchart (FIG. 6) showing the MIMO radio signal transmission method according to the second embodiment.
However, in the flowchart of FIG. 6, the processing in steps S21 to S26 and S29 is different in that the processing is performed in units of subcarriers, and the processing in steps S27 and S28 is different in that the processing is performed in all subcarriers.

Next, a MIMO radio signal transmission system according to the sixth embodiment of the present invention will be described. The configurations of the radio signal transmitting apparatus and the radio signal receiving apparatus constituting the MIMO radio signal transmission system according to the present embodiment are the same as the configuration of the MIMO radio signal transmission system according to the fourth embodiment (FIG. 8). The MIMO radio signal transmission system according to the sixth embodiment is an application of the OFDM scheme to the MIMO radio signal transmission system according to the third embodiment.
In FIG. 8, IFFT means 5-3-1 to 5-3 -N and FFT means 6-4-1 to 6-4 -N are arranged in the wireless signal transmitter 5 and the wireless signal receiver 6, respectively. Yes.

The processes from the error correction coding means 1-1-1 to 1-1-N to the likelihood weight coefficient multiplying means 2-7-1 to 2-7-N are the MIMO radio signal transmission system according to the third embodiment. The same processing as the processing from the error correction coding means 1-1-1 to 1-1-N to the likelihood weight coefficient multiplying means 2-7-1 to 2-7-N in FIG. To do.
The processes of the soft decision error correction decoding means 2-8-1 to 2-8 -N after the likelihood weight multiplication means 2-7-1 to 2-7 -N are the same as those in the fourth embodiment (FIG. 8). This is the same as the decision error correction decoding means 2-8-1 to 2-8-N.
In the MIMO radio signal transmission system according to the sixth embodiment, by applying the OFDM scheme to the MIMO radio signal transmission system according to the third embodiment, the likelihood weighting factor A × d in a frequency selective fading environment. soft decision gain enhancement according to the use of _{2 /} d ₁ can be further expected.

Next, a MIMO radio signal transmission method according to the sixth embodiment of the present invention will be described.
The flowchart showing the MIMO radio signal transmission method according to the sixth embodiment is the same as the flowchart (FIG. 7) showing the MIMO radio signal transmission method according to the third embodiment.
However, in the flowchart of FIG. 7, the processing in steps S31 to S36 and S39 is different in that the processing is performed in units of subcarriers, and the processing in steps S37 and S38 is different in that the processing is performed in all subcarriers.

In order to quantitatively show the effects of likelihood weighting factors A, A × √ (d ₂ ² −d ₁ ² ), A × d ₂ / d ₁ according to embodiments of the present invention, these and conventional likelihood weighting factors FIG. 9 is a graph showing a simulation evaluation of PER (Packet Error Rate) characteristics when (d ₂ ² −d ₁ ² ) and likelihood weighting factors are not used.
The simulation conditions in FIG. 9 were as follows. That is, the number of transmission antennas and the number of reception antennas are both two. In addition, the OFDM system was applied, and the OFDM system was compliant with the 5 GHz wireless LAN standard IEEE802.11a. The subcarrier modulation method used 64QAM and the coding rate used 3/4. The packet length was 64 bytes. In the MIMO channel model, each wireless transmission path is a Rayleigh fading channel specified by IEEE 802.11a and has a delay dispersion of 100 ns. In addition, the spatial correlation was made uncorrelated, and the simulation was performed under the conditions of channel estimation and synchronization ideal.

From FIG. 9, the PER characteristic is improved in the order of (no likelihood weight) <(d ₂ ² −d ₁ ² ) <A <A × √ (d ₂ ² −d ₁ ² ) <A × d ₂ / d _1. You can see that From this, the effect of the likelihood weighting coefficient of this embodiment can be confirmed.

  As mentioned above, although embodiment of this invention was described with reference to drawings, it is not limited to these embodiment about a concrete structure, The design change etc. of the range which does not deviate from the summary of this invention are possible. It is.

1 is a block diagram of a MIMO radio signal transmission system according to a first embodiment of the present invention. FIG. It is a figure which shows the structure of the pilot signal which implement | achieves estimation of the transfer response matrix H. It is the figure which showed the physical meaning of the likelihood weighting coefficient of this embodiment. It is a flowchart which shows the flow of a process of the MIMO radio signal transmission system by 1st Embodiment. It is a block diagram of a MIMO radio signal transmission system according to the second and third embodiments of the present invention. It is a flowchart which shows the flow of a process of the MIMO radio signal transmission system by 2nd Embodiment. It is a flowchart which shows the flow of a process of the MIMO radio signal transmission system by 3rd Embodiment. It is a block diagram of a MIMO radio signal transmission system according to a fourth embodiment of the present invention. It is a graph which shows the simulation evaluation result of the PER characteristic at the time of using the likelihood weighting coefficient of embodiment of this invention. It is the figure which showed the physical meaning of the conventional likelihood weighting coefficient.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Radio signal transmitter 1-1-1 to 1-1-N ... Error correction coding means 1-2-1 to 1-2-N ... Modulation means 1-3 ... Path Transmission response matrix estimation pilot signal generation means 1-4-1 to 1-4-N signal multiplexing means 1-5-1 to 1-5-N frequency conversion means 1-6-1 to 1-1 -6-N ... transmitting antenna 2 ... radio signal receiving apparatus 2-1-1 to 2-1-M ... receiving antenna 2-2-1-2-2-M ... frequency conversion means 2-3 ... Transmission response matrix estimation means 2-4, 4-4 ... MLD means 2-5-1 to 2-5-N ... Likelihood calculation means 2-6, 4-6. Likelihood weight coefficient calculation means 2-7-1 to 2-7-N ... Likelihood weight coefficient multiplication means 2-8-1 to 2-8-N ... Soft decision error correction decoding means 5-3 -1-5-3-N IFFT Stage 6-4-1~6-4-N ··· FFT means

Claims (12)

A radio signal T (T = (t ₁ , t ₂ ,...) Provided with N (N is an integer of 2 or more) transmission antennas and subjected to error correction coding using the same frequency from the transmission antennas. means t _N) ^t. also, ^t denotes a transposed vector. the, i of t _i is the radio signal transmission device for transmitting.) meaning transmission antenna number,
M reception antennas (M is an integer equal to or greater than 1) are provided, and transmission responses of N × M wireless transmission paths that are combinations of the N transmission antennas and the M reception antennas are used as elements. Transfer response matrix H (H (i, j) component h _ij means a transfer response between transmission antenna j and reception antenna i. J is an integer of 1 to N and i is an integer of 1 to M). ) Estimation means,
A replica H × (^ T _s ) of the received signal is generated from the transmission response matrix H estimated by the estimation means and the transmission signal candidate point (^ T _s ), and the replica H × (^ T _s ) and the M Euclidean distance from the received signal R (R = (r ₁ , r ₂ ,..., R _M ) ^t of j receiving antennas, and j of r _j means a receiving antenna number). the difference | and the minimizing ^(~ T) of the original transmitted signal and determines MLD _{unit, | R-H × (^} T s)
In MIMO radio signal transmission system comprising a radio signal receiving device having a soft-decision error correction decoding means for soft-decision error correction decoding was determined ^{(~ T)} by the MLD unit,
The radio signal receiving apparatus uses a left-side pseudo signal of a transfer response matrix H as a likelihood weighting factor for a transmission signal t _k (k is an integer of 1 or more and N or less) as soft decision information used by the soft decision error correction decoding means. Inverse matrix (H ^H H) ⁻¹ H ^H (sH and t column components of (H ^H H) ⁻¹ H ^H (s is an integer from 1 to N, t is an integer from 1 to M) h ′ _The sum of the squares of the absolute values of the k-th element component of ( _st )) in the column direction and then the reciprocal square root

A MIMO radio signal transmission system characterized by using the above. As a likelihood weighting factor for the transmission signal t _k used for the soft decision error correction decoding means, the difference between the Euclidean distance | R−H × (^ T) | at the A and the total transmission signal candidate points (^ T _s ) Of these, the product A × √ (d ₂ ²⁻ d ₁ ² ) of the square root √ (d ₂ ² −d ₁ ² ) of the square difference between the _second smallest distance d ₂ and the minimum distance d ₁ is used. The radio signal transmission system according to claim 1. As a likelihood weighting factor for the transmission signal t _k used for the soft decision error correction decoding means, the difference between the Euclidean distance | R−H × (^ T) | at the A and the total transmission signal candidate points (^ T _s ) of, MIMO radio signal transmission system according to claim 1, characterized by using the second a product a × _d 2 / _{d 1} of a small distance _{d 2} and the ratio of the minimum distance _{d 1.} The transmission signal T is OFDM signal, claim, wherein said a likelihood weighting factor for the sub-carrier component of the transmission signal t _k to be used for soft decision error correction decoding means, using said A subcarrier components f 2. The MIMO radio signal transmission system according to 1. The transmission signal T is OFDM signal, the soft as the likelihood weighting factor for the sub-carrier component of the transmission signal t _k used to determine the error correction decoding means, wherein the subcarrier components ^{_{f A × √ (d 2 2}} -d 1 The MIMO radio signal transmission system according to claim 2, wherein ² ) is used. The transmission signal T is OFDM signal, as the likelihood weighting factor for the sub-carrier component of the transmission signal t _k to be used for the soft-decision error correction decoding means, using said A × d _{2 /} d ₁ of the subcarrier components f The MIMO radio signal transmission system according to claim 3. Radio signal T (T = (t ₁ , t ₂ , T ₂ , T ₂ , T ₂ , T ₂ , T ₂ , T ₂ , T ₂ , T ₂ , T ₂ , T ₂ , T ₂ , T ₂ , T ₂ , _., refers to t ^{N) t. also, t} denotes a transposed vector. the, i of _{t i} denotes the transmission antenna number. the)
A MIMO radio that is received by a radio signal receiving apparatus having M (M is an integer of 1 or more) receiving antennas and having a transfer response matrix estimating means, likelihood weighting coefficient calculating means, MLD means, and soft decision error correction decoding means In the signal transmission method,
By the transfer response matrix estimation means, the transfer response matrix H ((i, j) component h _ij of H means a transfer response between the transmission antenna j and the reception antenna i. J is an integer of 1 to N, i Is an integer from 1 to M.)
The likelihood weighting coefficient calculation means uses the transmission response matrix H estimated by the transmission response matrix estimation means, and uses the likelihood weighting coefficient for N transmission signals t _j.

A step of calculating
Accumulating likelihood weighting factor A calculated by the likelihood weighting factor calculating means;
Determining N transmission signals t _j by the MLD means, and calculating a likelihood L _j for the determined t _j ;
Obtaining a value L _j × A obtained by multiplying the likelihood calculated by the MLD means and the accumulated A as a soft decision value;
Correcting and decoding the soft decision value obtained by the MLD means by the soft decision error correction decoding means;
A MIMO radio signal transmission method comprising: Of the Euclidean distance difference | R−H × (^ T) | between the A and all transmission signal candidate points (^ T) by the MLD means, the second smallest distance d ₂ , the minimum distance d _1, and the square difference And calculating the product A × √ (d ₂ ² −d ₁ ² ) of the square root √ (d ₂ ² −d ₁ ² ) of
The MIMO radio signal transmission method according to claim 7, wherein A × √ (d ₂ ² −d ₁ ² ) is used as the likelihood weighting factor. Of the Euclidean distance difference | R−H × (^ T) | between the A and all transmission signal candidate points (＾ T) by the MLD means, the ratio of the _second smallest distance d ₂ to the ratio of the minimum distance d ₁ Further comprising calculating a product A × d ₂ / d ₁ ,
The MIMO radio signal transmission method according to claim 7, wherein A × d ₂ / d ₁ is used as the likelihood weighting factor. The transmission signal T is an OFDM signal, and the transmission signal t _k used in the step of correcting and decoding the soft decision value obtained by the MLD means by the soft decision error correction decoding means (k is an integer of 1 to N) The MIMO radio signal transmission method according to claim 7, wherein A is used as a likelihood weighting factor for each subcarrier component. The transmission signal T is an OFDM signal, and the transmission signal t _k used in the step of correcting and decoding the soft decision value obtained by the MLD means by the soft decision error correction decoding means (k is an integer of 1 to N) The MIMO radio signal transmission method according to claim 8, wherein A × √ (d ₂ ² −d ₁ ² ) is used as a likelihood weighting factor for each subcarrier component. The transmission signal T is an OFDM signal, and the transmission signal t _k used in the step of correcting and decoding the soft decision value obtained by the MLD means by the soft decision error correction decoding means (k is an integer of 1 to N) The MIMO radio signal transmission method according to claim 9, wherein A × d ₂ / d ₁ is used as a likelihood weighting factor for each subcarrier component.

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