JP2005278127A - Micro-strip antenna and control method - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a small micro-strip antenna with high gain and advantages in portability and capable of controlling a radiation pattern and radiation directivity, and to provide its control method. <P>SOLUTION: Two or more almost λ/4 short-circuiting type micro-strip antenna elements arranged symmetrically in parallel on the same substrate, a power feeding means for feeding power to the micro-strip antenna elements, and a power feeding control means for managing the element fed with the power and a power feeding mode are provided. The power feeding mode is set so as to offer radiation directivity near to a desired radiation pattern. In addition, an angle or a width of a radiation beam may be adjusted, independently from a dielectric constant of the substrate, by changing the height or a setting angle of a metal sidewall. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a portable microstrip antenna and a radiation control method for the microstrip antenna.

  In recent years, there has been great interest in multimedia communication using a computer network, and the use of high-speed data communication has been promoted in public communication networks. With the rapid spread of mobile communications represented by mobile phones, the use of multimedia is also demanded for mobile satellite communications, and there is a need for high-speed and large-capacity communications systems that support multimedia communications. .

  For example, currently, mobile satellite communication systems using geostationary satellites capable of voice / data communication using mobile terminals using the S-band (2.6 / 2.5 GHz) frequency, and high quality similar to CD Studies are being conducted on satellite digital multimedia broadcasting systems for mobiles that can transmit voice and images. In addition, among the plans for the technical test satellite VIII (ETS-VIII), which will be launched soon, it is planned to carry out technology development and demonstration experiments necessary for realizing these systems.

  ETS-VIII has an S-band service link and a Ka-band feeder link, uses a high-gain S-band antenna with a size of 19m x 17m, and is used for voice / data communication services and in-vehicle receivers for mobile terminals. It contributes to high-quality audio broadcasting. The present inventors mount a high-speed data exchange repeater on this ETS-VIII and conduct an experiment of high-speed data communication for an on-vehicle or portable earth station.

  Satellite communication devices mounted on automobiles are limited in size and weight, power consumption, etc., and are sufficient systems to use high-gain antennas for high-quality voice communication and high-speed data communication. It is necessary to obtain the performance of. For this reason, the on-vehicle high gain antenna has become one of the most important elemental technologies in the construction of a land mobile satellite communication system, and research and development are being carried out in various organizations around the world.

  In S-band in-vehicle mobile satellite communication in the ETS-VIII communication system, the elevation angle is 48 °, which is the satellite direction in Tokyo, and the elevation angle is within the range of ± 10 ° taking into account errors due to vibrations of the running car. An antenna gain of 6 dBi or more is required in the direction. Conical (conical) beams are effective for satisfying such requirements, and various studies are currently being conducted on conical beams, but the coverage is generally narrow.

In recent years, the trend of electronic devices has been to make them smaller, lighter, and thinner. Reflecting these trends, research on flat antennas that can be configured thinly and compactly has been actively conducted. A typical example of such a small planar antenna suitable for portability such as in-vehicle is a microstrip antenna.
However, it has been difficult to obtain a sufficiently high gain with the microstrip antenna in the prior art. Further, it has been impossible to appropriately control the radiation pattern and the beam direction and width according to various demands.

  Therefore, an object of the present invention is to provide a microstrip antenna that is excellent in portability, is small and has high gain, and can control a radiation pattern and radiation directivity, and a method for controlling the microstrip antenna.

The microstrip antenna of the present invention that solves the above problems has the following configuration.
That is, a plurality of substantially λ / 4 short-circuited microstrip antenna elements arranged in parallel on the same substrate in a symmetrical arrangement, power feeding means for feeding power to the microstrip antenna elements, power feeding elements and power feeding And a power supply control means for controlling the style.

  Here, four or more approximately λ / 4 short-circuited microstrip antenna elements may be provided and arranged on the same arc.

  Alternatively, the feeding may be performed by a pin feeding method, and the feeding point may be set at a position offset by about 30% to match the microstrip antenna element and the feeding system.

  The shape of the approximately λ / 4 short-circuited microstrip antenna element may be approximately rectangular, and the dimensional ratio of the long and short sides may be approximately 1: 2.

  A metallic side wall whose height or installation angle can be adjusted may be erected on the peripheral edge of the substrate on which a plurality of substantially λ / 4 short-circuited microstrip antenna elements are arranged.

  The height of the metal side wall may be about 1 / 4λ, which is a value corresponding to the propagation wavelength λ.

  The attachment angle of the metal side wall may be about 90 °.

  The patch portion of the substantially λ / 4 short-circuited microstrip antenna element may be integrated with the metal conductor via a conductive tape.

  The method for controlling a microstrip antenna according to the present invention includes a plurality of substantially λ / 4 short-circuited microstrip antenna elements arranged in parallel on the same substrate in a symmetrical arrangement, and power feeding for feeding power to the microstrip antenna elements. The radiation pattern is adjusted by selecting the elements to be fed simultaneously using a microstrip antenna including the means, the feeding element and the feeding control means for controlling the feeding mode.

  Here, a metal side wall whose height or installation angle can be adjusted is erected on the peripheral portion of the substrate on which the plurality of substantially λ / 4 short-circuited microstrip antenna elements are arranged, and the height of the metal side wall or The angle or width of the radiation beam may be adjusted by changing the installation angle.

  The feed mode is set so that the radiation directivity is close to the desired radiation pattern, and then the angle or width of the radiation beam is adjusted independently of the dielectric constant of the substrate by changing the height or installation angle of the metal side wall. May be.

  When providing a beam having symmetry or performing fine beam switching, it is preferable to feed power to two substantially λ / 4 short-circuited microstrip antenna elements arranged opposite to each other in the same phase.

  When a strong beam is provided in one direction, it is preferable to feed power to one substantially λ / 4 short-circuited microstrip antenna element.

  According to the present invention, it is small and thin, easy to manufacture and inexpensive, has a clear feature as an element, uses a microstrip antenna with a narrow frequency band, has excellent portability, and is small and has high gain. In addition, a microstrip antenna that can freely control the radiation pattern and radiation directivity, and a method for controlling the microstrip antenna can be obtained.

  In this example, a microstrip antenna for beam tracking configured by combining λ / 4 short-circuited microstrip antennas is illustrated, but the present invention can be applied to other similar types of microstrip antennas. The design can be changed as appropriate.

The basic configuration of the microstrip antenna is that a circular or square planar circuit resonance element is formed by etching or the like on a dielectric substrate such as a thin copper foil-attached printed circuit board and used as a radiator. The element size of the microstrip antenna is usually configured to be about half a wavelength, and the fundamental mode (TM ₁₀₀ mode) is provided as the test excitation mode.

The microstrip antenna has the following characteristics.
Since the microstrip antenna uses a planar circuit resonant element formed on an extremely thin low dielectric constant substrate (usually about several millimeters) as a radiator, a small and thin antenna can be manufactured.

  In addition, the microstrip element is usually configured by a photoetching technique. Therefore, if the original drawing of the antenna element is performed by CAD or the like, a large number of microstrip antenna elements can be duplicated from the same original drawing, which is suitable for mass production. Furthermore, since an extremely low cost film substrate and foamed foam substrate have been developed recently, a low cost planar antenna can be realized even in a high frequency region such as the SHF band.

Furthermore, the microstrip antenna can analyze the most important characteristics as an antenna, that is, characteristics such as radiation efficiency η, input impedance Z _in , radiation pattern directivity gain, and the like, and the analysis results thereof. Agrees well with the measured values within a design-significant range.

  In normal communication, one antenna is used for both transmission and reception. Therefore, the antenna must have sufficient electrical characteristics at both transmission and reception frequencies. Generally, a band of about 4 to 10% of the center frequency is required, but the band of the microstrip antenna is usually 1% or less of the center frequency, and various studies for widening the band have been conducted. Yes.

  When a leakage electric field is formed at the edge of the microstrip antenna, radio waves are radiated therefrom. For example, what contributes to radiation in the front direction facing the plane of the antenna is a leakage electric field that can be generated in the left and right direction of the antenna element plane, and does not contribute to radiation because the leakage electric field in the vertical direction cancels each other.

Here, if the thickness t of the substrate is sufficiently small with respect to the propagation wavelength λ, the TM _mn0 mode is excited. The electromagnetic field of the TM _mn0 mode can be obtained from the following wave equation.

  Here, the vector n is a unit normal vector at the open boundary. Moreover, k is a plane wave wave number and is given by the following equation.

When Equation 1 is solved in consideration of the boundary condition Equation 2, the E _z component of the TM _mn0 mode is obtained as follows.

Here, E ₀ is an arbitrary constant, m and n are integers, V ₀ is the peak voltage at the end of the microstrip antenna, and can be expressed as V ₀ = tE ₀ . Substituting this E _z into Maxwell's equation and paying attention to the condition of the TM wave (H _z = 0), and obtaining other electromagnetic field components, the following equation is obtained.

From the above, the internal electromagnetic field of the TM ₁₀₀ mode, which is the basic mode, is displayed as follows.

Further, the resonance frequency can be obtained from Expression 4 and can be approximated by Expressions 13 to 15 below.

（ただし、Ｃ；光速、ａ_ｅｆｆ；実効パッチ幅、ａ；パッチ幅、ｔ；基板の厚さ、ε_ｅｆｆ；基板の実効比誘電率、ε_ｒ；基板の比誘電率）

(Where C: speed of light, a _eff : effective patch width, a: patch width, t: substrate thickness, ε _eff : effective relative dielectric constant of substrate, ε _r ; relative dielectric constant of substrate)

The following three typical power supply methods can be used for the microstrip antenna.
The first power supply method is a pin power supply method. This feeds power by attaching a power feed pin, which is an extension of the core of a coaxial line, to a patch through a dielectric substrate from the ground plane conductor side.
The second power supply method is a microstrip power supply method. In this configuration, a feeding system using a microstrip antenna element and a strip line is integrally formed on the same plane. In this method, a microwave semiconductor element or an MMIC (monolithic microwave integrated circuit) element can be integrated with the microstrip antenna element, and the multi-functionality of the power feeding system can be facilitated.
The third power feeding method is an electromagnetic coupling type power feeding method. This is because the ground conductor of the feeding system using a strip line or the like formed on another substrate is tightly loaded on the back of the patch, and an opening for electromagnetic coupling such as a slit is provided on the tangent plane. The antenna element is excited.
The pattern of the microstrip antenna excited by these various feeding methods exhibits almost equivalent characteristics except for the influence of unnecessary radiation from the feeding system.

In general, the input impedance Z _in of the microstrip antenna excited in the main mode differs depending on the feed point setting position F. Z _{in in} the resonance region and the frequency region in the vicinity thereof is proportional to F and increases to several hundreds Ω from the center to the open boundary. Accordingly, when power is supplied from the open boundary (F = a), Z _in at resonance usually exhibits a high impedance characteristic of 300 to 500Ω, and matching with a power supply system having a characteristic impedance of about 50Ω is required.

In the pin feeding method, matching can be achieved by offsetting the setting position F of the feeding point. Usually, the feeding point is set at a position offset by about 30% (F / a = 0.3) and matching is often performed.
Therefore, in this embodiment, a pin feeding method is adopted.

Antenna tracking in land mobile communications is difficult compared to tracking in aircraft and ships for the following reasons.
That is, shadowing or blocking occurs due to trees or buildings. The movement of the moving body is agile and frequent left and right turns occur. Magnetic disturbance due to the artificial structure occurs.

  Various antenna tracking control methods have been studied and reported for these problems. Typical examples include a closed-loop method that uses the reception level of signals transmitted from satellites, and tracking sensors such as azimuth sensors on mobile units, and the satellite direction is determined from azimuth information obtained from them. There is an open-loop system that is required, but when considering an in-vehicle antenna, an open-loop system that does not use a signal from a satellite is suitable.

  Examples of azimuth measuring sensors used in the open loop method include a magnetic azimuth meter and a gyro. A vibrating gyroscope or an optical fiber gyroscope (FOG) having no mechanical portion may be used.

An equatorial coordinate system is used as the coordinate system representing the motion of the artificial satellite. The equator coordinate system takes the center of the earth as the coordinate origin, the z axis on the earth's rotation axis (normal direction of the equator plane), and the x axis in the direction of the equinox that is the intersection of the equator and the ecliptic. It is a right-handed Cartesian coordinate system. That is, this coordinate system is expressed by a geocentric distance r representing an angle from the equator plane, an elevation angle φ (Elevation Angle), and an azimuth angle σ (Azimuth Angle).
On the other hand, the coordinate system normally used in the antenna is polar coordinates (R, θ, φ). Here, θ represents an inclination angle from the vertical direction in the vertical plane.
The relationship between the two coordinate systems is as follows. In this embodiment, the antenna characteristics will be described using the polar coordinate system.

  The electromagnetic field at a sufficiently far distance (R, θ, φ) radiated by an arbitrary antenna placed at the origin of the polar coordinate system can be expressed by the following equation.

（ただし、λは自由空間波長、ｋは平面波波数２π／λ）

(Where λ is the free space wavelength and k is the plane wave number 2π / λ)

U (θ, φ) is called a directivity function, and is determined only by θ and φ, that is, the direction, regardless of the distance from the antenna, and includes all the properties related to directivity.
In addition, what is measured by cutting the directivity U (θ, φ) on a certain plane is called an antenna pattern.
Further, a pattern measured in a plane containing an excited linearly polarized electric field vector is called E plane directivity, and a pattern measured in a plane perpendicular to it and containing a magnetic field is called H plane directivity. A cut plane is called a cut plane. Call.

  The gain of an antenna is defined as the ratio of the power radiated from an antenna per unit cubic angle in an arbitrary direction to the power radiated per unit solid angle from an isotropic antenna supplied with the same power. . The isotropic antenna here is a virtual omnidirectional antenna that is lossless and radiates an electromagnetic field of uniform strength in all directions, and is used as a reference for the gain of the antenna.

If the power intensity of a certain antenna in the (θ, φ) direction is F (θ, φ), the gain G (θ, φ) is given by the following equation. However, _{P 0} is the power supply.
Moreover, the gain of the following formula includes a loss at the antenna, and is also called a power gain.

When a load (antenna part) different from the characteristic impedance of the line is connected to the transmission line (circuit part), reflection of electromagnetic waves occurs, resulting in transmission loss. In order to evaluate the magnitude of this reflection, the return loss is defined by the following equation. However, S ₁₁ represents the reflection coefficient is a measurable parameter in the network analyzer.

The resonance frequency, S ₁₁ becomes small to take matching between the circuit portion and the antenna portion, it becomes small return loss. Using this fact, the frequency with the smallest return loss can be determined as the resonance frequency.

Next, defining the directivity gain as a gain that is determined only by the relative directivity, regardless of the loss at the antenna, it means that `` the power intensity in a specific direction and the total radiated power averaged per unit solid angle The ratio can be described. When the power radiated in the entire space from the antenna and P _t, expressed by the following equation. However, ω represents a solid angle.

Since the average power per unit solid angle is P _t / 4π, the directivity gain D (θ, φ) can be expressed by the following equation.

Here, the relationship between P ₀ and P _t can be expressed by the following equation because an antenna generally has a loss. Here, P ₁ is the internal power loss of the antenna. Also, η is called radiation efficiency and represents how much power supplied to the antenna is radiated into space.

  From Equations 18 and 22, the ratio of antenna gain (power gain) and directivity gain can be expressed by the following equation.

In general, the value G (θ, φ) of Equation 18 in the maximum radiation direction (θ ₀ , φ ₀ ) is called an absolute gain and is expressed by a decibel value dBi.

Based on the above, focusing on the characteristics of the λ / 4 short-circuited microstrip antenna (hereinafter referred to as λ / 4SMSA), which will be described later, the λ / 4 short-circuiting microstrip antenna for beam tracking (hereinafter referred to as λ / 4SMSA for BT) will be studied. did.
FIG. 1 is a perspective explanatory view showing the basic configuration of a λ / 4SMSA for BT.
First, the antenna portion has a structure in which a dielectric substrate (12) is placed on a metal conductor (11), and λ / 4 SMSA elements (20) are loaded in an annular shape at equal intervals on the upper surface thereof. For example, Non-Patent Document 1 discloses this.
Nakajo, Teshirogi: A λ / 4 short-circuited circularly polarized microstrip antenna with a good axial ratio over a wide angle, IEICE (B), J71-B, No. 7, pp. 1420-1424, Jul. 1999

  The surface toward the center of the λ / 4 SMSA element is short-circuited to the metal conductor (11) through the dielectric substrate (12). Here, the characteristic impedance is set to 50Ω in order to match the coaxial line and the antenna.

A microstrip antenna generally tends to have a lower gain as the number of operation modes becomes higher. However, since this antenna operates with TM ₁₀₀ waves, which is the basic mode of a microstrip antenna, a higher gain can be expected compared to a TM ₂₁₀ mode microstrip antenna generally used for beam tracking.

The resonance frequency of the TM ₂₁₀ mode microstrip antenna is derived from the following equation.

Where c is the speed of light, a _e is the effective radius of the patch, ε _r is the relative permittivity of the substrate, X _nm represents the derivative of the Bessel function, and X ₂₁₀ = 3.0542 in the TM ₂₁₀ mode.
From this formula, in the TM ₂₁₀ mode, when operating at 2.5 GHz, a patch having a diameter of about 78 mm is required, and the size of the substrate is about 200 mm in diameter.

On the other hand, the λ / 4SMSA for BT has a feature that the size derived from the resonance frequency equation of the rectangular microstrip antenna can be reduced to about ¼ for one element as will be described later. In this embodiment, the λ / 4SMSA for 6-element BT, which is the maximum size in this embodiment, has an element diameter of 130 mm and a substrate diameter of 200 mm, and can be configured to be almost the same size as a microstrip antenna excited in the TM ₂₁₀ mode. .

  This antenna has various feeding variations, and has an advantage that various beam patterns can be generated by using it together with a switch. In this embodiment, only the case of 1 power supply and 2 power supply is described, but other power supply modes can be used as appropriate.

Non-Patent Document 2 discloses λ / 4SMSA, which is the basis of λ / 4SMSA for beam tracking of this embodiment.
Haneishi, Matsui, Saito: A Study on Miniaturization of Microstrip Antenna, Shingaku J, J71-B, pp. 1378-1380, Nov. 1988

The basic configuration of λ / 4SMSA is the same as that of an ordinary microstrip antenna, but one end of the patch is short-circuited to a metal conductor. This antenna is less than half of the patch width derived from Equation 13 at the same resonance frequency by short-circuiting the zero-potential surface of a square microstrip antenna excited with TM ₁₀₀ waves, which is the fundamental mode of the microstrip antenna. It has the feature of resonating with the size.

When λ / 4SMSA is fed by a probe, a reverse-phase mode and a common-mode can be considered in the current path. The current from the power supply probe flows from the probe to the short circuit plate because the short circuit plate is near the power supply probe.
The currents in the reverse-phase mode paths cancel each other and do not contribute to the resonance of the antenna. On the other hand, the current of the common-mode path that passes from the probe to the lower side of the plate and folds back at the end and flows through the upper surface has a length of λ / 2, and the direction of the current on the probe is opposite. The phase is delayed by π, resulting in the same phase.
Thus, in λ / 4SMSA, in-phase current flows through the short-circuit plate and the probe, and when the current on the top surface of the patch is viewed, it appears that current flows from the short-circuit plate toward the end of the antenna. At this time, the path length of the current is λ / 2 and resonance occurs.

At this time, the resonance frequency f _r of the antenna is obtained from the following equation.

Here, L ₁ and L ₂ represent the vertical and horizontal widths of the patch, respectively, and f ₁ and f ₂ are the resonance frequencies of the modes excited in the L ₁ direction and the L ₂ direction, respectively.
In addition, the following equation holds.

From Equation 25, when the entire cross-section short-circuiting the width s of the short-circuit plane (s = L _2), short-circuit plane - since the open face direction is dominant, the influence of the resonant frequency be varied width L ₂ of the antenna Becomes smaller. It is possible to further reduce the size if shortened L ₁ direction by using this.
Further, when the width s of the short-circuit plane is reduced, this antenna is equivalently loaded with an inductance, and the resonance frequency is lowered, so that the resonance frequency can be easily controlled.

The parameters of λ / 4SMSA are as follows.
Dielectric substrate;
Relative permittivity ε _r = 2.17
Substrate thickness t = 1.524mm
Substrate size d = 60 × 60mm
patch;
Patch length L ₁ = 19.15 mm
Patch width L ₂ = 19.15 mm
Short face width s = 19.15mm

  As a result of obtaining return loss characteristics and vertical plane directivity for λ / 4SMSA, although return loss and directivity gain are greatly deteriorated, λ / 4SMSA is almost the same as 2 / λ microstrip antenna patch. It turns out that it resonates at frequency.

In BT / 4SMSA for BT, a method of supplying power in phase to two opposing patches was studied.
FIG. 1 shows a configuration of a λ / 4 short-circuited microstrip antenna that feeds two opposing patches (20) in the same phase.
The circuit unit includes a sensor unit for tracking the satellite, a plurality of power distributors, and a switch circuit for switching them.

  As a power feeding method, first, patch elements (20) are paired with opposing patch elements (20) and (20) such as P1-P5, P2-P6, P3-P7, and P4-P8, and this is switched by a switch. . The power from the power source is divided into two by the power distributor, and is fed to the opposing patches (20) and (20) by the pin feeding method.

By feeding the pair of patch elements (20) and (20) in the same phase, a strong M-shaped beam is generated in the element facing direction, and a weak beam is generated in the orthogonal direction.
In the figure, the beam direction when power is supplied to the patches (20) and (20) of P3 and P7 is shown as an example. In this case, in the YZ plane where the patches (20) and (20) of P3 and P7 are opposed to each other, The direction (θ = 0 °) is weak, and an M-type beam that emits strongly in the θ = ± 45 ° direction is generated, and a weak beam is generated in the XZ plane perpendicular to the YZ plane.

As the principle of M-type beam generation, the principle of a dipole antenna disclosed in Non-Patent Document 3 is used, and each patch element is excited by a TM ₁₀₀ wave which is a fundamental mode, and is synchronized with an opposing patch. By feeding in phase, radio waves weaken each other at the center of the antenna, so that an M-shaped beam is formed.
Yasuhito Mimeaki: Antenna Engineering Handbook, Ohmsha, 1980

FIG. 2 is an explanatory diagram showing a basic configuration of a two-element opposed feed type λ / 4 short-circuited microstrip antenna.
As shown in the figure, a patch (20) whose one side is short-circuited on the same substrate is configured to face each other with a distance c therebetween so that the short-circuit surfaces face each other.
About the characteristic of this antenna, it simulated using HFSS (High Frequency Simulation Soft) which performs an electromagnetic field analysis by the finite element method. The design frequency was 2.5 GHz (λ = 120 mm), and the parameters were set as follows.
Dielectric substrate;
ε _r = 2.17 (tan δ = 0.00085)
t = 1.524mm
d = 60 × 60 mm (1 / 2λ)
patch;
L ₁ = 19.15 mm
L ₂ = 38.3 mm

Here, the element interval c was examined. This antenna uses the principle of a dipole antenna, and the total length (L ₁ + c + L ₁ ) of the antenna including the element interval c greatly affects the radiation characteristics as in the case of the dipole antenna. When the total length exceeds about 1 wavelength, the shape of the electric field including the antenna is 8-shaped. However, when the total length exceeds 1 wavelength, the sidelobe characteristics are increased and the beam is broken.
Based on the above, the total length of the antenna under test was selected to be one wavelength or less.
Further, the element spacing c was set to 15 mm from the element spacing characteristics by simulation, and the feeding point F was set to 7 mm on the center line from the short-circuit plane.

  If the return loss characteristics of this antenna, the directivity in the vertical plane at cut plane φ = 0 °, 45 °, 90 °, and 135 ° and the directivity in the horizontal plane of cut surface θ = 42 ° are displayed, the return loss will resonate. It was found that the frequency was as high as -11 dB, and the directivity slightly exceeded 6 dB in the vicinity of θ = 32 °, φ = −90 °, and 90 °, but not in other ranges at all.

In order to improve this, a simulation was carried out on an element with four elements.
FIG. 3 is an explanatory diagram showing a basic configuration of a four-element opposed two-feed type λ / 4 short-circuited microstrip antenna.
In this manner, four patches (20) are arranged in a substantially annular shape at 90 ° intervals, equidistant from the center of the substrate so that the short-circuit surface is located on the opposing surface. In the four-element opposed two-feed type λ / 4SMSA, the patches (20) and (20) of P1 and P3 form a pair, and P2 and P4 form a pair, and P1-P3 and P2-P4 are switched by a switch in the circuit unit. As in the case of two elements, P1 and P3 are fed in phase, but at this time, P2 and P4 are not fed.
As a result, these two elements act as parasitic elements.

  This method requires two switches and four switching operations. Further, the element interval c is set to 0 point where the element interval cannot be further reduced, that is, the point where the ends of adjacent elements contact each other.

The patch width of this antenna was examined. Now, as shown in FIG. 4 for the element size ratio As, a patch having a size of L ₁ = L ₂ = 19.15 mm is defined as As = 1.0, As = 2.0, As = 1.0, As = The characteristics when the element interval c was changed were examined for each of 0.5. The feeding point F was a point 3 mm above the center line from the short-circuit plane, and the other parameters were the same as in the case of two elements.

As a result of comparison for each of the above, the relationship between the element spacing c and the return loss is shown in FIG. 5A, the relationship between c and the beam direction in the vertical plane is shown in FIG. 5B, and c is the maximum in the vertical plane. The relationship between the directivity gains is shown in FIG. 5C, and the relationship between c and the range exceeding 6 dB is shown in FIG.
Further, the vertical in-plane directivities of the cut planes φ = 0 °, 45 °, 90 °, and 135 ° with the element size ratios As = 2.0, As = 1.0, and As = 0.5 are shown in FIG. Shown in a) to (c). For FIG. 6, the element spacing c, which has the best characteristics in FIG. 5, is employed.

From these graphs, it was found that, by changing the element interval c, the beam direction hardly changed, but there was a position range showing good characteristics in the range exceeding the maximum directivity gain and 6 dB.
This is considered to be caused by the fact that if the element spacing c is too close, the characteristics are deteriorated by disturbing the electromagnetic field, and if it is too far, the action of the opposing patch is reduced, so that the beam width becomes narrow. It is done.

  As for the element size ratio As, the return loss is deteriorated as As is reduced. However, when As = 1.0, the radiation characteristics are not greatly deteriorated as compared with the case of As = 2.0. Rather, it was found that the range exceeding 6 dB increased as a result of the beam being directed to a low elevation angle and the beam in the direction of the cut plane φ = 90 ° being radiation in an unnecessary direction being suppressed. Further, when As = 0.5, the maximum gain is lowered and the cover range is narrowed.

However, what should be noted here is the radiation efficiency η. η is about 90% when As = 2.0, about 80% when As = 1.0, and about 80% when As = 0.5 when the relative dielectric constant ε _r = 2.17 and the substrate thickness t = 1.524 mm. The smaller the element is, the lower the value is 60%.

  From the above results, As = 1.0 was adopted as the element size ratio. FIG. 7 shows the directivity in the horizontal plane of this antenna. From this graph, it can be seen that the cover range is greatly expanded compared to the case of the two elements.

Unwanted radiation may be generated from the antenna side. In order to reduce this, a metal side wall (30) is attached to the side of the antenna to stand upright.
The metal side wall (30) is arranged so that the front direction of radiation is blown out, and by attaching this, unnecessary radiation that has leaked from the side surface of the dielectric substrate (12) to the rear of the antenna can be suppressed. .

FIG. 8A is a graph showing a comparison of return loss characteristics in a state where the metal side wall (30) is attached (dashed line) and a state where the metal side wall (30) is not attached (solid line).
The height l of the metal side wall (30) is 30 mm, and the other parameters are the same as in the previous embodiment. From this graph, it can be seen that the return loss is improved by 10 dB or more by providing the metal side wall (30).

  A comparison of the directivity in the vertical plane at the cut plane φ = 0 ° is shown in FIG. From this result, it was found that both the maximum gain and the beam width were improved, and the effect of attaching the metal side wall (30) appeared.

The height l of the metal side wall (30) was examined.
Regarding various characteristics when the height l is changed from 10 mm to 40 mm, FIG. 9A shows the return loss characteristic, FIG. 9B shows the directivity in the vertical plane, and FIG. 9C shows the directivity in the horizontal plane from (c) to (e). Show.
From FIG. 9B, although l = 40 mm (1 / 3λ) is the highest at the maximum gain, from the viewpoint of a 6 dB cover range at θ = 42 ° ± 10 °, l = 30 mm from FIG. (1 / 4λ) indicates the best characteristic. Therefore, l is 30 mm.

A study was made to control the beam direction by changing the attachment angle ζ of the metal side wall (30).
Here, the angle ζ is an angle from the antenna plane toward the zenith direction, the height l of the metal side wall (30) is kept constant, ζ is changed from −30 ° to 120 °, and power is supplied to P1 and P3. The radiation directivity was investigated.

FIG. 10A shows the vertical plane directivity of the cut surface φ = 0 ° when ζ is changed from −30 ° to 120 °, and the cut surfaces θ = 32 °, 42 °, and 52 °. Comparison of horizontal plane directivities is shown in FIGS.
From FIG. 10A, in the range of ζ = −30 ° to + 30 °, tilting ζ in the zenith direction tilts the beam in the zenith direction, increases the gain, and narrows the beam width. When tilted above ζ = ± 30 °, the beam is directed in the horizontal direction, the gain is decreased, and the beam width is expanded.
This is considered to be because the focal point of the radio wave traveling straight from the radiation surface and the radio wave reflected by the metal side wall is located between ζ = + 30 ° and + 60 °.

Also, in FIGS. 10C and 10D, the effect of attaching the metal surface is particularly good when the cut surface θ = 42 °, 52 °, which is a low elevation angle, according to the target elevation angle, peak gain, and beam width. It was confirmed that the metal side wall should be attached at an appropriate angle.
From this, it was shown that the radiation from the antenna can be adjusted to a desired beam angle and beam width within a certain range by changing the mounting angle of the metal side wall.

  From the above results, in this embodiment, since it is desired to obtain a high gain in the range of θ = 42 ° to ± 10 ° in the vertical plane, the attachment angle of the metal side wall (30) is set to ζ = 90 °.

In FIG. 11, in this antenna, the directivity in the horizontal plane when the switch is switched between P1-P3 and P2-P4 is as follows: cut plane θ = 32 ° (a), 42 ° (b), 52 ° (c). Shown for each case.
When FIG. 11 and FIG. 6 are compared, it can be seen that the cut surface of θ = 32 °, 42 °, or 52 ° is greatly improved. However, in the vicinity of φ = ± 45 °, ± 135 °, that is, in the vicinity of the angle at which switching is performed, none of the cut surfaces satisfies 6 dB. The difference is the largest at θ = 52 ° and a maximum of 3 dB lower.

Furthermore, we studied to cover all directions by increasing the number of elements.
The number of elements was further increased to 8 elements, and simulations were similarly performed.
Eight patches (30) were arranged in an annular shape by shifting by 45 °. Opposing patches P1-P5, P2-P6, P3-P7, and P4-P8 are fed as a pair, respectively.

This antenna requires four switches and eight switching operations to cover all directions. In the case of 8 elements, if the substrate is square, the symmetry between the elements cannot be maintained, so the shape of the substrate is a circle. Although the edge shape differs in the substrate shape, there is no influence on the characteristics if there is a certain distance from the element to the substrate edge. The parameters were set to L ₁ = L ₂ = 19.15 mm, t = 1.524 mm, 1 = 30 mm, ζ = 90 °, and d = 150 mm based on the results.

This antenna was also examined for characteristics when the element spacing c was changed. Here, the quality judgment is made based on the return loss and the beam switching point when θ = 52 ° (here, φ = ± 22.5 °, ± 67.5 °, ± 112.5 °, ± It was based on how much it exceeded 6 dB at 157.5 °).
As a result, the directivity in the vertical plane and the directivity in the horizontal plane when c = 5 mm, which had the best characteristics, are shown in FIGS.

  As is apparent from these graphs, the directivity in the horizontal plane exceeds approximately 7.5 dB and 6 dB at the beam switching points in the horizontal plane of θ = 32 ° and 42 °, respectively. Moreover, it was 6.27 dB also at θ = 52 °, and θ = 42 ° ± 10 ° was satisfied.

Next, changes in characteristics due to an increase in the number of elements were examined.
FIG. 13 shows the return loss characteristics, FIG. 14 shows the vertical in-plane directivity at φ = 0 °, and FIGS. 15A to 15C show θ = 32 °, 42 °, and the characteristics due to the increase in the number of elements. The directivity in the horizontal plane at 52 ° is shown. Note that these are all the characteristics when the metal side wall (30) is not attached.

  As apparent from FIG. 13, the return loss tends to be better as the number of elements is increased. Further, as shown in FIG. 14, the width of the beam is narrowed, and the direction of the beam is also directed to the zenith (θ = 0 °) direction. In the horizontal plane, the beam width increases when the number of elements is increased to a certain level. However, when the number of elements is increased beyond that, the beam is broken. This is because a patch that is not supplied power serves as a non-powered patch, and its interaction increases with an increase in the number of elements.

In the above description, the opposed two-feed type λ / 4SMSA has been described. Here, a method is described in which the same antenna is used to feed power to only one patch (30) and switch it with a switch.
In this method, the circuit is simpler than the opposed two-feed type, and no power distributor is required, and the beam is switched by switching (in this case, six times).
The beam in this system generates a strong beam in the opposite direction to the fed patch.

  In this method, the characteristics of 2 elements, 4 elements, 6 elements, and 8 elements were examined. As a result, it has been found that the specifications required in the present embodiment are satisfied with 6 elements or more, and therefore, a 6-element 1-feed type λ / 4SMSA will be described below.

Element spacing c is 7 mm, patch width L ₁ = L ₂ = 19.15 mm, substrate thickness t = 1.524 mm, ε _r = 2.17 (tan δ = 0.00085), substrate width d = 150 mm, feeding point F FIG. 16 (a) shows the return loss characteristics when 5 mm = 5 mm, and FIG. 16 (b) shows the vertical in-plane directivity when power is supplied to one patch (30) P1, from P1 to P6. The directivity in the horizontal plane when the power is supplied is shown in (c) to (e).
From this graph, it can be seen that in this method, the maximum gain is about 10 dB in the horizontal plane of φ = 32 °, and about 7 dB is obtained at the beam switching point at φ = 52 °, and a strong beam is widely obtained. It was.

FIG. 17 is a table comparing the characteristics of the opposed 2-feed system and the 1-feed system.
From the comparison table of FIG. 17, for the desired satellite communication of the present embodiment, the 1-feed system is simpler in circuit system and control and has higher directivity gain, and as a result, has a wide coverage. I understand.
On the other hand, although the two-feed method has the disadvantage that the circuit system and the control are complicated, it has a feature of having a symmetrical beam with respect to the Z axis, and is useful when fine beam switching is required.

Based on the above examination results, a 6-element 1-feed type λ / 4SMSA having a more suitable performance when actually used for ETS-VIII was manufactured.
The parameters were designed as follows based on the optimum values in the simulation.
Dielectric substrate;
ε _r = 2.17
t = 1.524mm
d = 150mm
tan δ = 0.00085
patch;
L ₁ = 19.15 mm
L ₂ = 19.15 mm
s = 19.15 mm
F = 5mm
c = 7mm
Metal sidewalls;
l = 30mm
ζ = 90 °

FIG. 18 shows the return loss characteristics of this antenna, and FIG. 19 shows the vertical in-plane directivity at φ = 0 ° when the patch (30) P1 is fed. In this graph, the simulation value is the directivity gain [dB], and the actual measurement value is the power gain [dBi].
In these graphs, although the resonance frequency is almost as designed, the pattern of directivity is similar, but the power gain is much inferior to the directivity gain of the simulation, and the maximum gain is attenuated by 5 dB. Be looked at. The cause of this was considered to be leakage loss of radio waves at the connection between the conductive tape and the vapor deposition patch.

  As shown in FIG. 20 (a), in this antenna, the conductive tape (15) is used when conducting the conduction between the deposited patch (14) and the conductor (11). In the tape (15), the conductive tape (15) and the conductor (14), there is a possibility that leakage or a flow of current to an unnecessary portion has occurred in each connection portion. Therefore, as shown in FIG. 20B, the patch portion is integrated with the short-circuit plate (11) by using the conductive tape (15) without vapor deposition, thereby preventing leakage.

  FIG. 21 shows the vertical in-plane directivity of the cut plane φ = 0 ° when the antenna manufactured with this structure is fed to the patch (30) P1. From this graph, the radiation pattern showed significant agreement with the simulation. The gain is greatly improved, which is 3 dB lower than the directional gain of the simulation value, and the radiation efficiency η is about 50% from Equation 22 based on the simulation value.

In order to increase the radiation efficiency further than this antenna, an antenna with As = 2.0 was produced. The parameters are as follows.
Dielectric substrate;
ε _r = 2.17
t = 1.524mm
d = 200mm
tan δ = 0.00085
patch;
L ₁ = 19.15 mm
L ₂ = 38.3 mm
s = 19.15 mm
F = 5mm
c = 15mm
Metal sidewalls;
l = 30mm
ζ = 90 °

With respect to this antenna, the return loss characteristic is shown in FIG. 22, and the cut plane φ = 0 ° and the switch switching point (φ = −30 °, φ = 30 °) in the vertical plane when power is supplied to the patch (30) P1. About directivity, it shows to Fig.23 (a)-(c).
The measured values and the simulated values are in good agreement with each other in the beam principal axis direction. In the φ = 0 ° plane, the maximum gain is 9.65 dBi at a low elevation angle of θ = 45 °, and the radiation efficiency η is 95.3%, which is a significant improvement over the As = 1.0 antenna. Yes. Moreover, it exceeds 6 dBi in a wide range of θ = 26 ° to 67 °. The reason why it does not match the simulation value outside the beam main axis is because the patch part is manufactured manually, so that each patch shape is slightly different, so there is a discrepancy especially outside the beam main axis. It seems that it is happening.

  Further, even in the φ = −30 ° plane, the maximum gain is 6.80 dBi at θ = 41 °, and the range exceeding 6 dBi is θ = 32 ° to 52 °, and this antenna satisfies the specifications of this embodiment over a wide range. It was proved.

According to the present invention, it is possible to obtain a microstrip antenna which has a small size and a high gain and can be freely tracked by a satellite or the like.
Since this antenna can be applied not only to satellite communication but also to various fields such as wireless LAN and ITS by changing the feeding mode, it has high industrial utility value.

Explanatory drawing which shows the structure of a λ / 4 short-circuited microstrip antenna that feeds two opposing patches in phase. Explanatory drawing which shows the basic composition of a λ / 4 short-circuiting type microstrip antenna with two elements facing feeding Explanatory drawing which shows the basic composition of a 4 element opposite 2 feeding type lambda / 4 short circuit type microstrip antenna Explanatory drawing which shows element size ratio in 4 element opposing 2 feeding type | formula (lambda) / 4 short circuit type microstrip antenna Relationship between element spacing c and return loss (a), relationship between c and vertical in-plane beam direction (b), relationship between c and vertical in-plane maximum directivity gain (c), and range exceeding c and 6 dB Graph showing relationship (d) Vertical planes with respective element cut ratios of As = 2.0 (a), As = 1.0 (b), As = 0.5 (c) φ = 0 °, 45 °, 90 °, 135 ° Graph showing internal directivity The graph which shows the directivity in a horizontal surface in case element dimension ratio As = 1.0 A graph comparing the return loss characteristic (a) and the directivity in the vertical plane (b) for the state with the metal side wall attached (dashed line) and the state without the metal side wall (solid line) A graph showing (a) return loss characteristics, (b) directivity in the vertical plane, and (c) to (e) directivity in the horizontal plane when the height of the metal side wall is changed. Graph showing horizontal plane directivity of cut surfaces φ = 0 ° (a), 32 ° (b), 42 ° (c), and 52 ° (d) when the attachment angle of the metal side wall is changed In an antenna with a metal side wall attached at an angle of 90 °, the directivity in the horizontal plane when the feed switch is switched is as follows: cut plane θ = 32 ° (a), 42 ° (b), 52 ° (c) Graph showing about A graph showing vertical in-plane directivity (a) and horizontal plane directivity θ = 32 ° (b), 42 ° (c), 52 ° (d) when the element spacing is 5 mm in the case of 8 elements. Graph showing the relationship between the number of elements and return loss characteristics Same as above, graph showing directivity in the vertical plane at φ = 0 ° Same as above, a graph showing the directivity in the horizontal plane at θ = 32 °, 42 °, 52 ° For the 6-element 1-feed type short-circuited microstrip antenna, the return loss characteristic (a), the directivity in the vertical plane when power is supplied to one patch P1, (b), and θ = 32 when power is supplied from P1 to P6, respectively. A graph showing the directivity in the horizontal plane at ° (c), 42 ° (d), and 52 ° (e) A table comparing the characteristics of the opposed 2-feed system and the 1-feed system Graph showing return loss in 6-element 1-feed type λ / 4 short-circuited microstrip antenna for ETS-VIII Same as above, vertical in-plane directivity graph Explanatory drawing (a) which shows the state which used the conductive tape when conducting the vapor-deposited patch and the conductor, and explanatory drawing which shows the state integrated with the short-circuit plate with the conductive tape without vaporizing the patch portion ( b) FIG. 20B is a graph showing the directivity in the vertical plane of the cut plane φ = 0 ° when the patch P1 is fed with respect to the antenna having the structure of FIG. The graph which shows the return loss characteristic of an antenna when element size ratio As = 2.0 Similarly, the vertical plane directivities of the cut plane φ = 0 ° (a) and the switch switching points (φ = −30 ° (b), φ = 30 ° (c)) when the patch P1 is supplied with power are shown. Graph

Explanation of symbols

DESCRIPTION OF SYMBOLS 11 Metal conductor 12 Dielectric board | substrate 13 Feed part 14 Evaporation patch 15 Conductive tape 20 (lambda) / 4 short circuit type microstrip antenna element 21 Short circuit part 30 Metal side wall

Claims (13)

A plurality of substantially λ / 4 short-circuited microstrip antenna elements arranged in parallel in a symmetrical arrangement on the same substrate;
A feeding means for feeding power to the microstrip antenna element;
A microstrip antenna, comprising: a power feeding element; and power feeding control means for controlling a power feeding mode. The microstrip antenna according to claim 1, wherein four or more substantially λ / 4 short-circuited microstrip antenna elements are provided and are arranged on the same arc. The microstrip antenna according to claim 1 or 2, wherein the power supply means is a pin power supply system, and the power supply point is set at a position offset by about 30% so that the microstrip antenna element and the power supply system are matched. The microstrip antenna according to any one of claims 1 to 3, wherein the substantially λ / 4 short-circuited microstrip antenna element is substantially rectangular and has a dimensional ratio of long and short sides of about 1: 2. 5. The microstrip according to claim 1, wherein a metal side wall whose height or installation angle can be adjusted is erected on a peripheral portion of a substrate on which a plurality of substantially λ / 4 short-circuited microstrip antenna elements are arranged. antenna. The microstrip antenna according to claim 5, wherein the height of the metal side wall is about ¼λ, which is a value corresponding to the propagation wavelength λ. The microstrip antenna according to claim 5 or 6, wherein the attachment angle of the metal side wall is about 90 °. The microstrip antenna according to any one of claims 1 to 7, wherein a patch portion of the substantially λ / 4 short-circuited microstrip antenna element is integrated with a metal conductor via a conductive tape. A plurality of substantially λ / 4 short-circuited microstrip antenna elements arranged in parallel in a symmetrical arrangement on the same substrate;
A feeding means for feeding power to the microstrip antenna element;
Using a microstrip antenna provided with a power feeding element and a power feeding control means for controlling a power feeding mode,
A method of controlling a microstrip antenna, characterized by adjusting a radiation pattern by selecting elements to be fed simultaneously. A metal side wall whose height or installation angle can be adjusted is erected on the peripheral edge of the substrate on which a plurality of substantially λ / 4 short-circuited microstrip antenna elements are arranged,
The method of controlling a microstrip antenna according to claim 9, wherein the angle or width of the radiation beam is adjusted by changing the height or installation angle of the metal side wall. The feed mode is set so that the radiation directivity is close to the desired radiation pattern, and then the angle or width of the radiation beam is adjusted independently of the dielectric constant of the substrate by changing the height or installation angle of the metal side wall. The method of controlling a microstrip antenna according to claim 9 or 10. 12. When supplying a beam having symmetry or performing fine beam switching, power is fed in phase to two substantially λ / 4 short-circuited microstrip antenna elements arranged opposite to each other. The control method of the microstrip antenna as described. The method for controlling a microstrip antenna according to any one of claims 9 to 11, wherein when a strong beam is provided in one direction, power is supplied to one substantially λ / 4 short-circuited microstrip antenna element.

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