JP2005261134A - Motor rotation information detecting apparatus - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a motor rotation information detecting apparatus having control consistency and reliably detecting motor rotation information even if a filter is used and has a changeable pass frequency band. <P>SOLUTION: A switched capacitor filter (SCF) 7 is used and extracts a ripple component from a motor current i detected by a motor current detecting section 4. The ripple component in the current is converted into a pulse signal by a binarization process of a comparator 8. A counter electromotive voltage estimating section 5 estimates a counter electromotive voltage Vg based on a voltage Vm across terminals of a motor M detected by a section 3 for detecting the voltage across the terminals of the motor and the motor current i detected by the motor current detecting section 4. The pass frequency band of the switched capacitor filter 7 is variably controlled in response to a rotation speed of the motor M based on the counter electromotive voltage Vg by utilizing a proportional relationship between the counter electromotive voltage Vg and the rotation speed of the motor M. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a motor rotation information detection device, and more particularly to a DC motor having a brush, which detects rotation information based on a motor drive waveform such as a current flowing through the motor or a voltage between terminals of the motor. .

  Many systems using DC brush motors are installed in automobiles for comfort and convenience. For example, the air conditioner is equipped with several direct current motors that move the outlet and the door for changing the air mix amount. In addition, a DC motor is used as an actuator for the operation of the position of the door mirror and the seat, and for the operation of the position of the window in the power window system. Furthermore, the applications in which these DC motors are used will continue to expand, such as a system that changes the optical axis of the headlight according to the turning angle of the steering.

  As described above, in these systems, the position of the door, the position of the mirror, the position of the seat, the position of the window, the position of the headlight, and the position of the driving object are controlled by a DC motor. For this reason, it is important to detect the position of the drive target, that is, the rotation information of the motor by some method.

  Heretofore, as a method for detecting the position of the drive target (rotation information of the motor), for example, as described in Patent Document 1, the position is measured by measuring the rotation amount of the motor using a hall sensor or the like. There are known a method for specifying the position and a method for specifying the position with a potentiometer. However, such a method requires a sensor such as the Hall sensor or potentiometer, so that the cost increases due to the sensor itself and sensor mounting, the number of signal lines increases, and the reliability decreases due to the life of the sensor. Cannot be ignored.

  Thus, conventionally, a method has been proposed in which such a sensor is not required, specifically, a ripple component included in the current flowing through the motor is extracted, and motor rotation information is obtained based on the extracted ripple component (for example, Patent Document 2). FIG. 11 and FIG. 12 illustrate the configuration for detecting the rotation information of the motor and the detection mode thereof.

  That is, in order to obtain the rotation information of the motor in the above mode, as shown in FIG. 11, a resistor Ro is inserted in series with the motor M in the power feeding circuit to the motor M, and between the terminals of the inserted resistor Ro. The motor current i flowing through the motor M is detected by extracting the voltage. At this time, the detected motor current i is caused by discontinuity when the brush and commutator segments are switched with respect to the ripple component of a predetermined period, as shown in FIG. Surge component is added. Then, by passing the motor current i flowing through the motor through a band-pass filter BPF corresponding to the period (frequency range) of the ripple component, as the filter output BPFo, as shown in FIG. A signal from which components and other noise components are removed is obtained. That is, a signal corresponding to the ripple component of the motor current i flowing through the motor M is obtained. Further, the signal BPFo corresponding to the ripple component of the motor current i is binarized under a predetermined threshold voltage Vth through the comparator CP, whereby the rotation of the motor is performed in the manner shown in FIG. A pulse signal CPo corresponding to the angle is obtained. Then, by counting the number of pulse signals CPo, rotation information such as the rotation speed (rotation speed) and rotation angle of the motor M is detected. Thus, if the ripple component contained in the motor current i flowing through the motor M can be extracted, the rotation information of the motor can be obtained based on this.

  However, in the case of this method, as described above, when the ripple component of the motor current i is extracted using the band pass filter BPF, the frequency pass band of the band pass filter BPF is set according to the frequency range. There is a need. That is, since the frequency of the ripple component of the motor current i is proportional to the number of rotations of the motor M, in the low rotation range such as when the motor M starts rotating or immediately before the rotation stops and in the high rotation range such as during steady rotation, the above The frequency of the ripple component is naturally different. Usually, in order to obtain sufficient filter characteristics as the bandpass filter BPF, it is desirable to set the frequency pass band to be narrow to some extent. Therefore, when the pass band is matched with the frequency at the time of low speed rotation of the motor M, the motor M The ripple component at the time of high speed rotation cannot be extracted. On the contrary, if the same pass band is matched with the frequency pass band at the time of high speed rotation of the motor M, the ripple component at the time of the low speed rotation can no longer be extracted. On the other hand, if such a band-pass filter BPF is used to cover the frequency from the low speed rotation of the motor M to the high speed rotation frequency, the attenuation characteristic of the filter itself is lowered, and the surge component cannot be sufficiently attenuated. . That is, when the filter output BPFo is binarized by the comparator CP, an extra pulse may be observed as the output CPo, which may result in erroneous rotation information.

Therefore, conventionally, in order to avoid such inconvenience, the waveform of the current flowing through the motor using a switched capacitor filter (SCF) whose frequency pass band can be changed according to the rotation speed (rotation speed) of the motor is further shown. An apparatus for shaping and extracting the ripple component has been proposed (see, for example, Patent Document 3).
JP 2003-049586 A Japanese Patent Laid-Open No. 2002-010667 Japanese Patent Application Laid-Open No. 2000-114962

  Thus, if the frequency passband as a filter can be made variable according to the number of rotations of the motor, basically, it becomes possible to accurately extract the ripple component in all operating regions of the motor, and consequently It is also easy to acquire motor rotation information based on such ripple components. However, the apparatus described in Patent Document 3 uses a fact that the frequency of the extracted ripple component of the current and the rotational speed of the motor are in a proportional relationship, and uses the switched capacitor filter from the frequency of the extracted ripple component. The clock frequency is determined. For this reason, the following inconvenience may be newly caused. That is, in this configuration, if the ripple component cannot be extracted, the clock frequency of the switched capacitor filter is not determined, so that it is more difficult to extract the ripple as the initial value. For this reason, if the frequency characteristic of the switched capacitor filter itself becomes a characteristic that causes an erroneous pulse for some reason, the control of the filter characteristic as the switched capacitor filter is started from such a characteristic, and thus an erroneous pulse is generated. It may lead to control contradiction such as convergence or divergence. In such a case, the reliability of the detected motor rotation information itself is greatly reduced.

  The present invention has been made in view of the above situation, and detects more reliable motor rotation information without causing a control contradiction, even when a filter capable of changing the frequency passband is used. An object of the present invention is to provide a motor rotation information detection device that can perform the above-described operation.

  In order to achieve such an object, in the motor rotation information detecting device according to claim 1, the periodic component waveform is extracted from the motor driving waveform of the DC brush motor by a filter whose frequency pass band can be changed, and the extracted periodic component As a motor rotation information detection device that detects rotation information of the motor based on a waveform, a motor terminal voltage detection unit that detects a voltage between terminals of the motor, and a motor current detection unit that detects a current flowing through the motor, A counter electromotive voltage estimation unit that estimates a counter electromotive voltage generated in the motor based on the detected inter-terminal voltage and current, and determines a frequency passband of the filter based on the estimated counter electromotive voltage. Configure as something to change.

  Usually, the back electromotive force is proportional to the rotation speed of the motor. For this reason, if the frequency pass band of the filter is changed based on the counter electromotive voltage, the periodic component waveform is accurately extracted from the motor current in the entire rotation range from the low-speed rotation range to the high-speed rotation range of the motor. Will be able to.

  Therefore, according to such a configuration, the rotational speed of the motor estimated from the periodic component waveform extracted by the filter is fed back to the filter, and compared with a conventional configuration in which the frequency pass band is changed, the control is more efficient. Thus, more reliable motor rotation information can be detected.

  According to a second aspect of the present invention, in the configuration of the first aspect of the present invention, the filter is a switched capacitor filter whose frequency pass band is variable based on the frequency of an applied clock. The frequency of the applied clock is determined based on the counter electromotive voltage estimated through the counter electromotive voltage estimation unit.

  By controlling the clock frequency of the switched capacitor filter in such a manner, the frequency pass band of the filter can be easily changed according to the rotation speed of the motor.

  Here, as the periodic component waveform extracted from the motor drive waveform in order to detect the rotation information of the motor, one of the ripple component waveform and the surge component waveform included in the motor drive waveform is provided, as in the third aspect of the invention. Is effective.

  According to a fourth aspect of the present invention, in the configuration of the first aspect of the present invention, the motor current detector is connected in parallel with a motor driving transistor that drives the motor. A current sense transistor having a current mirror configuration, and detecting the current flowing through the motor based on the current flowing through the current sense transistor.

  According to such a configuration as the motor current detection unit, the current flowing through the motor is detected in a manner reduced by an appropriate current mirror ratio. For this reason, the heat generation by the current detecting circuit elements such as resistors which have been conventionally used is greatly reduced, and the circuit elements can be formed as IC chips. That is, external parts are reduced, and a more compact circuit design can be expected.

According to a fifth aspect of the present invention, in the configuration of the first aspect of the present invention, the counter electromotive voltage estimation unit is used as the detected terminal voltage of the motor Vm. When the current flowing through the motor is i, the internal resistance specific to the motor is r, the internal inductance specific to the motor is L, and the counter electromotive voltage is Vg, the following differential equation:

Vg = Vm-r.i-L (di / dt)

As a solution, the back electromotive force Vg is estimated.

  According to such a configuration as the back electromotive voltage estimation unit, for example, the back electromotive voltage Vg can be estimated using an arithmetic device such as a CPU, so that the number of parts of circuit elements such as resistors is reduced, and A more flexible design can be realized.

In the above differential equation, the voltage drop due to the internal inductance L of the motor is sufficiently small and can be ignored. Therefore, as in the sixth aspect of the invention, the following simplified expression

Vg = Vm−r · i

The back electromotive force Vg can also be obtained as a solution of

  Here, the internal resistance r of the motor is treated as having a constant resistance value, but in reality, the resistance value may change due to the influence of manufacturing variations, temperature characteristics, and the like.

Therefore, particularly in the invention described in claim 6, in order to avoid the influence of the variation in the internal resistance r of the motor, for example, according to the invention described in claim 7,
(A) The voltage across the terminals is so small that the motor cannot rotate.
Alternatively, as in the invention according to claim 8,
(B) A state in which the rotation of the motor is mechanically stopped.
In the next operation

r = Vm / i

Is effective to obtain the internal resistance r of the motor.

In addition to the above, the invention described in the sixth aspect, in addition to these, for example, according to the invention described in the ninth aspect, the voltage Vm1 between the terminals and the motor current i1 during the steady rotation of the motor, and the braking operation of the motor. The inter-terminal voltage Vm2 and the motor current i2 immediately after switching are respectively obtained, and the following simultaneous equations

Vg = Vm1-r · i1
Vg = Vm2-r · i2

The internal resistance r of the motor may be obtained as a solution of

In any of these methods, it is possible to appropriately avoid the influence due to the variation in resistance value with respect to the internal resistance r of the motor, and to obtain the back electromotive voltage Vg more accurately.
On the other hand, in the invention described in claim 10, in the configuration of the invention described in any one of claims 1 to 4, this is detected as the counter electromotive voltage estimation unit by the motor current detection unit. A fixed resistance portion that causes a voltage drop equivalent to the internal resistance of the motor, and a voltage generated between the terminals of the fixed resistance portion connected in series to the fixed resistance portion in accordance with steady rotation of the motor and the fixed resistance portion And a variable resistance unit whose resistance value is variably controlled so that the sum of the voltage and the terminal voltage of the motor becomes equal to the voltage detected by the terminal voltage detection unit of the motor, and is generated between the terminals of the variable resistance unit. The counter electromotive voltage is estimated based on the voltage.

That is, here, the back electromotive force estimation unit is composed of a series circuit of the fixed resistance unit and the variable resistance unit, and during steady operation of the motor,
(A) The voltage drop generated in the fixed resistance portion is equal to the voltage drop generated in the internal resistance of the motor. and,
(B) The sum of the voltage drop generated in the fixed resistance portion and the voltage drop generated in the variable resistance portion (the voltage across the terminals of the series circuit of the fixed resistance portion and the variable resistance portion) becomes equal to the voltage between the motor terminals.
The resistance value of the variable resistance unit is controlled in such a manner that the above conditions are satisfied. Then, the back electromotive force is estimated based on the voltage generated between the terminals of the variable resistance unit controlled in this way. By adopting such a circuit during steady operation of the motor, it is possible to estimate the back electromotive voltage with high accuracy without using an arithmetic unit such as the CPU as the back electromotive voltage estimation unit.

  For this purpose, according to the invention described in claim 11, a series circuit of the fixed resistor portion and the variable resistor portion constituting the counter electromotive voltage estimating portion is connected to a motor drive circuit with a predetermined current mirror ratio. And the resistance value of the fixed resistor section is set to a value obtained by multiplying the internal resistance of the motor by the current mirror ratio, thereby satisfying the condition (a). Further, the terminal voltage of the series circuit of the fixed resistor part and the variable resistor part constituted by these current mirrors and the terminal voltage of the motor are respectively input to the operational amplifier, and the above-mentioned (b The configuration satisfying the condition (2) is also advantageous in making each of these circuits into an IC chip.

  Also, at this time, as in the invention described in claim 12, the variable resistance section has a field effect transistor, and the output of the virtually grounded operational amplifier is applied to the gate terminal of the field effect transistor. As a result, the resistance value control as the variable resistance part through the resistance value between the source and the drain can be easily realized.

  On the other hand, in the invention according to claim 13, in the configuration according to any one of claims 1 to 4, this is detected by the motor current detection unit as the counter electromotive voltage estimation unit. A fixed resistance portion that causes a voltage drop equivalent to the internal resistance of the motor with respect to the current, and a voltage generated between terminals of the motor during braking operation of the motor is connected in series to the fixed resistance portion. A variable resistance unit whose resistance value is variably controlled so as to be equal to the voltage detected by the inter-terminal voltage detection unit, and based on the voltage generated between the terminals of the series circuit of these fixed resistance unit and variable resistance unit Thus, the counter electromotive voltage is estimated.

That is, here, the back electromotive force estimation unit is composed of a series circuit of the fixed resistance unit and the variable resistance unit, and in the braking operation state of the motor,
(A) The voltage drop generated in the fixed resistance portion is equal to the voltage drop generated in the internal resistance of the motor. and,
(B) The voltage drop generated between the terminals of the variable resistance section becomes equal to the voltage between the terminals of the motor.
The resistance value of the variable resistance unit is controlled in such a manner that the above conditions are satisfied. Then, the back electromotive force is estimated based on the voltage generated between the terminals of the series circuit of the fixed resistor and the variable resistor controlled in this way. By adopting such a circuit during the braking operation of the motor, it is possible to estimate the counter electromotive voltage with high accuracy without using an arithmetic unit such as the CPU as the counter electromotive voltage estimation unit.

  In this case as well, according to the invention described in claim 14, the series circuit of the fixed resistor and variable resistor constituting the back electromotive force estimation unit has a predetermined current mirror ratio with respect to the motor drive circuit. Thus, the current mirror connection is established, and the resistance value of the fixed resistor section is set to a value obtained by multiplying the internal resistance of the motor by the current mirror ratio, thereby satisfying the condition (a). Furthermore, the configuration in which the terminal voltage of the variable resistor section and the terminal voltage of the motor configured by these current mirrors are respectively input to an operational amplifier, and the above condition (b) is satisfied using virtual grounding by the operational amplifier, As described above, these circuits are advantageous in forming an IC chip.

  Also in this case, as in the invention described in claim 15, the variable resistance portion has a field effect transistor, the drain voltage of the field effect transistor is Vd, the source voltage is Vs, and the voltage between the terminals of the motor is When Vm is applied, an output of Vd− (Vs−Vm) from the operational amplifier is applied to the gate terminal of the field effect transistor, so that the variable resistance portion through the resistance value between the drain and source can be obtained. Resistance value control is also easily realized.

  On the other hand, in the invention according to claim 16, in the configuration of the invention according to any one of claims 1 to 15, the periodic component waveform of the current extracted by the filter capable of changing the frequency pass band is: It is configured to be pulsed through binarization processing by a comparator.

In this way, the motor rotation information can be detected more easily by converting the periodic component waveform of the current extracted by the filter into a pulse signal through the binarization processing by the comparator.
In each of the above inventions, as the motor drive waveform of the DC brush motor described above, for example, as in the invention according to claim 17,
-Waveform of current flowing through the DC brush motor.
Or like invention of Claim 18,
• DC brush motor terminal voltage waveform.
and so on. Both of these are effective in extracting the above-described periodic component waveform.

(First embodiment)
Hereinafter, a first embodiment of a motor rotation information detection device according to the present invention will be described with reference to FIGS.

FIG. 1 is a block diagram showing an outline of the configuration of the motor rotation information detecting apparatus according to this embodiment.
As shown in FIG. 1, the DC motor M is connected to a motor drive power source E1 through a pair of brushes 2, and is supplied with power by the motor drive power source E1. The motor M is connected to the motor terminal voltage detector 3 for detecting the motor terminal voltage Vm in parallel with the motor M, and the motor M detects the current i flowing through the motor M in series with the motor M. A current detection unit 4 is connected.

  In the present embodiment, the motor terminal voltage Vm detected by the motor terminal voltage detector 3 and the motor current i detected by the motor current detector 4 are both taken into the counter electromotive voltage estimation unit 5, The counter electromotive voltage estimation unit 5 calculates a counter electromotive voltage Vg generated in the motor M.

  In general, as a method for detecting the back electromotive voltage Vg of a motor, as shown in FIG. 2, a bridge circuit is configured by the motor M and a resistor having a resistance value as shown in the figure, and between the output terminals of the bridge circuit. There is a method of detecting the back electromotive voltage Vg by detecting the voltage. In this circuit, when the motor M is not rotating, the divided voltage values of the resistances on both sides constituting the bridge are balanced, so the voltage between the output terminals becomes zero. However, when the motor M rotates, the counter electromotive voltage Vg is As a result, the bridge becomes unbalanced, and a voltage corresponding to the counter electromotive voltage Vg is generated between the output terminals. That is, the back electromotive voltage Vg of the motor is obtained from the voltage between the output terminals.

  However, when the bridge circuit is configured in this way to obtain the back electromotive voltage Vg, a large current flows through at least the resistance on the motor side and heat is generated. For this reason, such a resistor cannot be built in the IC chip, and the number of external parts increases after all.

Therefore, in the present embodiment, as a method for detecting the counter electromotive voltage Vg, as shown in FIG. 1, an arithmetic unit such as a CPU is used based on the motor terminal voltage Vm and the motor current i flowing through the motor. The counter electromotive voltage Vg is obtained. The motor terminal voltage Vm, the motor current i flowing through the motor, and the counter electromotive voltage Vg are represented by the following equation from the equivalent circuit of the motor M shown in FIG.

Vm = Vg + L (di / dt) + r · i (1)

Here, L is the internal inductance of the motor, and r is the internal resistance of the motor. Therefore, if the inter-terminal voltage Vm and the motor current i of the motor are detected, the back electromotive voltage Vg can be calculated (estimated) as a solution of the differential equation.

  In the present embodiment (FIG. 1), when the counter electromotive voltage Vg is thus estimated by the counter electromotive voltage estimation unit 5, the result is output to the clock frequency setting unit 6. Since the counter electromotive voltage Vg is proportional to the rotation speed of the motor M, the clock frequency setting unit 6 calculates the rotation speed of the motor M from the estimated counter electromotive voltage Vg and further determines the rotation of the motor M. The clock frequency f is set based on the number. Based on the set clock frequency f, the frequency pass band of the switched capacitor filter 7 is controlled.

  On the other hand, the motor current i detected by the previous motor current detector 4 is also input to the switched capacitor filter 7, and the periodic capacitor waveform is extracted by the switched capacitor filter 7. In the present embodiment, a ripple component (see FIG. 12) superimposed on the current waveform in proportion to the rotational speed of the motor M is extracted as the periodic component waveform of the motor current i.

  Note that when the motor rotation speed changes, the cycle (frequency) of the ripple component included in the motor current i also changes together with the back electromotive voltage Vg. Therefore, in order to change the frequency pass band of the switched capacitor filter 7 in accordance with the motor rotation speed, the frequency f corresponding to the back electromotive voltage Vg from the clock frequency setting unit 6 to the switched capacitor filter 7 is changed. A clock to be applied is applied. Thereby, the frequency pass band of the switched capacitor filter 7 is always set as a band corresponding to the rotational speed of the motor M.

  The ripple component of the motor current i thus extracted by the switched capacitor filter 7 is taken into the comparator 8 and is binarized by the comparator 8 under a predetermined threshold voltage Vth. Becomes a pulse signal having the same frequency. As described above, since the ripple component included in the motor current i is proportional to the rotational speed of the motor M, the frequency of this pulse signal is also proportional to the rotational speed of the motor M.

  Further, the pulse signal thus generated through the comparator 8 is input to the rotation amount calculation unit 9. The rotation amount calculation unit 9 obtains rotation information such as the rotation speed and rotation angle (rotation position) of the motor M based on the number of pulse signals, the cycle time, and the like. Then, a control command for the motor M is given through the motor rotation control unit 10 so that the rotation information of the motor M obtained by the rotation amount calculation unit 9 matches the control target value of the motor M.

  FIG. 4 is a block diagram showing in more detail the configuration of the motor rotation information detecting device of the present embodiment shown in FIG. In the present embodiment, a bridge circuit (H bridge) as a drive circuit for the motor M is formed by using four field effect transistors (FETs) M1, M2, M3, and M4 as switching elements. Each of these transistors M1 to M4 is controlled in its conduction state (ON or OFF) through the gate drive circuit 10B.

  That is, here, when the transistors M2 and M3 are turned on and the transistors M1 and M4 are turned off through the gate drive circuit 10B, the motor M rotates forward, and the transistors M2 and M3 are turned off. It is assumed that the motor M reverses when M4 is turned on. Further, when the transistors M1 and M2 are turned off and M3 and M4 are turned on from the steady operation state of the motor M, the motor M enters the braking operation state.

Next, the motor current detection unit 4 shown in FIG. 4 will be described in detail.
The motor current detection unit 4 employed in the present embodiment is configured to detect the motor current i using the current mirrors C1 and C2. That is, as shown in FIG. 4, the transistor M3 provided in the motor drive circuit and driving the motor M is provided with a sense transistor M5 in which the gate drive voltage and the source voltage are shared, and this transistor A resistor R1 is connected to the drain side of M5. Similarly, a sense transistor M6 in which a gate drive voltage and a source voltage are shared is provided for the transistor M4, and a resistor R2 is connected to the drain side of the transistor M6.

  Thus, the transistors M3 and M5 form a current mirror C1, and the transistors M4 and M6 form a current mirror C2. These current mirrors C1 and C2 reduce the motor current i at a predetermined current mirror ratio (for example, 1: m). For example, when the motor M is rotating forward, the motor current i is the drain current of the transistor M3, and the drain current of the transistor M5 is “i / m” with respect to the drain current of M3. Similarly, when the motor M is rotating in the reverse direction, the motor current i is the drain current of the transistor M4, and the drain current of the transistor M6 is “i / m” with respect to the drain current of the transistor M4.

  That is, by detecting the voltage across the resistor R1 or R2, the drain current of the transistor M5 or M6 is detected, and the motor current i is detected as m times the current. The current mirror ratio can be appropriately set to an arbitrary value as long as the current flowing through the resistor R1 or R2 is sufficiently small.

  In this embodiment, since the motor current i is detected using the current mirrors C1 and C2, the heat generation at the resistors R1 and R2 is greatly reduced, and the resistors R1 and R2 are integrated into the IC chip. It can also be built in.

  Thus, the motor current i detected using the resistors R1 and R2 is differentially amplified by the operational amplifiers A3 (when the motor M is rotating in the forward direction) and A2 (when the motor M is rotating in the reverse direction). It is taken into the multiplexer 41.

  Here, the multiplexer 41 selectively outputs the output signal from the operational amplifier A2 or A3 to the switched capacitor filter 7 in accordance with the rotation direction of the motor M recognized through the rotation control unit 10A. That is, when the motor M is rotating forward, the operational amplifier A3 and the switched capacitor filter 7 are connected, and when the motor M is rotating reversely, the operational amplifier A2 and the switched capacitor filter 7 are connected. become.

  On the other hand, the motor current i output from the operational amplifiers A2 and A3 is taken into the AD converter 51 together with the voltage Vm between the terminals of the motor M detected through the operational amplifier A1, and is converted into a digital signal. Based on this digital signal, the counter electromotive voltage calculation unit 52 calculates the above-described counter electromotive voltage Vg. The AD converter 51 and the counter electromotive voltage calculation unit 52 constitute the counter electromotive voltage estimation unit 5 (FIG. 1).

  The counter electromotive voltage Vg calculated by the counter electromotive voltage calculation unit 52 is input to a clock frequency setting unit 6 including a frequency division number setting unit 61 and a variable frequency divider 62. Among these, the frequency division number setting unit 61 sets the frequency division number based on the back electromotive voltage Vg. Further, the variable frequency divider 62 divides a reference clock input separately based on the set frequency dividing number, and generates a clock having a frequency f corresponding to the rotational speed of the motor M. The frequency pass band of the switched capacitor filter 7 is changed to a band corresponding to the rotation speed of the motor M by the clock thus generated.

  The switched capacitor filter 7 extracts a ripple component included in the motor current i detected by the motor current detection unit 4, and the extracted ripple component is binarized through the comparator 8 to generate a pulse. As described above, it becomes a signal. Then, the pulse signal is taken into the rotation amount calculation unit 9 realized by the CPU, and rotation information of the motor M is detected.

  In FIG. 4, the previous motor rotation control unit 10 (FIG. 1) is constituted by the rotation control unit 10A and the gate drive circuit 10B described above. Of these, the rotation control unit 10A is a part that gives a command to the gate drive circuit 10B to rotate or stop the motor M, or to perform forward or reverse rotation based on the detected rotation information of the motor M. The gate drive circuit 10B generates and outputs a gate drive signal for each transistor described above based on the given command. The rotation control unit 10A also instructs the multiplexer 41 which output of the operational amplifier A2 or A3 should be selected according to the rotation direction of the motor M.

As described above, according to the present embodiment, the following effects can be obtained.
(1) When the ripple component of the motor current i is extracted using the switched capacitor filter 7 and the rotation information of the motor M is detected based on the ripple component of the current, the counter electromotive voltage Vg proportional to the rotation speed of the motor M is obtained. The frequency pass band of the switched capacitor filter 7 was changed using this. For this reason, there is no contradiction in control, and the ripple component of the motor current i can be extracted with high accuracy, so that more reliable motor rotation information can be detected.

  (2) The motor current i is extracted using the current mirrors C1 and C2. As a result, the current detection resistors R1 and R2 can be built in the IC chip, and the design of the motor current detection unit 4 and, consequently, the motor rotation information detection device as a whole can be made compact.

  (3) In addition, the counter electromotive voltage Vg is estimated from the motor terminal voltage Vm and the motor current i using an arithmetic unit such as a CPU. For this reason, it is not necessary to use an external resistor or the like as in the prior art, and in this respect also, a more compact device design can be realized.

In addition, the said embodiment can also be implemented as the following forms, for example.
In the above embodiment, when calculating the back electromotive force Vg, both the internal resistance r and the internal inductance L of the motor M are taken into account, but the voltage drop due to the internal inductance L of the motor M is generally small. . For this reason, this term may be omitted in the above formula (1). That is, the following formula (2)

Vg = Vm−r · i (2)

May be executed to estimate the back electromotive voltage Vg. Thereby, the estimation of the back electromotive voltage Vg is further simplified.

In the above embodiment, the internal resistance r of the motor M is known and used as a fixed value. However, in practice, the resistance value may fluctuate due to manufacturing variations, temperature characteristics of resistance, and the like. There is also sex. Therefore, in order to correct such variation in resistance value, particularly when the above equation (2) is employed, the internal resistance r may be obtained individually. As a specific method,
(A) A state where the voltage Vm between the terminals is reduced to such an extent that the motor M cannot rotate. Or
(B) A state in which the rotation of the motor M is mechanically stopped.
And operation

r = Vm / i (3)

To calculate the internal resistance r. Further, the inter-terminal voltage Vm1 and the motor current i1 at the time of steady rotation of the motor and the inter-terminal voltage Vm2 and the motor current i2 immediately after switching to the braking operation of the motor are obtained, respectively, and the following simultaneous equations

Vg = Vm1-r · i1
Vg = Vm2-r · i2 (4)

Alternatively, the internal resistance r of the motor may be obtained. As a result, the influence of variations in the resistance value of the internal resistance r is suitably suppressed, and the back electromotive voltage Vg can be obtained more accurately. In addition, if the internal resistance r of the motor M is obtained in this way, the same rotation information detection device can be applied to another motor.

  In the above embodiment, the back electromotive voltage Vg is estimated as a solution of the above equation (1) or (2) during the steady operation of the motor M. However, the present invention is not limited to the case where the motor M is in the steady operation state. Even in the braking operation state, the back electromotive voltage Vg may be obtained using these equations. During such braking operation of the motor M, as shown in FIG. 5, the direction of the current i flowing through the motor changes from the direction of the solid line arrow in the figure to the direction of the broken line arrow, but this time is detected by the resistor R1. Since the current also takes a negative value, the back electromotive voltage Vg can be calculated using the formula (1) or the formula (2) as described above.

(Second Embodiment)
Next, a second embodiment in which the motor rotation information detecting device according to the present invention is embodied will be described in detail with reference to FIGS.

  The motor rotation detection device according to the present embodiment is reversely performed using hardware, specifically an analog circuit, without using an arithmetic device such as a CPU as in the first embodiment. The electromotive voltage Vg is obtained.

First, FIG. 6 shows an outline of a circuit for estimating the counter electromotive voltage Vg in a state where the motor M is rotating forward.
As shown in FIG. 6, the back electromotive voltage estimation unit 105 employed in the present embodiment is configured as a series circuit of a variable resistance unit VR1 and a fixed resistance unit R3. In the present embodiment, as shown in FIG. 6, the motor M is equivalently represented by a back electromotive voltage Vg and an internal resistance r. The motor M and the back electromotive force estimation unit 105 are connected in a current mirror manner by a motor driving transistor M3 and a sense transistor M7. As described above, the current mirror connection causes the counter electromotive voltage estimation unit 105 to flow a current of “1 / m” of the motor current i according to the current mirror ratio (1: m) of the current mirror. Become.

  Here, the resistance value of the fixed resistor portion R3 is set to a value obtained by multiplying the internal resistance r of the motor by the current mirror ratio, that is, “mr”. Therefore, the voltage drop “mr · i / m = r · i” generated in the fixed resistance portion R3 is equal to the voltage drop “r · i” generated in the internal resistance r of the motor.

  On the other hand, in the present embodiment, the inter-terminal voltage (voltage between T1 and T2) of the back electromotive force estimation unit 105 and the inter-terminal voltage Vm of the motor M (T3- (Voltage between T4) is held to be equal.

  According to such a circuit, the voltage drop generated in the variable resistor VR1 of the counter electromotive voltage estimating unit 105 is equal to the motor counter electromotive voltage Vg. Therefore, if the voltage generated in the variable resistor VR1 is detected, The counter electromotive voltage Vg can be obtained (estimated).

FIG. 7 shows more specifically a circuit for estimating the back electromotive voltage Vg in a state where the motor M whose outline is shown in FIG. 6 is rotating forward (steady rotation).
As shown in FIG. 7, in this circuit, transistors M9 and M7 that are current-mirror connected to the transistors M2 and M3 in the drive circuit of the motor M are provided. The back electromotive force estimation unit 105 is connected between the source terminal of the transistor M9 and the drain terminal of the transistor M7.

  Further, the terminal T1 of the back electromotive voltage estimation unit 105 and the terminal T3 of the motor M are respectively input to an operational amplifier (differential amplifier) A4, and the terminals T1 and T3 are connected to each other using the virtual ground of the operational amplifier A4. The terminal voltages are equal. Further, due to the current mirror configuration of the transistors M3 and M7, the drain voltage of these transistors M3 and M7, that is, the voltage at the terminal T2 of the back electromotive force estimation unit 105 and the voltage at the terminal T4 of the motor M are equal. In this way, the terminal voltage of the back electromotive voltage estimation unit 105 and the terminal voltage of the motor M are equal to each other.

  Further, as shown in FIG. 7, in the present embodiment, a field effect transistor M11 is used as the variable resistance portion VR1. That is, in the variable resistance portion VR1, the output voltage (differential voltage) of the operational amplifier A4 is applied to the gate terminal of the transistor M11, and the drain / The resistance value between the sources is variably controlled. For example, when the rotational speed of the motor M changes due to some factor such as load fluctuation, the back electromotive voltage Vg also changes in proportion to this rotational speed, and the voltage at the terminal T3 of the motor M also changes in conjunction with this. Now, assuming that the rotational speed of the motor M has changed, for example, in an increasing direction, the output voltage of the operational amplifier A4 increases to the negative side accordingly. At this time, since the resistance value between the drain and the source of the transistor M11 constituting the variable resistance portion VR1 also increases, the voltage drop between the drain and the source of the transistor M11 increases, and the back electromotive force The voltage at the terminal T1 of the voltage estimation unit 105 becomes equal to the voltage at the terminal T3 of the motor M again.

  In the circuit illustrated in FIG. 7, the resistance value of the variable resistor portion VR1 is variably controlled so as to follow the change in the back electromotive voltage Vg due to the change in the rotation speed of the motor M. As described above, if the voltage drop generated in the variable resistor VR1, that is, the drain-source voltage of the transistor M11 is detected, the counter electromotive voltage Vg of the motor M is obtained (estimated) each time. Will be able to.

  The circuits illustrated in FIGS. 6 and 7 are applied when the motor M rotates in the forward direction. When the motor M rotates in the reverse direction although it is in a steady rotation, the circuit for the motor M is used. On the other hand, it is necessary to provide a circuit that is symmetrical to the above circuit. However, even in this case, the detection principle of the back electromotive voltage Vg is basically the same as the above-described principle, and the detailed description thereof will be omitted.

Next, a circuit for estimating the back electromotive voltage Vg when the motor M changes from the steady rotation state to the braking operation state will be described.
In the steady operation (steady rotation) state of the motor M, the relationship between the motor terminal voltage Vm and the counter electromotive voltage Vg is as shown in the above equation (2), but the motor M is in the braking operation state from the steady operation state. , The direction of the motor current i is reversed, so the relationship between the motor terminal voltage Vm and the back electromotive voltage Vg is

Vm = Vg−r · i (5)

It becomes. Therefore, in this case, an estimation circuit for the back electromotive voltage Vg different from the circuits illustrated in FIGS. 6 and 7 is required.

FIG. 8 shows an outline of a circuit for estimating the counter electromotive voltage Vg when the motor M shifts from the steady operation state to the braking operation state.
In the braking operation state of the motor M, in the circuit shown in FIG. 7, the transistors M1 and M2 are turned off and the transistors M3 and M4 are turned on. In this case, in this embodiment, as shown in FIG. 8, a sense transistor M8 that is current-mirror connected to the transistor M4 with a predetermined current mirror ratio (1: m) is provided, and the drain of the transistor M8 is provided. The counter electromotive voltage estimation unit 205 is connected to the side. With such a current mirror configuration, a current “1 / m” of the motor current i flows through the back electromotive force estimation unit 205 in accordance with the current mirror ratio “1: m”.

  Again, the back electromotive force estimation unit 205 is configured as a series circuit of a fixed resistor unit R4 and a variable resistor unit VR2. The resistance value of the fixed resistor R4 is set to a value obtained by multiplying the internal resistance r of the motor by the current mirror ratio, that is, “mr”. For this reason, the voltage drop “mr · i / m = i · r” generated in the fixed resistor R4 is equal to the voltage drop “i · r” generated in the internal resistance r of the motor.

  As described above, when the motor M shifts from the steady operation state to the braking operation state, the direction of the motor current i is reversed, so that the counter electromotive voltage Vg is obtained based on the above equation (5). Become. In addition, since the voltage drop generated in the fixed resistor R4 of the back electromotive force estimation unit 205 is equal to the voltage drop generated in the motor internal resistance r, the voltage drop generated in the variable resistor VR2 and the motor terminal voltage Vm If the circuits are configured to be equal to each other, the counter electromotive voltage Vg of the motor M can be obtained as the terminal voltage of the counter electromotive voltage estimation unit 205.

FIG. 9 shows more specifically a circuit for estimating the back electromotive voltage Vg when the motor M whose outline is shown in FIG. 8 is in a braking operation state.
As shown in FIG. 9, in this circuit, a transistor M12 functioning as a variable resistor and the back electromotive force estimation unit 205 are connected in series to the drain terminal of a sense transistor M8 that is current-mirror connected to the transistor M4. It is connected. Among these, the output terminal of the operational amplifier A5 is connected to the gate terminal of the transistor M12, and the drain terminal (the terminal T3 of the motor M) of the transistor M4 is connected to the non-inverting input terminal and the inverting input terminal of the operational amplifier A5, respectively. And the drain terminal of the transistor M8. That is, here, using the virtual ground of the operational amplifier A5, the drain voltages of the transistors M8 and M4 are maintained to be equal to each other. Thereby, the current mirror ratio as a current mirror circuit constituted by the transistors M4 and M8 is also maintained constant.

  On the other hand, in this circuit, as shown in FIG. 9, a P-channel transistor M10 is provided as the variable resistor portion VR2. The source terminal of the transistor M10 is directly connected to the drive power supply E1 of the motor M. The voltage at the terminal T3 of the motor M is input to the inverting input terminal of the operational amplifier A6, and the voltage at the terminal T4 of the motor M and the source voltage Vs of the transistor M10 (the driving power supply E1 of the motor M) are the same operational amplifier. It is input to the non-inverting input terminal of A6. Therefore, “Vs−Vm” is output as an output voltage (differential voltage) to the output terminal of the operational amplifier A6, and the output voltage (differential voltage) of the operational amplifier A6 is further input to the inverting input of the operational amplifier A7. Input to the terminal. The drain terminal of the transistor M10 is connected to the non-inverting input terminal of the operational amplifier A7, and the output terminal of the operational amplifier A7 is connected to the gate terminal of the transistor M10. Therefore, in the variable resistance unit VR2, the drain voltage Vd of the transistor M10 is controlled to be equal to the output voltage “VS−Vm” of the operational amplifier A6 by the virtual ground of the operational amplifier A7. That is, the voltage “Vd−Vs” between the drain and source terminals of the transistor M10 is a voltage corresponding to the terminal voltage Vm of the motor M.

  In the circuit shown in FIG. 9, the voltage drops generated in the fixed resistor portion R4 and the variable resistor portion VR2 are “r · i” and “Vm”, respectively, and the back electromotive force estimating portion 205 is configured. The voltage between the terminals of the series circuit of the fixed resistor portion R4 and the variable resistor portion VR2 is the back electromotive voltage Vg (Vg = Vm + r · i).

Next, resistance value control of the transistor M10 constituting the variable resistance portion VR2 in this circuit will be described.
When the rotational speed of the motor M changes due to some factors such as load fluctuations even though it is in the braking operation state, the back electromotive force Vg also changes in proportion to this rotational speed, as described above. The voltage at the terminal T3 of the motor M also changes. For example, when the rotational speed of the motor M increases due to a load reduction or the like, the back electromotive voltage Vg increases accordingly, and the voltage at the terminal T3 of the motor M increases. At this time, the output voltage of the operational amplifier A6 decreases, and conversely, the output of the operational amplifier A7, that is, the voltage at the gate terminal of the transistor M10 increases. Here, since the transistor M10 is a P-channel transistor as described above, the resistance value between the source and the drain increases as the gate voltage increases. For this reason, the voltage drop between the source and drain of the transistor M10 also increases following the change in the back electromotive voltage Vg.

  The circuits illustrated in FIGS. 8 and 9 are applied when the motor M shifts from the normal rotation state to the braking operation state in the normal rotation, and the motor M is also in the normal rotation. When shifting from the reverse rotation state to the braking operation state, it is necessary to separately provide a circuit symmetrical to the above circuit with respect to the drive circuit of the motor M. However, even in this case, the detection principle of the back electromotive voltage Vg is basically the same as the above-described principle, and the detailed description thereof will be omitted.

  However, FIG. 10 shows the entire circuit including the above-described symmetrical circuits, including the circuit for obtaining (estimating) the back electromotive voltage Vg in the steady rotation state described above. Since all the configurations and operations are the same as those described above, overlapping description is omitted here. In FIG. 10, elements that are symmetrical to the circuit elements shown in FIG. 7 and FIG. Is attached, and “′” is attached to the reference numerals to achieve the correspondence.

Further, in the present embodiment, in addition to the part that controls the gate drive circuit 10B shown in FIGS. 7, 9, and 10, the following part, that is, based on the back electromotive voltage Vg obtained (estimated) above. 4 is a part for setting the frequency division number of the variable frequency divider 62 shown in FIG.
A part for calculating the rotation amount and the like from the rotation information of the motor M obtained through the comparator 8.
Etc. are also configured by hardware that does not use an arithmetic device such as a CPU.

  For example, with respect to the part for setting the frequency dividing number, a circuit for AD converting the obtained counter electromotive voltage Vg and then converting the required frequency dividing number into a table based on the converted digital code, etc. It can be adopted. In addition, a counter circuit that counts the pulse signal output from the comparator 8 can be employed as the part for calculating the rotation amount and the like. As a part for controlling the gate drive circuit 10B, for example, a digital comparator for comparing each count value of the counter circuit with a desired value is provided. A circuit that outputs a reverse or braking command can be employed. However, these parts are parts that are flexibly designed according to the specifications desired as the motor rotation information detection device, and these functions are appropriately realized as custom LSIs according to the desired specifications. It becomes.

  As described above, according to the present embodiment, the same effects as the effects (1) and (2) of the first embodiment can be obtained. Furthermore, this motor rotation information detection device can be realized as a simpler system by not using an arithmetic device such as a CPU.

(Other embodiments)
It should be noted that each of the above embodiments can be implemented with the following modifications.
In each of the above embodiments, the motor rotation information is detected based on the ripple component extracted from the motor current i. However, the motor is not limited to this ripple component but based on other periodic components such as a surge component. You may make it detect rotation information.

  In each of the above embodiments, the waveform of the periodic component extracted from the motor current i is binarized (converted into a pulse signal) through a comparator. However, in an analog circuit or the like, based on the waveform of the periodic component When the motor rotation information can be directly grasped, the arrangement of the comparator and the like can be omitted. Of course, when such rotation information is computer-processed or digitally processed, the binarization makes it easier to detect the rotation information.

  In each of the above embodiments, the ripple component is extracted from the waveform of the current flowing through the motor and the motor rotation information is detected. However, the present invention is not limited to this, and ripple from other motor drive waveforms such as the voltage across the motor terminals Periodic components such as components and surge components may be extracted to detect motor rotation information.

The block diagram which shows the outline of the structure about 1st Embodiment of the motor rotation information detection apparatus concerning this invention. The circuit diagram which shows the example of a bridge circuit which detects a back electromotive voltage. The circuit diagram which shows the equivalent circuit of the motor made into the detection object of rotation information in the apparatus of 1st Embodiment. The block diagram which shows the structure of the motor rotation information detection apparatus of the 1st embodiment in detail. The circuit diagram which shows the structure about the motor current detection part mainly of the motor rotation information detection apparatus of the said 1st Embodiment. The circuit diagram which shows the outline | summary of the circuit which detects the back electromotive voltage at the time of steady operation (steady rotation) of a motor in 2nd Embodiment of the motor rotation information detection apparatus concerning this invention. FIG. 7 is a circuit diagram showing a specific example of the circuit shown in FIG. 6. The circuit diagram which shows the outline | summary of the circuit which detects the back electromotive force at the time of the braking driving | running of a motor in 2nd Embodiment of the motor rotation information detection apparatus concerning this invention. FIG. 9 is a circuit diagram showing a specific example of the circuit shown in FIG. 8. The circuit diagram which shows the whole structure of the part which estimates the counter electromotive voltage of a motor about the motor rotation information detection apparatus of 2nd Embodiment. The circuit diagram which shows the example about the conventional method of extracting the periodic component from a motor current. (A), (b) and (c) is a time chart which shows the extraction aspect of the periodic component from the motor current by the circuit illustrated in FIG.

Explanation of symbols

  M: Motor, 2 ... Brush, 3 ... Motor terminal voltage detector, 4 ... Motor current detector, 41 ... Multiplexer, 5, 105, 205 ... Back electromotive force estimator, 51 ... AD converter, 52 ... Back electromotive Voltage calculation unit, 6 ... clock frequency setting unit, 61 ... frequency division number setting unit, 62 ... variable frequency divider, 7 ... switched capacitor filter, 8 ... comparator, 9 ... rotation amount calculation unit, 10 ... motor rotation control unit DESCRIPTION OF SYMBOLS 10A ... Rotation control part, 10B ... Gate drive circuit, M1-M12 ... Transistor (field effect transistor), C1, C2 ... Current mirror circuit, A1-A7 ... Operational amplifier, VR1, VR2 ... Variable resistance part, R3, R4 ... fixed resistance part.

Claims (18)

In the motor rotation information detecting device for extracting the periodic component waveform from the motor driving waveform of the DC brush motor by a filter capable of changing the frequency pass band, and detecting the rotation information of the motor based on the extracted periodic component waveform.
A motor inter-terminal voltage detection unit that detects a voltage between the terminals of the motor, a motor current detection unit that detects a current flowing through the motor, and a reverse generated in the motor based on the detected inter-terminal voltage and current. A motor rotation information detection device comprising: a counter electromotive voltage estimation unit that estimates an electromotive voltage; and changing a frequency pass band of the filter based on the estimated counter electromotive voltage. The filter is a switched capacitor filter whose frequency pass band is variable based on the frequency of the applied clock, and the frequency of the applied clock is changed to a counter electromotive voltage estimated through the counter electromotive voltage estimation unit. The motor rotation information detection device according to claim 1, which is determined based on the determination. The motor rotation information detection device according to claim 1, wherein the switched capacitor filter extracts one of a ripple component waveform and a surge component waveform as a periodic component waveform from the motor drive waveform. The motor current detection unit includes a current sense transistor having a current mirror configuration that is connected in parallel with a motor drive transistor that drives the motor, and detects a current that flows through the motor based on a current that flows through the current sense transistor. The motor rotation information detection device according to any one of Items 1 to 3. The back electromotive force estimation unit is configured such that the detected terminal voltage of the motor is Vm, the current flowing through the motor is i, the internal resistance specific to the motor is r, and the internal inductance specific to the motor is L, When the back electromotive force is Vg, the following differential equation:

Vg = Vm-r.i-L (di / dt)

The motor rotation information detection device according to claim 1, wherein the counter electromotive voltage Vg is estimated as a solution of The counter electromotive voltage estimation unit calculates the voltage between the terminals of the detected motor as Vm, the current flowing through the motor as i, the internal resistance specific to the motor as r, and the counter electromotive voltage as Vg.

Vg = Vm−r · i

The motor rotation information detecting device according to any one of claims 1 to 4, wherein the counter electromotive voltage Vg is estimated by executing. In the motor rotation information detection device according to claim 6,
The internal resistance r inherent to the motor is calculated based on the inter-terminal voltage Vm and the motor current i when the inter-terminal voltage is so small that the motor cannot rotate.

r = Vm / i

This is a value obtained by executing the motor rotation information detecting device. In the motor rotation information detection device according to claim 6,
The internal resistance r inherent to the motor is calculated based on the voltage Vm between the terminals and the motor current i in a state where the rotation of the motor is mechanically stopped.

r = Vm / i

This is a value obtained by executing the motor rotation information detecting device. In the motor rotation information detection device according to claim 6,
The internal resistance r inherent to the motor is represented by Vm1 as a voltage across the terminals during steady rotation of the detected motor, i1 as a current flowing through the motor at the same time, and during the braking operation of the detected motor. When the voltage between the terminals immediately after switching is Vm2, and the current flowing through the motor at that time is i2, the following simultaneous equations

Vg = Vm1-r · i1
Vg = Vm2-r · i2

A motor rotation information detection device characterized in that the value obtained as a solution of The back electromotive force estimation unit is connected in series to a fixed resistance unit that generates a voltage drop equivalent to the internal resistance of the motor with respect to the current detected by the motor current detection unit. The resistance value is variable so that the sum of the voltage generated between the terminals during steady rotation of the motor and the voltage between the terminals of the fixed resistance unit is equal to the voltage detected by the voltage detection unit between the terminals of the motor. The motor rotation information detection device according to claim 1, further comprising: a variable resistance unit to be controlled, wherein the counter electromotive voltage is estimated based on a voltage generated between terminals of the variable resistance unit. A series circuit of the fixed resistor unit and the variable resistor unit constituting the back electromotive voltage estimation unit is connected to the motor drive circuit with a current mirror ratio with a predetermined current mirror ratio, and the fixed resistor unit The resistance value of the unit is set to a value obtained by multiplying the internal resistance of the motor by the current mirror ratio, and the voltage between the terminals of the series circuit of the fixed resistance unit and the variable resistance unit configured by these current mirrors and the motor The inter-terminal voltage is equalized by using a virtual ground of an operational amplifier to which a terminal voltage of a series circuit of the fixed resistance unit and the variable resistance unit and a terminal voltage of the motor are respectively input. The motor rotation information detection device according to 10. The variable resistance portion is configured to include a field effect transistor, and the resistance value between the source and drain terminals is based on the fact that the output of the virtual grounded operational amplifier is applied to the gate terminal of the field effect transistor. The motor rotation information detection device according to claim 11, which is variably controlled. The back electromotive force estimation unit is connected in series to a fixed resistance unit that generates a voltage drop equivalent to the internal resistance of the motor with respect to the current detected by the motor current detection unit. A variable resistance unit whose resistance value is variably controlled so that a voltage generated between the terminals in accordance with the braking operation of the motor becomes equal to a voltage detected by the voltage detection unit between the terminals of the motor. The motor rotation information detection device according to any one of claims 1 to 4, wherein the back electromotive force is estimated based on a voltage generated between terminals of a series circuit of a resistance portion and a variable resistance portion. A series circuit of the fixed resistor unit and the variable resistor unit constituting the back electromotive voltage estimating unit is connected to the motor drive circuit with a current mirror ratio with a predetermined current mirror ratio, and the fixed resistor unit The resistance value of the motor is set to a value obtained by multiplying the internal resistance of the motor by the current mirror ratio, and the voltage between the terminals of the variable resistance unit and the voltage between the terminals of the motor configured by these current mirrors are the variable resistance unit. The motor rotation information detection device according to claim 13, wherein the terminal voltage of the motor and the terminal voltage of the motor are equalized by using virtual ground of an operational amplifier to which the terminal voltage is input. The variable resistance portion is configured to include a field effect transistor. When the drain voltage of the field effect transistor is Vd, the source voltage of the field effect transistor is Vs, and the terminal voltage of the motor is Vm, an operational amplifier is used. The motor rotation information detecting device according to claim 14, wherein a resistance value between the source and drain terminals is variably controlled based on an output of Vd- (Vs-Vm) being applied to a gate terminal of the field effect transistor. . The motor rotation information detection according to any one of claims 1 to 15, wherein a periodic component waveform of the current extracted by the filter capable of changing the frequency pass band is pulsed through a binarization process by a comparator. apparatus. The motor rotation information detection device according to any one of claims 1 to 16, wherein a motor driving waveform of the DC brush motor is a waveform of a current flowing through the DC brush motor. The motor rotation information detection device according to any one of claims 1 to 16, wherein a motor driving waveform of the DC brush motor is a waveform of a voltage between terminals of the DC brush motor.

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