JP2005210016A - Frequency selecting device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a frequency selecting device in which frequency selectivity can be easily adjusted and which is excellent in the selection performance. <P>SOLUTION: The frequency selecting device includes a plurality of layers, each of which has a plurality of metal conductors 10 that are regularly arranged in an x-y plane, while the layers are arranged at prescribed intervals in z-axis direction. In a frequency range higher than a resonance frequency determined by the shape and the intervals in the arrangement of the metal conductors 10, the frequency selecting device exhibits negative equivalent dielectric constants, and shows the property of total reflection for the electromagnetic waves. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a frequency selection device, and more particularly to a structure for selectively transmitting or reflecting an electromagnetic wave having a desired frequency.

  A frequency selective surface (FSS) is widely used as a maintenance member for a radio wave environment in an office or the like by utilizing a characteristic of selectively blocking or absorbing electromagnetic waves of a specific frequency.

  FIG. 22 is a configuration diagram for explaining the principle of the frequency selective surface. The configuration shown in FIG. 22 is well known as an example of a frequency selective surface as described in Non-Patent Document 1, for example.

  Referring to FIG. 22, the frequency selective surface is formed by mixing metal conductor 100 with dielectric plate 110 such as polymer or glass. The shape of the metal conductor 100 includes a strip shape and a cross shape in addition to a square ring shape as shown in FIG. 22, and is arranged at equal intervals on the xz plane. FIG. 22 shows a configuration in the case of a single layer. However, in the case of a multilayer configuration, the xz plane is arranged as one layer, and a plurality of layers are arranged in parallel with a predetermined interval in the y-axis direction. Is done.

  In this structure, the frequency selective surface has a resonant frequency that depends on the shape of the individual metal conductors 100 and the spacing at which the metal conductors 100 are arranged. In the vicinity of this resonance frequency, as shown in FIG. 22, the transmission coefficient τ of the entire frequency selective surface shows 0, while the reflection coefficient Γ shows 1. Thereby, only the electromagnetic waves corresponding to the resonance frequency among the incident electromagnetic waves are selectively reflected, and the frequency selective surface functions as a band rejection filter that reflects only the electromagnetic waves of a specific frequency.

  FIG. 23 is an equivalent circuit diagram of a frequency selective surface having a multilayer structure.

  Referring to FIG. 23, the frequency selective surface has a structure in which three single-layer frequency selective surfaces shown in FIG. 22 are laminated. The equivalent circuit of each layer is represented by an inductance L, a capacitance C, and a resistance element R coupled in series. Further, the one layer and the other layer are coupled in parallel on the transmission line with a distance d. The distance d actually corresponds to the thickness of the spacer inserted between the layers.

  Here, when focusing on a certain layer, the inductance L, the capacitance C, and the resistance element R constitute a series resonance circuit, the impedance of the layer becomes minimum at the resonance frequency, and the current flowing through each metal conductor becomes maximum. Since each element parameter of the resonance circuit changes depending on the shape and arrangement interval of the metal conductors, it can be said that the resonance frequency uniquely determined by these element parameters also varies depending on the shape of the metal conductor and the arrangement interval.

Furthermore, in a multilayer structure, it can be considered that the band rejection filter which has a predetermined resonant frequency for every single layer was arranged on the transmission line for every space | interval d. Therefore, the filter characteristics over the entire frequency selective surface will be determined by the resonant frequency of each layer and the spacing between layers. In other words, by adjusting the resonance frequency of each layer and the distance between the layers, a frequency selective surface having a desired frequency selectivity can be formed.
Ben A. Munk, “Frequency Selective Surfaces,” John Willey & Sons, Inc., pp.1-6, 2000.

  As described above, in the conventional frequency selective surface, the frequency selectivity is determined by the resonance frequency of each constituting layer and the distance between the layers. Further, the resonance frequency for each layer is determined by the shape and arrangement interval of the individual metal conductors. Therefore, desired frequency selectivity can be obtained by adjusting the shape and arrangement interval of metal conductors and the interval between layers.

  However, in practice, the frequency selective surface is formed by forming a metal conductor by etching or the like on a printed circuit board or a resin film, and holding it at a predetermined interval with a dielectric material such as a foam material or a resin. Arrange the conductors in space. For this reason, it is practically impossible to change the shape and arrangement interval of the metal conductors after the manufacture.

  In addition, even if the desired frequency selectivity cannot be obtained due to an error between the finished shape and the design value due to variations in the manufacturing process, it is time-consuming and costly to adjust the frequency by the tuning. It has also been considered difficult.

  Therefore, the present invention has been made to solve such problems of the frequency selective surface, and an object thereof is to provide a frequency selective device capable of easily adjusting the frequency selectivity. It is to be.

  Another object of the present invention is to provide a frequency selection device having excellent frequency selection performance.

  According to one aspect of the present invention, a layer in which a plurality of metal conductors are regularly arranged on a plane is defined as one layer, and a single layer is provided with a plurality of layers that are stacked in parallel at a constant interval. In the cut-off frequency band including the resonance frequency of the metal conductor, electromagnetic waves incident on the plurality of layers are blocked, and in the pass frequency band equal to or lower than the cut-off frequency band, the electromagnetic waves incident on the plurality of layers are transmitted.

  Preferably, the equivalent relative permittivity when a plurality of layers are viewed as a macroscopically homogeneous dielectric exhibits a value of 1 or more in a frequency band below the resonance frequency, and a negative value in a cutoff frequency band above the resonance frequency. Indicates.

  Preferably, the plurality of layers transmit electromagnetic waves having a frequency that is an integral multiple of one half of the wavelength of the electromagnetic waves propagating through a dielectric having an equivalent dielectric constant.

  Preferably, an electromagnetic wave including a frequency at which the thickness of the plurality of layers is not more than one fifth of the wavelength of the electromagnetic wave propagating through a dielectric having an equivalent relative dielectric constant is transmitted.

  More preferably, the plurality of layers further includes a plurality of first and second dielectric parallel plates that are joined either alone or together before and after the plurality of metal conductors. The thickness of the plurality of layers is the sum of the thicknesses of the plurality of first and second dielectric parallel plates.

  According to another aspect of the present invention, a frequency-selective surface joined to a plurality of layers at a predetermined interval is further provided, and electromagnetic waves of some frequencies are blocked in the pass frequency band.

  According to another aspect of the present invention, a dielectric parallel plate or a frequency-selective surface joined to a plurality of layers at a predetermined interval is further provided, and an electromagnetic wave having a specific frequency in a pass frequency band. To cancel the reflection.

  Preferably, the dielectric parallel plate is one of the dielectric plate and the second plurality of layers.

  According to another aspect of the present invention, the apparatus further includes a frequency selective surface or a second plurality of layers bonded to the plurality of layers at a predetermined interval, and the frequency selective surface or the second plurality of layers is further provided. This layer has a characteristic of blocking electromagnetic waves having a frequency not more than 1.1 times the upper limit frequency of the cutoff frequency band or not less than 0.9 times the lower limit frequency.

  According to an aspect of the present invention, by applying a conventional artificial dielectric technology to a frequency selection device and expanding it to a frequency range equal to or higher than the resonance frequency, an electromagnetic wave in a cutoff frequency band including the resonance frequency is selectively selected. In addition to blocking, it is possible to easily and accurately realize frequency selectivity of transmitting an electromagnetic wave in a transmission frequency band lower than the cutoff frequency band with low loss.

  Furthermore, since the current induced in the metal conductor at the transmission frequency is small, the loss can be suppressed as compared with the conventional frequency selective surface.

  According to another aspect of the invention, frequency selection is performed by joining a conventional frequency selective surface to the frequency selective device with a predetermined interval and adjusting the resonant frequency of the frequency selective surface and the interval. It is possible to easily realize frequency selectivity that blocks a part of the transmission frequency of the device.

  According to another aspect of the present invention, the frequency selection devices are arranged opposite to each other with a predetermined interval on one frequency selection device, and the reflection of a desired frequency is reduced by adjusting the interval. Can do.

  According to another aspect of the present invention, a conventional frequency selective surface or frequency selection device is joined to the frequency selection device, and these cutoff frequency bands are adjusted to be close to the cutoff frequency band of the frequency selection device. Thus, the cut-off frequency band of the frequency selection device can be expanded.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. In the drawings, the same reference numerals indicate the same or corresponding parts.

[Embodiment 1]
First, the basic principle of the frequency selection device according to the first embodiment of the present invention will be described. In summary, the present invention is intended to facilitate the adjustment of the frequency selectivity of the conventional frequency selective surface and improve the selection performance by expanding and applying the existing artificial dielectric technology. .

  FIG. 1 is a diagram schematically showing the structure of an existing artificial dielectric. In addition, this structure is an example of an artificial dielectric, by Klaus, “Aerial Line”, Modern Science Co., pp. It is well known as described in 454-458.

  Referring to FIG. 1, an artificial dielectric 200 is a conventional non-metallic dielectric composed of microscopic sized molecular particles, whereas a macroscopic sized separate metal conductor 10. Made from. In addition to the sphere, the metal conductor 10 includes a strip shape or a cylinder. As shown in FIG. 1, the metal conductors 10 are arranged in a three-dimensional lattice structure to form a convex lens. Such an arrangement is similar to the crystal lattice of a normal dielectric material, only with much larger dimensions.

  When each of the metal conductors 10 receives an electromagnetic wave from a wave source, a polarization phenomenon is caused by electrostatic induction, and positive and negative charges are induced in the direction of the electric field E. These charge effects can be represented by point charges with + q and -q separated by a distance l. Thus, each metal conductor 10 constitutes an electric dipole having a dipole efficiency ql. That is, when viewed macroscopically, the arrangement of the metal conductors 10 shown in FIG. 1 is continuously distributed at intervals shorter than the wavelength, and thus can be regarded as a dielectric that is polarized by the electric field E.

  Here, for each metal conductor 10, when the magnitude l in the direction parallel to the electric field E is equal to one half of the wavelength of the electromagnetic wave, the metal conductor 10 is in a resonance state and a large current is induced. Are known. Furthermore, since the movement of positive and negative charges in the metal conductor 10 cannot catch up with the speed of the electromagnetic wave, the phase is delayed. As a result, the equivalent dielectric constant of the artificial dielectric as a whole greatly varies depending on the frequency of the electromagnetic wave when the size 1 of the metal conductor is in the vicinity of half the wavelength.

  Therefore, in order to avoid the influence of this resonance, the size l of the metal conductor 10 must be smaller than half the wavelength at the operating frequency. In particular, it has already been found that when the size l of the metal conductor 10 is equal to or less than a quarter of the wavelength, the dielectric constant is stable regardless of the frequency. For this reason, in the conventional artificial dielectric, the size of the metal conductor is generally designed to be about one quarter or less of the wavelength.

  In addition, the interval s between the adjacent metal conductors 10 in the propagation direction of electromagnetic waves is shorter than one wavelength in order to avoid the influence of diffraction.

  As described above, the conventional artificial dielectric is intended to stably maintain the equivalent dielectric constant as a whole by using a shape and frequency that avoids resonance that may occur in individual metal conductors. This time, focusing on the variation of the equivalent permittivity when the artificial dielectric that had been avoided until now was resonated, and applying this to the frequency selection device, the frequency selection performance could be improved. In the following, the equivalent dielectric constant of the frequency selection device is obtained, and the frequency selectivity derived therefrom will be described in detail.

  For the calculation of the equivalent dielectric constant when the frequency selection device according to the present invention is macroscopically regarded as a dielectric, a model in which metal conductors of finite length shown in FIG. 2 are arranged is used. As shown in FIG. 2, each metal conductor has a strip shape, a length l = 15 mm, a width w = 0.75 mm, and a thickness t that is infinitely small. The metal conductors are regularly arranged at intervals of Δx = 2.25 mm and Δy = 3 mm in the x-axis direction and the y-axis direction, respectively. Further, the metal conductors on the xy plane are arranged as one layer, and five layers are arranged in the z-axis direction. At this time, the interval Δz between the layers is 3 mm.

  The incident wave incident on the frequency selection device having the above structure is a plane wave traveling in the negative direction of the z-axis, has an electric field component only in the y-axis direction, and has an electric field strength E of 1 V / m.

  The equivalent dielectric constant is calculated by first obtaining the reflected wave of the frequency selection device from the radiation of the current induced in each metal conductor when a plane wave is incident on the model shown in FIG. 2, and calculating the equivalent dielectric constant from these. I used the technique to do.

  First, with respect to the current induced in the metal conductor, as shown in FIG. 2, the length l of the metal conductor is divided into m elements (m is a natural number of 2 or more), and the x-axis direction in each element Two surface currents flowing in the y-axis direction are derived by the moment method. As for the calculation of the induced current this time, Kashiyama and Hayashi, “Numerical Solution of Electromagnetic Reflection / Transmission by Regular Arrangement of Finite-Length Metal Cylinder”, IEICE Transactions B-II, vol. J81-B-II, no. 10, pp. 983-989, October 1998, is applied. In this solution, a single-layer model with metal conductors arranged on a plane is used. In this calculation, this model is further expanded to a model consisting of a plurality of layers (for example, five layers). The only difference is that it was executed. Therefore, a detailed description of the calculation method is omitted.

  3 to 7 show the frequency characteristics of the induced current calculated by the above method for each layer of the frequency selection device. The induced currents shown in FIGS. 3 to 7 correspond to currents flowing through the centers of the metal conductors arranged in the first to fifth layers, respectively.

  As is clear from FIGS. 3 to 7, although the absolute values of the induced currents of the respective layers are different from each other, a maximum result is obtained in the vicinity of 9 GHz in common to all the layers. Here, the length l = 15 mm of the metal conductor shown in FIG. 2 corresponds to a length of about one half of the wavelength of the electromagnetic wave at a frequency of 9 GHz. Therefore, it can be determined that a large current is induced in each metal conductor by resonance.

  FIG. 8 is a change along the z-axis of the electric field distribution at 5 GHz obtained from the induced current. Here, the solid line indicates the scattered wave electric field from the metal conductor, and the broken line indicates the total electric field obtained by combining the scattered wave electric field and the incident wave.

  Referring to FIG. 8, in the region of z> 0 (incident side), the solid line represents only the reflected wave, and the broken line represents the standing wave generated by the combination of the reflected wave and the incident wave. It can be seen that the reflected wave level is small at 5 GHz.

  On the other hand, in the region of z <0 (transmission side), the broken line is the transmitted wave. At 5 GHz, the electric field strength of the transmitted wave is 1.0 V / m, and it can be seen that it is almost transmitted.

  FIG. 9 shows changes along the z-axis of the electric field distribution at 9.5 GHz. Similarly to FIG. 8, the solid line indicates the scattered wave electric field, and the broken line indicates the total electric field.

  Referring to FIG. 9, in the region of z> 0, the electric field intensity of the reflected wave is as high as 1 V / m, and it can be considered that the total reflection is performed. On the other hand, in the region of z <0, the transmitted wave indicated by the broken line is almost 0 V / m, and the incident wave is not transmitted.

  Note that the periodic fluctuation depending on the location of the transmitted wave and reflected wave seen in FIGS. 8 and 9 is a phenomenon caused by the calculation method of the electric field. The reflection coefficient shown below is calculated by removing this influence. Was done.

  Next, an equivalent relative dielectric constant of the frequency selection device is obtained from the obtained electric field distribution. In obtaining the equivalent dielectric constant, the complex reflection coefficient Γ is calculated using the transmission line model when the frequency selection device is assumed to be a parallel plate of a dielectric placed in space, and the obtained complex reflection coefficient Γ is calculated. Based on this, the equivalent dielectric constant ε is calculated.

  In the transmission line model shown in FIG. 10, the complex reflection coefficient Γ is represented by the ratio of the complex amplitude between the incident wave and the reflected wave. Of the scattered wave electric field shown in FIGS. 8 and 9, the electric field present on the incident side is a reflected wave by this structure, and therefore the amplitude and phase of the reflected wave are required. Since the incident wave is known in the previous electromagnetic field analysis, the complex reflection coefficient Γ can be obtained from these.

  FIG. 11 is a frequency characteristic of the absolute value of the complex reflection coefficient Γ in the frequency selection device of FIG. 2 derived from the above calculation.

  Referring to FIG. 11, the absolute value of complex reflection coefficient Γ is 1 or less in a frequency band of 9 GHz or less with 9 GHz as a boundary. On the other hand, almost 1 is maintained in the frequency band of 9 GHz or higher. Note that the value of 1 or more is due to a calculation error. Further, when focusing on the frequency band of 9 GHz or less, the absolute value of the complex reflection coefficient Γ is discretely reduced to near 0 in the vicinity of 4.5 GHz, 7.5 GHz, and 8.5 GHz. On the other hand, in the frequency band of 9 GHz or more, the absolute value of the complex reflection coefficient Γ is fixed to 1, indicating total reflection.

  Further, the equivalent relative dielectric constant of the frequency selection device shown in FIG. 2 is obtained using the frequency characteristic of the obtained complex reflection coefficient Γ.

Referring to the transmission line model of FIG. 10 again, the normalized input impedance Z _in viewed from the surface of the dielectric plate is expressed by the relative permittivity of the dielectric plate ε _r , the thickness d, and the free space of the incident wave If the wavelength at λ is λ, Equation (1) is obtained.

On the other hand, the following relational expression is established between the complex reflection coefficient Γ and the relative dielectric constant ε _r of the dielectric plate using the normalized input impedance Z _in .

Therefore, if the normalized input impedance Z _in is obtained from the complex reflection coefficient Γ, the relative dielectric constant ε _r can be numerically derived from the equation (1) by the Newton method.

  FIG. 12 shows the frequency characteristic of the equivalent dielectric constant ε (real part) of the frequency selection device of FIG. 2 obtained by the above calculation.

  Referring to FIG. 12, equivalent dielectric constant ε maintains a constant value of 1 or more in a low frequency band. From such behavior of the equivalent dielectric constant ε, the frequency selection device according to the present invention can be regarded as a conventional artificial dielectric in a low frequency band. That is, the frequency selection device can be considered as a thin dielectric parallel plate. Here, considering the reflection of the parallel plate, the combination of the wave reflected at the first interface and the wave that emerges through the first interface once it enters the plate and undergoes multiple reflection inside the plate. Becomes the reflected wave of this plate. If the thickness of the parallel plate is sufficiently thin relative to the wavelength, see Morita et al., “Microwave Antenna”, Ohm, pp. As described in 127-128, the two reflected waves are out of phase with each other and cancel each other, thereby reducing reflection. Therefore, in this frequency selection device, the complex reflection coefficient Γ in the low frequency band is kept low.

  Furthermore, when it reaches 9 GHz from a low frequency band, the equivalent relative dielectric constant ε increases rapidly to the positive side. This is because the induced current increased rapidly due to the resonance of the individual metal conductors shown in FIG.

  Further, the equivalent relative dielectric constant ε is discontinuous around 10 GHz and increases from negative infinity in a frequency band of 10 GHz or more. The frequency at which the equivalent relative dielectric constant ε becomes discontinuous is set as the resonance frequency of the frequency selection device.

Such a phenomenon when the equivalent relative permittivity ε of the frequency selection device becomes negative can be compared to the ionosphere located in the upper atmosphere of the earth. In the ionosphere, since the relative dielectric constant ε is always smaller than 1, the refractive index is lower than the refractive index 1 of the atmosphere. Therefore, in wave propagation between the atmosphere and the ionosphere, total reflection occurs at oblique incidence. become. Furthermore, when the plasma frequency is a critical frequency and the frequency is equal to or lower than the critical frequency, the relative dielectric constant ε becomes negative. As a result, the wave number k = ω (ε ₀ εμ ₀ ) ^{1/2 of} the propagating plane wave becomes a pure imaginary number, and even if it is normal incidence, it cannot propagate at all, and the electromagnetic wave is totally reflected.

  When this phenomenon is applied to the present frequency selection device, it can be determined that the incident electromagnetic wave is totally reflected without being propagated in a frequency range of 10 GHz or more where the equivalent relative dielectric constant ε is negative.

  The reason why the equivalent relative dielectric constant ε is negative in a frequency band higher than the resonance frequency can be explained as follows. In a frequency band higher than the resonance frequency, the phase of the current flowing through the metal conductor is delayed, and the dipole moment generated by the movement of the current, that is, the charge, is in phase opposite to the incident electric field. For this reason, when viewed macroscopically, the polarization of the dielectric is directed in the opposite direction to the incident electric field, and the equivalent relative dielectric constant ε takes a negative value.

  Further, in the frequency band from 9 GHz to 10 GHz, the equivalent relative dielectric constant ε becomes very large, so that reflection at the boundary surface increases and almost total reflection occurs.

  Next, paying attention to the low frequency band below the resonance frequency, the frequency selection device shows the same behavior as the conventional artificial dielectric as described above, and is apparent from the absolute value of the complex reflection coefficient Γ in FIG. As described above, at specific frequencies such as 4.5 GHz, 7.5 GHz, and 8.5 GHz, the reflection coefficient is discretely reduced to near zero.

  Here, the characteristic of the reflection coefficient which is discretely near 0 at such a specific frequency will be considered.

  According to the above-mentioned document (Morita et al., “Microwave Antenna”), the reflection coefficient Γ of the parallel plate of dielectric is

It becomes. Here, γ is a reflection coefficient when a plane wave is perpendicularly incident on an interface with a substance having a relative dielectric constant ε _r , and d is a thickness of the parallel plate.

In equation (3), the reflection coefficient Γ is 0 when sin φ = 0, and φ at this time is 0, π, 2π,. Further, according to the equation (4), the thickness d of the parallel plate when φ = 0, π, 2π... Is ₀ of the wavelength λ (= λ ₀ / ε _r ^1/2 ) in the parallel plate. .5, 1.0, 1.5... That is, when the thickness d of the parallel plate is an integral multiple of the half wavelength λ / 2, the reflection coefficient Γ of the dielectric plate is 0, and all incident waves are transmitted. The wavelength λ corresponds to the wavelength in the wavelength shortening effect of a dielectric in the dielectric constant epsilon _r.

  When the frequency selection device according to the present invention is viewed as a macroscopically homogeneous dielectric plate, the thickness d of the dielectric plate has a frequency corresponding to an integral multiple of the half wavelength λ / 2, as in an actual dielectric plate. Electromagnetic waves are expected to be transmitted with low loss and low loss.

Here, in the conventional artificial dielectric, the relative dielectric constant ε _r is expressed by the following equation, and it can be seen that the change is proportional to the number of metal conductors in the unit volume.

Here, ε _m is a relative dielectric constant of a material surrounding the metal conductor, n is a density of the number of metal conductors, and α is a coefficient. If the frequency selection device has a certain number of layers, the density of the number of metal conductors is inversely proportional to the thickness d of the dielectric plate, so that the relative dielectric constant ε _r is converted into Equation (6).

On the other hand, the relative permittivity epsilon _r due to the wavelength shortening effect is inversely proportional to the square root of the dielectric constant epsilon _r, the electromagnetic wave velocity v of the internal frequency selection device,

  And the speed v is a function of the thickness d of the dielectric plate. Here, c represents the speed of light in vacuum. On the other hand, as described above, when the thickness d is n / 2 times the wavelength, an increase in the transmitted wave occurs. Therefore, the following relationship is established between the transmission frequency and the thickness d of the dielectric plate.

  As a result, it can be seen that the frequency at which the transmission loss of the frequency selection device is small varies depending on the thickness d of this structure. In other words, the transmission frequency can be adjusted by changing the thickness d of this structure. This is simpler than the conventional frequency selective surface in which the transmission frequency is adjusted by changing the shape and arrangement interval of individual metal conductors arranged in each layer and the interval between layers. Obviously, it is highly accurate.

  In this calculation by electromagnetic field analysis, the electrical conductivity of the metal conductor is infinite, but since it is actually finite, heat is generated due to the electrical resistance due to this electrical conductivity, and the electromagnetic field energy is Converted into energy. This corresponds to a loss when viewed from an electromagnetic wave. In the conventional frequency selective surface, since the resonance frequency of the metal conductor is matched with the transmission frequency, a large current is induced in the metal conductor at the time of resonance as shown in FIGS. Loss will increase. On the other hand, in this frequency selection device, since the resonance frequency is not selected as the transmission frequency, the current induced in the metal conductor is small, and the loss during transmission can be suppressed.

  In the frequency selection device of this time, the thickness d of the structure is transmitted in a very low frequency band, specifically, in a frequency corresponding to one fifth or less of the wavelength in the wavelength shortening effect by the equivalent dielectric constant. Characteristics were obtained. This is because the electromagnetic wave is almost transmitted because the thickness d of this structure is thinner than the wavelength of the electromagnetic wave.

  FIG. 13 is a configuration diagram showing an example of a frequency selection device that realizes the above basic principle.

  Referring to FIG. 13, the frequency selection device has a multilayer structure formed by stacking a plurality of layers. Each layer includes a printed circuit board 20, metal conductors 10 arranged on the printed circuit board, and spacers. The foam material 30 is provided.

  The metal conductor 10 is formed on the printed board 20 by etching or the like. In addition to the strip shape shown in FIG. 13, the shape of the metal conductor 10 includes a rectangular or circular ring shape, a cross shape, and the like. The spacer includes a honeycomb material in addition to the foam material 30. An adhesive is used for joining the one layer and the other layer.

  In the frequency selection device having the multilayer structure described above, the resonance frequency is determined by the shape and arrangement interval of the metal conductors 10 of each layer as described above. In the cut-off frequency band including this resonance frequency, there is a selective cut-off property that electromagnetic waves are totally reflected. On the other hand, in the pass frequency band equal to or lower than the cut-off frequency band, reflection can be reduced with respect to a frequency corresponding to an integral multiple of a half wavelength of the entire structure. The thickness of the entire structure can be adjusted by the thickness of the foam material 30 serving as a spacer, whereby a desired frequency can be transmitted with low loss.

  As described above, according to the first embodiment of the present invention, the conventional artificial dielectric technology is applied to the frequency selection device, and the cutoff frequency including the resonance frequency is expanded to a frequency range equal to or higher than the resonance frequency. In addition to selectively blocking electromagnetic waves in a band, it is possible to easily and accurately realize frequency selectivity of transmitting with low loss in a frequency band lower than the cutoff frequency band.

  Furthermore, since the current induced in the metal conductor at the transmission frequency is small, the loss can be suppressed as compared with the conventional frequency selective surface.

[Embodiment 2]
In the first embodiment, a configuration for adjusting the transmission frequency according to the thickness of the frequency selection device has been proposed. The transmission frequency corresponds to a frequency at which the thickness of the frequency selection device is an integral multiple of a half wavelength (= n · λ / 2).

  Here, depending on the application in which the frequency selection device is used, it may be required to block a certain frequency in these pass frequency bands.

  Therefore, in this embodiment, a frequency selection device having a frequency selectivity of blocking a specific frequency in the pass frequency band is proposed. Hereinafter, as a specific example, a case where a design that cuts off only about 7.5 GHz out of the transmission frequencies shown in FIG. 11 will be described.

  FIG. 14 is a schematic configuration diagram of a frequency selection device according to the second embodiment of the present invention.

  Referring to FIG. 14, the frequency selection device includes a frequency selection device 40 having a multilayer structure, and a conventional frequency selective surface 60 bonded with a spacer 50 interposed therebetween.

  The frequency selection device 40 is the same as that described in the first embodiment. As shown in FIG. 13, a plurality of metal conductors 10 and dielectric plates 30 arranged on the printed circuit board 20 are used as one layer. The layers are joined to form a dielectric plate of thickness d.

  The conventional frequency selective surface 60 has a single layer structure, and square annular metal conductors are arranged on the printed circuit board 60 at predetermined intervals.

  The spacer 50 includes a foam material, a honeycomb material, or the like, and an air layer, and has a thickness t.

  FIG. 15 is an equivalent circuit diagram of the frequency selection device shown in FIG.

Referring to FIG. 15, the frequency selection device 40 has a relative dielectric constant ε = 7.26 and a thickness d = 15 mm. A conventional frequency selective surface 60 is coupled across the frequency selector 40 and a spacer 50 having a thickness l = 5 mm. The conventional frequency selective surface 60 is designed to have a resonance frequency of 7.5 GHz. The design includes an equivalent circuit model (LC series) described in the literature (RJLangley and EAParker, “Equivalent circuit model for arrays of square Loop,” Electorn. Lett. Vol.18, No.7 pp.294-296, 1982). Circuit). In the following calculation, all parameters were normalized so that the characteristic impedance was 1Ω. According to this document, the shape and arrangement interval of metal conductors on a conventional frequency selective surface (p = 14.178 mm, s = 0.906 mm, d = 12.4714 mm, g = 2.0066 mm) are equivalent. As the circuit constants of the circuit model, L = 0.01065 nH and C = 0.04261 nF are calculated. From these circuit constants, the admittance of the equivalent circuit model is Y = −4.9802 × 10 ⁸ jS (j: imaginary unit). Thus, the longitudinal continuation line of the conventional frequency selective surface 60 only is

It becomes. In addition, the longitudinal continuation row of the spacer 50 of length l is

It is represented by Furthermore, since the frequency selection device 40 can be regarded as a dielectric plate having a relative dielectric constant ε (= 7.26) and a thickness d (= 15 mm),

It becomes. Substituting a numerical value for each parameter in the vertical continuation column,

Is obtained. The transmission coefficient τ is

Therefore, the absolute value of the transmission coefficient τ is 4.352 × 10 ⁻⁹ using Equation (12). From the above calculation results, it was confirmed that the frequency selection device shown in FIG. 15 has frequency selectivity that is cut off at a frequency of 7.5 GHz.

  As described above, according to the second embodiment of the present invention, the conventional frequency selective surface is joined to the frequency selective device with a predetermined interval, and the resonance frequency of the frequency selective surface and the interval are adjusted. By doing so, it is possible to easily realize the frequency selectivity of blocking a part of the transmission frequency of the frequency selection device.

[Embodiment 3]
In the present embodiment, as a second design example for improving the frequency selection performance, a design that adds a characteristic that reflection at 6 GHz is reduced to the frequency selectivity of the frequency selection device of the first embodiment shown in FIG. Do.

  FIG. 16 is a schematic structural diagram of a frequency selection device according to the third embodiment of the present invention.

  Referring to FIG. 16, the frequency selection device includes two frequency selection devices 40 facing each other at a predetermined interval.

  Each of the two frequency selection devices 40 is the same as that shown in FIG. 13 of the first embodiment. That is, the printed circuit board 20 on which the metal conductors 10 are arranged and the dielectric plate 30 on the board are used as one layer, and a plurality of layers are stacked. In the following, the thicknesses of the frequency selection device 40 are d1 and d2, respectively. In addition, the reflection coefficient in 6 GHz with this frequency selection apparatus alone is 0.634.

  Between the two frequency selection devices, a foam material or honeycomb material having a thickness s is disposed as the spacer 50. The relative dielectric constant of the spacer 50 is ε = 1.0.

  FIG. 17 is an equivalent circuit diagram of the frequency selection device shown in FIG. The interval s between the two frequency selection devices 40 was adjusted so that the reflection at 6 GHz was reduced. As a result, the reflection coefficient was 0.0883 (= −21 dB) at the interval s = 21 mm, indicating that the reflection was sufficiently small. Detailed calculation is shown below.

  The vertical continuation column of the frequency selector is

The vertical continuation line of the space with a spacing of 21 mm is given by

Given in. The continuation sequence representing the frequency selection device shown in FIG. 16 can be obtained from the product of these continuation sequences,

It becomes. The reflection coefficient Γ is

Therefore, it can be seen that the reflection coefficient Γ in this structure has an absolute value of 0.083 at a frequency of 6 GHz, and the reflection becomes small.

[Modification 1]
In the above-described embodiment, the reflection of a specific frequency is reduced by arranging two frequency selection devices at a predetermined interval. This specific frequency is adjusted by changing the interval between the two frequency selection devices.

  Here, it is expected that such frequency selectivity can be obtained not only by using two identical frequency selection devices but also by using any one of them as a normal dielectric plate.

  Therefore, as a first modification of the frequency selection device shown in FIG. 16, one of the configurations is a normal dielectric plate, and a design that realizes the same frequency selectivity (reducing 6 GHz reflection) as described above is performed. It was. For the design, a glass fiber reinforced epoxy resin plate (relative dielectric constant ε = 4.5) used for a printed circuit board was adopted as a normal dielectric plate. From the calculation results, when an ordinary dielectric plate having a thickness of 18 mm is disposed facing the frequency selection device with a spacing of 12 mm, the reflection coefficient at 6 GHz is 0.056 (= −25 dB), and the reflection is small. It was confirmed that

  Specifically, the longitudinal continuation row of the frequency selection device is the same as described above,

Given in. In addition, the vertical continuation line of the space of 12 mm spacing is

Given in. Furthermore, the vertical continuation row of normal dielectric plates is

It becomes. Using these, the longitudinal continuation line of the entire frequency selection device is

It becomes. The reflection coefficient Γ is

Therefore, when the equation (20) is substituted, a reflection coefficient Γ = 0.056 at 6 GHz is obtained.

[Modification 2]
Further, as a second modification of the frequency selection device shown in FIG. 16, one of the configurations is a conventional frequency selective surface, and a design that realizes the same frequency selectivity (reducing 6 GHz reflection) as described above. The result of having performed is shown. Note that the equivalent circuit model used in the second embodiment was used for calculating the admittance for the conventional frequency selective surface.

First, the input impedance Z _in 0 of one of the frequency selection devices facing each other is Z _in 0 = 0.243975-0.182336 jΩ. The input impedance Zin and the input admittance Yin viewed from a position 5.1 mm away are Zin = 0.28699 + 0.45064 jΩ and Yin = 1.0054-1.5787 jS, respectively.

  The conventional frequency-selective surface is the same as the frequency-selective surface 60 shown in FIG. 14, and the admittance Y of the equivalent circuit model at 6 GHz is Y = 1.60909 jS (where p = 23.144 mm, s = 0.9906 mm, d = 21.074 mm, g = 2.0066 mm).

  For this reason, the input admittance Y seen from the conventional frequency selective surface is Y = 1.0054 + 0.0222 jS. The absolute value of the reflection coefficient at this time is 0.0114 (= −38.9 dB), and it can be said that the reflection is sufficiently offset.

  As described above, according to the third embodiment of the present invention, a frequency selection device is arranged opposite to each other with a predetermined interval on one frequency selection device, and the desired frequency is adjusted by adjusting the interval. Thus, the frequency selection performance can be easily improved.

  Note that the same effect can be obtained not only by the frequency selection device but also by using a normal dielectric plate or a conventional frequency selective surface.

[Embodiment 4]
In the previous embodiment, the frequency selective device showed that the equivalent relative permittivity was negative at a frequency higher than the resonance frequency, and had frequency selectivity that totally reflected electromagnetic waves.

  However, it has been experimentally confirmed that as the frequency becomes higher than the resonance frequency, the influence of resonance becomes smaller, so that the equivalent relative dielectric constant becomes positive again and begins to transmit electromagnetic waves.

  Here, depending on the application of the frequency selection device, it may be required to increase the frequency at which transmission begins, that is, to widen the band of the cutoff frequency.

  Therefore, in this embodiment, a study for expanding the cut-off frequency band is conducted experimentally, and a frequency selection device having new frequency selectivity is proposed.

  FIG. 18 is an overall configuration diagram schematically showing a frequency selection device used in the present embodiment. FIG. 18A is a diagram showing the shape of the metal conductor 10, and FIG. 18B is an overall configuration diagram.

  Referring to FIG. 18B, the frequency selection device 40 includes a plurality of metal conductors 10 arranged with a distance c on one printed circuit board (glass epoxy board) 20 as a total of four layers. However, the dielectric plate (foamed polypropylene plate) 30 is stacked between layers. The printed board 20 has a thickness of 1.6 mm, and the dielectric plate 30 has a thickness of 3 mm.

  As shown in FIG. 18A, each metal conductor 10 has a square ring shape with a side length a = 4.741 mm and a width b = 0.289 mm. The arrangement interval c of the metal conductors 10 is set to 5.927 mm.

  FIG. 19 shows the result of actually measuring the transmission coefficient for the frequency selection device 40 having the above configuration.

  Referring to FIG. 19, it can be seen that the transmission coefficient sharply decreases from around 9 GHz and is affected by the resonance frequency. Furthermore, although the transmission frequency gradually decreased in the frequency range of 9 GHz or higher, it rapidly increased again in the vicinity of 16 GHz and approached the characteristics in the low frequency band of 9 GHz or lower. That is, in this frequency selection device, it can be said that the cut-off frequency band is a frequency range from 9 GHz to 16 GHz.

  Here, it is considered that the cut-off frequency band is further expanded to the high frequency side in the frequency characteristics of FIG. For this purpose, a study was made to join a conventional frequency selective surface to the frequency selective device of FIG.

  FIG. 20 is a schematic overall configuration diagram of the frequency selection device according to the present embodiment.

  Referring to FIG. 20, a conventional frequency selective surface 60 is bonded to the upper portion of the frequency selection device 40 via a spacer 50. As the spacer 50, a foamed polypropylene plate having a thickness of 3 mm is used. Further, the conventional frequency selective surface 60 is not shown, but a square annular metal conductor (a = 3.533 mm, b = 0.275 mm) similar to FIG. They are arranged with an arrangement interval c = 4.426 mm. The single transmission characteristic of this conventional frequency selective surface has a frequency at which the transmission coefficient is minimized at 16.64 GHz.

  FIG. 21 shows the results of actually measuring the transmission coefficient for the frequency selection device shown in FIG.

  As can be seen from FIG. 21, in this structure, the cut-off frequency band extends from 9 GHz to around 18 GHz, and the rising of the transmission coefficient at 16 GHz seen in FIG.

  In this embodiment, the structure for expanding the cutoff frequency band to the high frequency side has been described. However, a conventional frequency selection in which a frequency lower than the cutoff frequency band of the frequency selection device 40 is set as the center frequency of the blocking characteristic is described. It is also possible to extend the cutoff frequency band to the low frequency side by bonding the conductive surface 60. Further, instead of a conventional frequency selective surface, a frequency selection device having a different cutoff frequency may be used. The cut-off frequency of the frequency-selective surface to be joined or the frequency selection device is 1.1 times or less of the upper limit frequency of the cut-off frequency band of the frequency selection device 40 or 0.9 times or more of the lower limit frequency. It is desirable to have a blocking characteristic.

  As described above, according to the fourth embodiment, the conventional frequency selective surface or the frequency selective device is joined to the frequency selective device so that these cutoff frequency bands are close to the cutoff frequency band of the frequency selective structure. By adjusting to, the cut-off frequency band of the frequency selection device can be expanded.

  The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

It is a figure which shows the structure of the existing artificial dielectric material roughly. This is a model in which finite-length metal conductors used for electromagnetic field analysis of the frequency selection device are arranged. It is a frequency characteristic of the electric current induced | guided | derived to the metal conductor arranged in the 1st layer. It is a frequency characteristic of the electric current induced | guided | derived to the metal conductor arranged in a 2nd layer. It is a frequency characteristic of the electric current induced | guided | derived to the metal conductor arranged in the 3rd layer. It is a frequency characteristic of the electric current induced | guided | derived to the metal conductor arranged in the 4th layer. It is a frequency characteristic of the electric current induced | guided | derived to the metal conductor arranged in the 5th layer. It is a change along the z-axis of the electric field distribution at 5 GHz obtained from the induced current. It is a change along the z-axis of the electric field distribution at 9.5 GHz. 3 is a transmission line model of the frequency selection device of FIG. It is the frequency characteristic of the absolute value of the complex reflection coefficient (GAMMA) in the frequency selection apparatus of FIG. 2 derived | led-out from calculation. 3 is a frequency characteristic of an equivalent dielectric constant ε (real part) of the frequency selection device of FIG. 2 obtained by calculation. It is a whole block diagram which shows an example of the frequency selection apparatus which implement | achieves the basic principle of this Embodiment. It is a typical whole block diagram of the frequency selection apparatus according to Embodiment 2 of this invention. FIG. 15 is an equivalent circuit diagram of the frequency selection device shown in FIG. 14. It is a typical whole structure figure of the frequency selection apparatus according to Embodiment 3 of this invention. FIG. 17 is an equivalent circuit diagram of the frequency selection device shown in FIG. 16. It is a whole block diagram which shows typically the frequency selection apparatus used for this Embodiment. It is the result of having measured the transmission coefficient about the frequency selection apparatus 40 shown in FIG. It is a typical whole block diagram of the frequency selection apparatus according to this Embodiment. It is the result of having measured the transmission coefficient about the frequency selection apparatus shown in FIG. It is a block diagram for demonstrating the principle of a frequency selective surface. It is an equivalent circuit diagram of the frequency selective surface which consists of a multilayer structure.

Explanation of symbols

  10,100 metal conductor, 20 printed circuit board, 30,110 dielectric plate, 40 frequency selection device, 50 spacer, 60 frequency selective surface.

Claims (9)

A layer in which a plurality of metal conductors are regularly arranged on a plane is defined as one layer, and the one layer includes a plurality of layers stacked in parallel at a constant interval,
In the cut-off frequency band including the resonance frequency of the metal conductor, the electromagnetic wave incident on the plurality of layers is cut off, and in the pass frequency band equal to or lower than the cut-off frequency band, the electromagnetic wave incident on the plurality of layers is transmitted. Frequency selection device.   The equivalent relative permittivity when the plurality of layers are viewed as a macroscopically homogeneous dielectric exhibits a value of 1 or more in the frequency band below the resonance frequency, and is negative in the cutoff frequency band above the resonance frequency. The frequency selection device according to claim 1, which indicates a value.   3. The frequency selection according to claim 2, wherein a thickness of the plurality of layers transmits an electromagnetic wave including a frequency that is an integral multiple of a half of a wavelength of the electromagnetic wave propagating through the dielectric having the equivalent relative dielectric constant. apparatus.   4. The frequency according to claim 2, wherein a thickness of the plurality of layers transmits an electromagnetic wave including a frequency that is not more than one fifth of the wavelength of the electromagnetic wave propagating through the dielectric having the equivalent relative dielectric constant. Selection device. The plurality of layers are:
A plurality of first and second dielectric parallel plates that are joined either alone or together before and after the plurality of metal conductors;
5. The frequency selection device according to claim 2, wherein a thickness of the plurality of layers is a sum of thicknesses of the plurality of first and second dielectric parallel plates. A frequency selective surface bonded to the plurality of layers at a predetermined interval;
The frequency selection device according to claim 2, wherein an electromagnetic wave having a part of the frequency is cut off in the pass frequency band. A dielectric parallel plate or a frequency selective surface joined to the plurality of layers at a predetermined interval;
The frequency selection device according to claim 2, wherein reflection of electromagnetic waves having a specific frequency is canceled in the pass frequency band.   The frequency selecting device according to claim 7, wherein the parallel plate of the dielectric is one of a dielectric plate and the second plurality of layers. A frequency selective surface joined to the plurality of layers at a predetermined interval or the second plurality of layers;
The frequency selective surface or the second plurality of layers has a characteristic of blocking electromagnetic waves having a frequency of 1.1 times or less of an upper limit frequency of the cut-off frequency band or 0.9 times or more of a lower limit frequency. The frequency selection device according to 2.

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