JP2005124175A - Amplifier and frequency converter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a negative feedback amplifier with a wide dynamic range suitable for integration into a semiconductor integrated circuit. <P>SOLUTION: An amplifier circuit 10 amplifies a signal inputted from an input terminal P1. A first feedback circuit 20 is placed across an emitter of a bipolar transistor 101 and an input of the amplifier circuit 10. A second feedback circuit 30 is placed across the input and an output of the amplifier circuit 10 for feeding the output of the amplifier circuit 10 back to the input. A phase change amount in the first feedback circuit 20 is determined by the values of an inductor 201 and a capacitor 202. A phase change amount in the second feedback circuit 30 is determined by the values of a resistor 301 and a capacitor 302. The values of these elements are selected so that the phase of a signal in which fundamental waves included in two feedback signals are combined and the phase of a signal in which second harmonics included in the two feedback signals are combined are shifted by approximately 180 degrees from the phase of a fundamental wave of an input signal. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to an amplifying device for amplifying an input signal and a frequency converting device for amplifying an input signal and converting the frequency thereof, and more specifically, has a wide dynamic range and is integrated in a semiconductor integrated circuit. The present invention relates to a suitable amplification device and frequency conversion device.

  In a receiver of a wireless system typified by a cellular phone, a signal received by an antenna is amplified by a first stage amplifier circuit. The first-stage amplifier circuit is required to have low noise and high gain characteristics when receiving weak signals, and to have low distortion and low gain characteristics when receiving large signals. In particular, in recent mobile communication, the characteristics of the received electric field greatly change according to the distance between the base station and the mobile station. For this reason, the receiving system is required to have a wider dynamic range than before.

  As a method for stably operating an amplifier circuit, a method of inserting a resistor between a signal line and ground (ground) at the input or output of the amplifier circuit is widely used. However, when the resistor is inserted on the input side, the noise characteristic is greatly deteriorated, and when the resistor is inserted on the output side, the distortion characteristic is greatly deteriorated. As a method other than the above, a method of applying negative feedback to the input, that is, a method of feeding back a signal whose phase has been converted by 180 degrees to the input is known. According to this method, although both noise characteristics and distortion characteristics are slightly deteriorated, an amplifier circuit having a wide dynamic range as a whole circuit can be realized. Further, since the negative feedback circuit also operates as a distortion compensation circuit, the dynamic range can be further expanded by devising the circuit.

  Hereinafter, a conventional negative feedback amplifier will be described with reference to FIGS. 15 to 20. As a first conventional example, a “negative feedback power amplification device” described in Patent Document 1 is known (see FIG. 15). 15 includes a negative feedback circuit that operates as a distortion compensation circuit, and is used in the microwave band. In FIG. 15, transistors 601 and 602 are all field effect transistors, and inductors 603, 604, and 605 and capacitors 606 and 607 constitute a matching circuit of the transistors 601 and 602. The microstrip line 608 serves as a phase shifter. The power supply voltage Vcc is applied to the amplification device via the microstrip line 608.

  Part of the output signal of the transistor 602 is fed back to the input of the transistor 601 via the inductor 605, the microstrip line 608, and the inductor 604. At this time, the length L of the microstrip line 608 is adjusted so that the phase difference between the feedback signal and the output signal of the transistor 602 is 180 degrees. Thus, by inverting a part of the output signal including the distortion component and feeding it back to the input, the distortion characteristic in the high frequency band can be improved.

  As a second conventional example, a “high output amplifier” described in Patent Document 2 is known (see FIG. 16). The amplifier shown in FIG. 16 includes signal line strip lines 701a and 701b, a signal amplification transistor 702, a directional coupler 703, a feedback strip line 704, a stub 705, feedback amount changing resistors 706a and 706b, and a level detection circuit 707. A harmonic suppression control circuit 708 and a termination resistor 709.

  In FIG. 16, an input is supplied from a signal line strip line 701a and amplified by a signal amplification transistor 702. The output of the signal amplifying transistor 702 is fed back to the input of the signal amplifying transistor 702 via a feedback strip line 704 having a predetermined line length and a directional coupler 703. As a result, a signal having a phase opposite to that of the second harmonic is fed back to the input of the signal amplification transistor 702. In this manner, the linearity of the signal amplification transistor 702 can be improved by canceling the distortion of the second harmonic.

  As a third conventional example, an “amplifier” described in Patent Document 3 is known (see FIG. 17). 17 includes a transistor 801, signal sources 802 and 803, a signal source resistor 804, input matching circuits 805, 806, 807, 808, and 809, output matching circuits 810, 811 and 815, a band-pass filter 812, and a phase shifter. 813, a variable attenuator 814, and a load resistor 816.

  In FIG. 17, a band pass filter 812 passes the second harmonic of the output of the transistor 801. The phase shifter 813 and the variable attenuator 814 adjust the phase and amplitude of the second harmonic, respectively. In this amplifier, the third-order intermodulation product of the amplifier can be reduced by feeding back the second harmonic of the output signal to the input as in the second conventional example.

  As a fourth conventional example, a “power amplifying device” described in Patent Document 4 is known (see FIG. 18). 18 includes a synthesizer 901, a power amplifier 902, a distributor 903, a filter 904, a variable phase shifter 905, and a variable attenuator 906. This amplifying device phase-converts the fundamental wave of the output of the power amplifier 902 and the second harmonic, the third harmonic, the fourth harmonic, and the like in a wide band by 180 degrees and feeds back to the input of the power amplifier 902. Thus, by negatively feeding back both the fundamental wave and the second harmonic of the output of the power amplifier 902 to the input, distortion of the output signal can be compensated.

  As a fifth conventional example, a “broadband feedback amplifier” described in Patent Document 5 is known (see FIG. 19). The amplifier shown in FIG. 19 includes an amplifying element 1001, a signal input 1002, a signal output 1003, a slot line ground plane 1004, a slot line release surface 1005, a strip line 1006, a micro strip line 1007 of a slot line conversion unit, a slot line 1008, a via. A hole 1009, a through hole 1010, and a feedback amount determining resistor 1011 are provided. As in the fourth conventional example, this amplifier performs 180-degree phase conversion of the output of the amplifying element 1001 over a wide band and feeds it back to the input of the amplifying element 1001. Thus, by negatively feeding back both the fundamental wave and the second harmonic of the output of the amplifying element 1001 to the input, distortion of the output signal can be compensated. Patent Document 5 discloses a specific example of a feedback circuit that performs 180-degree phase conversion over a wide band.

As a sixth conventional example, “broadband amplification with high linearity and low power consumption” described in Patent Document 6 is known (see FIG. 20). The amplifier shown in FIG. 20 includes an input transistor 1101, an output transistor 1102, a series reactive feedback network 1103, and a shunt reactive feedback network 1104. Input transistor 1101 and output transistor 1102 are coupled in a cascode configuration, with input transistor 1101 determining the input of the amplifier and output transistor 1102 determining the output of the amplifier. Between the input and output, a shunt reactive feedback network 1104 is provided having a substantially zero resistance and non-zero reactance impedance. With this circuit configuration, distortion characteristics can be improved without deteriorating noise characteristics.
Japanese Patent Laid-Open No. 10-22751 JP-A-6-216670 Japanese National Patent Publication No. 11-500027 JP-A-7-94954 JP-A-10-335554 JP 2002-536859 A

  However, each of the first to sixth conventional examples has only one type of shunt feedback path, and the feedback circuit is complicated and large-scaled because the phase of the feedback signal with respect to the input signal is approximately 180 degrees. There is a problem of becoming. Further, the amplification device of the first conventional example has a problem that the second harmonic is phase-converted by nearly 360 degrees and fed back to the input, so that distortion compensation by negative feedback of the second harmonic is not performed. Further, the amplifiers of the second and third conventional examples have a problem that distortion compensation is not performed by negative feedback of a third-order intermodulation wave generated at a frequency near the fundamental wave because the fundamental wave is hardly fed back. . Further, the amplification devices of the fourth and fifth conventional examples have a problem that the feedback circuit is complicated and large-scale because the phase adjustment of both the fundamental wave and the harmonic is performed only by the feedback circuit.

  Therefore, the present invention has a simple configuration, can negatively feedback the phase of the fundamental wave of the input signal and the phase of the third-order intermodulation wave to the input, and is suitable for integration in a semiconductor integrated circuit and a negative feedback frequency. An object is to provide a conversion device. Another object of the present invention is to provide a negative feedback amplification device and a negative feedback frequency conversion device that can negatively feed back the phase of the second harmonic to the input in addition to the fundamental wave and the third order intermodulation wave of the input signal. To do.

  An amplifying device of the present invention includes an amplifying circuit that amplifies an input signal, a first feedback circuit that feeds back a feedback output of the amplifying circuit to an input of the amplifying circuit while changing the phase, and an amplifying circuit that changes the phase. And a second feedback circuit that feeds back the output of the output to the input of the amplifier circuit. Since this increases the degree of freedom of the circuit, the phase of the signal obtained by synthesizing the fundamental wave included in the two feedback signals with a simple circuit configuration is configured to be approximately 180 degrees away from the phase of the fundamental wave of the input signal. Can do. Furthermore, the phase of the signal obtained by synthesizing the second harmonic included in the two feedback signals can be configured to be approximately 180 degrees away from the phase of the fundamental wave of the input signal.

  The amplifying device of the present invention includes an amplifying device to which a signal of one system is input and an amplifying device to which a differential signal composed of an in-phase signal and a reverse-phase signal is input.

  In an amplifying apparatus to which one system signal is input, the amplifier circuit, the first feedback circuit, and the second feedback circuit all input and output one system signal. In this case, the first feedback circuit includes an inductor connected to the feedback terminal of the amplifier circuit and the ground (or a parallel connection circuit of the inductor and the capacitor), and the feedback terminal and the input terminal. A circuit including a capacitor is used. The amplifier circuit includes a first bipolar transistor having a base connected to the input, an emitter connected to the feedback terminal of the amplifier circuit, a second connected to the collector of the first bipolar transistor, and a collector connected to the output. And a circuit including a bipolar transistor in which a base is connected to an input, an emitter is connected to a feedback terminal of an amplifier circuit, and a collector is connected to an output. In the second feedback circuit, a circuit in which a capacitor is connected in series to a parallel connection circuit of a resistor and a capacitor, a circuit in which a capacitor is connected in parallel to a series connection circuit of a resistor and a capacitor, or two capacitors in series And a circuit in which a resistor or an inductor is disposed between the connection point and the ground.

  In an amplifying apparatus to which a differential signal is input, an amplifier circuit has an in-phase input terminal and an in-phase feedback terminal, and operates based on the in-phase signal, and has an anti-phase input terminal and an anti-phase feedback terminal. A first phase feedback circuit including one or more feedback units, and a second feedback circuit configured to operate based on the common phase signal, and a negative phase signal. And a negative-phase feedback unit that operates based on In this case, the first feedback circuit includes a first feedback section connected to the ground, the common-mode feedback terminal and the common-mode input terminal, and a second connected to the ground, the negative-phase feedback terminal and the negative-phase input terminal. A circuit including the feedback section (circuit A), a first feedback section connected to a certain node, the common-mode feedback terminal, and the common-mode input terminal, and the node, the negative-phase feedback terminal, and the negative-phase input terminal A circuit including the second feedback unit and the third feedback unit connected to the node and the ground (circuit B), the connection point of the common-mode feedback terminal and the negative-phase feedback terminal, and the common-mode input terminal A circuit including a first feedback section to be connected, a second feedback section connected to the connection point and the negative phase input terminal, and a third feedback section connected to the connection point and the ground ( Circuit C) or the like is used. Such a feedback unit is configured by an inductor or a parallel connection circuit of an inductor and a capacitor.

  The in-phase amplifying unit and the anti-phase amplifying unit include a first bipolar transistor having a base connected to the input and an emitter connected to one of the in-phase feedback terminal and the anti-phase feedback terminal, and an emitter connected to the first bipolar amplifier. A circuit including a transistor collector and a second bipolar transistor whose collector is connected to the output, a base connected to the input, an emitter connected to either the common-mode feedback terminal or the negative-phase feedback terminal, and a collector connected to the output A circuit including a bipolar transistor is used. The common-phase feedback unit and the negative-phase feedback unit include a circuit in which a capacitor is connected in series to a parallel connection circuit of a resistor and a capacitor, a circuit in which a capacitor is connected in parallel to a series connection circuit of a resistor and a capacitor, and two capacitors May be included, and a circuit in which a resistor or an inductor is disposed between the connection point and the ground may be included.

  Furthermore, the frequency converter of the present invention can be obtained by adding a frequency converter circuit for converting the frequency of the amplified signal to the amplifier of the present invention. Each component included in the frequency conversion device of the present invention is configured in the same manner as each component included in the amplification device of the present invention. In addition, the wireless receiver of the present invention includes amplifying device of the present invention that amplifies the received signal received by the antenna, a frequency converting device of the present invention that converts the frequency of the output of the amplifying device, and a determination regarding an interference signal for the received signal. An interference signal determination unit to perform, and a control unit that changes current consumption in the amplification device and the frequency conversion device based on a determination result in the interference signal determination unit.

  According to the amplifying device of the present invention, by appropriately designing the first and second feedback circuits, the fundamental wave, the third intermodulation wave and the second harmonic of the input signal are negatively fed back with a simple configuration. be able to. Therefore, an amplifying device with a wide dynamic range can be provided.

  According to the amplifying apparatus to which a signal of one system is input, the basic wave, the third intermodulation wave, and the second harmonic of the input signal are negatively fed back with a simple configuration when a non-differential signal is input. be able to. In this case, if the first feedback circuit is used, the passing phase in the amplifier circuit can be adjusted by suitably selecting the value of the inductor (or the inductor and the capacitor). In particular, if a first feedback circuit having an inductor and a capacitor is used, the degree of freedom of phase control for the fundamental wave and the second harmonic can be increased. If the amplifier circuit is used, an input signal can be amplified using a cascode (or single) amplifier circuit. In particular, if a single format is used, an amplifier circuit with good noise characteristics can be realized. Further, if the second feedback circuit is used, the pass phase in the second feedback circuit can be adjusted by suitably selecting the characteristics of the elements constituting the second feedback circuit.

  According to the amplifying apparatus to which the differential signal is input, the fundamental wave, the third intermodulation wave, and the second harmonic of the input signal can be negatively fed back with a simple configuration when the differential signal is input. it can. In this case, if the first feedback circuit, the in-phase feedback unit, and the negative-phase feedback unit are used, the same effect as that of the amplifying apparatus to which one system of signals is input appears depending on the components used. In particular, if the first feedback circuit configured as the circuit B is used, the fundamental wave, the third intermodulation wave, and the second harmonic of the input signal can be negatively fed back with a high degree of freedom. In addition, if the first feedback circuit configured as the circuit C is used, the basic wave of the input signal, the third-order intermodulation wave, and the like can be obtained with a simple configuration while maintaining good pairing of the differential amplifier circuit. The second harmonic can be negatively fed back.

  According to the frequency conversion device of the present invention, by appropriately designing the first and second feedback circuits, the basic wave, the third intermodulation wave, and the second harmonic of the input signal can be negatively fed back with a simple configuration. can do. Therefore, it is possible to provide a frequency converter having a wide dynamic range. Further, according to the wireless reception device of the present invention, the dynamic range of the amplification device and the frequency conversion device can be increased while suppressing an increase in current consumption to the minimum necessary.

(First embodiment)
FIG. 1 is a circuit diagram of an amplifying apparatus according to the first embodiment of the present invention. The amplifying device shown in FIG. 1 includes an amplifying circuit 10, a first feedback circuit 20, a second feedback circuit 30, DC cut capacitors 501, 503, and a choke inductor 502. In this amplifying device, the phase of the signal obtained by synthesizing the fundamental wave included in the two feedback signals and the two feedback signals included in the two feedback signals by the action of the first feedback circuit 20 and the second feedback circuit 30. The phase of a signal obtained by synthesizing the second-order harmonic is characterized by being approximately 180 degrees away from the phase of the fundamental wave of the input signal, and is mainly used in a high frequency band.

  The amplifier circuit 10 includes bipolar transistors 101 and 102, a bypass capacitor 103, and bias circuits 104 and 105, and amplifies a signal input from the input terminal P1. The first feedback circuit 20 includes an inductor 201 and a capacitor 202, and feeds back the output from the emitter of the bipolar transistor 101 included in the amplifier circuit 10 (the feedback terminal of the amplifier circuit 10) to the input of the amplifier circuit 10. The inductor 201 and the capacitor 202 are used for adjusting the passing phase of the first feedback circuit 20. The second feedback circuit 30 includes a resistor 301, a capacitor 302, and a DC cut capacitor 303, and outputs the output from the collector of the bipolar transistor 102 included in the amplifier circuit 10 (the output terminal of the amplifier circuit 10). Return to the input. The resistor 301 and the capacitor 302 are used for adjusting the passing phase of the second feedback circuit 30.

  In the amplifying device shown in FIG. 1, the input terminal P <b> 1 is connected to the base of the bipolar transistor 101 via the DC cut capacitor 501. The collector of the bipolar transistor 101 is connected to the emitter of the bipolar transistor 102. The collector of the bipolar transistor 102 is connected to the output terminal P2 via the DC cut capacitor 503. The emitter of the bipolar transistor 101 is grounded via the inductor 201. A capacitor 202 is inserted between the base and emitter of the bipolar transistor 101. Base currents are supplied from the bias circuits 104 and 105 to the bases of the bipolar transistors 101 and 102, respectively. The resistor 301 and the capacitor 302 are connected in parallel. The parallel connection circuit and the DC cut capacitor 303 are inserted in series between the base of the bipolar transistor 101 and the collector of the bipolar transistor 102. The base of the bipolar transistor 102 is grounded via the bypass capacitor 103. The power supply voltage Vcc is supplied to the collector of the bipolar transistor 102 via the choke inductor 502.

  The operation of the amplifying device shown in FIG. 1 will be described with reference to FIG. Here, the circuit shown in FIG. 2A (a circuit including the bipolar transistor 101, the inductor 201, and the capacitor 202), and the circuit shown in FIG. 2B (the bipolar transistors 101 and 102, the bypass capacitor 103, the inductor 201, A pass phase characteristic of a circuit including a resistor 301 and a capacitor 302) and a circuit illustrated in FIG. 2C (a circuit in which the capacitor 202 is added to the circuit illustrated in FIG. 2B) is examined.

FIG. 2D is a feather display diagram of the input signal (fundamental wave) and the output signal (fundamental wave and second harmonic) in the circuit shown in FIG. As shown in FIG. 2D, the phase of the fundamental wave S _1a included in the feedback signal output from the terminal P4 advances by Φ _{a with} respect to the fundamental wave S _{0 of the} signal input to the terminal P3. Further, the phase of the second harmonic S _2a included in the feedback signal output from the terminal P4 advances by 2Φ _{a with} respect to the fundamental wave S ₀ of the input signal. The rate at which the passing phase advances is determined by the values of inductor 201 and capacitor 202.

FIG. 2E is a feather display diagram of the input signal (fundamental wave) and the output signal (fundamental wave and second harmonic) in the circuit shown in FIG. As shown in FIG. 2 (e), the fundamental wave S _1b of the phase included in the feedback signal output from the terminal P5 is delayed by [Phi _b to the fundamental wave S ₀ of the signal inputted to the terminal P3. Further, the phase of the second harmonic S _2b included in the feedback signal output from the terminal P5 is delayed by 2Φ _{b with} respect to the fundamental wave S ₀ of the input signal. The rate at which the passing phase is delayed is determined by the values of the resistor 301 and the capacitor 302.

FIG. 2F is a feather display diagram of the input signal (fundamental wave) and the output signal (fundamental wave and second harmonic) in the circuit shown in FIG. FIG. 2 (f) shows how the combined signal of the feedback signals shown in FIGS. 2 (d) and 2 (e) (hereinafter referred to as the combined feedback signal) is obtained by vector addition. According to FIG. 2F, the combined feedback signal includes (S _1a + S _1b ) as the fundamental wave component and (S _2a + S _2b ) as the second harmonic component.

The circuit shown in FIG. 2C has two feedback paths. Therefore, by appropriately selecting the values of the inductor 201, the capacitors 202 and 302, and the resistor 301, as shown in FIG. 2 (f), the fundamental wave (S _1a + S _1b ) included in the composite feedback signal can be obtained. Both the phase and the phase of the second harmonic (S _2a + S _2b ) included in the combined feedback signal can be separated from the phase of the fundamental wave S ₀ of the input signal by approximately 180 degrees.

  Therefore, the amplifying apparatus shown in FIG. 1 can feed back the second harmonic whose phase is converted by 180 degrees to the input. That is, the second harmonic can be negatively fed back to the input. The frequency of the third order intermodulation wave is close to the frequency of the fundamental wave. Therefore, this amplifying apparatus can feed back the third-order intermodulation wave whose phase is converted by 180 degrees to the input. That is, the third-order intermodulation wave can be negatively fed back to the input.

  1 adjusts the passing phase of the first feedback circuit 20 with the inductor 201 and the capacitor 202, and adjusts the passing phase of the second feedback circuit 30 with the resistor 301 and the capacitor 302. Thus, the fundamental wave, the third-order intermodulation wave, and the second-order harmonic of the input signal can be negatively fed back to the input as a combined vector of the outputs of the two feedback circuits. Further, this amplifying device does not use a strip line or a slot line as compared with the conventional amplifying device. Therefore, even those used in the microwave band can be easily integrated into a semiconductor integrated circuit.

  As described above, according to the amplifying apparatus according to the present embodiment, the first feedback circuit 20 and the second feedback circuit 30 are used to appropriately adjust the phase of the feedback signal, so that the entire circuit is input. The fundamental wave, third-order intermodulation wave, and second-order harmonic of the signal can be negatively fed back. Therefore, it is possible to realize a high frequency negative feedback amplifying device having a wide dynamic range with a simple configuration.

(Second Embodiment)
FIG. 3 is a circuit diagram of an amplifying apparatus according to the second embodiment of the present invention. The amplifying device shown in FIG. 3 is configured by using the differential pair of the amplifying device according to the first embodiment. The operation and effect of this amplifying device are the same as those of the amplifying device according to the first embodiment. Therefore, in the following description, the operation and effect of the circuit will be omitted, and only the circuit configuration will be described.

  3 includes an amplifier circuit 15, a first feedback circuit 25, a second feedback circuit 35, DC cut capacitors 501a, 501b, 503a, and 503b, and choke inductors 502a and 502b. The amplifier circuit 15, the first feedback circuit 25, and the second feedback circuit 35 are respectively the amplifier circuit 10, the first feedback circuit 20, and the second feedback circuit 30 according to the first embodiment. It is configured using a differential pair.

  The amplifier circuit 15 includes bipolar transistors 101a, 101b, 102a, 102b, and bias circuits 104, 105. The amplifier circuit 15 amplifies the differential signal input from the input terminal pair {P1 +, P1-}. The first feedback circuit 25 includes inductors 201a and 201b and capacitors 202a and 202b. The inductors 201 a and 201 b and the capacitors 202 a and 202 b are used to adjust the passing phase of the first feedback circuit 25. The second feedback circuit 35 includes resistors 301a and 301b, capacitors 302a and 302b, and DC cut capacitors 303a and 303b, and feeds back the output of the amplifier circuit 15 to the input. The resistors 301 a and 301 b and the capacitors 302 a and 302 b are used to adjust the passing phase of the second feedback circuit 35. In the differential circuit, the same effect as that of the bypass capacitor in the non-differential circuit can be obtained by simply connecting a pair of transistors. Therefore, the amplifier circuit 15 does not need to include a separate bypass capacitor.

  In the amplifying apparatus shown in FIG. 3, the input terminals P1 + and P1− are connected to the bases of the bipolar transistors 101a and 101b via the DC cut capacitors 501a and 501b, respectively. The collectors of bipolar transistors 101a and 101b are connected to the emitters of bipolar transistors 102a and 102b, respectively. The collectors of bipolar transistors 102a and 102b are connected to output terminals P2 + and P2- through DC cut capacitors 503a and 503b, respectively. The emitters of bipolar transistors 101a and 101b are connected to one ends of inductors 201a and 201b, respectively. The other ends of the inductors 201a and 201b are connected, and the connection point Q1 is grounded. A capacitor 202a is inserted between the base and emitter of the bipolar transistor 101a, and a capacitor 202b is inserted between the base and emitter of the bipolar transistor 101b. A base current is supplied from the bias circuit 104 to the bases of the bipolar transistors 101a and 101b. A base current is supplied from the bias circuit 105 to the bases of the bipolar transistors 102a and 102b.

  The resistor 301a and the capacitor 302a are connected in parallel. The parallel connection circuit and the DC cut capacitor 303a are inserted in series between the base of the bipolar transistor 101a and the collector of the bipolar transistor 102a. The resistor 301b and the capacitor 302b are connected in parallel. The parallel connection circuit and the DC cut capacitor 303b are inserted in series between the base of the bipolar transistor 101b and the collector of the bipolar transistor 102b. The power supply voltage Vcc is supplied to the collectors of the bipolar transistors 102a and 102b via the choke inductors 502a and 502b, respectively.

(Third embodiment)
FIG. 4 is a circuit diagram of an amplifying apparatus according to the third embodiment of the present invention. The amplifying apparatus shown in FIG. 4 is obtained by replacing the first feedback circuit 25 with a first feedback circuit 26 in the amplifying apparatus according to the second embodiment. Among the constituent elements of the present embodiment, the same constituent elements as those of the second embodiment are denoted by the same reference numerals, and the description thereof is omitted.

  The first feedback circuit 26 includes inductors 201a, 201b, 203 and capacitors 202a, 202b. Similar to the second embodiment, the emitters of the bipolar transistors 101a and 101b are connected to one ends of the inductors 201a and 201b, respectively. The other ends of the inductors 201a and 201b are connected, and the connection point Q2 is grounded through the inductor 203. A capacitor 202a is inserted between the emitter and base of the bipolar transistor 101a, and a capacitor 202b is inserted between the emitter and base of the bipolar transistor 101b. The inductor 203 is used for adjusting the passing phase of the second harmonic of the first feedback circuit 26.

  The operation of the amplifying device shown in FIG. 4 will be described with reference to FIG. A differential signal is input to the first feedback circuit 26. For this reason, the two input signals cancel each other at the connection point Q2, and the potential at the connection point Q2 is always 0 V even if the input signal is an alternating current. Therefore, the pass phase of the fundamental wave component of the output signal is not affected by the inductor 203. On the other hand, when the second harmonic component of the output signal is expressed using the angular velocity ω, the non-inverted signal is cos (2ωt), the inverted signal is cos {2 (ωt + π)} = cos (2ωt), and both are in-phase signals. It can be seen that it is. Therefore, the passing phase of the second harmonic component of the output signal becomes a delayed phase due to the influence of the inductor 203.

  Here, as in the first embodiment, the circuit shown in FIG. 5A (a circuit including bipolar transistors 101a and 101b, inductors 201a, 201b, and 203 and capacitors 202a and 202b) is shown in FIG. The circuit shown (bipolar transistors 101a, 101b, 102a, 102b, inductors 201a, 201b, 203, resistors 301a, 301b, and capacitors 302a, 302b) and the circuit shown in FIG. 5C (FIG. 5 ( The passing phase characteristics of a circuit in which capacitors 202a and 202b are added to the circuit shown in b) are examined.

FIG. 5D is a feather display diagram of the input signal (fundamental wave) and the output signal (fundamental wave and second harmonic) in the circuit shown in FIG. As shown in FIG. 5D, the phase of the fundamental wave S _1c included in the feedback signals output from the terminals P4 + and P4- is relative to the fundamental wave S _{0 of the} signals input to the terminals P3 + and P3-. It advances by Φ _c. Further, the phase of the second harmonic S _2c included in the feedback signal output from the terminals P4 + and P4- advances by 2Φ _c ′ (<2Φ _c ) with respect to the fundamental wave S ₀ of the input signal. Note that the phase advance 2Φ _c ′ is smaller than 2Φ _c , and the second harmonic S _2c included in the feedback signal has a phase lag behind the signal whose phase advance is 2Φ _c . This is because the pass phase of the fundamental wave is determined by the values of the inductors 201a and 201b, whereas the pass phase of the second harmonic is determined by taking into account the value of the inductor 203 in addition to the values of the inductors 201a and 201b. Because.

FIG. 5E is a feather display diagram of the input signal (fundamental wave) and the output signal (fundamental wave and second harmonic) in the circuit shown in FIG. 5B. As shown in FIG. 5E, the phase of the fundamental wave S _1d included in the feedback signals output from the terminals P5 + and P5- is relative to the fundamental wave S _{0 of the} signals input to the terminals P3 + and P3-. Delayed by Φ _d . The phase of the second harmonic S _2d included in the feedback signals output from the terminals P5 + and P5- is delayed by 2Φ _d ′ (<2Φ _d ) with respect to the fundamental wave S ₀ of the input signal. Note that the phase delay 2Φ _d ′ is smaller than 2Φ _d , and the second harmonic S _2d included in the feedback signal has a leading phase compared to the signal whose phase advance is 2Φ _d . For this reason, the pass phase of the fundamental component is determined by the values of the inductors 201a and 201b, whereas the pass phase of the second harmonic component takes into account the value of the inductor 203 in addition to the values of the inductors 201a and 201b. It is because it is decided.

FIG. 5F is a feather display diagram of the input signal (fundamental wave) and the output signal (fundamental wave and second harmonic) in the circuit shown in FIG. FIG. 5 (f) shows how the combined signal (synthesized feedback signal) of the feedback signals shown in FIGS. 5 (d) and 5 (e) is obtained by vector addition. According to FIG. 5F, the composite feedback signal includes (S _1c + S _1d ) as the fundamental wave component and (S _2c + S _2d ) as the second harmonic component.

The circuit shown in FIG. 5C has two feedback paths. Therefore, by appropriately selecting the values of the inductors 201a, 201b, 203, capacitors 202a, 202b, 302a, 302b, and resistors 301a, 301b, as shown in FIG. 5 (f), they are included in the composite feedback signal. The phase of the fundamental wave (S _1c + S _1d ) and the phase of the second harmonic (S _2c + S _2d ) included in the combined feedback signal are both approximately 180 degrees from the phase of the fundamental wave S ₀ of the input signal. Can be separated.

  Therefore, the amplifying apparatus shown in FIG. 4 can negatively feed back the second harmonic to the input, similarly to the amplifying apparatus according to the second embodiment. Further, since the frequency of the third-order intermodulation wave is close to the frequency of the fundamental wave, this amplifying apparatus can negatively feed back the third-order intermodulation wave to the input.

  4 adjusts the passing phase of the first feedback circuit 26 with the inductors 201a, 201b, and 203 and the capacitors 202a and 202b, and the passing phase of the second feedback circuit 35 with the resistor 301a, By adjusting with 301b and capacitors 302a and 302b, the fundamental wave, the third intermodulation wave, and the second harmonic of the input signal can be negatively fed back to the input with a simple configuration.

  In addition, this amplifying device includes an inductor 203 for independently adjusting the second harmonic component as compared with the amplifying device according to the second embodiment. Therefore, it is possible to independently adjust the pass phases of the fundamental wave, the third intermodulation wave, and the second harmonic of the input signal with a simple configuration. Therefore, for example, it is possible to select the value of the inductor 203 after selecting the values of the inductors 201a and 201b so that other high-frequency characteristics such as noise characteristics are optimized.

  As described above, according to the amplifying device according to the present embodiment, by providing the inductor 203, it is possible to realize a high-frequency negative feedback amplifying device having a wide dynamic range with a higher degree of freedom than the second embodiment. Can do.

(Fourth embodiment)
FIG. 6 is a circuit diagram of an amplifying apparatus according to the fourth embodiment of the present invention. The amplifying device shown in FIG. 6 is obtained by replacing the first feedback circuit 25 with a first feedback circuit 27 in the amplifying device according to the second embodiment. Among the constituent elements of the present embodiment, the same constituent elements as those of the second embodiment are denoted by the same reference numerals, and the description thereof is omitted.

  The first feedback circuit 27 includes capacitors 202 a and 202 b and an inductor 204. Unlike the second embodiment, the emitters of the bipolar transistors 101a and 101b are directly connected, and the connection point Q3 thereof is grounded via the inductor 204. The inductor 204 is used to adjust the passing phase of the second harmonic of the amplifier circuit.

  The amplifying apparatus shown in FIG. 6 can feed back a fundamental wave whose phase is converted by 180 degrees by the action of the resistors 301a and 301b and the capacitors 202a, 202b, 302a and 302b. That is, the fundamental wave can be negatively fed back to the input. Also, this amplifying device can feed back the second harmonic whose phase is converted by 180 degrees by the action of the inductor 204 to the input. That is, the second harmonic can be negatively fed back. The frequency of the third order intermodulation wave is close to the frequency of the fundamental wave. Therefore, this amplifying apparatus can feed back the third-order intermodulation wave whose phase is converted by 180 degrees to the input. That is, the third-order intermodulation wave can be negatively fed back to the input.

  6 adjusts the passing phase of the first feedback circuit 27 with the capacitors 202a and 202b and the inductor 204, and adjusts the passing phase of the second feedback circuit 35 with the resistors 301a and 301b and the capacitor 302a. , 302b, the fundamental wave, the third-order intermodulation wave, and the second-order harmonic of the input signal can be negatively fed back to the input with a simple configuration.

  In general, a differential amplifier circuit is required to have very close characteristics (DC characteristics and AC characteristics) of a pair of amplifier circuits, that is, good pair characteristics. Since the amplifying apparatus according to the third embodiment uses the inductors 201a and 201b, the differential circuit pairability may deteriorate due to variations in stray capacitance and parasitic resistance. On the other hand, since the amplifying apparatus shown in FIG. 6 does not use the inductors 201a and 201b, there is an effect that the pair property of the amplifying circuit does not deteriorate.

  As described above, according to the amplifying apparatus according to the present embodiment, by removing the inductors 201a and 201b, a high frequency negative feedback amplifying apparatus having a wide dynamic range can be realized while maintaining the pairability of the differential amplifier circuit. can do.

(Fifth embodiment)
FIG. 7 is a circuit diagram of a frequency conversion apparatus according to the fifth embodiment of the present invention. The frequency converter shown in FIG. 7 includes an amplifier circuit 10, a first feedback circuit 20, a second feedback circuit 30, a frequency converter circuit 40, DC cut capacitors 501, 503a, 503b, 504a, 504b, and a choke inductor 502a. , 502b. This frequency conversion device is obtained by adding a frequency conversion circuit 40 to the amplification device according to the first embodiment. Among the constituent elements of the present embodiment, the same constituent elements as those of the first embodiment are denoted by the same reference numerals and description thereof is omitted.

  The frequency conversion circuit 40 includes bipolar transistors 401 and 402, a bias circuit 403, and a capacitor 404, and converts the frequency of the output signal of the amplifier circuit 10.

  In the frequency converter shown in FIG. 7, the input terminal P <b> 1 is connected to the base of the bipolar transistor 101 via the DC cut capacitor 501. The collector of the bipolar transistor 101 is connected to the emitter of the bipolar transistor 102. The collector of the bipolar transistor 102 is connected to both the emitter of the bipolar transistor 401 and the emitter of the bipolar transistor 402. The collectors of the bipolar transistors 401 and 402 are connected to output terminals P3 + and P3- through DC cut capacitors 503a and 503b, respectively. The emitter of the bipolar transistor 101 is grounded via the inductor 201. A capacitor 202 is inserted between the base and emitter of the bipolar transistor 101. Base currents are supplied from the bias circuits 104 and 105 to the bases of the bipolar transistors 101 and 102, respectively. A base current is supplied from the bias circuit 403 to the bases of the bipolar transistors 401 and 402.

  The resistor 301 and the capacitor 302 are connected in parallel. The parallel connection circuit and the DC cut capacitor 303 are inserted in series between the base of the bipolar transistor 101 and the collector of the bipolar transistor 102. The base of the bipolar transistor 102 is grounded via the bypass capacitor 103. The power supply voltage Vcc is supplied to the collectors of the bipolar transistors 401 and 402 via the choke inductors 502a and 502b, respectively.

  Input terminals P2 + and P2- are connected to the bases of bipolar transistors 401 and 402 via DC cut capacitors 504a and 504b, respectively. The capacitor 404 is inserted between the collectors of the bipolar transistors 401 and 402, and reduces leakage of the second harmonic of the local signal to the output terminals P3 + and P3-.

  In a normal usage pattern, an RF (Radio Frequency) signal received by an antenna and amplified by a low noise amplifier is input to the input terminal P1. An LO (Local Oscillator) signal output from the local oscillator is input to the input terminals P2 + and P2-. From the output terminals P3 + and P3-, an IF (Intermediate Frequency) signal mainly including an intermediate frequency signal is output.

  The frequency conversion device shown in FIG. 7 is fed back to the fundamental wave, the second harmonic, and the input of the RF signal input from the input terminal P1, with the phase converted by 180 degrees, as in the amplification device according to the first embodiment. can do. That is, the fundamental wave and the second harmonic can be negatively fed back to the input. The frequency of the third order intermodulation wave is close to the fundamental frequency. Therefore, this frequency conversion device can feed back the third-order intermodulation wave whose phase is converted by 180 degrees to the input. That is, the third-order intermodulation wave can be negatively fed back to the input.

  7 adjusts the passing phase of the first feedback circuit 20 with the inductor 201 and the capacitor 202, and adjusts the passing phase of the second feedback circuit 30 with the resistor 301 and the capacitor 302. Thus, the fundamental wave, the third-order intermodulation wave, and the second-order harmonic of the input signal can be negatively fed back to the input with a simple configuration. Further, this frequency conversion device does not use a strip line or a slot line as compared with the conventional frequency conversion device. Therefore, even those used in the microwave band can be easily integrated into a semiconductor integrated circuit.

  In the conventional frequency converter, the second harmonic of the LO signal generated in the frequency converter circuit adversely affects the operation of the amplifier circuit. On the other hand, in the frequency conversion device shown in FIG. 7, by adjusting the first feedback circuit 20, the second harmonic of the LO signal generated in the frequency conversion circuit 40 is converted into the first feedback circuit 20 and the amplification circuit. 10, the phase can be inverted 180 degrees and output to the frequency conversion circuit 40. As a result, the level of the second harmonic of the LO signal generated in the frequency conversion circuit 40 can be reduced.

  As described above, according to the frequency conversion device according to the present embodiment, by appropriately adjusting the phase of the feedback signal using the first feedback circuit 20 and the second feedback circuit 30, the circuit as a whole is obtained. The fundamental wave, third-order intermodulation wave, and second-order harmonic of the input signal can be negatively fed back. Therefore, a high frequency negative feedback frequency converter with a wide dynamic range can be realized with a simple configuration.

(Sixth embodiment)
FIG. 8 is a circuit diagram of a frequency conversion apparatus according to the sixth embodiment of the present invention. The frequency converter illustrated in FIG. 8 is configured by using the differential pair of the frequency converter according to the fifth embodiment. The operation and effect of this frequency converter are the same as those of the frequency converter according to the fifth embodiment. Therefore, in the following description, the operation and effect of the circuit will be omitted, and only the circuit configuration will be described.

  The frequency converter shown in FIG. 8 includes an amplifier circuit 15, a first feedback circuit 25, a second feedback circuit 35, a frequency converter circuit 45, DC cut capacitors 501a, 501b, 503a, 503b, 504a, 504b, and a choke. Inductors 502a and 502b are provided. The amplifier circuit 15, the first feedback circuit 25, the second feedback circuit 35, and the frequency conversion circuit 45 are the amplifier circuit 10, the first feedback circuit 20, and the second feedback, respectively, according to the fifth embodiment. The circuit 30 and the frequency conversion circuit 40 are configured using a differential pair.

  Since the amplifier circuit 15, the first feedback circuit 25, and the second feedback circuit 35 are the same as those in the second embodiment, description thereof is omitted. The frequency conversion circuit 45 includes bipolar transistors 401a, 401b, 402a, and 402b, a bias circuit 403, and a capacitor 404, and converts the frequency of the output signal of the amplifier circuit 15.

  In the frequency converter shown in FIG. 8, the input terminals P1 + and P1- are connected to the bases of the bipolar transistors 101a and 101b via the DC cut capacitors 501a and 501b, respectively. The collectors of bipolar transistors 101a and 101b are connected to the emitters of bipolar transistors 102a and 102b, respectively. The collector of the bipolar transistor 102a is connected to both the emitter of the bipolar transistor 401a and the emitter of the bipolar transistor 402a. The collector of the bipolar transistor 102b is connected to both the emitter of the bipolar transistor 401b and the emitter of the bipolar transistor 402b. The collectors of the bipolar transistors 401a and 401b are connected at the connection point R1, the collectors of the bipolar transistors 402a and 402b are connected at the connection point R2, and the connection points R1 and R2 are output via the DC cut capacitors 503a and 503b, respectively. Connected to terminals P3 + and P3-. The emitters of bipolar transistors 101a and 101b are connected to one ends of inductors 201a and 201b, respectively. The other ends of the inductors 201a and 201b are connected, and the connection point Q1 is grounded. A capacitor 202a is inserted between the base and emitter of the bipolar transistor 101a, and a capacitor 202b is inserted between the base and emitter of the bipolar transistor 101b. A base current is supplied from the bias circuit 104 to the bases of the bipolar transistors 101a and 101b. A base current is supplied from the bias circuit 105 to the bases of the bipolar transistors 102a and 102b. A base current is supplied from the bias circuit 403 to the bases of the bipolar transistors 401a, 401b, 402a, and 402b.

  The resistor 301a and the capacitor 302a are connected in parallel. The parallel connection circuit and the DC cut capacitor 303a are inserted in series between the base of the bipolar transistor 101a and the collector of the bipolar transistor 102a. The resistor 301b and the capacitor 302b are connected in parallel. The parallel connection circuit and the DC cut capacitor 303b are inserted in series between the base of the bipolar transistor 101b and the collector of the bipolar transistor 102b. The power supply voltage Vcc is supplied to the connection points R1 and R2 via the choke inductors 502a and 502b, respectively.

  The input terminal P2 + is connected to the bases of the bipolar transistors 401a and 402b via the DC cut capacitor 504a. The input terminal P2- is connected to the bases of the bipolar transistors 401b and 402a via the DC cut capacitor 504b. The capacitor 404 is inserted between the connection points R1 and R2, and reduces leakage of the second harmonic of the local signal to the output terminals P3 + and P3-.

(Seventh embodiment)
FIG. 9 is a circuit diagram of a frequency conversion device according to the seventh embodiment of the present invention. The frequency converter illustrated in FIG. 9 is obtained by replacing the first feedback circuit 25 with the first feedback circuit 26 in the frequency converter according to the sixth embodiment.

  The configuration and effects of the frequency conversion device shown in FIG. 9 are apparent from the description of the amplification devices according to the first and third embodiments and the frequency conversion devices according to the fifth and sixth embodiments. The description is omitted here.

  According to the frequency converter according to the present embodiment, by providing the inductor 203, a high-frequency negative feedback frequency converter having a wide dynamic range with a simple configuration and a higher degree of freedom than that of the sixth embodiment is realized. be able to.

(Eighth embodiment)
FIG. 10 is a circuit diagram of a frequency conversion device according to the eighth embodiment of the present invention. The frequency converter illustrated in FIG. 10 is obtained by replacing the first feedback circuit 25 with a first feedback circuit 27 in the frequency converter according to the sixth embodiment.

  The configuration and effects of the frequency conversion device shown in FIG. 10 are apparent from the description of the amplification devices according to the first and fourth embodiments and the frequency conversion devices according to the fifth and sixth embodiments. The description is omitted here.

  According to the frequency conversion device according to the present embodiment, by removing the inductors 201a and 201b, it is possible to realize a high-frequency negative feedback frequency conversion device having a wide dynamic range while maintaining good pair characteristics of the differential amplifier circuit. .

(Modification of each embodiment)
Hereinafter, modifications of the amplifying device according to the first to fourth embodiments and the frequency conversion device according to the fifth to eighth embodiments will be described. All of the devices shown below have the same effects as the devices described so far.

  First, instead of the second feedback circuits 30 and 35 shown in the embodiments, other feedback circuits may be used. For example, any of the three types of feedback circuits shown in FIG. 11 may be used. A feedback circuit 31 illustrated in FIG. 11A includes a resistor 311 and capacitors 312 and 313. The resistor 311 and the capacitor 312 are connected in series, and the capacitor 313 is connected in parallel with the series connection circuit. The feedback circuit 32 illustrated in FIG. 11B includes capacitors 321 and 322 and a resistor 323. The capacitors 321 and 322 are connected in series, and the connection point S1 is grounded via the resistor 323. A feedback circuit 33 illustrated in FIG. 11C includes capacitors 331 and 332 and an inductor 333. The capacitors 331 and 332 are connected in series, and the connection point S2 is grounded via the inductor 333.

  For example, when the feedback circuit shown in FIG. 11A is used in the amplifying device according to the first embodiment, the terminal P7 is connected to the base of the bipolar transistor 101, and the terminal P8 is connected to the collector of the bipolar transistor 102. The same applies to the case where other feedback circuits are used in devices according to other embodiments. However, in the case of the second feedback circuit 35 configured using a differential pair, one of the three types of feedback circuits shown in FIG. 11 is used as a pair.

  In the feedback circuit shown in each embodiment, the amplitude and phase of the feedback signal may be changed in accordance with conditions such as the magnitude of the input signal. For example, a variable capacitance diode (see FIG. 12) may be used instead of the capacitor included in the feedback circuit. In the amplifying device shown in FIG. 12, the first feedback circuit 29 includes a variable capacitance diode 252, and the second feedback circuit 39 includes a variable capacitance diode 352. By changing the voltages of the impedance control signals Vctl1 and Vctl2 according to the conditions such as the magnitude of the input signal, the capacitance values of the variable capacitance diodes 252 and 352 change, and as a result, the amplitude and phase of the feedback signal change. To do. As a result, when the input power is small, the gain is controlled so that the gain is high and the IIP3 (3rd Input Intercept Point) is low. When the input power is large, the gain is low, and The amplifier circuit can be controlled so that IIP3 becomes high.

  Further, instead of the cascode amplifier circuits 10 and 15 shown in the embodiments, amplifier circuits having other configurations may be used. For example, a single-type amplifier circuit (see FIG. 13) may be used. In the amplifying apparatus shown in FIG. 13, the amplifying circuit 11 includes a bipolar transistor 101 and a bias circuit 104, and the collector of the bipolar transistor 101 is connected to the output terminal P <b> 2 via a DC cut capacitor 503. The points other than the above are the same as those of the amplification device according to the first embodiment. By using such a single-type amplifier circuit, an amplifier device with good noise characteristics can be realized. The same applies to the case where a single-type amplifier circuit is used in an apparatus according to another embodiment.

  Further, instead of the bipolar transistor shown in each embodiment, a heterojunction bipolar transistor using SiGe / Si, AlGaAs / GaAs, GaInP / GaAs, or the like may be used. As a result, a device having good high frequency characteristics such as noise characteristics and distortion characteristics can be realized. Further, a MOS FET may be used in place of the bipolar transistor. Thereby, since a low-cost CMOS process can be used, the device according to each embodiment can be manufactured at a low cost. The bipolar transistors that can be replaced in this way are 101 and 102 in FIG. 1, 101a, 101b, 102a, and 102b in FIG. 3, and 101, 102, 401, 402, FIG. 9 and 10, reference numerals 101a, 101b, 102a, 102b, 401a, 401b, 402a, and 402b are given.

  In each embodiment, the first feedback circuits 20, 25, 26, and 27 are configured using an inductor and a capacitor. However, a circuit in which an inductor and a capacitor are connected in parallel may be used instead of the inductor. Thereby, the freedom degree of the phase control with respect to a fundamental wave and a 2nd harmonic can be enlarged. 1 and FIG. 7, 201a, 201b in FIGS. 3 and 8, 201a, 201b, 203 in FIGS. 4 and 9, and 204 in FIGS. 6 and 10. A reference numeral is attached.

  In addition, it is desirable to use polysilicon for the resistor shown in each embodiment, use a MOS capacitor or a MIM (Metal Insulator Metal) capacitor for the capacitor, and form the inductor with a wiring layer such as aluminum, copper, or gold. Thereby, the device according to each embodiment can be easily integrated in a semiconductor integrated circuit. Note that the resistors to be used for polysilicon are given the reference numerals 301 in FIGS. 1 and 7, and 301 a and 301 b in FIGS. 3, 4, 6, 8, 9, and 10. Capacitors to be used such as MOS capacitors are 103, 202, 302, 303 in FIG. 1, 202a, 202b, 302a, 302b, 303a, 303b in FIG. 3, and 103, 202, 302 in FIG. 303, 404, FIG. 8, FIG. 9, and FIG. 10, the reference numerals 202a, 202b, 302a, 302b, 303a, 303b, 404 are given. Inductors to be formed in the wiring layer are denoted by reference numerals 201 in FIGS. 1 and 7, 201a and 201b in FIGS. 3 and 8, 201a, 201b and 202 in FIGS. 4 and 9, and 204 in FIGS. It is attached.

  Further, the phase of the feedback signal of the device according to each embodiment is affected by the impedance when the signal source side is viewed from the device and the impedance when the load side is viewed from the device. Therefore, when a matching circuit is used for the input or output of the device according to each embodiment, the phase difference between the input signal and the feedback signal of the fundamental wave and the second harmonic is 180 degrees in consideration of the impedance of the matching circuit. Therefore, it is necessary to design the first and second feedback circuits.

  As described above, according to the amplification device according to the first to fourth embodiments, the frequency conversion device according to the fifth to eighth embodiments, and the device according to the modification of each embodiment, By adjusting both the passing phase of the feedback circuit and the passing phase of the second feedback circuit, the fundamental wave, the third intermodulation wave, and the second harmonic of the input signal are negatively fed back to the input as a whole circuit. Thus, a high frequency negative feedback amplification device and a high frequency negative feedback frequency conversion device having a wide dynamic range can be realized with a simple configuration.

(Ninth embodiment)
FIG. 14 is a diagram illustrating a configuration of a wireless terminal device according to the ninth embodiment of the present invention. In a wireless terminal device that performs transmission and reception at the same time, a part of the transmission signal generated by the transmission circuit leaks to the reception circuit during transmission. At this time, a very large dynamic range is required for the amplification device and the frequency conversion device included in the reception circuit. Further, the amplification device and the frequency conversion device included in the reception circuit are required to have low current consumption when not transmitting.

  Therefore, the wireless terminal device according to the present embodiment includes, as the amplifying device 42 and the frequency converting device 80 included in the receiving circuit, the amplifying devices according to the first to fourth embodiments and their modifications, respectively, The frequency converter which concerns on 5th-8th embodiment and those modifications is provided. The wireless terminal device also includes a control unit 90 that changes current consumption in the amplification device 42 and the frequency conversion device 80.

  When the control unit 90 receives a signal indicating the start of transmission from the transmission baseband circuit 12, the control unit 90 puts the modulator 13 and the amplification device 41 into an operating state. Thereafter, the level of the leak signal that leaks from the transmission circuit to the reception circuit increases. Therefore, when the control unit 90 receives a signal indicating the start of transmission from the transmission baseband circuit 12, the control unit 90 performs control to increase current consumption in the amplification device 42 and the frequency conversion device 80.

  Since the amplifying device 42 and the frequency converting device 80 have a negative feedback configuration, even when the current consumption is increased, the gains of the amplifying device 42 and the frequency converting device 80 do not increase so much. Accordingly, it is possible to realize a wireless terminal device having a large dynamic range without greatly changing the level diagram configuration of the wireless terminal device.

  Hereinafter, a detailed configuration of the wireless communication terminal illustrated in FIG. 14 will be described. In FIG. 14, a transmission baseband circuit 12, a modulator 13, an amplification device 41, an isolator 50, and a filter 51 constitute a transmission circuit, and filters 52 to 54, amplification devices 42 and 70, a signal generator 60, and a frequency conversion device. 80, the demodulator 21, and the reception baseband circuit 22 constitute a reception circuit. A leak signal leaking from the transmission circuit to the reception circuit becomes an interference signal for the reception circuit. The transmission baseband circuit 12 functions as a transmission signal generation unit, and the control unit 90 functions as an interference signal determination unit and control unit.

  The transmission baseband circuit 12 generates a transmission signal, and the modulator 13 modulates the transmission signal output from the transmission baseband circuit 12. The demodulator 21 demodulates the reception signal output from the filter 54, and the reception baseband circuit 22 performs predetermined processing on the reception signal output from the demodulator 21. The amplifying devices 41, 42, and 70 amplify the input signal, and the filters 51 to 54 filter the input signal. The isolator 50 allows signals to pass only in one direction. The signal generator 60 generates a signal having a predetermined frequency. The frequency converter 80 mixes two input signals. The antenna 9 transmits and receives radio waves.

  As described above, when receiving a signal indicating the start of transmission from the transmission baseband circuit 12, the control unit 90 puts the modulator 13 and the amplifying device 41 into the operating state and consumes current in the amplifying device 42 and the frequency converting device 80. Do more control. For example, the control unit 90 performs control to increase the bias voltage of the bias circuit included in the amplification device illustrated in FIG. 1 and control to increase the bias voltage of the bias circuit included in the frequency conversion device illustrated in FIG. Thereby, the consumption current in the amplification device 42 and the frequency conversion device 80 is increased, and the distortion level of the amplification device 42 and the frequency conversion device 80 is increased. As described above, when the level of the leak signal leaking from the transmission circuit to the reception circuit is high, the current consumption in the amplification device 42 and the frequency conversion device 80 is controlled to be high.

  Hereinafter, the operation of the wireless communication terminal shown in FIG. 14 will be described to describe the wireless reception device and the wireless reception method of the present invention. Here, in order to facilitate understanding, the operation of the wireless communication terminal will be described in three parts: transmission, reception, and control, but these three operations are actually executed simultaneously.

(1) Transmission Operation The transmission baseband circuit 12 generates a transmission signal and outputs it to the modulator 13. The modulator 13 modulates the transmission signal output from the transmission baseband circuit 12 and outputs it to the amplifying device 41. The amplifying device 41 amplifies the transmission signal output from the modulator 13 and outputs it to the isolator 50. The isolator 50 outputs the transmission signal output from the amplifying device 41 to the filter 51 and prevents the reflected signal from the filter 51 from entering the amplifying device 41. The filter 51 filters the transmission signal output from the isolator 50 and outputs it to the antenna 9. The antenna 9 transmits the transmission signal output from the filter 51 using radio waves.

(2) Reception Operation The antenna 9 receives a reception signal using radio waves and outputs it to the filter 52. The filter 52 filters the received signal output from the antenna 9 and outputs it to the amplifying device 42. The amplification device 42 amplifies the reception signal output from the filter 52 and outputs the amplified signal to the filter 53. The filter 53 filters the reception signal output from the amplification device 42 and outputs the filtered signal to the frequency conversion device 80. The frequency converter 80 mixes the received signal output from the filter 53 and the local signal (the signal generated by the signal generator 60 and amplified by the amplifier 70), and outputs the mixed signal to the filter 54. The filter 54 filters the reception signal output from the frequency conversion device 80 and outputs the filtered signal to the demodulator 21. The demodulator 21 demodulates the reception signal output from the filter 54 and outputs the demodulated signal to the reception baseband circuit 22. The reception baseband circuit 22 performs predetermined processing on the reception signal output from the demodulator 21.

(3) Control Operation When the control unit 90 receives a signal indicating the start of transmission from the transmission baseband circuit 12, the control unit 90 puts the modulator 13 and the amplifying device 41 into an operating state and consumes current in the amplifying device 42 and the frequency converting device 80. Do more control. In addition, when the control unit 90 receives a signal indicating the end of transmission from the transmission baseband circuit 12, the control unit 90 puts the modulator 13 and the amplifying device 41 into a non-operating state and further reduces the current consumption in the amplifying device 42 and the frequency converting device 80. Control to reduce. Thereby, the current consumption in the amplifier 42 and the frequency converter 80 is controlled to increase when the level of the leak signal leaking from the transmission circuit to the reception circuit is high.

  As described above, according to the wireless terminal device according to the present embodiment, even when a communication method (for example, W-CDMA method) that performs transmission and reception at the same time is adopted, an increase in current consumption is minimized. The dynamic range of the amplification device and the frequency conversion device can be increased while suppressing the noise. In this way, an expansion of the dynamic range and a reduction in current consumption can be achieved at the same time.

  Since the amplification device and the frequency conversion device of the present invention have a simple configuration and a wide dynamic range, they are used for various amplification devices and frequency conversion devices including receivers of wireless communication systems. be able to.

1 is a circuit diagram of an amplifying device according to a first embodiment of the present invention. The figure for demonstrating operation | movement of the amplifier which concerns on the 1st Embodiment of this invention. Circuit diagram of an amplifying device according to a second embodiment of the present invention Circuit diagram of an amplifying apparatus according to a third embodiment of the present invention The figure for demonstrating operation | movement of the amplifier which concerns on the 3rd Embodiment of this invention. Circuit diagram of an amplifier according to the fourth embodiment of the present invention The circuit diagram of the frequency converter which concerns on the 5th Embodiment of this invention The circuit diagram of the frequency converter which concerns on the 6th Embodiment of this invention Circuit diagram of frequency converter according to seventh embodiment of the present invention The circuit diagram of the frequency converter which concerns on the 8th Embodiment of this invention Another circuit diagram of the second feedback circuit according to the first to eighth embodiments of the present invention The circuit diagram of the amplifier which concerns on the modification of the 1st Embodiment of this invention The other circuit diagram of the amplifier which concerns on the modification of the 1st Embodiment of this invention The figure which shows the structure of the radio | wireless terminal apparatus which concerns on the 9th Embodiment of this invention. The figure which shows the negative feedback amplifier of the 1st prior art example The figure which shows the negative feedback amplifier of the 2nd prior art example The figure which shows the negative feedback amplifier of the 3rd prior art example The figure which shows the negative feedback amplifier of the 4th prior art example The figure which shows the negative feedback amplifier of the 5th prior art example The figure which shows the negative feedback amplifier of a 6th prior art example

Explanation of symbols

10, 11, 15... Amplifier circuits 20, 25, 26, 27, 29... First feedback circuits 30, 35, 39... Second feedback circuits 31, 32, 33. , 102, 401, 402 ... Bipolar transistor 103 ... Bypass capacitors 104, 105, 403 ... Bias circuits 201, 203, 204, 333 ... Inductors 202, 302, 312, 313, 321, 322, 331, 332, 404 ... Capacitors 252 352, variable capacitance diodes 301, 311, 323, resistors 303, 501, 503, 504, DC cut capacitors 502, choke inductors.

Claims (39)

An amplifying apparatus that amplifies and outputs an input signal,
An amplifier circuit that is disposed on a path from the input terminal to the output terminal and amplifies the signal input from the input terminal;
A first feedback circuit that is disposed between the input of the amplifier circuit and a feedback terminal and feeds back the feedback output of the amplifier circuit to the input of the amplifier circuit while changing the phase of a signal passing therethrough;
A second feedback circuit that is disposed between the input and output of the amplifier circuit and feeds back the output of the amplifier circuit to the input of the amplifier circuit while changing the phase of the signal passing therethrough,
The phase of the signal obtained by synthesizing the fundamental wave included in the two feedback signals fed back by the first and second feedback circuits is approximately 180 degrees away from the phase of the fundamental wave of the signal input from the input terminal. An amplifying apparatus characterized by comprising:   The phase of the signal obtained by synthesizing the second harmonic included in the two feedback signals fed back by the first and second feedback circuits is approximately 180 degrees from the phase of the fundamental wave of the signal input from the input terminal. The amplifying device according to claim 1, wherein the amplifying device is separated. One signal is input from the input terminal,
2. The amplification device according to claim 1, wherein each of the amplification circuit, the first feedback circuit, and the second feedback circuit inputs and outputs one system of signals. The first feedback circuit includes:
An inductor having one end connected to the feedback terminal and the other end grounded;
The amplifying apparatus according to claim 3, further comprising: a capacitor having one end connected to the feedback terminal and the other end connected to the input terminal. The first feedback circuit includes:
A parallel connection circuit of an inductor and a capacitor, with one end connected to the feedback terminal and the other end grounded;
The amplifying apparatus according to claim 3, further comprising: a capacitor having one end connected to the feedback terminal and the other end connected to the input terminal. The amplifier circuit is
A first bipolar transistor having a base connected to the input and an emitter connected to the feedback terminal;
The amplifying apparatus according to claim 3, further comprising: a second bipolar transistor having an emitter connected to the collector of the first bipolar transistor and a collector connected to the output.   The amplifying device according to claim 3, wherein the amplifying circuit includes a bipolar transistor having a base connected to an input, an emitter connected to the feedback terminal, and a collector connected to an output. The second feedback circuit includes:
Resistance,
A first capacitor connected in parallel with the resistor;
The amplification device according to claim 3, further comprising: a second capacitor connected in series with a parallel connection circuit of the resistor and the first capacitor. The second feedback circuit includes:
Resistance,
A first capacitor connected in series with the resistor;
The amplification device according to claim 3, further comprising: a second capacitor connected in parallel with a series connection circuit of the resistor and the first capacitor. The second feedback circuit includes:
First and second capacitors connected in series;
The amplifying apparatus according to claim 3, comprising a resistor disposed between a connection point of the first and second capacitors and a ground. The second feedback circuit includes:
First and second capacitors connected in series;
The amplifying apparatus according to claim 3, further comprising an inductor disposed between a connection point of the first and second capacitors and a ground. A differential signal consisting of an in-phase signal and a reverse-phase signal is input from the input terminal,
The amplifier circuit is
An in-phase input unit to which the in-phase signal is input, and an in-phase amplifier having one in-phase feedback terminal that is one of the feedback terminals and operating based on the in-phase signal;
A negative phase input terminal to which the negative phase signal is input, and a negative phase feedback terminal that is another one of the feedback terminals, and a negative phase amplification unit that operates based on the negative phase signal;
The first feedback circuit includes one or more feedback units,
The second feedback circuit includes:
A common-mode feedback unit that operates based on the common-mode signal;
The amplification device according to claim 1, further comprising: a negative phase feedback unit that operates based on the negative phase signal. The first feedback circuit includes first and second feedback units having one end grounded,
The other end of the first feedback section is connected to the common-mode feedback terminal,
Still another end of the first feedback section is connected to the common-mode input terminal,
The other end of the second feedback unit is connected to the negative phase feedback terminal,
The amplification device according to claim 12, wherein the other end of the second feedback section is connected to the negative phase input terminal. The first feedback circuit includes first to third feedback units having one end connected to one connection point,
The other end of the first feedback section is connected to the common-mode feedback terminal,
Still another end of the first feedback section is connected to the common-mode input terminal,
The other end of the second feedback unit is connected to the negative phase feedback terminal,
Still another end of the second feedback unit is connected to the negative phase input terminal,
The amplifying apparatus according to claim 12, wherein the other end of the third feedback unit is grounded. The in-phase feedback terminal and the negative-phase feedback terminal are directly connected,
The first feedback circuit includes first to third feedback units, one end of which is connected to the in-phase feedback terminal and the negative-phase feedback terminal.
The other end of the first feedback unit is connected to the in-phase input terminal,
The other end of the second feedback unit is connected to the negative phase input terminal,
The amplifying apparatus according to claim 12, wherein the other end of the third feedback unit is grounded.   The amplifying apparatus according to claim 12, wherein the feedback unit includes an inductor.   The amplification device according to claim 12, wherein the feedback unit includes a parallel connection circuit of an inductor and a capacitor. The in-phase amplification unit and the reverse-phase amplification unit are both
A first bipolar transistor having a base connected to an input and an emitter connected to one of the common-mode feedback terminal and the negative-phase feedback terminal;
The amplifying apparatus according to claim 12, further comprising: a second bipolar transistor having an emitter connected to a collector of the first bipolar transistor and a collector connected to an output.   In each of the in-phase amplification unit and the negative-phase amplification unit, the base is connected to the input, the emitter is connected to one of the common-mode feedback terminal and the negative-phase feedback terminal, and the collector is connected to the output. The amplifying device according to claim 12, comprising a bipolar transistor. The in-phase feedback unit and the negative-phase feedback unit are both
Resistance,
A first capacitor connected in parallel with the resistor;
The amplifying apparatus according to claim 12, further comprising: a second capacitor connected in series with a parallel connection circuit of the resistor and the first capacitor. The in-phase feedback unit and the negative-phase feedback unit are both
Resistance,
A first capacitor connected in series with the resistor;
The amplification device according to claim 12, further comprising a second capacitor connected in parallel with a series connection circuit of the resistor and the first capacitor. The in-phase feedback unit and the negative-phase feedback unit are both
First and second capacitors connected in series;
The amplifying apparatus according to claim 12, further comprising a resistor disposed between a connection point of the first and second capacitors and a ground. The in-phase feedback unit and the negative-phase feedback unit are both
First and second capacitors connected in series;
The amplifying apparatus according to claim 12, comprising an inductor disposed between a connection point of the first and second capacitors and ground.   The amplifying apparatus according to claim 1, wherein impedances of the first and second feedback circuits change in accordance with first and second impedance control signals, respectively. A frequency converter that amplifies an input signal, converts a frequency, and outputs the converted signal.
An amplifier circuit that is disposed on a path from the input terminal to the output terminal and amplifies the signal input from the input terminal;
A frequency conversion circuit for converting the frequency of the signal amplified by the amplification circuit;
A first feedback circuit that is disposed between the input of the amplifier circuit and a feedback terminal and feeds back the feedback output of the amplifier circuit to the input of the amplifier circuit while changing the phase of a signal passing therethrough;
A second feedback circuit that is disposed between the input and output of the amplifier circuit and feeds back the output of the amplifier circuit to the input of the amplifier circuit while changing the phase of the signal passing therethrough,
The phase of the signal obtained by synthesizing the fundamental wave included in the two feedback signals fed back by the first and second feedback circuits is approximately 180 degrees away from the phase of the fundamental wave of the signal input from the input terminal. A frequency converter characterized by comprising:   The phase of the signal obtained by synthesizing the second harmonic contained in the two feedback signals fed back by the first and second feedback circuits is approximately 180 degrees from the phase of the fundamental wave of the signal input from the input terminal. The frequency converter according to claim 25, wherein the frequency converter is separated. One signal is input from the input terminal,
26. The frequency converter according to claim 25, wherein each of the amplifier circuit, the first feedback circuit, and the second feedback circuit inputs and outputs one system signal. The first feedback circuit includes:
An inductor having one end connected to the feedback terminal and the other end grounded;
28. The frequency converter according to claim 27, further comprising a capacitor having one end connected to the feedback terminal and the other end connected to the input terminal. The first feedback circuit includes:
A parallel connection circuit of an inductor and a capacitor, with one end connected to the feedback terminal and the other end grounded;
28. The frequency converter according to claim 27, further comprising a capacitor having one end connected to the feedback terminal and the other end connected to the input terminal. A differential signal consisting of an in-phase signal and a reverse-phase signal is input from the input terminal,
The amplifier circuit is
An in-phase input unit to which the in-phase signal is input, and an in-phase amplifier having one in-phase feedback terminal that is one of the feedback terminals and operating based on the in-phase signal;
A negative phase input terminal to which the negative phase signal is input, and a negative phase feedback terminal that is another one of the feedback terminals, and a negative phase amplification unit that operates based on the negative phase signal;
The first feedback circuit includes one or more feedback units,
The second feedback circuit includes:
A common-mode feedback unit that operates based on the common-mode signal;
The frequency converter according to claim 25, further comprising a negative phase feedback unit that operates based on the negative phase signal. The first feedback circuit includes first and second feedback units having one end grounded,
The other end of the first feedback section is connected to the common-mode feedback terminal,
Still another end of the first feedback section is connected to the common-mode input terminal,
The other end of the second feedback unit is connected to the negative phase feedback terminal,
The frequency conversion device according to claim 30, wherein the other end of the second feedback unit is connected to the negative phase input terminal. The first feedback circuit includes first to third feedback units having one end connected to one connection point,
The other end of the first feedback section is connected to the common-mode feedback terminal,
Still another end of the first feedback section is connected to the common-mode input terminal,
The other end of the second feedback unit is connected to the negative phase feedback terminal,
Still another end of the second feedback unit is connected to the negative phase input terminal,
The frequency converter according to claim 30, wherein the other end of the third feedback unit is grounded. The in-phase feedback terminal and the negative-phase feedback terminal are directly connected,
The first feedback circuit includes first to third feedback units, one end of which is connected to the in-phase feedback terminal and the negative-phase feedback terminal.
The other end of the first feedback unit is connected to the in-phase input terminal,
The other end of the second feedback unit is connected to the negative phase input terminal,
The frequency converter according to claim 30, wherein the other end of the third feedback unit is grounded.   The frequency converter according to claim 30, wherein the feedback unit includes an inductor.   The frequency converter according to claim 30, wherein the feedback unit includes a parallel connection circuit of an inductor and a capacitor.   26. The frequency conversion device according to claim 25, wherein impedances of the first and second feedback circuits change in accordance with first and second impedance control signals, respectively. The amplifying device according to claim 1, which amplifies a received signal received by an antenna;
The frequency converter according to claim 25, which converts the frequency of the output of the amplifier.
An interfering signal determination unit that performs determination regarding an interfering signal with respect to the received signal;
A wireless reception device comprising: a control unit that changes current consumption in the amplification device and the frequency conversion device based on a determination result in the interference signal determination unit.   The radio reception apparatus according to claim 37, wherein the control unit increases current consumption in the amplification device and the frequency conversion device when it is determined that the level of the interference signal is high. A transmission signal generation unit that generates a transmission signal transmitted from the antenna;
The radio reception apparatus according to claim 38, wherein the interference signal determination unit performs the determination based on whether or not the transmission signal is generated.

Images