JP2005039912A - Control unit of ac motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a torque control unit of a permanent magnet synchronous motor capable of realizing precision torque control even in the entire speed range, from a low speed range to a high speed range. <P>SOLUTION: So as for the current detection values of a d-axis and a q-axis in a rotating coordinate system to agree with the current instruction value of a first d-axis and that of a first q-axis provided from a superior position, second current instruction values of the d-axis and the q-axis are calculated. The output voltage of a power converter is controlled according to the second current instruction values of the d-axis and q-axis. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

【００１４】
数１において、Ｒ_１ ^＊は電動機の抵抗の設定値であり、Ｌｄ^＊ はｄ軸インダクタンスの設定値であり、Ｌｑ^＊はｑ軸インダクタンスの設定値であり、Ｋｅ^＊は誘起電圧定数の設定値であり、ω_１ ^＊：周波数指令値である。また、磁極位置検出器４では、電気角６０度毎の磁極位置を把握することができる。この時の位置検出値θ_ｉを本実施例では、（数２）のように示す。
【００１５】
【数２】

【００１６】
（数２）において、ｉ＝０，１，２，３，４，５である。周波数演算部５においては、この位置検出値θ_ｉ から、最短で６０度区間における平均の回転速度である周波数指令値ω_１ ^＊を（数３）に従い算出する。
【００１７】
【数３】

【００１８】
（数３）において、Δθはθ_ｉ−θ（ｉ−１）であり、Δｔは６０度区間の位置検出信号を検出するまでの時間である。位相演算部６では、位置検出値θ_ｉ と周波数指令値ω_１ ^＊を用いて、回転位相指令θｃ^＊を（数４）のように演算して、電動機１の基準位相を制御する。
【００１９】
【数４】

【００２０】
以上が、本発明の永久磁石同期電動機のベクトル制御装置での電圧制御と位相制御の基本動作である。
【００２１】
次に、本発明の一つ特徴であるｄ軸電流指令演算部８およびｑ軸電流指令演算部９の構成について説明する。最初に、ｄ軸電流指令演算部８の構成を図２に示す。構成は、上位装置から与えられる第１のｄ軸電流指令値Ｉｄ^＊ と電流検出値Ｉｄｃの偏差に比例ゲインＫｐｄを乗じる比例演算部８Ａの出力信号と、積分ゲインＫｉｄを乗じて積分処理を行う積分演算部８Ｂの出力信号を加算して、（数５）に従い、第２のｄ軸電流指令値Ｉｄ^＊＊を出力する。
【００２２】
【数５】

【００２３】
次に、ｑ軸電流指令演算部９の構成を図３に示す。構成は、上位装置から与えられる第１のｑ軸電流指令値Ｉｑ^＊ と電流検出値Ｉｑｃの偏差に比例ゲインＫｐｑを乗じる比例演算部９Ａの出力信号と、積分ゲインＫｉｑを乗じて積分処理を行う積分演算部９Ｂの出力信号を加算して、（数６）に従い、第２のｑ軸電流指令Ｉｑ^＊＊を出力する。
【００２４】
【数６】

【００２５】
ここでは、ｄ軸電流指令演算部８およびｑ軸電流指令演算部９において、比例＋積分演算の処理を行っているが、比例あるいは積分演算のみでも同様の効果を得ることはできる。
【００２６】
次に、本発明のもたらす一つの作用効果について、本実施例により説明する。
【００２７】
図１の制御装置において、第１のｄ軸およびｑ軸の電流指令値Ｉｄ^＊，Ｉｑ^＊を電圧ベクトル演算部１０に入力した場合について考える（第２の電流指令値Ｉｄ^＊＊，Ｉｑ^＊＊は演算に使用しない）。電圧ベクトル演算部１０においては、（数７）に従い電圧指令値Ｖｄ^＊，Ｖｑ^＊を演算した場合である。
【００２８】
【数７】

【００２９】
（数７）に示すベクトル演算で、トルク制御運転を行った場合、高トルクが要求されると、トルクに見合った大きな電流を流す必要がある。連続した時間で高トルクが要求される場合には、電動機電流による発熱により、時間と共に電動機内部の巻線抵抗値Ｒが増加する。すると、電圧ベクトル演算部１０で演算する抵抗設定値Ｒ^＊ と実抵抗値Ｒが一致しなくなるため、永久磁石同期電動機１に必要な電圧を供給することができなくなり、その結果、特に低速域では、トルク発生に必要な電流が流れず、トルク不足が発生する。
【００３０】
図９は、（数７）でベクトル演算を行った場合のモータトルクと回転数の実測結果を示す。破線がトルク指令値、実線が測定したモータトルク値である。丸の破線で囲んでいる低速の高トルク側（Ａの領域）では、指令値通りのトルクが発生していないことがわかる。
【００３１】
そこで、本実施例では、ｄ軸およびｑ軸の電流検出値Ｉｄｃ，Ｉｑｃが、上位装置から与えられる各々の指令値に一致するように、電流指令演算部８，９において第２の電流指令値Ｉｄ^＊＊，Ｉｑ^＊＊を演算し、（数１）により、変換器の出力電圧を演算するようにしている。
【００３２】
この結果、電圧ベクトル演算部１０で設定するＲ^＊ と実抵抗値Ｒが一致していなくとも、電動機電流を電流指令値に一致させるように出力電圧が制御され、全速度域において「高精度なトルク制御」を実現することができる。
【００３３】
図１０には、本実施例を用いた場合の一つの実測結果を示す。図９と図１０の破線は、トルク指令値を示しており、又、実線はそれぞれのトルク指令値で実験した実際のトルク値を示している。図９と図１０を比較すると、全体的に本実施例を用いた図１０の方が精度良く追従していることがわかる。特に、トルク指令値が約２５［Ｎｍ］の低速領域において、図１０の方が最大で８［Ｎｍ］ほど精度よく追従していることがわかるように、本実施例を用いた実験結果である図１０では、低速の高トルク側において、指令値通りのトルクが発生していることがわかる。
【００３４】
これより、本実施例を用いると全領域において高精度に追従することが可能であり、そして、特に低速領域において、高トルクの出力を実現することが可能であることがわかる。
【００３５】
＜実施例２＞
図４は、本発明の実施例２を示す。実施例２は、第１のｄ軸およびｑ軸の電流指令値Ｉｄ^＊，Ｉｑ^＊と、電流指令演算部８，９の出力値Ｉｄ^＊＊，Ｉｑ^＊＊との加算値により、第２の電流指令値Ｉｄ^＊＊＊，Ｉｑ^＊＊＊を得る方式の永久磁石同機電動機の制御装置である。
【００３６】
図４において、１〜１１，２１は図１のものと同一物である。１２は第１のｄ軸電流指令値Ｉｄ^＊とｄ軸の電流指令演算部８の出力値Ｉｄ^＊＊とを加算する加算部、１３は第１のｑ軸電流指令値Ｉｑ^＊とｑ軸の電流指令演算部９の出力値Ｉｑ^＊＊とを加算する加算部である。この方法で演算される電流指令値Ｉｄ^＊＊＊，Ｉｑ^＊＊＊を用いて、（数８）に示す電圧指令値Ｖｄ^＊＊＊，Ｖｑ^＊＊＊を演算し、変換器出力電圧を制御する。
【００３７】
【数８】

【００３８】
この方式では、トルク発生に比例する電流指令値は、基本的にＩｄ^＊ およびＩｑ^＊により供給される。
【００３９】
また、電圧ベクトル演算部１０で設定するモータ定数とモータの実際値が一致していなくとも、電流指令演算部８，９により電動機電流を電流指令値に一致させるように（過不足電流分を補うように）電流指令値が演算されるため、全速度域において高精度なトルク制御を実現することができる。Ｉｄ^＊とＩｄｃ，Ｉｑ^＊とＩｑｃが各々一致することを考慮すれば、前記実施例と同様に動作し、同様の効果が得られることは明らかである。
【００４０】
また、サンプリング演算周期が長い場合は、制御ゲインを上げることができず高応答を実現することができないが、本実施例のようなフィードフォワード制御をすることにより、即応性を高めることが可能となる。
【００４１】
＜実施例３＞
図５は、実施例３を示す。本実施例は、第１のｄ軸およびｑ軸電流指令値Ｉｄ^＊，Ｉｑ^＊ の一次進み遅れ信号と、前記電流指令値Ｉｄ^＊，Ｉｑ^＊の一次遅れ信号と電流検出値Ｉｄｃ，Ｉｑｃから演算される電流指令値Ｉｄ^＊＊′，Ｉｑ^＊＊′との加算値より、第２の電流指令値Ｉｄ^＊＊＊′，Ｉｑ^＊＊＊′を得る方式の永久磁石同機電動機の制御装置である。
【００４２】
図５において、１〜１１，２１は図１のものと同一物である。１２はｄ軸の電流指令演算部８の出力値Ｉｄ^＊＊とｄ軸の第１の電流指令値Ｉｄ^＊ とを加算する加算部、１３はｑ軸の電流指令演算部９の出力値Ｉｑ^＊＊とｑ軸の第１の電流指令値Ｉｑ^＊ とを加算する加算部、１４は遅れ時定数Ｔ１ｄのゲインを持つ一次遅れフィルタ１５は遅れ時定数Ｔ１ｄ，進み時定数Ｔ２ｄのゲインを持つ一次進み遅れフィルタ１６は遅れ時定数Ｔ１ｑのゲインを持つ一次遅れフィルタは遅れ時定数Ｔ１ｑ，進み時定数Ｔ２ｑのゲインを持つ一次進み遅れフィルタである。この方法で演算される電流指令値Ｉｄ^＊＊＊′，Ｉｑ^＊＊＊′を用いて、（数９）に示す電圧指令値Ｖｄ^＊＊＊′，Ｖｑ^＊＊＊′を演算し、変換器出力電圧を制御する。
【００４３】
【数９】

【００４４】
ここで、ｄ軸およびｑ軸電流指令演算部８，９の比例ゲイン（Ｋｐｄ，Ｋｐｑ），積分ゲイン（Ｋｉｄ，Ｋｉｑ）を、（数１０）のように設定し、
【００４５】
【数１０】

【００４６】
ここに、ωｃｄ，ωｃｑはｄ軸およびｑ軸の制御応答角周波数［ｒａｄ／ｓ］ である。また、Ｌｄ，Ｌｑは電動機のインダクタンス分、Ｒは電動機の抵抗分である。１４〜１７の演算部において、Ｔ１ｄ，Ｔ２ｄ，Ｔ１ｑ，Ｔ２ｑを、（数１１）のように設定すると、
【００４７】
【数１１】

【００４８】
電流指令値Ｉｄ^＊，Ｉｑ^＊から、電流検出値Ｉｄｃ，Ｉｑｃまでの電流制御応答を（数１２）のように、一次遅れで定義することができるので、オーバーシュート・レスなトルク制御系を構成することが可能である。
【００４９】
【数１２】

【００５０】
このような方式でも、Ｉｄ^＊とＩｄｃ，Ｉｑ^＊とＩｑｃが各々一致することを考慮すれば、前記実施例と同様に動作し、同様の効果が得られることは明らかであり、さらにオーバーシュート・レスなトルク制御系を構築することができる。
【００５１】
＜実施例４＞
上記の実施例１〜３までは、高価な電流検出器３で３相の交流電流Ｉｕ〜Ｉｗを検出する方式であったが、電流検出器を用いずに電流検出を行うことができる。図６に、この実施例４を示す。図６において、１，２，４〜１１，２１は、図１のものと同一物である。１７は電力変換器の入力母線（直流シャント抵抗）に流れる直流電流ＩＤＣから、同期電動機に流れる３相の交流電流Ｉｕ，Ｉｖ，Ｉｗを推定する電流推定部である。
【００５２】
この推定電流値Ｉｕ＾，Ｉｖ＾，Ｉｗ＾を用いて、座標変換部７において、ｄ軸及びｑ軸の電流検出値Ｉｄｃ，Ｉｑｃを演算する。このような方式でも、Ｉｄ^＊とＩｄｃ，Ｉｑ^＊とＩｑｃが各々一致することから、前記実施例と同様に動作し、同様の効果が得られることは明らかである。
【００５３】
また、これにより、電流検出器の代わりにあらかじめ過電流防止のために組み込まれている直流シャント抵抗からＩｄｃ，Ｉｑｃを求めるため電流検出器を少なく制御することが可能である。
【００５４】
＜実施例５＞
実施例５は、安価な電流検出を行う制御装置に、図４の制御装置を適用したものである。図７に、実施例５の構成を示す。図７において、１，２，４〜１３，２１は、図４のものと同一物である。１７は電力変換器の入力母線に流れる直流電流ＩＤＣから、同期電動機に流れる３相の交流電流Ｉｕ，Ｉｖ，Ｉｗを推定する電流推定部である。この推定電流値Ｉｕ＾，Ｉｖ＾，Ｉｗ＾を用いて、座標変換部７において、ｄ軸及びｑ軸の電流検出値Ｉｄｃ，Ｉｑｃを演算する。このような方式でも、Ｉｄ^＊とＩｄｃ，Ｉｑ^＊とＩｑｃが各々一致することから、前記実施例と同様に動作し、同様の効果が得られることは明らかである。また、これにより、電流検出器の代わりにあらかじめ過電流防止のために組み込まれている直流シャント抵抗からＩｄｃ，Ｉｑｃを求めるため電流検出器を少なく制御することが可能である。
【００５５】
＜実施例６＞
実施例６は、安価な電流検出を行う制御装置に、図５の制御装置を適用したものである。図８に、実施例６の構成を示す。図８において、１，２，４〜１７，２１は、図５のものと同一物である。１７は電力変換器の入力母線に流れる直流電流ＩＤＣから、同期電動機に流れる３相の交流電流Ｉｕ，Ｉｖ，Ｉｗを推定する電流推定部である。この推定電流値Ｉｕ＾，Ｉｖ＾，Ｉｗ＾を用いて、座標変換部７において、ｄ軸及びｑ軸の電流検出値Ｉｄｃ，Ｉｑｃを演算する。このような方式でも、Ｉｄ^＊とＩｄｃ，Ｉｑ^＊とＩｑｃが各々一致することから、前記実施例と同様に動作し、同様の効果が得られることは明らかである。
【００５６】
また、これにより、電流検出器の代わりにあらかじめ過電流防止のために組み込まれている直流シャント抵抗からＩｄｃ，Ｉｑｃを求めるため電流検出器を少なく制御することが可能である。
【００５７】
＜実施例７＞
図１１は、本発明の実施例７を示す。実施例７は、ｄ軸側では第１の電流指令値を、ｑ軸側では第２の電流指令値Ｉｑ^＊＊を使って電圧ベクトル演算を行う方式の永久磁石同機電動機の制御装置である。図において、１〜７，８〜１１，２１は図１のものと同一物である。
【００５８】
この方法でも、ｄ軸電流指令値が零（Ｉｄ^＊＝０）であれば、Ｉｑ^＊とＩｑｃが一致することから、前記実施例と同様に動作し、同様の効果が得られることは明らかである。
【００５９】
＜実施例８＞
図１２を用いて本発明をモジュールに適用した例について説明する。本実施例は、第１実施例の実施形態を示すものである。ここで、周波数演算部５，位相演算部６，電圧ベクトル演算部１０，ｄ軸電流指令演算部８，ｑ軸電流指令演算部９，座標変換部１１は１チップマイコンを用いて構成している。また、１チップマイコンと電力変換器２は、同一基板上で構成される１モジュール内に納められている形態となっている。ここでいうモジュールとは「規格化された構成単位」という意味であり、分離可能なハードウェア／ソフトウェアの部品から構成されているものである。尚、製造上、同一基板上で構成されていることが好ましいが、同一基板に限定はされない。これより、同一筐体に内蔵された複数の回路基板上に構成されても良い。なお、他の実施例においても同様の形態構成をとることができる。
【００６０】
【発明の効果】
本発明は、高精度なトルク制御を実現できる交流電動機の制御装置を提供することにある。
【図面の簡単な説明】
【図１】本発明の一実施例を示す永久磁石同期電動機の制御装置の構成図。
【図２】図１の制御装置におけるｄ軸電流指令演算部８の説明図。
【図３】図１の制御装置におけるｑ軸電流指令演算部９の説明図。
【図４】本発明の他の実施例を示す永久磁石同期電動機の制御装置の構成図。
【図５】本発明の他の実施例を示す永久磁石同期電動機の制御装置の構成図。
【図６】本発明の他の実施例を示す永久磁石同期電動機の制御装置の構成図。
【図７】本発明の他の実施例を示す永久磁石同期電動機の制御装置の構成図。
【図８】本発明の他の実施例を示す永久磁石同期電動機の制御装置の構成図。
【図９】本発明を用いない場合のモータの回転数−トルク実測特性。
【図１０】本発明を用いた場合のモータの回転数−トルク実測特性。
【図１１】本発明の他の実施例を示す永久磁石同期電動機の制御装置の構成図。
【図１２】本発明の制御装置を適用したモジュールの実施例の説明図。
【符号の説明】
１…永久磁石同期電動機、２…電力変換器、３…電流検出器、４…磁極位置検出器、５…周波数演算部、６…位相演算部、７…座標変換部、８…ｄ軸電流指令演算部、９…ｑ軸電流指令演算部、１０…電圧ベクトル演算部、１７…電流推定部、２１…直流電源、Ｉｄ^＊ …第１のｄ軸電流指令値、Ｉｑ^＊ …第１のｑ軸電流指令値、Ｉｄ^＊＊，Ｉｄ^＊＊＊，Ｉｄ^＊＊＊′…第２のｄ軸電流指令値、Ｉｑ^＊＊，Ｉｑ^＊＊＊，Ｉｑ^＊＊＊′…第２のｑ軸電流指令値。[0001]
BACKGROUND OF THE INVENTION
The present invention relates to an AC motor control device and a module.
[0002]
[Prior art]
As a conventional technique provided with a magnetic pole position detector and an electric motor current sensor, there is a control device described in Japanese Patent Application Laid-Open No. 2000-324881. This control device switches the calculation means of the voltage command value for controlling the power converter in the low speed range and the high speed range, detects the motor current in the low speed range, and detects the current command value and the current detection value in the rotating coordinate system. The voltage command value is calculated so as to match, and in the high speed range, the voltage command value is calculated so as to be approximately proportional to the frequency command value.
[0003]
[Patent Document 1]
Japanese Patent Laid-Open No. 2000-324881
[Problems to be solved by the invention]
The objective of this invention is providing the control apparatus of the alternating current motor which can implement | achieve highly accurate torque control.
[0005]
[Means for Solving the Problems]
One feature of the present invention is that in the control apparatus for an AC motor that controls the output voltage of the power converter that drives the AC motor, the q axis current detection value of the rotating coordinate system is the first of the q axis of the rotating coordinate system. The second current command value of the q-axis of the rotating coordinate system is created so as to approach the current command value of the rotating coordinate system, and the output voltage of the power converter is calculated from the second current command value of the q-axis of the rotating coordinate system. Is to create.
[0006]
The other features of the present invention are as described in the claims.
[0007]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.
[0008]
<Example 1>
FIG. 1 shows a configuration example of a control device for a permanent magnet synchronous motor which is one of AC motors according to an embodiment of the present invention.
[0009]
1 is a permanent magnet synchronous motor, 21 is a DC power supply, 2 is a power converter having an input of a DC power supply 21 that outputs an output voltage proportional to three-phase AC voltage command values Vu ^* , Vv ^* , and Vw ^*. A current detector capable of detecting the three-phase AC currents Iu, Iv, Iw, 4 a magnetic pole position detector capable of detecting a position detection value θ _i every 60 ° of electrical angle of the motor, and 5 a frequency command value from the position detection value θ _i a frequency calculation unit for calculating ω ₁ ^* , 6 a phase calculation unit for calculating the rotational phase command θc ^* of the motor from the position detection value θ _i and the frequency command value ω ₁ ^* , and 7 for the three-phase alternating currents Iu, Iv, A coordinate conversion unit for outputting the detected current values Idc and Iqc of the d-axis and the q-axis from the detected values Iuc, Ivc and Iwc of the Iw and the rotation phase command θc ^* , and 8 denotes the first d-axis current command values Id ^* and d Depending on the deviation of the detected shaft current value Idc, the second d-axis A d-axis current command calculation unit 9 that outputs a current command value Id ^** , a second q-axis current command value Iq according to a deviation between the first q-axis current command value Iq ^* and the q-axis current detection value Iqc. q-axis current calculation unit for outputting a ^**, 10 motor constants and the second current command value ^Id **, ^{Iq **} and the voltage command value ^Vd ** based on the frequency command value omega ₁ ^{*, Vq **} Is a voltage conversion unit for outputting three-phase AC voltage command values Vu ^* , Vv ^* and Vw ^* from the voltage command values Vd ^** and Vq ^** and the rotation phase command θc ^*. .
[0010]
First, the basic operation of the d-axis current command calculation unit 8 and the q-axis current calculation command calculation unit 9, which is one feature of the present invention, will be described.
[0011]
In the coordinate conversion unit 7, the current value Iuc of the three-phase alternating current detected by the current detector 3.
Current detection values Idc and Iqc for the d-axis and q-axis are calculated from Ivc and Iwc and the rotational phase command θc ^* , and the current detection values Idc and Iqc for the d-axis and q-axis current command calculation units 8 and 9 are calculated. Are matched with the first d-axis and q-axis current command values Id ^* and Iq ^* given from the host device, respectively, so that the second d-axis and q-axis current command values Id ^** and Iq ^** Is calculated.
[0012]
The voltage vector calculation unit 10 calculates the voltage command values Vd ^** and Vq ^** using the calculated current command values Id ^** and Iq ^** and the motor constants as shown in (Formula 1) in advance. Control the converter output voltage.
[0013]
[Expression 1]

[0014]
In Equation 1, R ₁ ^* is the set value of the resistance of the motor, Ld ^* is the set value of the d-axis inductance, Lq ^* is the set value of the q-axis inductance, and Ke ^* is the set value of the induced voltage constant Ω ₁ ^* : a frequency command value. In addition, the magnetic pole position detector 4 can grasp the magnetic pole position for every electrical angle of 60 degrees. In this embodiment, the position detection value θ _i at this time is expressed as (Equation 2).
[0015]
[Expression 2]

[0016]
In (Expression 2), i = 0, 1, 2, 3, 4, and 5. The frequency calculation unit 5 calculates a frequency command value ω ₁ ^* , which is the average rotation speed in the shortest 60-degree section, from the position detection value θ _i according to (Equation 3).
[0017]
[Equation 3]

[0018]
In (Equation 3), Δθ is θ _i −θ (i−1), and Δt is the time until the position detection signal in the 60-degree section is detected. The phase calculation unit 6 uses the position detection value θ _i and the frequency command value ω ₁ ^* to calculate the rotation phase command θc ^* as shown in (Equation 4) and controls the reference phase of the electric motor 1.
[0019]
[Expression 4]

[0020]
The above is the basic operation of the voltage control and phase control in the vector control device of the permanent magnet synchronous motor of the present invention.
[0021]
Next, the configurations of the d-axis current command calculation unit 8 and the q-axis current command calculation unit 9 which are one feature of the present invention will be described. First, the configuration of the d-axis current command calculation unit 8 is shown in FIG. The configuration is obtained by multiplying the deviation between the first d-axis current command value Id ^* and the detected current value Idc given from the host apparatus by the proportional gain Kpd and the integral gain Kid to perform the integral processing. The output signals of the integral calculation unit 8B are added, and the second d-axis current command value Id ^** is output according to (Equation 5).
[0022]
[Equation 5]

[0023]
Next, the configuration of the q-axis current command calculation unit 9 is shown in FIG. The configuration is obtained by multiplying the deviation between the first q-axis current command value Iq ^* and the detected current value Iqc given from the host device by the proportional gain Kpq and the integral gain Kiq for integration processing. The output signals of the integral calculation unit 9B are added, and the second q-axis current command Iq ^** is output according to (Equation 6).
[0024]
[Formula 6]

[0025]
Here, the d-axis current command calculation unit 8 and the q-axis current command calculation unit 9 perform the process of proportional + integral calculation, but the same effect can be obtained only by proportional or integral calculation.
[0026]
Next, one working effect brought about by the present invention will be described with reference to this embodiment.
[0027]
Consider the case where the first d-axis and q-axis current command values Id ^* and Iq ^* are input to the voltage vector calculation unit 10 in the control device of FIG. 1 (second current command values Id ^** and Iq ^**). Is not used in calculations). In the voltage vector calculation unit 10, the voltage command values Vd ^* and Vq ^* are calculated according to (Equation 7).
[0028]
[Expression 7]

[0029]
When a torque control operation is performed by the vector calculation shown in (Expression 7), when a high torque is required, it is necessary to flow a large current commensurate with the torque. When high torque is required for a continuous time, the winding resistance value R inside the motor increases with time due to heat generated by the motor current. Then, since the resistance set value R ^* calculated by the voltage vector calculation unit 10 and the actual resistance value R do not coincide with each other, the necessary voltage cannot be supplied to the permanent magnet synchronous motor 1, and as a result, particularly in the low speed range. The current required for torque generation does not flow and torque shortage occurs.
[0030]
FIG. 9 shows the actual measurement results of the motor torque and the number of rotations when the vector calculation is performed in (Expression 7). The broken line is the torque command value, and the solid line is the measured motor torque value. It can be seen that on the low-speed, high-torque side (A region) surrounded by a circular broken line, torque according to the command value is not generated.
[0031]
Therefore, in the present embodiment, the second current command value is set in the current command calculation units 8 and 9 so that the detected current values Idc and Iqc of the d-axis and the q-axis coincide with the command values given from the host device. Id ^** and Iq ^** are calculated, and the output voltage of the converter is calculated by (Equation 1).
[0032]
As a result, even if R ^* set by the voltage vector calculation unit 10 and the actual resistance value R do not match, the output voltage is controlled so that the motor current matches the current command value. Torque control "can be realized.
[0033]
FIG. 10 shows one actual measurement result when this embodiment is used. The broken lines in FIG. 9 and FIG. 10 indicate the torque command values, and the solid lines indicate the actual torque values obtained by experimenting with the respective torque command values. When FIG. 9 and FIG. 10 are compared, it can be seen that FIG. 10 using the present embodiment generally follows with higher accuracy. In particular, in the low speed region where the torque command value is about 25 [Nm], it is an experimental result using this embodiment so that FIG. 10 can follow the maximum accurately by 8 [Nm]. In FIG. 10, it can be seen that the torque according to the command value is generated on the low-speed, high-torque side.
[0034]
From this, it can be seen that when this embodiment is used, it is possible to follow the entire region with high accuracy, and it is possible to realize a high torque output particularly in the low speed region.
[0035]
<Example 2>
FIG. 4 shows a second embodiment of the present invention. In the second embodiment, the first d-axis and q-axis current command values Id ^* and Iq ^* and the output values Id ^** and Iq ^** of the current command calculation units 8 and 9 are used as the second value. This is a control device for a permanent-magnet electric motor of a type that obtains current command values Id ^*** and Iq ^*** .
[0036]
In FIG. 4, reference numerals 1-11, 21 are the same as those in FIG. 12 is an adder that adds the first d-axis current command value Id ^* and the output value Id ^{** of the} d-axis current command calculation unit 8, and 13 is the first q-axis current command value Iq ^* and the q-axis current value. It is an adder that adds the output value Iq ^** of the current command calculator 9. Using the current command values Id ^*** and Iq ^*** calculated by this method, the voltage command values Vd ^*** and Vq ^*** shown in (Equation 8) are calculated to control the converter output voltage. To do.
[0037]
[Equation 8]

[0038]
In this system, a current command value proportional to torque generation is basically supplied by Id ^* and Iq ^* .
[0039]
Further, even if the motor constant set by the voltage vector calculation unit 10 and the actual value of the motor do not match, the current command calculation units 8 and 9 make the motor current match the current command value (compensating for excess / deficient current) Since the current command value is calculated, highly accurate torque control can be realized in the entire speed range. Considering that Id ^* and Idc, and Iq ^* and Iqc match each other, it is obvious that the operation is the same as in the above-described embodiment and the same effect can be obtained.
[0040]
In addition, when the sampling calculation cycle is long, the control gain cannot be increased and a high response cannot be realized. However, it is possible to improve the responsiveness by performing the feedforward control as in this embodiment. Become.
[0041]
<Example 3>
FIG. 5 shows a third embodiment. In this embodiment, the first d-axis and q-axis current command values Id ^* and Iq ^* are calculated from the primary advance / delay signals, the current command values Id ^* and Iq ^* and the current detection values Idc and Iqc. is the current command value ^{Id **} ', ^Iq **' than the sum of the second current command value ^{Id *** ', Iq} ***' is the control apparatus for a permanent magnet synchronous motor of the type to obtain .
[0042]
In FIG. 5, reference numerals 1-11, 21 are the same as those in FIG. 12 adder for adding the first current command value Id ^* output value Id ^** and the d-axis current command calculating unit 8 of the d-axis, 13 an output value of the current command calculation unit 9 of the q-axis Iq ^{* *} a first current command value Iq ^* and the adder for adding the q-axis, 14 the filter 15 a first-order lag having a gain of delay time constant T1d proceeds primary having delay time constant T1d, the gain of the advance time constant T2d The delay filter 16 is a first-order lag filter having a delay time constant T1q and a first-order lag filter having a delay time constant T1q and a lead time constant T2q. Using the current command values Id ^*** 'and Iq ^*** ' calculated by this method, the voltage command values Vd ^*** 'and Vq ^*** ' shown in (Equation 9) are calculated, and the converter Control the output voltage.
[0043]
[Equation 9]

[0044]
Here, the proportional gain (Kpd, Kpq) and integral gain (Kid, Kiq) of the d-axis and q-axis current command calculation units 8 and 9 are set as in (Equation 10),
[0045]
[Expression 10]

[0046]
Here, ωcd and ωcq are control response angular frequencies [rad / s] of the d-axis and the q-axis. Ld and Lq are the inductance of the motor, and R is the resistance of the motor. In the calculation units 14 to 17, when T1d, T2d, T1q, and T2q are set as in (Equation 11),
[0047]
[Expression 11]

[0048]
Since the current control response from the current command values Id ^* and Iq ^* to the current detection values Idc and Iqc can be defined with a first-order lag as shown in (Equation 12), an overshoot-less torque control system is configured. Is possible.
[0049]
[Expression 12]

[0050]
Even in such a system, considering that Id ^* and Idc, and Iq ^* and Iqc match each other, it is clear that the operation is the same as in the above-described embodiment, and the same effect can be obtained. A less torque control system can be constructed.
[0051]
<Example 4>
In the first to third embodiments, the expensive current detector 3 detects the three-phase alternating currents Iu to Iw. However, the current can be detected without using the current detector. FIG. 6 shows the fourth embodiment. In FIG. 6, 1, 2, 4 to 11, 21 are the same as those in FIG. Reference numeral 17 denotes a current estimation unit that estimates three-phase AC currents Iu, Iv, and Iw that flow to the synchronous motor from a DC current IDC that flows to the input bus (DC shunt resistor) of the power converter.
[0052]
Using the estimated current values Iu ^, Iv ^, Iw ^, the coordinate conversion unit 7 calculates d-axis and q-axis current detection values Idc, Iqc. Even in such a system, since Id ^* and Idc, and Iq ^* and Iqc coincide with each other, it is obvious that the operation is the same as in the above-described embodiment and the same effect can be obtained.
[0053]
In addition, this makes it possible to control the current detector with a small amount in order to obtain Idc and Iqc from a DC shunt resistor incorporated in advance to prevent overcurrent instead of the current detector.
[0054]
<Example 5>
In the fifth embodiment, the control device of FIG. 4 is applied to a control device that performs inexpensive current detection. FIG. 7 shows the configuration of the fifth embodiment. In FIG. 7, 1, 2, 4 to 13, 21 are the same as those in FIG. Reference numeral 17 denotes a current estimation unit that estimates the three-phase AC currents Iu, Iv, and Iw that flow through the synchronous motor from the DC current IDC that flows through the input bus of the power converter. Using the estimated current values Iu ^, Iv ^, Iw ^, the coordinate conversion unit 7 calculates d-axis and q-axis current detection values Idc, Iqc. Even in such a system, since Id ^* and Idc, and Iq ^* and Iqc coincide with each other, it is obvious that the operation is the same as in the above-described embodiment and the same effect can be obtained. In addition, this makes it possible to control the current detector with a small amount in order to obtain Idc and Iqc from a DC shunt resistor incorporated in advance to prevent overcurrent instead of the current detector.
[0055]
<Example 6>
In the sixth embodiment, the control device of FIG. 5 is applied to a control device that performs inexpensive current detection. FIG. 8 shows the configuration of the sixth embodiment. In FIG. 8, 1, 2, 4 to 17, 21 are the same as those in FIG. Reference numeral 17 denotes a current estimation unit that estimates the three-phase AC currents Iu, Iv, and Iw that flow through the synchronous motor from the DC current IDC that flows through the input bus of the power converter. Using the estimated current values Iu ^, Iv ^, Iw ^, the coordinate conversion unit 7 calculates d-axis and q-axis current detection values Idc, Iqc. Even in such a system, since Id ^* and Idc, and Iq ^* and Iqc coincide with each other, it is obvious that the operation is the same as in the above-described embodiment and the same effect can be obtained.
[0056]
In addition, this makes it possible to control the current detector with a small amount in order to obtain Idc and Iqc from a DC shunt resistor incorporated in advance to prevent overcurrent instead of the current detector.
[0057]
<Example 7>
FIG. 11 shows a seventh embodiment of the present invention. The seventh embodiment is a control device for a permanent-magnet electric motor of a type in which a voltage vector calculation is performed using a first current command value on the d-axis side and a second current command value Iq ^** on the q-axis side. In the figure, 1 to 7, 8 to 11 and 21 are the same as those in FIG.
[0058]
Even in this method, if the d-axis current command value is zero (Id ^* = 0), Iq ^* and Iqc coincide with each other. Therefore, it is clear that the operation is the same as in the above embodiment and the same effect can be obtained. is there.
[0059]
<Example 8>
An example in which the present invention is applied to a module will be described with reference to FIG. This example shows an embodiment of the first example. Here, the frequency calculation unit 5, the phase calculation unit 6, the voltage vector calculation unit 10, the d-axis current command calculation unit 8, the q-axis current command calculation unit 9, and the coordinate conversion unit 11 are configured using a one-chip microcomputer. . Further, the one-chip microcomputer and the power converter 2 are housed in one module configured on the same substrate. The module here means “standardized structural unit” and is composed of separable hardware / software components. In addition, although it is preferable that it is comprised on the same board | substrate on manufacture, it is not limited to the same board | substrate. From this, it may be configured on a plurality of circuit boards built in the same housing. In the other embodiments, the same configuration can be adopted.
[0060]
【The invention's effect】
An object of the present invention is to provide a control device for an AC motor that can realize highly accurate torque control.
[Brief description of the drawings]
FIG. 1 is a configuration diagram of a control device for a permanent magnet synchronous motor according to an embodiment of the present invention.
FIG. 2 is an explanatory diagram of a d-axis current command calculation unit 8 in the control device of FIG.
3 is an explanatory diagram of a q-axis current command calculation unit 9 in the control device of FIG. 1. FIG.
FIG. 4 is a configuration diagram of a permanent magnet synchronous motor control apparatus according to another embodiment of the present invention.
FIG. 5 is a configuration diagram of a controller for a permanent magnet synchronous motor showing another embodiment of the present invention.
FIG. 6 is a configuration diagram of a control device for a permanent magnet synchronous motor according to another embodiment of the present invention.
FIG. 7 is a block diagram of a permanent magnet synchronous motor control apparatus according to another embodiment of the present invention.
FIG. 8 is a block diagram of a permanent magnet synchronous motor control apparatus according to another embodiment of the present invention.
FIG. 9 shows motor rotation speed-torque actual measurement characteristics when the present invention is not used.
FIG. 10 shows the actual rotational speed-torque characteristics of a motor when the present invention is used.
FIG. 11 is a block diagram of a permanent magnet synchronous motor control apparatus according to another embodiment of the present invention.
FIG. 12 is an explanatory diagram of an embodiment of a module to which the control device of the present invention is applied.
[Explanation of symbols]
DESCRIPTION OF SYMBOLS 1 ... Permanent magnet synchronous motor, 2 ... Power converter, 3 ... Current detector, 4 ... Magnetic pole position detector, 5 ... Frequency calculating part, 6 ... Phase calculating part, 7 ... Coordinate converting part, 8 ... d-axis current command Calculation unit, 9 ... q-axis current command calculation unit, 10 ... voltage vector calculation unit, 17 ... current estimation unit, 21 ... DC power supply, Id ^* ... first d-axis current command value, Iq ^* ... first q-axis Current command value, Id ^** , Id ^*** , Id ^*** '... Second d-axis current command value, Iq ^** , Iq ^*** , Iq ^*** ' ... Second q-axis current command value.

Claims (17)

In the control device for the AC motor that controls the output voltage of the power converter that drives the AC motor,
Creating a q-axis second current command value of the rotating coordinate system so that the q-axis current detection value of the rotating coordinate system approaches the first current command value of the q-axis of the rotating coordinate system;
A control device for an AC motor, wherein a command voltage for the power converter is created based on a second current command value for the q-axis of the rotating coordinate system. In claim 1,
The command voltage of the power converter is
A control apparatus for an AC motor, which is created based on the second current command value, a motor constant, and a frequency command value of a power converter. In claim 1,
The control apparatus for an AC motor, wherein the current detection value is created based on a DC current detection value of the power converter. In the control device for the AC motor that controls the output voltage of the power converter that drives the AC motor,
A second current command value is created so that the current detection value approaches the first current command value, and based on an addition value of the first current command value and the second current command value, A control device for an AC motor, characterized by creating a command voltage for a power converter. In claim 4,
The command voltage of the power converter is
A control apparatus for an AC motor, which is created based on an addition value of the first current command value and the second current command value, a motor constant, and a frequency command value of a power converter. In claim 5,
The control apparatus for an AC motor, wherein the current detection value is created based on a DC current detection value of the power converter. In the control device for the AC motor that controls the output voltage of the power converter that drives the AC motor,
Creating a second current command value for the d-axis of the rotating coordinate system such that the detected current value for the d-axis of the rotating coordinate system approaches the first current command value for the d-axis of the rotating coordinate system;
Creating a second current command value for the q-axis of the rotating coordinate system such that the detected current value for the q-axis of the rotating coordinate system approaches the first current command value for the q-axis of the rotating coordinate system;
A first addition value of the d-axis first current command value and the d-axis second current command value; the q-axis first current command value; and the q-axis second current command. A control device for an AC motor, wherein a command voltage for the power converter is created based on a second addition value to the value. In claim 7,
In order for the current detection value of the d-axis of the rotating coordinate system to approach the first current command value of the d-axis of the rotating coordinate system,
The d-axis current detection value of the rotating coordinate system approaches the first-order lag signal of the first current command value of the d-axis of the rotating coordinate system.
In order for the detected current value of the q-axis of the rotating coordinate system to approach the first current command value of the q-axis of the rotating coordinate system,
A control apparatus for an AC motor, wherein the detected current value of the q-axis of the rotating coordinate system approaches a first-order lag signal of the first current command value of the q-axis of the rotating coordinate system. In claim 8,
The first order lag time constant required to create the first order lag signal is:
A control device for an AC motor, wherein the control device is determined based on a control response angular frequency or a control gain. In claim 7,
The first addition value is an addition value of a primary advance / delay signal of the first current command value of the d-axis and a second current command value of the d-axis,
The control apparatus for an AC motor, wherein the second addition value is an addition value of a primary advance / delay signal of the q-axis first current command value and the second current command value of the q-axis. . In claim 10,
The primary advance / delay time constant required to create the primary advance / delay signal is:
The primary time constant is determined based on the ratio between the d-axis and q-axis inductance values and the motor resistance value,
The control apparatus for an AC motor, wherein the first-order lag time constant is determined based on a control response angular frequency or a control gain. In claim 7,
In order for the current detection value of the d-axis of the rotating coordinate system to approach the first current command value of the d-axis of the rotating coordinate system,
The d-axis current detection value of the rotating coordinate system approaches the first-order lag signal of the first current command value of the d-axis of the rotating coordinate system.
In order for the detected current value of the q-axis of the rotating coordinate system to approach the first current command value of the q-axis of the rotating coordinate system,
The q axis current detection value of the rotating coordinate system approaches the first order lag signal of the first current command value of the q axis of the rotating coordinate system,
The first addition value is an addition value of a primary advance / delay signal of the first current command value of the d-axis and a second current command value of the d-axis,
The control apparatus for an AC motor, wherein the second addition value is an addition value of a primary advance / delay signal of the q-axis first current command value and the second current command value of the q-axis. . In claim 7,
The command voltage of the power converter is
A control apparatus for an AC motor, which is created based on the first addition value, the second addition value, a motor constant, and a frequency command value of a power converter. In claim 13,
The current detection value is created based on a DC current detection value of the power converter. A module comprising the control device for an AC motor according to claim 1 and the power converter. 5. A module comprising the control device for an AC motor according to claim 4 and the power converter. A module comprising the control device for an AC motor according to claim 7 and the power converter.

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