JP2004301550A

JP2004301550A - Operation device of power related quantity and phase angle

Info

Publication number: JP2004301550A
Application number: JP2003092019A
Authority: JP
Inventors: Atsufumi Kuroda; 淳文黒田; Masaru Shindoi; 賢新土井; Tomohiko Akiyama; 智彦秋山
Original assignee: Mitsubishi Electric Corp
Current assignee: Mitsubishi Electric Corp
Priority date: 2003-03-28
Filing date: 2003-03-28
Publication date: 2004-10-28

Abstract

<P>PROBLEM TO BE SOLVED: To provide a highly-accurate and noise-resistant operation device of a power related quantity and a phase angle, not using a zero cross point. <P>SOLUTION: This device has a constitution equipped with power operation means 17A, 18A, 19A for operating the power (for example, W1) of each phase based on a voltage value (for example, V1) and a current value (for example, I1) of each phase, Hilbert transformation means 15, 16 for inputting the voltage value and the current value, and acquiring a voltage output and a current output having a phase angle of 90° between them, reactive power operation means 17B, 18B, 19B for operating a reactive power (for example, var1) of each phase based on the voltage output and the current output, and a phase angle operation means for operating the phase angle between the voltage and the current of each phase based on an operation result of the power operation means and the reactive power operation means. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

【０００１】
【発明の属する技術分野】
この発明は、電圧電流値の同期検波を利用した電力関連量、即ち、電力、無効電力、電力量、無効電力量およびこれらの高調波成分の少なくとも１つ並びに位相角を演算する電力関連量および位相角演算装置に関するものである。
【０００２】
【従来の技術】
従来の位相角の演算は、所定相の電圧値Ｖ１と、その他の信号、例えば他相の電圧値Ｖ２、Ｖ３または電流値Ｉ１、Ｉ２、Ｉ３との位相角を演算する場合、所定相の電圧値Ｖ１の１周期の時間Ａおよび所定相の電圧値Ｖ１と位相角を演算する他相の電圧または電流信号Ｖ２、Ｖ３、Ｉ１、Ｉ２、Ｉ３との０クロス点または事前に決められた一定レベルでのクロス点の時間のずれＢを検出し、（Ｂ／Ａ）×３６０°として算出していた。（例えば特許文献１参照）。
【０００３】
【特許文献１】
特開平７−１２０５１１号公報（段落０００２、図４）
【特許文献２】
特開平６−３００７９６号公報
【特許文献３】
特許第３２０６２７３号公報
【０００４】
【発明が解決しようとする課題】
従来の位相角の演算装置では、Ａ／Ｄ変換器のサンプリング周波数が例えば４０００Ｈｚの場合、１／４０００秒の精度でしか時間を検出できないため、０クロス点を正確に演算するために直線補間などの処理が必要であった。しかし、直線補間などの処理を行なっても、実際の時間ずれは正確に算出できないため、高精度の位相角算出は不可能であった。また、上記の時間の算出は０クロス点の時間のずれが必要であるため、リアルタイムで位相角を算出することが不可能であるという欠点も生じていた。さらに、０クロス点については、信号に基本波以外のノイズ信号が含まれる場合には、０クロス点が複数点発生することがあるため、正確な演算ができないという問題点があった。
【０００５】
この発明は、上述のような問題点を解決するためになされたもので、第１の目的は、０クロス点を用いずに高精度な位相角を求める電力関連量および位相角演算装置を提供することである。
【０００６】
また、第２の目的は、信号に基本波以外のノイズ信号が含まれる場合でも、ノイズ信号の影響がほとんどない電力関連量および位相角演算装置を提供することである。
【０００７】
さらに、第３の目的は、リアルタイムで演算可能な電力関連量および位相角演算装置を提供することである。
【０００８】
【課題を解決するための手段】
この発明に係る電力関連量および位相角演算装置は、各相の電圧値および電流値にもとづいて各相の電力を演算する電力演算手段、上記電圧値および電流値を入力してそれぞれの間に９０°の位相角を有する電圧出力および電流出力を得るヒルベルト変換手段、上記電圧出力および電流出力にもとづいて各相の無効電力を演算する無効電力演算手段、上記電力演算手段および無効電力演算手段の演算結果にもとづいて各相の電圧、電流間の位相角を演算する位相角演算手段を備えたものである。
【０００９】
【発明の実施の形態】
実施の形態１．
以下、この発明の実施の形態１を図にもとづいて説明する。図１は、実施の形態１による電力関連量演算装置の構成を示すブロック図である。この図において、入力信号である各相の入力電圧をＶｉｎ、各相の入力電流をＩｉｎで示している。
所定相の入力電圧である１側入力電圧Ｖｉｎ−１は、１側電圧センサ１１ＡでＡ／Ｄ変換可能な電圧レベルに降圧し、さらに、１側電圧Ａ／Ｄ変換器１２Ａでデジタル信号である１側電圧値Ｖ１に変換される。同様に、２側入力電圧Ｖｉｎ−２は２側電圧センサ２１Ａ、２側電圧Ａ／Ｄ変換器２２Ａを介して２側電圧値Ｖ２に変換され、３側入力電圧Ｖｉｎ−３は３側電圧センサ３１Ａ、３側電圧Ａ／Ｄ変換器３２Ａを介して３側電圧値Ｖ３に変換される。また、所定相の入力電流である１側入力電流Ｉｉｎ−１は、１側電流センサ１１Ｂで電圧値に変換され、さらに、１側電流Ａ／Ｄ変換器１２Ｂでデジタル信号である１側電流値Ｉ１に変換される。同様に、２側入力電流Ｉｉｎ−２は２側電流センサ２１Ｂ、２側電流Ａ／Ｄ変換器２２Ｂを介して２側電流値Ｉ２に変換され、３側入力電流Ｉｉｎ−３は３側電流センサ３１Ｂ、３側電流Ａ／Ｄ変換器３２Ｂを介して３側電流値Ｉ３に変換される。
【００１０】
また、１側電圧値Ｖ１と２側電圧値Ｖ２は積算器４７Ａに入力され、それぞれの信号が掛け合わされる。掛け合わされた信号は、ローパスフィルタ（以下、ＬＰＦという）４８Ａと加算平均器４９Ａで高周波成分とホワイトノイズの除去を行ない、Ｖ１とＶ２の仮想電力（以下、Ｖ１Ｖ２間同相乗算値という）４１が得られる。同様に、１側電圧値Ｖ１と３側電圧値Ｖ３を積算器５７Ａで掛け合わせ、ＬＰＦ５８Ａ、加算平均器５９Ａを経てＶ１とＶ３の仮想電力（以下、Ｖ１Ｖ３間同相乗算値という）５１が得られる。また、１側電圧値Ｖ１と１側電流値Ｉ１を積算器１７Ａで掛け合わせ、ＬＰＦ１８Ａ、加算平均器１９Ａを経て１側電力値Ｗ１を得、同様に、２側電圧値Ｖ２と２側電流値Ｉ２を積算器２７Ａで掛け合わせて２側電力値Ｗ２を得、３側電圧値Ｖ３と３側電流値Ｉ３を積算器３７Ａで掛け合わせて３側電力値Ｗ３を得ている。
【００１１】
さらに、この実施の形態は２つの入力信号の位相を９０°ずらせるための手段として、ヒルベルト直交相変換器１５、２５、３５とヒルベルト同相変換器１６、２６、３６、４６、５６を有し、ヒルベルト直交相変換器１５を通った１側電圧値Ｖ１とヒルベルト同相変換器４６を通った２側電圧値Ｖ２を積算器４７Ｂに入力し、ここで掛け合わされた信号をＬＰＦ４８Ｂと加算平均器４９Ｂでノイズ除去することにより、Ｖ１とＶ２の仮想無効電力（以下、Ｖ１Ｖ２間直交相乗算値という）４２を得ている。
同様に、ヒルベルト直交相変換器１５を通った１側電圧値Ｖ１とヒルベルト同相変換器５６を通った３側電圧値Ｖ３を積算器５７Ｂに入力し、ＬＰＦ５８Ｂ、加算平均器５９Ｂを経てＶ１とＶ３の仮想無効電力（以下、Ｖ１Ｖ３間直交相乗算値という）５２を得ている。また、ヒルベルト直交相変換器１５を通った１側電圧値Ｖ１とヒルベルト同相変換器１６を通った１側電流値Ｉ１を積算器１７Ｂに入力し、ＬＰＦ１８Ｂ、加算平均器１９Ｂを経て１側無効電力値ｖａｒ１を得、ヒルベルト直交相変換器２５を通った２側電圧値Ｖ２とヒルベルト同相変換器２６を通った２側電流値Ｉ２を積算器２７Ｂに入力し、ＬＰＦ２８Ｂ、加算平均器２９Ｂを経て２側無効電力値ｖａｒ２を得、ヒルベルト直交相変換器３５を通った３側電圧値Ｖ３とヒルベルト同相変換器３６を通った３側電流値Ｉ３を積算器３７Ｂに入力し、ＬＰＦ３８Ｂ、加算平均器３９Ｂを経て３側無効電力値ｖａｒ３を得ている。
この実施の形態では、説明をわかりやすくするため、２側電圧値Ｖ２および３側電圧値Ｖ３にヒルベルト同相変換器４６，５６を通す例について説明したが、１側電圧値Ｖ１に図示しないヒルベルト同相変換器を通し、２側電圧値Ｖ２および３側電圧値Ｖ３は無効電力演算のために通すヒルベルト直交相変換器２５，３５の結果を用いれば、同様の機能をさらに演算量を削減して実現できる。
【００１２】
この電力関連量演算装置においては、ホワイトノイズは、ＬＰＦ１８Ａ〜５８Ｂと加算平均器１９Ａ〜５９Ｂを通すことにより除去することができ、また、ｎ次高周波は、電流にのりやすく電圧にのりにくい性質であるため、積算器１７Ａ〜５７Ｂにおいて電圧値Ｖｎと電流値Ｉｎ（仮想のときは、電圧値Ｖｍ）の積をとることによってカットすることができ、ノイズを含んだ信号に対しても高精度な電力（仮想電力も含む）および無効電力（仮想無効電力も含む）の演算が可能である。
【００１３】
次に、位相角を求める手順について説明する。図２は、電圧、電流の位相角を説明するためのベクトル図、図３は、電力と無効電力から位相角を求める場合の説明図、図４は、仮想電力と仮想無効電力から位相角を求める場合の説明図である。
【００１４】
所定相の電圧値Ｖ１とその他の信号、例えば他相の電圧値Ｖ２、Ｖ３または電流値Ｉ１、Ｉ２、Ｉ３との位相角を演算する場合には、上述した電力関連量演算装置で算出ずみの各相の電力Ｗｎおよび無効電力ｖａｒｎを利用して、図２に示すＶ１とＩ１間の位相角、Ｖ２とＩ２間の位相角、Ｖ３とＩ３間の位相角を算出する。
この場合、電力Ｗｎおよび無効電力ｖａｒｎを（式１）に代入することにより、図３に示すように、Ｖｎを基準としたＶｎＩｎ間の位相角θを求めることができる。
しかし、（式１）で求めたＶｎＩｎ間の位相角θは、０°≦θ≦１８０°の範囲に限られるので、位相角θが１８０°＜θ＜３６０°、つまりｖａｒｎ＜０の場合には、（式１）による演算結果を３６０°から減算する形で位相角を求める。
【００１５】
【数４】

【００１６】
また、Ｖ１とＶ２間の位相角およびＶ１とＶ３間の位相角は、上述した電力関連量演算装置にて算出したＶ１Ｖ２間同相乗算値、Ｖ１Ｖ２間直交相乗算値、Ｖ１Ｖ３間同相乗算値、Ｖ１Ｖ３間直交相乗算値を用いて位相角を演算する。即ち、Ｖ１Ｖｎ間同相乗算値、Ｖ１Ｖｎ間直交相乗算値を（式２）に代入することにより、図４に示すように、Ｖ１を基準としたＶ１Ｖｎ間位相角θを求めることができる。しかし、（式２）で求めたＶ１Ｖｎ間の位相角θは、０°≦θ≦１８０°の範囲に限られるので、位相角θが１８０°＜θ＜３６０°、つまりｖａｒｎ＜０の場合には、（式２）による演算結果を３６０°から減算する形で位相角を求める。
【００１７】
【数５】

【００１８】
なお、図２は、三相４線式の場合の入力電圧Ｖｎおよび入力電流Ｉｎの関係を示すものであるため、Ｖ１Ｖ２間位相角、Ｖ１Ｖ３間位相角およびＶ１Ｉ１間位相角は、（式２）および（式１）で求めた位相角をそのまま利用することができる。また、Ｖ１Ｉ２間位相角は、Ｖ１Ｖ２間位相角とＶ２Ｉ２間位相角を加算することで求めることができ、Ｖ１Ｉ３間位相角は、Ｖ１Ｖ３間位相角とＶ３Ｉ３間位相角を減算することで求めることができる。さらに、位相角が３６０°を超えた場合は、３６０°を減算することで、出力範囲を０〜３６０°とすることができる。
【００１９】
実施の形態２．
次に、この発明の実施の形態２について説明する。電圧値Ｖ１、Ｖ２、Ｖ３と電流値Ｉ１、Ｉ２、Ｉ３はセンサ、例えば電圧トランスとＣＴなどの種類および基板のアナログ回路特性により、実際の信号とＡ／Ｄ変換器で取得する信号に位相ずれが発生することがあるため、この実施の形態は以下の処理をすることにより上述の位相ずれを調整するようにしたものである。
【００２０】
例えば、１側電力Ｗ１を求める過程でＬＰＦ１８Ａから出力された値をＷ１ｉｎ、加算平均器１９Ａに入力する値をＷ１ｏｕｔとし、１側無効電力ｖａｒ１を求める過程でＬＰＦ１８Ｂから出力された値をｖａｒ１ｉｎ、加算平均器１９Ｂに入力する値をｖａｒ１ｏｕｔとし、ＬＰＦ１８Ａ、１８Ｂと加算平均器１９Ａ、１９Ｂの間で
【数６】

の処理を行なうことにより位相ずれを調整することができる。φは位相ずれ角であるが、センサの種類や基板のアナログ特性にもとづく位相ずれ角φを初期設定時に予め求めておき、初期位相ずれ角φとして保持しておけば容易に演算を行なうことができる。他の各相についても同様の処理を行なうことにより、各相ごとに位相ずれを調整することができる。また、Ｖ１Ｖｎ間同相乗算値とＶ１Ｖｎ間直交相乗算値についても同様の処理を行なうことができる。
なお、各電圧値の位相については、同一のセンサ、例えば抵抗分圧回路などを用いる場合、位相ずれはほとんど生じないため問題はない。
【００２１】
実施の形態３．
次に、この発明の実施の形態３について説明する。この実施の形態は、実施の形態１で求めた位相角の精度を上げるために、求める位相角の角度によって演算式を変えるようにしたものである。即ち、図３、図４において、位相角θが４５°≦θ≦１３５°および２２５°≦θ≦３１５°のいずれかの領域にある場合、つまり｜Ｗｎ｜≦｜ｖａｒｎ｜の場合には、位相角θが微小に変化したとき、Ｗｎの変化量の方がｖａｒｎの変化量より大きいため、
【数７】

で求めた方が
【数８】

で求めるよりも精度がよくなる。
【００２２】
逆に、位相角θが０°＜θ＜４５°、１３５°＜θ＜２２５°および３１５°＜θ＜３６０°のいずれかの領域にある場合、つまり｜Ｗｎ｜＞｜ｖａｒｎ｜の場合には、位相角θが微小に変化したとき、ｖａｒｎの変化量の方がＷｎの変化量より大きいため、
【数９】

で求めた方が
【数１０】

で求めるよりも精度がよくなる。
【００２３】
ここで、
【数１１】

によって求めた位相角θは、０°≦θ≦１８０°の範囲に限られるので、位相角θが１８０°＜θ＜３６０°、つまりｖａｒｎ（またはＶ１Ｖｎ間直交相乗算値）＜０の場合には、３６０°から演算結果を減算して位相角を求める。また、
【数１２】

によって求めた位相角θは、−９０°≦θ≦９０°の範囲に限られるので、位相角θが９０°＜θ＜２７０°、つまりＷｎ（またはＶ１Ｖｎ間同相乗算値）＜０の場合には、演算結果に１８０°を加算して位相角を求め、さらに、位相角θが２７０°≦θ＜３６０°、つまりＷｎ（またはＶ１Ｖｎ間同相乗算値）≧０で演算結果が負の場合には、演算結果に３６０°を加算して位相角を求める。
【００２４】
よって、ＶｎＩｎ間の位相角θは、（式３）、（式４）によって求めることができる。
【数１３】

【００２５】
なお、（式３）の演算結果は表１の項目ｂとｃ、（式４）の演算結果は表１の項目ａとｄのように操作して補正される。
【表１】

【００２６】
また、Ｖ１Ｖｎ間位相角θは、上述したＶｎＩｎ間位相角と同様に（式５）、（式６）を用いて求めることができる。
【数１４】

【００２７】
なお、（式５）の演算結果は表２の項目ｆとｇ、（式６）の演算結果は表２の項目ｅとｈのように操作して補正される。この表で「同相」は同相乗算値を、「直交相」は直交相乗算値をそれぞれ示す。
【表２】

【００２８】
また、図２は、三相４線式の場合の入力電圧Ｖｎおよび入力電流Ｉｎの関係を示すものであるため、Ｖ１Ｖ２間位相角、Ｖ１Ｖ３間位相角およびＶ１Ｉ１間位相角は、（式２）および（式１）で求めた位相角をそのまま利用することができる。また、Ｖ１Ｉ２間位相角は、Ｖ１Ｖ２間位相角とＶ２Ｉ２間位相角を加算することで求めることができ、Ｖ１Ｉ３間位相角は、Ｖ１Ｖ３間位相角とＶ３Ｉ３間位相角を減算することで求めることができる。さらに、位相角が３６０°を超えた場合は、３６０°を減算することで、出力範囲を０〜３６０°とすることができる。
【００２９】
実施の形態４．
次に、この発明の実施の形態４を図にもとづいて説明する。図５は、実施の形態４による電力関連量演算装置の構成を示すブロック図である。この図において、図１と同一または相当部分には同一符号を付して説明を省略する。図１と異なる点は、図１のデジタル信号（１側電圧値Ｖ１、２側電圧値Ｖ２、３側電圧値Ｖ３、１側電流値Ｉ１、２側電流値Ｉ２、３側電流値Ｉ３）に対し、８点移動平均器１３Ａ、１３Ｂ、２３Ａ、２３Ｂ、３３Ａ、３３Ｂ、で８点の移動平均をとり、その信号を１／４リサンプリング１４Ａ、１４Ｂ、２４Ａ、２４Ｂ、３４Ａ、３４Ｂに入力することにより、入力信号を４回に１回しか通さず、電力（仮想電力を含む）および無効電力（仮想無効電力を含む）の演算量を１／４に間引いている点である。
【００３０】
実施の形態１では、全てのサンプリング点で位相角演算を行なっているのに対し、実施の形態４では上述のように、サンプリング間隔を間引くことにより、演算量を削減することができ、位相角演算にかかる負荷を減らすことができる。
【００３１】
【発明の効果】
この発明に係る電力関連量および位相角演算装置は、以上のように構成されているため、０クロス点を用いず各相の電力および無効電力とその演算ルーチンを利用して各相の電圧値から算出した仮想電力および仮想無効電力を利用し高精度で位相角を求めることができる。また、信号に基本波以外のノイズ信号が含まれる場合でも、ノイズ信号の影響をほとんど受けることなく位相角の演算ができる。
【図面の簡単な説明】
【図１】この発明の実施の形態１の構成を示すブロック図である。
【図２】この発明の実施の形態１における電圧間および電圧電流間の位相差の関係を示すベクトル図である。
【図３】この発明の実施の形態１において、電力と無効電力から位相角を求める場合の説明図である。
【図４】この発明の実施の形態１において、仮想電力と仮想無効電力から位相角を求める場合の説明図である。
【図５】この発明の実施の形態４の構成を示すブロック図である。
【符号の説明】
１１Ａ，１１Ｂ，２１Ａ，２１Ｂ，３１Ａ，３１Ｂ電圧センサ，
１２Ａ，１２Ｂ，２２Ａ，２２Ｂ，３２Ａ，３２Ｂ電圧Ａ／Ｄ変換器，
１３Ａ，１３Ｂ，２３Ａ，２３Ｂ，３３Ａ，３３Ｂ８点移動平均器，
１４Ａ，１４Ｂ，２４Ａ，２４Ｂ，３４Ａ，３４Ｂ１／４リサンプリング，
１５，２５，３５ヒルベルト直交相変換器，
１６，２６，３６，４６，５６ヒルベルト同相変換器，
１７Ａ，１７Ｂ，２７Ａ，２７Ｂ，３７Ａ，３７Ｂ，４７Ａ，４７Ｂ，５７Ａ，５７Ｂ積算器，
１８Ａ，１８Ｂ，２８Ａ，２８Ｂ，３８Ａ，３８Ｂ，４８Ａ，４８Ｂ，５８Ａ，５８Ｂローパスフィルタ，
１９Ａ，１９Ｂ，２９Ａ，２９Ｂ，３９Ａ，３９Ｂ，４９Ａ，４９Ｂ，５９Ａ，５９Ｂ加算平均器，
４１Ｖ１Ｖ２間同相乗算値，４２Ｖ１Ｖ２間直交相乗算値，
５１Ｖ１Ｖ３間同相乗算値，５２Ｖ１Ｖ３間直交相乗算値。[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to a power-related amount utilizing synchronous detection of a voltage / current value, that is, a power-related amount for calculating at least one of power, reactive power, power amount, reactive power amount and harmonic components thereof and a phase angle, and The present invention relates to a phase angle calculation device.
[0002]
[Prior art]
Conventional phase angle calculation involves calculating the phase angle between a predetermined phase voltage value V1 and another signal, for example, a voltage value V2, V3 or a current value I1, I2, I3 of another phase. The time A of one cycle of the value V1, the voltage value V1 of the predetermined phase, and the zero cross point of the voltage or current signal V2, V3, I1, I2, I3 of the other phase for calculating the phase angle or a predetermined fixed level , The time lag B of the cross point at the time is detected and calculated as (B / A) × 360 °. (See, for example, Patent Document 1).
[0003]
[Patent Document 1]
JP-A-7-120511 (paragraph 0002, FIG. 4)
[Patent Document 2]
JP-A-6-300796 [Patent Document 3]
Japanese Patent No. 3206273
[Problems to be solved by the invention]
In a conventional phase angle calculation device, when the sampling frequency of the A / D converter is, for example, 4000 Hz, time can be detected only with an accuracy of 1/4000 second. Was required. However, even if processing such as linear interpolation is performed, the actual time lag cannot be calculated accurately, so that it is impossible to calculate the phase angle with high accuracy. Further, since the above calculation of the time requires a time lag of the zero cross point, there is a disadvantage that it is impossible to calculate the phase angle in real time. In addition, when the signal includes a noise signal other than the fundamental wave, a plurality of zero cross points may occur at the zero cross point, so that there is a problem that accurate calculation cannot be performed.
[0005]
SUMMARY OF THE INVENTION The present invention has been made to solve the above-described problems, and a first object of the present invention is to provide a power-related quantity and a phase angle calculation device for obtaining a highly accurate phase angle without using a zero cross point. It is to be.
[0006]
It is a second object of the present invention to provide a power-related amount and phase angle calculation device that is hardly affected by a noise signal even when the signal includes a noise signal other than the fundamental wave.
[0007]
Further, a third object is to provide a power-related amount and phase angle calculation device that can be calculated in real time.
[0008]
[Means for Solving the Problems]
A power-related quantity and phase angle calculation device according to the present invention includes: a power calculation means for calculating power of each phase based on a voltage value and a current value of each phase; A Hilbert transforming means for obtaining a voltage output and a current output having a phase angle of 90 °, a reactive power calculating means for calculating the reactive power of each phase based on the voltage output and the current output, a power calculating means and a reactive power calculating means. A phase angle calculating means for calculating a phase angle between a voltage and a current of each phase based on the calculation result is provided.
[0009]
BEST MODE FOR CARRYING OUT THE INVENTION
Embodiment 1 FIG.
Hereinafter, a first embodiment of the present invention will be described with reference to the drawings. FIG. 1 is a block diagram illustrating a configuration of a power-related amount calculation device according to the first embodiment. In this figure, the input voltage of each phase as an input signal is indicated by Vin, and the input current of each phase is indicated by Iin.
The one-side input voltage Vin-1, which is an input voltage of a predetermined phase, is reduced to a voltage level that can be A / D converted by the one-side voltage sensor 11A, and is a digital signal by the one-side voltage A / D converter 12A. It is converted to the one-side voltage value V1. Similarly, the second-side input voltage Vin-2 is converted into the second-side voltage value V2 via the second-side voltage sensor 21A and the second-side voltage A / D converter 22A, and the third-side input voltage Vin-3 is converted to the third-side voltage sensor. The voltage is converted to a third-side voltage value V3 via a third-side voltage A / D converter 32A. The one-side input current Iin-1 which is an input current of a predetermined phase is converted to a voltage value by the one-side current sensor 11B, and further converted to a one-side current value which is a digital signal by the one-side current A / D converter 12B. Converted to I1. Similarly, the second-side input current Iin-2 is converted to a second-side current value I2 via a second-side current sensor 21B and a second-side current A / D converter 22B, and the third-side input current Iin-3 is converted to a third-side current sensor. 31B is converted to a third-side current value I3 via a three-side current A / D converter 32B.
[0010]
The first-side voltage value V1 and the second-side voltage value V2 are input to an integrator 47A, and the respective signals are multiplied. The multiplied signal is subjected to removal of high-frequency components and white noise by a low-pass filter (hereinafter, referred to as LPF) 48A and an averaging device 49A, and a virtual power of V1 and V2 (hereinafter, referred to as an in-phase multiplication value between V1 and V2) 41. can get. Similarly, the first-side voltage value V1 and the third-side voltage value V3 are multiplied by an integrator 57A, and a virtual power (hereinafter, referred to as an in-phase multiplication value between V1 and V3) 51 of V1 and V3 is obtained through an LPF 58A and an averaging device 59A. Can be The 1-side voltage value V1 and the 1-side current value I1 are multiplied by an integrator 17A to obtain a 1-side power value W1 through an LPF 18A and an averaging device 19A. I2 is multiplied by an integrator 27A to obtain a second-side power value W2, and a third-side voltage value V3 and a third-side current value I3 are multiplied by an integrator 37A to obtain a third-side power value W3.
[0011]
Furthermore, this embodiment has Hilbert

quadrature converters

15, 25, 35 and Hilbert in-

phase converters

16, 26, 36, 46, 56 as means for shifting the phases of the two input signals by 90 °. , The one-side voltage value V1 passed through the Hilbert quadrature converter 15 and the second-side voltage value V2 passed through the Hilbert in-phase converter 46 are input to an integrator 47B, and the signal obtained by multiplication is input to an LPF 48B and an averaging device 49B. , A virtual reactive power (hereinafter referred to as a V1V2 quadrature multiplication value) 42 of V1 and V2 is obtained.
Similarly, the one-side voltage value V1 passing through the Hilbert quadrature phase converter 15 and the three-side voltage value V3 passing through the Hilbert in-phase converter 56 are input to the integrator 57B, and then V1 and V3 via the LPF 58B and the averaging device 59B. (Hereinafter, referred to as V1V3 quadrature phase multiplication value) 52 is obtained. Also, the one-side voltage value V1 passed through the Hilbert quadrature converter 15 and the one-side current value I1 passed through the Hilbert in-phase converter 16 are input to the integrator 17B, and are passed through the LPF 18B and the averaging unit 19B to the one-side reactive power. A value var1 is obtained, and a two-side voltage value V2 passing through the Hilbert quadrature converter 25 and a two-side current value I2 passing through the Hilbert in-phase converter 26 are input to an integrator 27B, and are passed through an LPF 28B and an averaging unit 29B. The side reactive power value var2 is obtained, and the three-side voltage value V3 passed through the Hilbert quadrature phase converter 35 and the three-side current value I3 passed through the Hilbert in-phase converter 36 are input to the integrator 37B, and the LPF 38B and the averaging device 39B , The three-side reactive power value var3 is obtained.
In this embodiment, an example has been described in which the Hilbert in-

phase converters

46 and 56 are passed through the second-side voltage value V2 and the third-side voltage value V3 for easy understanding. By using the results of the Hilbert quadrature converters 25 and 35, which pass through the converter for the second-side voltage value V2 and the third-side voltage value V3 for the calculation of the reactive power, the same function is realized by further reducing the calculation amount. it can.
[0012]
In this power-related amount calculation device, white noise can be removed by passing through LPFs 18A to 58B and averaging devices 19A to 59B, and the n-th high frequency has a property that it is easy to carry current and difficult to carry voltage. Therefore, it can be cut by taking the product of the voltage value Vn and the current value In (or the voltage value Vm in the virtual case) in the integrators 17A to 57B. Calculation of power (including virtual power) and reactive power (including virtual reactive power) is possible.
[0013]
Next, a procedure for obtaining the phase angle will be described. FIG. 2 is a vector diagram for explaining the phase angles of voltage and current, FIG. 3 is an explanatory diagram for obtaining a phase angle from power and reactive power, and FIG. 4 is a diagram showing a phase angle from virtual power and virtual reactive power. FIG. 9 is an explanatory diagram in the case of obtaining.
[0014]
When calculating the phase angle between the voltage value V1 of the predetermined phase and other signals, for example, the voltage values V2, V3 or the current values I1, I2, I3 of the other phases, the above-described power-related amount calculation device has already calculated the phase angle. The phase angle between V1 and I1, the phase angle between V2 and I2, and the phase angle between V3 and I3 shown in FIG. 2 are calculated using the power Wn and the reactive power varn of each phase.
In this case, by substituting the power Wn and the reactive power var into (Equation 1), the phase angle θ between VnIn with reference to Vn can be obtained as shown in FIG.
However, since the phase angle θ between VnIn obtained by (Equation 1) is limited to the range of 0 ° ≦ θ ≦ 180 °, when the phase angle θ is 180 ° <θ <360 °, that is, when varn <0, Finds the phase angle by subtracting the result of the operation according to (Equation 1) from 360 °.
[0015]
(Equation 4)

[0016]
The phase angle between V1 and V2 and the phase angle between V1 and V3 are the in-phase multiplied value between V1V2, the quadrature-phase multiplied value between V1V2, and the in-phase multiplied value between V1V3 calculated by the above-described power-related amount calculating device. , V1V3, the phase angle is calculated using the multiplication value. That is, by substituting the in-phase multiplication value between V1Vn and the quadrature phase multiplication value between V1Vn into (Equation 2), the phase angle θ between V1Vn with reference to V1 can be obtained as shown in FIG. However, since the phase angle θ between V1Vn obtained by (Equation 2) is limited to the range of 0 ° ≦ θ ≦ 180 °, when the phase angle θ is 180 ° <θ <360 °, ie, when varn <0, Calculates the phase angle by subtracting the result of the operation according to (Equation 2) from 360 °.
[0017]
(Equation 5)

[0018]
Since FIG. 2 shows the relationship between the input voltage Vn and the input current In in the case of the three-phase four-wire system, the phase angle between V1V2, the phase angle between V1V3, and the phase angle between V1I1 are represented by (Equation 2). And the phase angle obtained by (Equation 1) can be used as it is. Further, the phase angle between V1I2 can be obtained by adding the phase angle between V1V2 and the phase angle between V2I2, and the phase angle between V1I3 can be obtained by subtracting the phase angle between V1V3 and the phase angle between V3I3. it can. Further, when the phase angle exceeds 360 °, the output range can be set to 0 to 360 ° by subtracting 360 °.
[0019]
Embodiment 2 FIG.
Next, a second embodiment of the present invention will be described. The voltage values V1, V2, V3 and the current values I1, I2, I3 are out of phase between the actual signal and the signal obtained by the A / D converter due to the type of sensor, for example, voltage transformer and CT, and the analog circuit characteristics of the board. Therefore, in this embodiment, the above-described phase shift is adjusted by performing the following processing.
[0020]
For example, the value output from the LPF 18A in the process of obtaining the one-side power W1 is W1in, the value input to the averaging unit 19A is W1out, and the value output from the LPF 18B in the process of obtaining the one-side reactive power var1 is var1in. The value input to the averager 19B is var1out, and the value between the LPFs 18A and 18B and the averagers 19A and 19B is given by

By performing the above processing, the phase shift can be adjusted. φ is the phase shift angle, but if the phase shift angle φ based on the type of sensor and the analog characteristics of the board is determined in advance at the time of initial setting, and it is retained as the initial phase shift angle φ, the calculation can be easily performed. it can. By performing the same processing for the other phases, the phase shift can be adjusted for each phase. Similar processing can be performed on the in-phase multiplication value between V1Vn and the quadrature-phase multiplication value between V1Vn.
When the same sensor, for example, a resistance voltage dividing circuit, is used, there is no problem about the phase of each voltage value because there is almost no phase shift.
[0021]
Embodiment 3 FIG.
Next, a third embodiment of the present invention will be described. In this embodiment, in order to improve the accuracy of the phase angle obtained in the first embodiment, the arithmetic expression is changed according to the angle of the obtained phase angle. That is, in FIGS. 3 and 4, when the phase angle θ is in any of the regions of 45 ° ≦ θ ≦ 135 ° and 225 ° ≦ θ ≦ 315 °, that is, when | Wn | ≦ | varn | When the phase angle θ slightly changes, the change amount of Wn is larger than the change amount of varn.
(Equation 7)

The one obtained with [Expression 8]

The accuracy is better than that obtained by
[0022]
Conversely, when the phase angle θ is in any of the regions of 0 ° <θ <45 °, 135 ° <θ <225 °, and 315 ° <θ <360 °, that is, when | Wn |> | varn | Is that when the phase angle θ is slightly changed, the change amount of var is larger than the change amount of Wn.
(Equation 9)

The one obtained in [Expression 10]

The accuracy is better than that obtained by
[0023]
here,
[Equation 11]

Is limited to the range of 0 ° ≦ θ ≦ 180 °, the phase angle θ is 180 ° <θ <360 °, that is, when varn (or the quadrature multiplication value between V1Vn) <0 Calculates the phase angle by subtracting the operation result from 360 °. Also,
(Equation 12)

Is limited to the range of -90 ° ≦ θ ≦ 90 °, the phase angle θ is 90 ° <θ <270 °, that is, when Wn (or the in-phase multiplication value between V1Vn) <0 , 180 ° is added to the operation result to obtain a phase angle. Further, if the phase angle θ is 270 ° ≦ θ <360 °, that is, Wn (or the in-phase multiplication value between V1Vn) ≧ 0, the operation result is negative. In this case, 360 ° is added to the calculation result to determine the phase angle.
[0024]
Therefore, the phase angle θ between VnIn can be obtained by (Equation 3) and (Equation 4).
(Equation 13)

[0025]
The calculation result of (Equation 3) is corrected by operating as shown in items b and c of Table 1, and the calculation result of (Equation 4) is corrected by operating as shown in items a and d of Table 1.
[Table 1]

[0026]
Further, the phase angle θ between V1 and Vn can be obtained by using (Equation 5) and (Equation 6) in the same manner as the above-described phase angle between VnIn.
[Equation 14]

[0027]
The calculation result of (Equation 5) is corrected by operating as shown in items f and g of Table 2, and the calculation result of (Equation 6) is corrected by operating as shown in items e and h of Table 2. In this table, “in-phase” indicates an in-phase multiplication value, and “quadrature” indicates a quadrature-phase multiplication value.
[Table 2]

[0028]
FIG. 2 shows the relationship between the input voltage Vn and the input current In in the case of the three-phase four-wire system. Therefore, the phase angle between V1V2, the phase angle between V1V3, and the phase angle between V1I1 are represented by (Equation 2). And the phase angle obtained by (Equation 1) can be used as it is. Further, the phase angle between V1I2 can be obtained by adding the phase angle between V1V2 and the phase angle between V2I2, and the phase angle between V1I3 can be obtained by subtracting the phase angle between V1V3 and the phase angle between V3I3. it can. Further, when the phase angle exceeds 360 °, the output range can be set to 0 to 360 ° by subtracting 360 °.
[0029]
Embodiment 4 FIG.
Next, a fourth embodiment of the present invention will be described with reference to the drawings. FIG. 5 is a block diagram showing a configuration of a power-related amount calculating device according to the fourth embodiment. In this figure, the same or corresponding parts as those in FIG. The difference from FIG. 1 is that the digital signal (1 side voltage value V1, 2 side voltage value V2, 3 side voltage value V3, 1 side current value I1, 2 side current value I2, 3 side current value I3) of FIG. On the other hand, an 8-

point moving averager

13A, 13B, 23A, 23B, 33A, 33B takes a moving average of 8 points and inputs the signal to 1/4

resampling

14A, 14B, 24A, 24B, 34A, 34B. Thus, the input signal is passed only once in four times, and the amount of calculation of power (including virtual power) and reactive power (including virtual reactive power) is reduced to 1/4.
[0030]
In the first embodiment, the phase angle calculation is performed at all sampling points, whereas in the fourth embodiment, as described above, the amount of calculation can be reduced by thinning out the sampling interval, and the phase angle can be reduced. The load on the calculation can be reduced.
[0031]
【The invention's effect】
Since the power-related quantity and phase angle calculation device according to the present invention is configured as described above, the power and reactive power of each phase and the voltage value of each phase are calculated using the calculation routine without using the zero cross point. The phase angle can be obtained with high accuracy using the virtual power and the virtual reactive power calculated from. Further, even when a signal includes a noise signal other than the fundamental wave, the phase angle can be calculated with little influence of the noise signal.
[Brief description of the drawings]
FIG. 1 is a block diagram showing a configuration of a first embodiment of the present invention.
FIG. 2 is a vector diagram showing a relationship between phase differences between voltages and between currents and voltages in the first embodiment of the present invention.
FIG. 3 is an explanatory diagram in a case where a phase angle is obtained from power and reactive power in the first embodiment of the present invention.
FIG. 4 is an explanatory diagram in a case where a phase angle is obtained from virtual power and virtual reactive power in the first embodiment of the present invention.
FIG. 5 is a block diagram showing a configuration of a fourth embodiment of the present invention.
[Explanation of symbols]
11A, 11B, 21A, 21B, 31A, 31B Voltage sensor,
12A, 12B, 22A, 22B, 32A, 32B voltage A / D converter,
13A, 13B, 23A, 23B, 33A, 33B 8-point moving averager,
14A, 14B, 24A, 24B, 34A, 34B 1/4 resampling,
15, 25, 35 Hilbert quadrature phase converter,
16, 26, 36, 46, 56 Hilbert in-phase converter,
17A, 17B, 27A, 27B, 37A, 37B, 47A, 47B, 57A, 57B
18A, 18B, 28A, 28B, 38A, 38B, 48A, 48B, 58A, 58B low-pass filter,
19A, 19B, 29A, 29B, 39A, 39B, 49A, 49B, 59A, 59B arithmetic averager,
41 V1V2 in-phase multiplication value, 42 V1V2 quadrature-phase multiplication value,
51 In-phase multiplication value between V1V3, 52 Quadrature-phase multiplication value between V1V3.

Claims

Power calculating means for calculating the power of each phase based on the voltage and current values of each phase, inputting the voltage and current values to obtain a voltage output and a current output having a 90 ° phase angle between each; Hilbert transforming means, reactive power calculating means for calculating reactive power of each phase based on the voltage output and current output, phase between voltage and current of each phase based on calculation results of the power calculating means and reactive power calculating means A power-related quantity and phase angle calculation device comprising phase angle calculation means for calculating an angle.

Virtual power calculating means for calculating a virtual power, which is a common-mode multiplication value between the two voltages, based on the voltage values of the predetermined phase and the other phases; a voltage having a phase angle of 90 ° between the two input voltage values; Hilbert transforming means for obtaining an output, virtual reactive power calculating means for calculating virtual reactive power which is a quadrature multiplication value between the two voltages based on the two voltage outputs, calculation results of the virtual power calculating means and virtual reactive power calculating means. A power-related quantity and phase angle calculation device comprising a phase angle calculation means for calculating a phase angle between both voltages based on the calculated value.

When the power (including virtual power) is Wn and the reactive power (including virtual reactive power) is var,

The power-related quantity and phase angle calculation device according to claim 1 or 2, wherein the calculation is performed by:

A phase angle between a voltage of a predetermined phase and a current of another phase (a phase angle between V1I2 and a phase angle between V1I3) is obtained by using a phase angle between a voltage and a current of the same phase and a phase angle between the voltages. The power-related quantity and phase angle calculation device according to claim 3.

4. The power-related quantity and phase angle calculation device according to claim 3, wherein a phase shift angle based on analog characteristics is calculated and corrected in advance when calculating the phase angle.

When the phase angle θ is 45 ° ≦ θ ≦ 135 ° or 225 ° ≦ θ ≦ 315 °,

And for other phase angles,

The power-related quantity and phase angle calculation device according to claim 3, wherein the calculation is performed by:

The amount of calculation of power (including virtual power) and reactive power (including virtual reactive power) is reduced by thinning out the sampling interval of the voltage value and current value of each phase or the voltage value of a predetermined phase and the voltage value of another phase. The power-related quantity and phase angle calculation device according to claim 1 or 2, wherein: