JP2004219288A

JP2004219288A - Device for measuring power-related amount

Info

Publication number: JP2004219288A
Application number: JP2003007778A
Authority: JP
Inventors: Atsufumi Kuroda; 淳文黒田; Masaru Shindoi; 賢新土井; Yoshikuni Kondou; 桂州近藤
Original assignee: Mitsubishi Electric Corp
Current assignee: Mitsubishi Electric Corp
Priority date: 2003-01-16
Filing date: 2003-01-16
Publication date: 2004-08-05
Anticipated expiration: 2023-01-16
Also published as: JP4310113B2; CN1249443C; AU2003204811B2; CN1517715A; AU2003204811A1

Abstract

<P>PROBLEM TO BE SOLVED: To solve the problem wherein the calculation for power related amounts has errors, if the voltage signal input, current signal input, and A/D converter are influenced by the magnitudes of voltage and current to be detected, the frequency of the voltage and current to be detected, and an operating temperature. <P>SOLUTION: This power-related amount measuring device is set with primary straight lines, beforehand; changes in the characteristics of a voltage signal input; a current signal input whose parameters are the effective value of voltage Vrms_1; the effective value of current Irms_1, and the power supply frequency, and changes in the characteristic of the A/D converter, whose parameter is its temperature. The device is so formed as to correct the changes in the characteristics, by updating the coefficients of an amplitude phase correction matrix being used at regular operations, at a fixed time intervals. <P>COPYRIGHT: (C)2004,JPO&NCIPI

Description

【０００１】
【発明の属する技術分野】
この発明は、電力（有効、無効）、電力量（有効、無効）の少なくとも一つを計測する計測装置（以下電力関連量計測装置）に関するもので、測定精度を向上する補正機能を有するものに関する。
【０００２】
【従来の技術】
従来の電力関連量計測装置は、電圧信号および電流信号をオーバーサンプリング周波数によりそれぞれ量子化するデルタシグマＡＤ変調器と、量子化された電圧信号および電流信号をディジタルフィルタによりそれぞれ移動平均する移動平均処理手段と、移動平均処理された電圧信号および電流信号をサンプリング周波数毎に乗算する乗算手段と、乗算値の高周波成分を除去するディジタルローパスフィルタ手段とを備えている。
【０００３】
また、前記のオーバーサンプリング周波数により量子化するデルタシグマＡＤ変調器から、後段の移動平均手段へのデータ移行タイミングを変更可能として電圧信号−電流信号間の位相を調整するシフトレジスタ位相補正手段、および各相の電力関連量のバランスを調整するバランス補正手段とを備えている（特許文献１）。
また、電圧信号入力手段および電流信号入力手段により生ずる信号のバラツキを、各相の電圧信号−電流信号間の位相および各相の電力関連量のバランスを調整して抑制することにより、要求される計測精度を達成させていた（特許文献２）。
【０００４】
【特許文献１】
特許第３０８０２０７号公報。
【０００５】
【特許文献２】
特許第３３３０５１９号公報。
【０００６】
【発明が解決しようとする課題】
電圧信号入力手段および電流信号入力手段により生ずる信号のバラツキは、検出しようとする電圧および電流の大きさ、検出しようとする電圧および電流の周波数、演算中の温度に依存する。また温度によりＡＤ変換器の変換特性も影響を受ける。すなわち、検出しようとする電圧および電流の大きさ、周波数、演算中の温度が変化すると、電圧信号入力手段および電流信号入力手段およびＡＤ変換器により生ずる信号のバラツキが変化してしまう。
しかしながら、従来の電力関連量計測装置は、装置の出荷時に、定格電圧、定格電流、定格周波数、常温にて補正を行い、その補正値は一定としていた。つまり、初期設定以外に計測中に電圧、電流、周波数または温度によって演算値を補正していない。
したがって、広い入力範囲、広い温度範囲にわたって満足できる精度を得るために、電圧信号入力手段および電流信号入力手段、ＡＤ変換器は、検出しようとする電圧および電流の大きさや周波数、演算中の温度などによる誤差の変動が小さい、高精度なものを採用する必要があり、結果として高価なものとなっていた。
【０００７】
この発明は、かかる問題点を解決するためになされたものであり、検出しようとする電圧および電流の大きさ、周波数、演算中の温度の影響を受けることが少ない電力関連量計測装置を得ることを目的としている。
【０００８】
【課題を解決するための手段】
この発明に係る電力関連量計測装置は、電力線に装架された電流センサ及び電圧センサにより検出された電流および電圧を夫々ディジタル変換するＡＤ変換回路、
前記ＡＤ変換回路の出力から前記電圧の周波数を求めるとともに、前記ＡＤ変換回路の出力に基づいて前記電力線の電力関連量を演算する電力関連量演算部、前記電圧、電流、周波数のうち少なくともいずれか一つを助変数として、上記電力関連量を補正する補正手段を備えたものである。
【０００９】
また、電力線に装架されたＣＴ及びＰＴにより検出された電流および電圧を夫々ディジタル変換するＡＤ変換回路、
前記ＣＴ、ＰＴおよびＡＤ変換回路のすくなくとも１つに装着された温度検出器、
前記ＡＤ変換回路の出力から前記電圧の周波数を求めるとともに、前記ＡＤ変換回路の出力に基づいて前記電力線の電力関連量を演算する電力関連量演算部、前記温度を助変数として、上記電力関連量を補正する補正手段を備えたものである。
【００１０】
【発明の実施の形態】
実施の形態１．
以下、この発明の実施の形態１の電力関連量計測装置の構成を図１、図２により説明する。なお、図１，図２では図をわかりやすくするため１相、及び３相の場合のみを図示しており、２相の場合の図は省略している。
なお、以下の説明の内、特に符号については出願手続上、大文字、小文字の区別は厳密にはしないで記載するものとする。
図１において、図示しない電力線に装架した電圧センサ（以下ＰＴと言う場合もある）で検出した電力線の電圧信号１０１，１０３と、前記電力線に装架した図示しない電流センサ（以下ＣＴと言う場合もある）で検出した電流信号２０１，２０３を、各信号ごとに設けたデルタシグマＡＤ変換器（以下ＡＤ変換器）１２１、１２３、２２１、２２３がオーバーサンプリング周波数によりそれぞれ量子化し、電力関連量演算部１００に入力する。また、ＡＤ変換器の電圧出力から周波数演算手段５４が周波数を求め、また、各ＡＤ変換器の温度が温度計測手段５６により求められて電力関連量演算部１００に入力されている。
電力関連量演算部１００の詳細は図２に示す。図２において図示しない移動平均処理手段が量子化された電圧信号および電流信号をディジタルフィルタによりそれぞれ移動平均した後、乗算手段３０１、３０３に入力する。乗算手段３０１、３０３が移動平均処理された電圧信号および電流信号をサンプリング周波数毎に乗算する。ディジタルローパスフィルタ手段３２１，３２３が前記乗算値の高周波成分を除去する。デジタルローバスフィルタ手段３２１，３２３の出力を補正前の電力Ｗ＿ｉｎ＿Ｎ（Ｎ＝１，２，３）と言う。なおＷ＿ｉｎ＿Ｎは受電の場合に正の値、送電の場合に負の値となる。
なお、図には特に示さないが、図１のＡＤ変換器１２１，１２３，２２１，２２３によりＡＤ変換された後の部分の処理は全てソフトウェア／ハードウェアのいずれによる処理であってもかまわない。
【００１１】
ヒルベルト変換直交相１９１（１９３）とヒルベルト変換同相２９１（２９３）とからなるヒルベルト変換手段１９１，１９３，２９１，２９３は、電圧信号−電流信号間の位相を９０度回転する。乗算手段３４１、３４３は、ヒルベルト変換手段から出力された電圧信号および電流信号をサンプリング周波数毎に乗算する。ディジタルローパスフィルタ手段３６１，３６３は乗算値の高周波成分を除去する。ディジタルローパスフィルタ手段３６１，３６３の出力を補正前の無効電力ｖａｒ＿ｉｎ＿Ｎ（Ｎ＝１，２，３）と言う。なおｖａｒ＿ｉｎ＿Ｎは受電遅れおよび送電進みの場合に正の値、受電進みおよび送電遅れの場合に負の値となる。
【００１２】
振幅位相補正行列演算手段３８１，３８３は、補正前の電力Ｗ＿ｉｎ＿Ｎおよび無効電力ｖａｒ＿ｉｎ＿Ｎを入力とし、以下に説明する補正演算を行う。そしてその出力を補正後の電力Ｗ＿ｏｕｔ＿Ｎ（Ｎ＝１，２，３）、補正後の無効電力ｖａｒ＿ｏｕｔ＿Ｎ（Ｎ＝１，２，３）と言う。振幅位相補正行列演算手段３８１、３８３による補正演算は式（１）で示される。
【００１３】
【数２】

【００１４】
式（１）、式（２）において、Ｎ（Ｎ＝１，２，３）は各相を表す。式（１）、（２）は、θ＿Ｎが正の場合に左回転を行うので、θ＿Ｎが正の場合に遅れ方向の回転、負の場合に進み方向の回転を行う。理解を助けるため、この関係を図３に示す。
【００１５】
定格電圧、定格電流、定格周波数、常温時に、上記式（１）、式（２）の行列のＧａｉｎ＿Ｎおよびθ＿Ｎを求め、本発明の電力関連量計測装置の初期調整時に、これら式（１）、式（２）に基づいて、電圧信号−電流信号間の位相の調整、各相の電力関連量のバランス調整を行う。初期調整時に決定されたＧａｉｎ＿Ｎおよびθ＿Ｎは、調整時以外は変更せず、実際の演算に使用する振幅位相補正行列とは別に初期調整時の振幅位相補正行列として記憶しておく。
【００１６】
【数３】

【００１７】
本発明の補正のフローを図４に示す。電源周波数の変化による位相変化量の補正の概念を図５に示す。図５は、ＣＴの電流センシングにおける周波数と位相の関連並びに補正直線を説明する図である。図５（ａ）図は、横軸が周波数（４０Ｈｚ〜７０Ｈｚ）、縦軸が位相誤差（−１度〜＋０．４度）とした場合の特性図であり、ＣＴの実際の特性９０と、実際の特性９０を一次式（以下一次直線と言う）により近似させた補正用の一次直線９１とを示している。図５（ｂ）図は、図５（ａ）図と同じ横軸、同じ縦軸で補正用の一次直線９１により周波数を補正した後の、各周波数における位相誤差の誤差量を示しており、４５Ｈｚ〜６０Ｈｚの範囲で０．１度の誤差内に入っていることを示している。
【００１８】
実施例１．
実効値電圧演算手段１６１，１６３より、現在の実効値電圧Ｖｒｍｓ＿１，Ｖｒｍｓ＿３を取得して、実効値電圧による振幅補正率（振幅変化率の逆数）と位相変化量を補正用の一次直線により算出する。なお、電圧の振幅補正率と位相変化量の補正用の一次直線は、いずれも図５（ａ）図、（ｂ）図に示す周波数による誤差例と類似であるのでその特性を図に示すことは省略する。補正式については以下にさらに詳細に説明する。以下の説明においてＮは各相を表す。即ち、ここではＮ＝１，２，３である。
実効値電圧Ｖｒｍｓ＿１，Ｖｒｍｓ＿３による振幅補正率をＧａｉｎ＿Ｖｒｍｓ＿Ｎ、
位相変化量をＰｈａｓｅ＿Ｖｒｍｓ＿Ｎ、
振幅補正率の一次直線の傾きをＡ＿Ｇａｉｎ＿Ｖｒｍｓ、
切片をＢ＿Ｇａｉｎ＿Ｖｒｍｓ、
位相変化量の一次直線の傾きをＡ＿Ｐｈａｓｅ＿Ｖｒｍｓ、
切片をＢ＿Ｐｈａｓｅ＿Ｖｒｍｓ、
現在の実効値電圧をＶｒｍｓ＿Ｎとすると、補正式は以下のように表される。
【００１９】
Ｇａｉｎ＿Ｖｒｍｓ＿Ｎ＝Ａ＿Ｇａｉｎ＿Ｖｒｍｓ×Ｖｒｍｓ＿Ｎ＋Ｂ＿Ｇａｉｎ＿Ｖｒｍｓ・・・・（Ｖ１）
Ｐｈａｓｅ＿Ｖｒｍｓ＿Ｎ＝Ａ＿Ｐｈａｓｅ＿Ｖｒｍｓ×Ｖｒｍｓ＿Ｎ＋Ｂ＿Ｐｈａｓｅ＿Ｖｒｍｓ・・（Ｖ２）
振幅補正率Ｇａｉｎ＿Ｖｒｍｓ＿Ｎは、実効値電圧が初期調整時の電圧である場合に１となるように定めておく。また、Ｐｈａｓｅ＿Ｖｒｍｓ＿Ｎは、実効値電圧が初期調整時の電圧の場合に０、進み方向の変化量を正、遅れ方向の変化量を負となるように設定する。ここでいう初期調整時の電圧とは調整を行う際に基準（即ち補正を要しない）と定めた電圧のことを言い、必ずしも実際の調整時の電圧に限定するものではない。理解を助けるため上記の式（Ｖ１）（Ｖ２）を図６に示す。
【００２０】
なお、前記定格電圧を記憶しておき、定格電圧と直線の傾きから、切片を逆算して自動的に求めてもよく、その場合、切片の設定を省略できる。なお１次直線の傾きと切片は、自由に設定し、また変更することができるものとする。
【００２１】
したがって、電圧信号入力手段の回路構成が変更された場合（例えばＰＴ回路から抵抗分圧回路などに）、前記補正のための１次直線の傾きと切片の値を変更するのみでよい。
【００２２】
実施例２．
実効値電流演算手段２６１，２６３より、現在の実効値電流Ｉｒｍ＿１，Ｉｒｍ＿３を取得して、実効値電流による振幅補正率（振幅変化率の逆数）と位相変化量を一次直線により算出する。
実効値電流による振幅補正率をＧａｉｎ＿Ｉｒｍｓ＿Ｎ、
位相変化量をＰｈａｓｅ＿Ｉｒｍｓ＿Ｎ、
振幅補正率の一次直線の傾きをＡ＿Ｇａｉｎ＿Ｉｒｍｓ、
切片をＢ＿Ｇａｉｎ＿Ｉｒｍｓ、
位相変化量の一次直線の傾きをＡ＿Ｐｈａｓｅ＿Ｉｒｍｓ、
切片をＢ＿Ｐｈａｓｅ＿Ｉｒｍｓ、現在の実効値電流をＩｒｍｓ＿Ｎとすると補正式は次のようになる。
Ｇａｉｎ＿Ｉｒｍｓ＿Ｎ＝Ａ＿Ｇａｉｎ＿Ｉｒｍｓ×Ｉｒｍｓ＿Ｎ＋Ｂ＿Ｇａｉｎ＿Ｉｒｍｓ・・・・（Ｉ１）
Ｐｈａｓｅ＿Ｉｒｍｓ＿Ｎ＝Ａ＿Ｐｈａｓｅ＿Ｉｒｍｓ×Ｉｒｍｓ＿Ｎ＋Ｂ＿Ｐｈａｓｅ＿Ｉｒｍｓ・・・（Ｉ２）
Ｇａｉｎ＿Ｉｒｍｓ＿Ｎは、現在の実効値電流が初期調整時の電流となった場合に１となるように設定する。また、Ｐｈａｓｅ＿Ｉｒｍｓ＿Ｎは、進み方向の変化量を正、遅れ方向の変化量を負とし、現在の実効値電流が初期調整時の電流となった場合に０となるように設定する。よって初期調整時の電流を記憶しておき、現在の電流値と直線の傾きから、切片を逆算してもよく、その場合、切片の設定量を削減できる。なお１次直線の傾きと切片は、自由に設定変更可能とする。したがって、電流信号入力手段の回路構成が変更された場合（例えばＣＴ回路から分割形ＣＴ回路）、１次直線の傾きと切片の値を変更するのみでよい。式（Ｉ１）（Ｉ２）は図６と類似した特性図となるので、図示説明は省略する。
【００２３】
実施例３．
電源周波数演算手段５４より、現在の電源周波数Ｆｒｅｑを取得して、電源周波数による電圧信号入力および電流信号入力の振幅補正率（振幅変化率の逆数）と位相変化量を補正のための一次直線により算出する。
電源周波数による電圧信号入力の振幅補正率をＧａｉｎ＿ＦｒｅｑＶ、
位相変化量をＰｈａｓｅ＿ＦｒｅｑＶ、
振幅補正率の一次直線の傾きをＡ＿Ｇａｉｎ＿ＦｒｅｑＶ、
切片をＢ＿Ｇａｉｎ＿ＦｒｅｑＶ、
位相変化量の一次直線の傾きをＡ＿Ｐｈａｓｅ＿ＦｒｅｑＶ、
切片をＢ＿Ｐｈａｓｅ＿ＦｒｅｑＶ、
電源周波数による電流信号入力の振幅補正率をＧａｉｎ＿ＦｒｅｑＩ、
位相変化量をＰｈａｓｅ＿ＦｒｅｑＩ、
振幅補正率の一次直線の傾きをＡ＿Ｇａｉｎ＿ＦｒｅｑＩ、
切片をＢ＿Ｇａｉｎ＿ＦｒｅｑＩ、
位相変化量の一次直線の傾きをＡ＿Ｐｈａｓｅ＿ＦｒｅｑＩ、
切片をＢ＿Ｐｈａｓｅ＿ＦｒｅｑＩ、
現在の電源周波数をＦｒｅｑとすると補正の一次直線は以下のようになる。
Ｇａｉｎ＿ＦｒｅｑＶ＝Ａ＿Ｇａｉｎ＿ＦｒｅｑＶ×Ｆｒｅｑ＋Ｂ＿Ｇａｉｎ＿ＦｒｅｑＶ・・・・（ＦＶ１）
Ｇａｉｎ＿ＦｒｅｑＩ＝Ａ＿Ｇａｉｎ＿ＦｒｅｑＩ×Ｆｒｅｑ＋Ｂ＿Ｇａｉｎ＿ＦｒｅｑＩ・・・・（ＦＩ１）
Ｐｈａｓｅ＿ＦｒｅｑＶ＝Ａ＿Ｐｈａｓｅ＿ＦｒｅｑＶ×Ｆｒｅｑ＋Ｂ＿Ｐｈａｓｅ＿ＦｒｅｑＶ・・・（ＦＶ２）
Ｐｈａｓｅ＿ＦｒｅｑＩ＝Ａ＿Ｐｈａｓｅ＿ＦｒｅｑＩ×Ｆｒｅｑ＋Ｂ＿Ｐｈａｓｅ＿ＦｒｅｑＩ・・・（ＦＩ２）
【００２４】
Ｇａｉｎ＿ＦｒｅｑＶおよびＧａｉｎ＿ＦｒｅｑＩは、現在の電源周波数が初期調整時の電源周波数となった場合に１となるように１次直線の傾きと切片を設定する。また、Ｐｈａｓｅ＿ＦｒｅｑＶおよびＰｈａｓｅ＿ＦｒｅｑＩは、進み方向の変化量を正、遅れ方向の変化量を負とし、現在の電源周波数が初期調整時の電源周波数となった場合に０となるように１次直線の傾きと切片を設定する。よって初期調整時の電源周波数を記憶しておき、現在の周波数と直線の傾きから、切片を逆算してもよく、その場合、切片の設定量を削減できる。なお１次直線の傾きと切片は、自由に設定変更可能とする。したがって、電圧信号入力手段および電流信号入力手段の回路構成が変更された場合（例えばＰＴ回路から抵抗分圧回路、ＣＴ回路から分割形ＣＴ回路）、１次直線の傾きと切片の値を変更するのみでよい。補正式（ＦＶ１）（ＦＩ１）（ＦＶ２）（ＦＩ２）はいずれも図６と類似した特性であるから図示説明を省略する。
【００２５】
また、電圧信号入力手段と電流信号入力手段を合計した振幅補正率をＧａｉｎ＿Ｆｒｅｑ、位相変化量をＰｈａｓｅ＿Ｆｒｅｑとして、
Ｇａｉｎ＿Ｆｒｅｑ＝（Ａ＿Ｇａｉｎ＿ＦｒｅｑＶ＋Ａ＿Ｇａｉｎ＿ＦｒｅｑＩ）×Ｆｒｅｑ＋
Ｂ＿Ｇａｉｎ＿ＦｒｅｑＶ＋Ｂ＿Ｇａｉｎ＿ＦｒｅｑＩ
Ｐｈａｓｅ＿Ｆｒｅｑ＝（Ａ＿Ｐｈａｓｅ＿ＦｒｅｑＶ＋Ａ＿Ｐｈａｓｅ＿ＦｒｅｑＩ）×Ｆｒｅｑ＋
Ｂ＿Ｇａｉｎ＿ＦｒｅｑＶ＋Ｂ＿Ｐｈａｓｅ＿ＦｒｅｑＩ
として、演算量および直線傾きと切片の設定量を削減してもよい。
【００２６】
実施例４．
温度計測手段５６より、現在のＡＤ変換器の温度Ｔｅｍｐを取得して（ここではどのＡＤ変換器もほぼ同一の温度であると仮定している）、温度による電圧信号入力および電流信号入力およびＡＤ変換器の振幅補正率（振幅変化率の逆数）と位相変化量を一次直線により算出する。
１）温度による電圧信号入力の補正については
振幅補正率をＧａｉｎ＿ＴｅｍｐＶ、その位相変化量をＰｈａｓｅ＿ＴｅｍｐＶ、
振幅補正率の一次直線の傾きをＡ＿Ｇａｉｎ＿ＴｅｍｐＶ、切片をＢ＿Ｇａｉｎ＿ＴｅｍｐＶ、
位相変化量の一次直線の傾きをＡ＿Ｐｈａｓｅ＿ＴｅｍｐＶ、切片をＢ＿Ｐｈａｓｅ＿ＴｅｍｐＶ、
２）温度による電流信号入力の補正については
振幅補正率をＧａｉｎ＿ＴｅｍｐＩ、位相変化量をＰｈａｓｅ＿ＴｅｍｐＩ、
振幅補正率の一次直線の傾きをＡ＿Ｇａｉｎ＿ＴｅｍｐＩ、切片をＢ＿Ｇａｉｎ＿ＴｅｍｐＩ、
位相変化量の一次直線の傾きをＡ＿Ｐｈａｓｅ＿ＴｅｍｐＩ、切片をＢ＿Ｐｈａｓｅ＿ＴｅｍｐＩ、
３）温度によるＡＤ変換器の補正については
振幅補正率をＧａｉｎ＿ＴｅｍｐＡＤ、位相変化量をＰｈａｓｅ＿ＴｅｍｐＡＤ、
振幅補正率の一次直線の傾きをＡ＿Ｇａｉｎ＿ＴｅｍｐＡＤ、切片をＢ＿Ｇａｉｎ＿ＴｅｍｐＡＤ、
位相変化量の一次直線の傾きをＡ＿Ｐｈａｓｅ＿ＴｅｍｐＡＤ、切片をＢ＿Ｐｈａｓｅ＿ＴｅｍｐＡＤ、
とし、現在の温度をＴｅｍｐとすると補正式は次のようになる。
Ｇａｉｎ＿ＴｅｍｐＶ＝Ａ＿Ｇａｉｎ＿ＴｅｍｐＶ×Ｔｅｍｐ＋Ｂ＿Ｇａｉｎ＿ＴｅｍｐＶ・・・（ＴＶ１）
Ｇａｉｎ＿ＴｅｍｐＩ＝Ａ＿Ｇａｉｎ＿ＴｅｍｐＩ×Ｔｅｍｐ＋Ｂ＿Ｇａｉｎ＿ＴｅｍｐＩ・・・（ＴＩ１）
Ｇａｉｎ＿ＴｅｍｐＡＤ＝Ａ＿Ｇａｉｎ＿ＴｅｍｐＡＤ×Ｔｅｍｐ＋Ｂ＿Ｇａｉｎ＿ＴｅｍｐＡＤ・・・（ＴＡ１）
Ｐｈａｓｅ＿ＴｅｍｐＶ＝Ａ＿Ｐｈａｓｅ＿ＴｅｍｐＶ×Ｔｅｍｐ＋Ｂ＿Ｐｈａｓｅ＿ＴｅｍｐＶ・・・（ＴＶ２）
Ｐｈａｓｅ＿ＴｅｍｐＩ＝Ａ＿Ｐｈａｓｅ＿ＴｅｍｐＩ×Ｔｅｍｐ＋Ｂ＿Ｐｈａｓｅ＿ＴｅｍｐＩ・・・（ＴＩ２）
Ｐｈａｓｅ＿ＴｅｍｐＡＤ＝Ａ＿Ｐｈａｓｅ＿ＴｅｍｐＡＤ×Ｔｅｍｐ＋Ｂ＿Ｐｈａｓｅ＿ＴｅｍｐＡＤ・・（ＴＡ２）
【００２７】
Ｇａｉｎ＿ＴｅｍｐＶおよびＧａｉｎ＿ＴｅｍｐＩおよびＧａｉｎ＿ＴｅｍｐＡＤは、現在の温度が初期調整時の温度となった場合に１となるように１次直線の傾きと切片を設定する。また、Ｐｈａｓｅ＿ＴｅｍｐＶおよびＰｈａｓｅ＿ＴｅｍｐＩおよびＰｈａｓｅ＿ＴｅｍｐＡＤは、進み方向の変化量を正、遅れ方向の変化量を負とし、現在の温度が初期調整時の温度となった場合に０となるように１次直線の傾きと切片を設定する。よって初期調整時の温度を記憶しておき、現在の温度と直線の傾きから、切片を逆算してもよく、その場合、切片の設定量を削減できる。なお１次直線の傾きと切片は、自由に設定変更可能とする。したがって、電圧信号入力手段および電流信号入力手段の回路構成が変更された場合（例えばＰＴ回路から抵抗分圧回路、ＣＴ回路から分割形ＣＴ回路）、１次直線の傾きと切片の値を変更するのみでよい。
【００２８】
また、電圧信号入力手段と電流信号入力手段とＡＤ変換器を合計した振幅補正率をＧａｉｎ＿Ｔｅｍｐ、位相変化量をＰｈａｓｅ＿Ｔｅｍｐとして、
Ｇａｉｎ＿Ｔｅｍｐ＝
（Ａ＿Ｇａｉｎ＿ＴｅｍｐＶ＋Ａ＿Ｇａｉｎ＿ＴｅｍｐＩ＋Ａ＿Ｇａｉｎ＿ＴｅｍｐＡＤ）×
Ｔｅｍｐ＋Ｂ＿Ｇａｉｎ＿ＴｅｍｐＶ＋Ｂ＿Ｇａｉｎ＿ＴｅｍｐＩ＋Ｂ＿Ｇａｉｎ＿ＴｅｍｐＡＤ
Ｐｈａｓｅ＿Ｔｅｍｐ＝
（Ａ＿Ｐｈａｓｅ＿ＴｅｍｐＶ＋Ａ＿Ｐｈａｓｅ＿ＴｅｍｐＩ＋Ａ＿Ｐｈａｓｅ＿ＴｅｍｐＡＤ）×
Ｔｅｍｐ＋Ｂ＿Ｇａｉｎ＿ＴｅｍｐＶ＋Ｂ＿Ｐｈａｓｅ＿ＴｅｍｐＩ＋Ｂ＿Ｐｈａｓｅ＿ＴｅｍｐＡＤ
として、演算量および直線傾きと切片の設定量を削減してもよい。
【００２９】
前記で算出した振幅補正率および位相変化量を、電力、無効電力の演算に反映させるために、初期調整時の振幅位相補正行列（前述）に振幅補正率および位相変化量補正行列を乗算する。その結果を実際の演算に使用する振幅位相補正行列とする。
全体の振幅補正率および位相変化量は、実効値電圧、実効値電流、電源周波数、温度のうち、補正したいパラメータを任意に選択して、選択したパラメータによる影響の合計とする。全パラメータを選択した場合、全体の振幅補正率をＧａｉｎ＿ａｌｌ＿Ｎ、位相変化量をＰｈａｓｅ＿ａｌｌ＿Ｎとすると、式（３）のようになる。
Ｇａｉｎ＿ａｌｌ＿Ｎ＝Ｇａｉｎ＿Ｖｒｍｓ＿Ｎ×Ｇａｉｎ＿Ｉｒｍｓ＿Ｎ×Ｇａｉｎ＿Ｆｒｅｑ×Ｇａｉｎ＿Ｔｅｍｐ
Ｐｈａｓｅ＿ａｌｌ＿Ｎ＝Ｐｈａｓｅ＿Ｖｒｍｓ＿Ｎ＋Ｐｈａｓｅ＿Ｉｒｍｓ＿Ｎ＋Ｐｈａｓｅ＿Ｆｒｅｑ＋Ｐｈａｓｅ＿Ｔｅｍｐ
【００３０】
【数４】

【００３１】
となる。
したがって、実際の演算に使用する振幅位相補正行列は以下の式（４）のとおりとなる。
【００３２】
【数５】

【００３３】
また、前記で算出した振幅補正率および位相変化量を、実効値電圧、実効値電流に反映させる場合、実効値は位相に無関係であるため、実効値電圧Ｖｒｍｓ−Ｎ、実効値電流Ｉｒｍｓ−Ｎの振幅補正率を、初期調整時の変換係数１８１，１８３，２８１，２８３に乗算した結果を実際の演算に使用する変換係数とすればよい。
また、実際の演算に使用する振幅位相補正行列の係数の算出はサンプリング周波数毎に行うのではなく、例えば各実効値や温度の変化が有意の差として認められる時間間隔、例えば０．５秒〜数秒毎に算出して結果を更新すればよい。よって、演算量は更新した時に増加するのみであり、常時の電力関連量を演算する演算量は増加せず、演算負荷が増加するということもない。
【００３４】
以上のように、電力関連量の計測時に、検出された実効値電圧、実効値電流、電源周波数およびＡＤ変換器の温度により補正するので、電力関連量を常に高精度に計測することができる。なお、実効値電圧、実効値電流、電源周波数および温度により補正する例について説明したが、少なくともいずれか一つのパラメータにより、補正するものであってもよく、これについては実施の形態２で説明する。
【００３５】
また、回転行列により、位相を補正するので、短時間で補正ができる。
また、電流に対する電圧、または電圧に対する電流の位相を９０度直位相変換する９０度位相変換手段と、９０度位相変換手段の出力により無効電力を求める無効電力演算手段と、電流と電圧から有効電力を求める有効電力演算手段とを有するものに適用するので、調整時の振幅位相補正行列を得るための回転行列演算機能と上述の補正のための回転行列演算機能を兼用することができる。
なお、実施の形態１では温度はＡＤ変換器の温度であると説明しているが、以上に説明した演算を行うための演算装置の温度を用いても良い。
【００３６】
実施の形態２．
上記実施の形態１では、実効値電圧、実効値電流、電源周波数、ＡＤ変換器の温度の４つのパラメータにて、振幅補正率を７つおよび位相変化量を７つ算出したが、このうち任意の項目のみ選択して算出してもよい。例えば、電流信号入力の位相変化量のみが、実効値電流、電源周波数に大きく依存するため、この補正のみ実施したいのであれば、Ｐｈａｓｅ＿Ｉｒｍｓ＿Ｎ、Ｐｈａｓｅ＿ＦｒｅｑＩについてのみ行えばよい。他の項目についても同様である。
【００３７】
実施の形態３．
上記実施の形態２では、任意の項目のみ選択して算出したが、全ての項目を選択した上で、特に補正を要しない項目について、その補正のための１次直線の傾きと切片を以下のように設定することでも同様な動作を実現可能である。
即ち、不要な項目の振幅補正率は、一次直線の傾きに０、切片に１を設定する。不要な項目の位相変化量は、一次直線の傾きに０、切片に０を設定する。
【００３８】
実施の形態４．
実際の演算に使用する振幅位相補正行列の係数を浮動小数点演算にて算出した後、振幅位相補正行列を更新する際に浮動小数点から固定小数点に変換してセットしてもよい。これにより常時の演算は固定小数点の乗算と加算のみとなり、演算速度が向上する。
【００３９】
実施の形態５
上記の実施の形態４では、振幅位相補正行列を更新する際に固定小数点としたが、実際の演算に使用する振幅位相補正行列の係数の算出を固定小数点にて行うことも可能である。
実際には、一次直線の傾きおよび切片を固定小数点の設定とする。また、振幅補正率および位相変化量を算出する際の、実効値電圧、実効値電流、電源周波数、温度のパラメータも固定小数点にて取得する。これにより、振幅補正率および位相変化量を算出する際には、固定小数点の乗算と加算のみとなる。
さらに、位相変化量補正行列を演算する際に必要となる、正弦値、余弦値の浮動小数点による演算量を削除するため、例えば０．０１度左回転行列、０．１度左回転行列、１度左回転行列の係数を固定小数点のデフォルトで予め持っておいて、位相変化量を算出後、上記の左回転行列を乗算することで位相変化量補正行列を算出してもよい。
上記の例で、位相変化量が１．２４度（進み）である場合、位相変化量補正行列は、１度左回転行列×（０．１度左回転行列）^２×（０．０１度左回転行列）^４を演算することで算出できる。
【００４０】
実施の形態６
分割形ＣＴのように、検出しようとする電流の大きさによる位相変化量が非線形となり一次直線にて補正が困難である場合には、２つもしくは３つ等、傾斜および切片の異なる多数の１次直線を設定して、直線の交点を境界として補正量を算出する一次直線を切替えることにより補正する。
例えば、電流信号入力において分割形ＣＴを採用しており、検出しようとする電流の大きさによる位相変化量が非線形となる場合を例として説明する。このときの特性の一例９５を図７（ａ）図に示す。図７（ａ）の特性９５では負荷が約１０％を境として、これより軽負荷では位相誤差がそれより重負荷部分に比べて急に大きくなっている。このような場合、位相変化量Ｐｈａｓｅ＿Ｉｒｍｓ＿Ｎを算出する一次直線を例えば２つ用意する。位相変化量を算出するための、一つ目の一次直線９６の傾きをＡ１＿Ｐｈａｓｅ＿Ｉｒｍｓ、切片をＢ１＿Ｐｈａｓｅ＿Ｉｒｍｓ、２つ目の一次直線９７の傾きをＡ２＿Ｐｈａｓｅ＿Ｉｒｍｓ、切片をＢ２＿Ｐｈａｓｅ＿Ｉｒｍｓとする。２つの一次直線が設定された時点で、直線９６と直線９７との交点のＩｒｍｓを算出する。
【００４１】
【数６】

【００４２】
したがって、現在のＩｒｍｓ＿Ｎが直線交点のＩｒｍｓ以上の場合に１つ目の一次直線９６を使用して、
Ｐｈａｓｅ＿Ｉｒｍｓ＿Ｎ＝Ａ１＿Ｐｈａｓｅ＿Ｉｒｍｓ×Ｉｒｍｓ＿Ｎ＋Ｂ１＿Ｐｈａｓｅ＿Ｉｒｍｓ
とし、現在のＩｒｍｓ＿Ｎが、直線交点のＩｒｍｓ未満の場合に２つ目の一次直線９７を使用して、
Ｐｈａｓｅ＿Ｉｒｍｓ＿Ｎ＝Ａ２＿Ｐｈａｓｅ＿Ｉｒｍｓ×Ｉｒｍｓ＿Ｎ＋Ｂ２＿Ｐｈａｓｅ＿Ｉｒｍｓ
とする。
このようにすることにより、図７（ｂ）に示す位相誤差９８のように、電力関連量の計測をより高い精度で行うことができる。
特性９５の曲線に合わせてより多くの直線を用いて良いことは言うまでもない。
【００４３】
【発明の効果】
この発明に係る電力関連量計測装置は、検出しようとする電圧および電流の大きさ、周波数、演算中のＡＤ変換器の温度による電圧信号入力手段および電流信号入力手段、ＡＤ変換器の各特性が変化しても、それを少ない演算量、少ないメモリ容量で高速に補正して常に高精度に計測することができるので、電力関連量をより高精度に計測することができる。
【図面の簡単な説明】
【図１】本発明の実施の形態１に係る電力関連量計測装置の構成を示す図である。
【図２】図１の部分詳細図である。
【図３】補正前後の電力のベクトル図である。
【図４】補正フローを示す図である。
【図５】ＣＴの電流センシングにおける周波数と位相の関連並びに補正直線を説明する特性図である。
【図６】補正直線を説明する図である。
【図７】本発明の実施の形態５に係るＣＴの電流センシングにおける周波数と位相の関連並びに補正直線を説明する図である。
【符号の説明】
１０１，１０３電圧信号、
１２１，１２３，２２１，２２３デルタシグマＡＤ変換器、
１４１，１４３，２４１，２４３実効値電圧演算手段、
１６１，１６３，２６１，２６３実効値電流演算手段、
１９１，１９３，２９１，２９３ヒルベルト変換手段、
２０１，２０３電流信号、３０１，３０３乗算手段、
３２１，３２３ディジタルローパスフィルタ手段、
３４１，３４３乗算手段、
３６１，３６３ディジタルローパスフィルタ手段、
５４電源周波数演算手段、５６温度計測手段、
Ｗ＿ｉｎ＿Ｎ補正前の電力、Ｗ＿ｏｕｔ＿Ｎ補正後の電力、
ｖａｒ＿ｉｎ＿Ｎ補正前の無効電力、ｖａｒ＿ｏｕｔ＿Ｎ補正後の無効電力、
Ｖｒｍｓ＿１，Ｖｒｍｓ＿３実効値電圧、Ｉｒｍ＿１，Ｉｒｍ＿３実効値電流、
Ｆｒｅｑ電源周波数、Ｔｅｍｐ温度。[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to a measuring device for measuring at least one of electric power (valid, invalid) and electric energy (valid, invalid) (hereinafter referred to as a power-related amount measuring device), and to a device having a correction function for improving measurement accuracy. .
[0002]
[Prior art]
A conventional power-related amount measuring device includes a delta-sigma AD modulator that quantizes a voltage signal and a current signal using an oversampling frequency, respectively, and a moving average process that uses a digital filter to perform a moving average of the quantized voltage signal and the current signal. Means, multiplying means for multiplying the moving average processed voltage signal and current signal for each sampling frequency, and digital low-pass filter means for removing high frequency components of the multiplied value.
[0003]
A shift register phase correcting means for adjusting a phase between a voltage signal and a current signal by changing a data transfer timing from a delta-sigma AD modulator quantizing by the oversampling frequency to a moving average means at a subsequent stage, and There is provided a balance correction unit for adjusting the balance of the power-related amount of each phase (Patent Document 1).
Further, it is required to suppress the variation of the signal generated by the voltage signal input means and the current signal input means by adjusting the balance between the phase between the voltage signal and the current signal of each phase and the power related amount of each phase. Measurement accuracy was achieved (Patent Document 2).
[0004]
[Patent Document 1]
Japanese Patent No. 3080207.
[0005]
[Patent Document 2]
Japanese Patent No. 3330519.
[0006]
[Problems to be solved by the invention]
The variation of the signal generated by the voltage signal input means and the current signal input means depends on the magnitude of the voltage and current to be detected, the frequency of the voltage and current to be detected, and the temperature during the operation. The conversion characteristics of the AD converter are also affected by the temperature. That is, when the magnitude and frequency of the voltage and current to be detected and the temperature during the calculation change, the variation of the signal generated by the voltage signal input means, the current signal input means, and the AD converter changes.
However, the conventional power-related amount measuring device performs correction at rated voltage, rated current, rated frequency, and normal temperature when the device is shipped, and the correction value is constant. That is, the calculated value is not corrected based on the voltage, current, frequency, or temperature during measurement other than the initial setting.
Therefore, in order to obtain satisfactory accuracy over a wide input range and a wide temperature range, the voltage signal input means, the current signal input means, and the A / D converter are used to determine the magnitude and frequency of the voltage and current to be detected, the temperature during calculation, and the like. Therefore, it is necessary to employ a high-precision one in which the variation of the error due to the fluctuation is small, and as a result, it is expensive.
[0007]
SUMMARY OF THE INVENTION The present invention has been made to solve such a problem, and it is an object of the present invention to obtain a power-related amount measuring device which is less affected by the magnitude and frequency of a voltage and a current to be detected and a temperature during calculation. It is an object.
[0008]
[Means for Solving the Problems]
The power-related amount measuring device according to the present invention includes an AD conversion circuit that digitally converts a current and a voltage detected by a current sensor and a voltage sensor mounted on a power line, respectively.
A power-related amount calculation unit that calculates a frequency of the voltage from an output of the AD conversion circuit and calculates a power-related amount of the power line based on an output of the AD conversion circuit, at least one of the voltage, the current, and the frequency. A correction means for correcting the power-related amount using one as an auxiliary variable is provided.
[0009]
An AD conversion circuit for digitally converting the current and the voltage detected by the CT and the PT mounted on the power line, respectively;
A temperature detector mounted on at least one of the CT, PT and AD conversion circuits;
A power-related amount calculation unit that calculates the frequency of the voltage from the output of the AD conversion circuit and calculates the power-related amount of the power line based on the output of the AD conversion circuit; Is provided.
[0010]
BEST MODE FOR CARRYING OUT THE INVENTION
Embodiment 1 FIG.
Hereinafter, the configuration of the power-related amount measuring apparatus according to the first embodiment of the present invention will be described with reference to FIGS. Note that FIGS. 1 and 2 show only one-phase and three-phase cases for simplification of the drawings, and omit the two-phase cases.
It should be noted that, in the following description, in particular, reference numerals are not strictly distinguished in capital letters and small letters in the application procedure.
In FIG. 1,

voltage signals

101 and 103 of a power line detected by a voltage sensor (hereinafter, also referred to as PT) mounted on a power line (not shown) and a current sensor (hereinafter referred to as CT) mounted on the power line. ), The delta-sigma A / D converters (hereinafter, A / D converters) 121, 123, 221, and 223 quantize the current signals 201 and 203, respectively, based on the oversampling frequency, and calculate the power-related amount. Input to the unit 100. The frequency calculating means 54 obtains the frequency from the voltage output of the AD converter, and the temperature of each AD converter is obtained by the temperature measuring means 56 and input to the power-related amount calculating section 100.
The details of the power-related amount calculation unit 100 are shown in FIG. A moving average processing means (not shown in FIG. 2) performs moving average of the quantized voltage signal and current signal by a digital filter, respectively, and then inputs them to multiplication means 301 and 303. Multiplying means 301 and 303 multiply the moving average processed voltage signal and current signal for each sampling frequency. Digital low pass filter means 321 and 323 remove high frequency components of the multiplied value. Outputs of the digital low-

pass filter units

321 and 323 are referred to as uncorrected power W_in_N (N = 1, 2, 3). Note that W_in_N has a positive value in the case of power reception and a negative value in the case of power transmission.
Although not particularly shown in the figure, the processing of the part after the AD conversion by the

AD converters

121, 123, 221, and 223 in FIG. 1 may be all processing by software or hardware.
[0011]
The Hilbert transform means 191, 193, 291, 293 comprising the Hilbert transform quadrature phase 191 (193) and the Hilbert transform in-phase 291 (293) rotate the phase between the voltage signal and the current signal by 90 degrees. The

multiplying units

341 and 343 multiply the voltage signal and the current signal output from the Hilbert transform unit for each sampling frequency. Digital low-pass filter means 361 and 363 remove high frequency components of the multiplied value. Outputs of the digital low-pass filter means 361 and 363 are referred to as reactive power var_in_N (N = 1, 2, 3) before correction. Note that var_in_N is a positive value in the case of power reception delay and power transmission advance, and a negative value in the case of power reception advance and power transmission delay.
[0012]
The amplitude and phase correction matrix calculation means 381 and 383 receive the power W_in_N before the correction and the reactive power var_in_N as inputs and perform the correction calculation described below. The output is referred to as corrected power W_out_N (N = 1, 2, 3) and corrected reactive power var_out_N (N = 1, 2, 3). The correction calculation by the amplitude and phase correction matrix calculation means 381 and 383 is represented by Expression (1).
[0013]
(Equation 2)

[0014]
In equations (1) and (2), N (N = 1, 2, 3) represents each phase. Equations (1) and (2) perform left rotation when θ_N is positive, so that rotation is performed in the delay direction when θ_N is positive, and rotation in the forward direction is performed when θ_N is negative. This relationship is shown in FIG. 3 to aid understanding.
[0015]
At the time of rated voltage, rated current, rated frequency, and normal temperature, Gain_N and θ_N of the matrix of the above formulas (1) and (2) are obtained. Based on the equation (2), the phase between the voltage signal and the current signal is adjusted, and the balance of the power related amount of each phase is adjusted. Gain_N and θ_N determined at the time of the initial adjustment are not changed except at the time of the adjustment, and are stored as the amplitude-phase correction matrix at the time of the initial adjustment separately from the amplitude-phase correction matrix used for the actual calculation.
[0016]
[Equation 3]

[0017]
FIG. 4 shows a flow of the correction according to the present invention. FIG. 5 shows the concept of correcting the amount of phase change due to a change in the power supply frequency. FIG. 5 is a diagram illustrating the relationship between the frequency and the phase and the correction straight line in the CT current sensing. FIG. 5A is a characteristic diagram in the case where the horizontal axis represents frequency (40 Hz to 70 Hz) and the vertical axis represents phase error (−1 degree to +0.4 degree). A primary straight line 91 for correction is shown in which the actual characteristic 90 is approximated by a linear equation (hereinafter referred to as a primary straight line). FIG. 5B shows the error amount of the phase error at each frequency after the frequency is corrected on the same horizontal axis and the same vertical axis as in FIG. It shows that the error is within 0.1 degree in the range of 45 Hz to 60 Hz.
[0018]
Embodiment 1 FIG.
The current effective value voltages Vrms_1 and Vrms_3 are obtained from the effective value voltage calculating means 161 and 163, and the amplitude correction rate (the reciprocal of the amplitude change rate) and the phase change amount based on the effective value voltage are calculated by a primary straight line for correction. . Note that both the voltage amplitude correction rate and the primary straight line for correcting the phase change amount are similar to the frequency error examples shown in FIGS. 5A and 5B, and the characteristics thereof are shown in the figure. Is omitted. The correction formula will be described in more detail below. In the following description, N represents each phase. That is, N = 1, 2, 3 here.
The amplitude correction rate based on the effective value voltages Vrms_1 and Vrms_3 is Gain_Vrms_N,
The amount of phase change is Phase_Vrms_N,
A_Gain_Vrms represents the slope of the primary straight line of the amplitude correction rate,
The section is B_Gain_Vrms,
A_Phase_Vrms represents the slope of the primary straight line of the phase change amount,
The section is B_Phase_Vrms,
Assuming that the current effective value voltage is Vrms_N, the correction formula is expressed as follows.
[0019]
Gain_Vrms_N = A_Gain_Vrms × Vrms_N + B_Gain_Vrms (V1)
Phase_Vrms_N = A_Phase_Vrms × Vrms_N + B_Phase_Vrms (V2)
The amplitude correction rate Gain_Vrms_N is set to be 1 when the effective value voltage is the voltage at the time of the initial adjustment. Further, Phase_Vrms_N is set such that the effective value voltage is 0 at the time of the initial adjustment, the change amount in the leading direction is positive, and the change amount in the delay direction is negative. Here, the voltage at the time of the initial adjustment refers to a voltage determined as a reference (that is, no correction is required) at the time of the adjustment, and is not necessarily limited to the voltage at the time of the actual adjustment. The equations (V1) and (V2) are shown in FIG. 6 to facilitate understanding.
[0020]
The rated voltage may be stored, and the intercept may be automatically calculated from the rated voltage and the slope of the straight line. In this case, the setting of the intercept can be omitted. Note that the inclination and intercept of the primary straight line can be freely set and changed.
[0021]
Therefore, when the circuit configuration of the voltage signal input means is changed (for example, from a PT circuit to a resistive voltage dividing circuit), it is only necessary to change the slope and intercept of the primary straight line for the correction.
[0022]
Embodiment 2. FIG.
The current effective value currents Irm_1 and Irm_3 are obtained from the effective value current calculation means 261 and 263, and the amplitude correction rate (the reciprocal of the amplitude change rate) and the phase change amount based on the effective value current are calculated by a primary straight line.
Gain_Irms_N,
The amount of phase change is Phase_Irms_N,
A_Gain_Irms is the slope of the primary straight line of the amplitude correction rate,
The section is B_Gain_Irms,
A_Phase_Irms represents the slope of the primary straight line of the phase change amount,
Assuming that the intercept is B_Phase_Irms and the current effective current is Irms_N, the correction equation is as follows.
Gain_Irms_N = A_Gain_Irms × Irms_N + B_Gain_Irms (I1)
Phase_Irms_N = A_Phase_Irms × Irms_N + B_Phase_Irms (I2)
Gain_Irms_N is set to be 1 when the current effective current becomes the current at the time of the initial adjustment. Phase_Irms_N is set such that the amount of change in the leading direction is positive and the amount of change in the lagging direction is negative, and becomes zero when the current effective value current becomes the current at the time of the initial adjustment. Therefore, the current at the time of the initial adjustment may be stored, and the intercept may be calculated backward from the current value and the slope of the straight line. In this case, the set amount of the intercept can be reduced. The inclination and intercept of the primary straight line can be freely changed. Therefore, when the circuit configuration of the current signal input means is changed (for example, from a CT circuit to a divided CT circuit), it is only necessary to change the slope and intercept of the primary line. Equations (I1) and (I2) are characteristic diagrams similar to those in FIG.
[0023]
Embodiment 3 FIG.
The current power supply frequency Freq is obtained from the power supply frequency calculating means 54, and the amplitude correction rate (reciprocal of the amplitude change rate) and the phase change amount of the voltage signal input and the current signal input by the power supply frequency are corrected by a primary straight line. calculate.
The amplitude correction rate of the voltage signal input depending on the power supply frequency is Gain_FreqV,
The amount of phase change is Phase_FreqV,
A_Gain_FreqV represents the slope of the primary straight line of the amplitude correction rate,
The section is B_Gain_FreqV,
A_Phase_FreqV represents the slope of the primary straight line of the phase change amount,
Sections are B_Phase_FreqV,
Gain_FreqI, the amplitude correction rate of the current signal input according to the power supply frequency,
The phase change amount is represented by Phase_FreqI,
A_Gain_FreqI represents the slope of the primary straight line of the amplitude correction rate,
The section is B_Gain_FreqI,
A_Phase_FreqI represents the slope of the primary straight line of the phase change amount,
The section is B_Phase_FreqI,
Assuming that the current power supply frequency is Freq, the primary straight line of the correction is as follows.
Gain_FreqV = A_Gain_FreqV × Freq + B_Gain_FreqV (FV1)
Gain_FreqI = A_Gain_FreqI × Freq + B_Gain_FreqI (FI1)
Phase_FreqV = A_Phase_FreqV × Freq + B_Phase_FreqV (FV2)
Phase_FreqI = A_Phase_FreqI × Freq + B_Phase_FreqI (FI2)
[0024]
Gain_FreqV and Gain_FreqI set the slope and intercept of the linear line so that they become 1 when the current power supply frequency becomes the power supply frequency at the time of the initial adjustment. In addition, Phase_FreqV and Phase_FreqI are such that the amount of change in the leading direction is positive, and the amount of change in the lagging direction is negative, and the slope of the primary straight line becomes zero when the current power supply frequency becomes the power supply frequency at the time of initial adjustment. And set the intercept. Therefore, the power frequency at the time of the initial adjustment may be stored, and the intercept may be calculated backward from the current frequency and the slope of the straight line. In this case, the set amount of the intercept can be reduced. The inclination and intercept of the primary straight line can be freely changed. Therefore, when the circuit configuration of the voltage signal input means and the current signal input means is changed (for example, from the PT circuit to the resistance voltage dividing circuit, from the CT circuit to the divided CT circuit), the values of the slope and intercept of the linear line are changed. Only need. The correction equations (FV1), (FI1), (FV2), and (FI2) have characteristics similar to those in FIG.
[0025]
Also, the amplitude correction rate obtained by summing the voltage signal input means and the current signal input means is Gain_Freq, and the phase change amount is Phase_Freq.
Gain_Freq = (A_Gain_FreqV + A_Gain_FreqI) × Freq +
B_Gain_FreqV + B_Gain_FreqI
Phase_Freq = (A_Phase_FreqV + A_Phase_FreqI) × Freq +
B_Gain_FreqV + B_Phase_FreqI
Alternatively, the amount of calculation and the set amount of the linear slope and intercept may be reduced.
[0026]
Embodiment 4. FIG.
The current temperature Temp of the AD converter is obtained from the temperature measuring means 56 (here, it is assumed that all the AD converters have substantially the same temperature), and the voltage signal input, the current signal input, and the AD based on the temperature are input. The amplitude correction rate (reciprocal of the amplitude change rate) and the phase change amount of the converter are calculated by a linear line.
1) Correction of voltage signal input by temperature
The amplitude correction rate is Gain_TempV, the phase change amount is Phase_TempV,
A_Gain_TempV is the slope of the primary straight line of the amplitude correction rate, B_Gain_TempV is the intercept,
A_Phase_TempV is the slope of the primary line of the phase change amount, B_Phase_TempV is the intercept,
2) Correction of current signal input by temperature
The amplitude correction rate is Gain_TempI, the phase change amount is Phase_TempI,
A_Gain_TempI is the slope of the primary straight line of the amplitude correction rate, B_Gain_TempI is the intercept,
A_Phase_TempI represents the slope of the primary straight line of the phase change amount, B_Phase_TempI represents the intercept,
3) About correction of AD converter by temperature
The amplitude correction rate is Gain_TempAD, the phase change amount is Phase_TempAD,
A_Gain_TempAD is the slope of the primary straight line of the amplitude correction rate, B_Gain_TempAD is the intercept,
A_Phase_TempAD is the slope of the primary line of the phase change amount, B_Phase_TempAD is the intercept,
If the current temperature is Temp, the correction equation is as follows.
Gain_TempV = A_Gain_TempV × Temp + B_Gain_TempV (TV1)
Gain_TempI = A_Gain_TempI × Temp + B_Gain_TempI (TI1)
Gain_TempAD = A_Gain_TempAD × Temp + B_Gain_TempAD (TA1)
Phase_TempV = A_Phase_TempV × Temp + B_Phase_TempV (TV2)
Phase_TempI = A_Phase_TempI × Temp + B_Phase_TempI (TI2)
Phase_TempAD = A_Phase_TempAD × Temp + B_Phase_TempAD (TA2)
[0027]
Gain_TempV, Gain_TempI, and Gain_TempAD set the slope and intercept of the primary straight line so that they become 1 when the current temperature becomes the temperature at the time of the initial adjustment. Further, Phase_TempV, Phase_TempI and Phase_TempAD have positive values for the change amount in the leading direction and negative values for the change amount in the lagging direction. And set the intercept. Therefore, the temperature at the time of the initial adjustment may be stored, and the intercept may be calculated backward from the current temperature and the slope of the straight line. In this case, the set amount of the intercept can be reduced. The inclination and intercept of the primary straight line can be freely changed. Therefore, when the circuit configuration of the voltage signal input means and the current signal input means is changed (for example, from the PT circuit to the resistance voltage dividing circuit, from the CT circuit to the divided CT circuit), the values of the slope and intercept of the linear line are changed. Only need.
[0028]
Also, the sum of the amplitude correction rate of the voltage signal input means, the current signal input means, and the AD converter is Gain_Temp, and the phase change amount is Phase_Temp,
Gain_Temp =
(A_Gain_TempV + A_Gain_TempI + A_Gain_TempAD) ×
Temp + B_Gain_TempV + B_Gain_TempI + B_Gain_TempAD
Phase_Temp =
(A_Phase_TempV + A_Phase_TempI + A_Phase_TempAD) ×
Temp + B_Gain_TempV + B_Phase_TempI + B_Phase_TempAD
Alternatively, the amount of calculation and the set amount of the linear slope and intercept may be reduced.
[0029]
In order to reflect the calculated amplitude correction rate and phase change amount in the calculation of power and reactive power, the amplitude / phase correction matrix (described above) at the time of initial adjustment is multiplied by the amplitude correction rate and phase change amount correction matrix. The result is used as an amplitude / phase correction matrix used for actual calculation.
The overall amplitude correction rate and phase change amount are obtained by arbitrarily selecting a parameter to be corrected from among the effective value voltage, the effective value current, the power supply frequency, and the temperature, and calculating the total effect of the selected parameter. When all parameters are selected, Expression (3) is obtained assuming that the overall amplitude correction rate is Gain_all_N and the phase change amount is Phase_all_N.
Gain_all_N = Gain_Vrms_N × Gain_Irms_N × Gain_Freq × Gain_Temp
Phase_all_N = Phase_Vrms_N + Phase_Irms_N + Phase_Freq + Phase_Temp
[0030]
(Equation 4)

[0031]
It becomes.
Therefore, the amplitude / phase correction matrix used for the actual calculation is as shown in the following equation (4).
[0032]
(Equation 5)

[0033]
When the amplitude correction rate and the phase change amount calculated as described above are reflected on the effective value voltage and the effective value current, the effective value is independent of the phase, and therefore, the effective value voltage Vrms-N and the effective value current Irms-N The result obtained by multiplying the

conversion coefficients

181, 183, 281 and 283 at the time of the initial adjustment by the amplitude correction rate of may be used as the conversion coefficient used for the actual calculation.
In addition, the calculation of the coefficient of the amplitude / phase correction matrix used in the actual calculation is not performed for each sampling frequency, but for example, a time interval in which a change in each effective value or temperature is recognized as a significant difference, for example, 0.5 seconds to The calculation may be performed every few seconds and the result may be updated. Therefore, the amount of calculation only increases when updated, the amount of calculation for constantly calculating the power-related amount does not increase, and the calculation load does not increase.
[0034]
As described above, when the power-related amount is measured, the power-related amount is corrected with the detected effective value voltage, effective value current, power supply frequency, and AD converter temperature, so that the power-related amount can always be measured with high accuracy. Although an example in which the correction is performed based on the effective value voltage, the effective value current, the power supply frequency, and the temperature has been described, the correction may be performed using at least one of the parameters. This will be described in Embodiment 2. .
[0035]
Further, since the phase is corrected by the rotation matrix, the correction can be performed in a short time.
A 90-degree phase conversion unit for performing a 90-degree direct phase conversion of a voltage with respect to the current or a phase of the current with respect to the voltage; a reactive power calculation unit that obtains a reactive power from an output of the 90-degree phase conversion unit; Since the present invention is applied to a device having an active power calculating means for obtaining the correction value, the rotation matrix calculation function for obtaining the amplitude / phase correction matrix at the time of adjustment and the rotation matrix calculation function for the above-described correction can be shared.
In the first embodiment, the temperature is described as the temperature of the AD converter. However, the temperature of the arithmetic device for performing the above-described calculation may be used.
[0036]
Embodiment 2 FIG.
In the first embodiment, seven amplitude correction rates and seven phase change amounts are calculated using the four parameters of the effective value voltage, the effective value current, the power supply frequency, and the temperature of the AD converter. May be selected and calculated. For example, since only the phase change amount of the current signal input largely depends on the effective value current and the power supply frequency, if it is desired to perform only this correction, it is sufficient to perform only Phase_Irms_N and Phase_FreqI. The same applies to other items.
[0037]
Embodiment 3 FIG.
In the second embodiment, only arbitrary items are selected and calculated. However, after selecting all items, for items that do not particularly require correction, the slope and intercept of the primary straight line for the correction are calculated as follows. The same operation can be realized by setting as described above.
That is, the amplitude correction rate of an unnecessary item is set to 0 for the slope of the primary straight line and 1 for the intercept. The phase change amount of an unnecessary item is set to 0 for the slope of the primary straight line and to 0 for the intercept.
[0038]
Embodiment 4 FIG.
After the coefficients of the amplitude / phase correction matrix used for the actual calculation are calculated by the floating-point calculation, when updating the amplitude / phase correction matrix, the floating-point may be converted to a fixed point and set. As a result, the usual calculation is only multiplication and addition of the fixed point, and the calculation speed is improved.
[0039]
Embodiment 5
In the fourth embodiment, the fixed-point is used when updating the amplitude-phase correction matrix. However, it is also possible to calculate the coefficient of the amplitude-phase correction matrix used in the actual calculation at the fixed point.
In practice, the slope and intercept of the linear line are set as fixed-point values. In addition, the parameters of the effective value voltage, the effective value current, the power supply frequency, and the temperature at the time of calculating the amplitude correction rate and the phase change amount are also acquired in a fixed point. Accordingly, when calculating the amplitude correction rate and the phase change amount, only multiplication and addition of fixed points are performed.
Furthermore, in order to eliminate the operation amount of the sine value and the cosine value by the floating point required when calculating the phase change amount correction matrix, for example, a 0.01-degree left rotation matrix, a 0.1-degree left rotation matrix, Alternatively, the phase change amount correction matrix may be calculated by holding the coefficients of the degree left rotation matrix in advance as fixed point defaults, calculating the phase change amount, and multiplying the left rotation matrix.
In the above example, when the phase change amount is 1.24 degrees (lead), the phase change amount correction matrix is 1 degree left rotation matrix × (0.1 degree left rotation matrix) ² × (0.01 degree left rotation matrix) ⁴ Can be calculated.
[0040]
Embodiment 6
When the amount of phase change due to the magnitude of the current to be detected is non-linear and it is difficult to correct it with a linear line as in the case of the split type CT, a large number of 1s having different slopes and intercepts, such as two or three, are used. The correction is performed by setting the next straight line and switching the primary straight line for calculating the correction amount with the intersection of the straight lines as a boundary.
For example, a case will be described as an example in which a split type CT is adopted for current signal input, and the amount of phase change due to the magnitude of the current to be detected is non-linear. An example 95 of the characteristic at this time is shown in FIG. In the characteristic 95 of FIG. 7A, the phase error is sharply larger at a light load than at a load of about 10% as compared with a heavy load portion. In such a case, for example, two primary straight lines for calculating the phase change amount Phase_Irms_N are prepared. The slope of the first linear line 96 for calculating the amount of phase change is A1_Phase_Irms, the intercept is B1_Phase_Irms, the inclination of the second primary line 97 is A2_Phase_Irms, and the intercept is B2_Phase_Irms. When two primary straight lines are set, Irms at the intersection of the straight line 96 and the straight line 97 is calculated.
[0041]
(Equation 6)

[0042]
Therefore, when the current Irms_N is equal to or greater than the linear intersection point Irms, the first primary straight line 96 is used,
Phase_Irms_N = A1_Phase_Irms × Irms_N + B1_Phase_Irms
When the current Irms_N is smaller than the linear intersection point Irms, the second primary straight line 97 is used,
Phase_Irms_N = A2_Phase_Irms × Irms_N + B2_Phase_Irms
And
In this manner, the measurement of the power-related amount can be performed with higher accuracy as in the case of the phase error 98 shown in FIG. 7B.
It goes without saying that more straight lines may be used in accordance with the curve of the characteristic 95.
[0043]
【The invention's effect】
The power-related amount measuring apparatus according to the present invention is characterized in that the characteristics of the voltage signal input means, the current signal input means, and the AD converter according to the magnitude and frequency of the voltage and current to be detected and the temperature of the AD converter being calculated are Even if it changes, it can be corrected at high speed with a small amount of calculation and a small memory capacity and can always be measured with high accuracy, so that the power-related amount can be measured with higher accuracy.
[Brief description of the drawings]
FIG. 1 is a diagram showing a configuration of a power-related amount measuring device according to a first embodiment of the present invention.
FIG. 2 is a partial detailed view of FIG.
FIG. 3 is a vector diagram of power before and after correction.
FIG. 4 is a diagram showing a correction flow.
FIG. 5 is a characteristic diagram illustrating a relationship between a frequency and a phase and a correction straight line in CT current sensing.
FIG. 6 is a diagram illustrating a correction straight line.
FIG. 7 is a diagram illustrating a relationship between a frequency and a phase and a correction straight line in CT current sensing according to the fifth embodiment of the present invention.
[Explanation of symbols]
101,103 voltage signal,
121,123,221,223 Delta-sigma AD converter,
141, 143, 241, 243 effective value voltage calculating means,
161,163,261,263 Effective value current calculating means,
191, 193, 291, 293 Hilbert conversion means,
201, 203 current signal, 301, 303 multiplication means,
321,323 digital low-pass filter means,
341 and 343 multiplication means,
361, 363 digital low-pass filter means,
54 power supply frequency calculating means, 56 temperature measuring means,
W_in_N power before correction, W_out_N power after correction,
var_in_N reactive power before correction, var_out_N reactive power after correction,
Vrms_1, Vrms_3 RMS voltage, Irm_1, Irm_3 RMS current,
Freq Power frequency, Temp temperature.

Claims

An AD conversion circuit for digitally converting the current signal and the voltage signal detected by the current sensor and the voltage sensor mounted on the power line, respectively.
A power-related amount calculating unit that calculates a frequency of the voltage signal and calculates a power-related amount of the power line based on an output of the AD conversion circuit;
A power-related amount measuring device, comprising: a correction unit that corrects the power-related amount using at least one of the voltage signal, the current signal, and the frequency as an auxiliary variable.

An AD conversion circuit for digitally converting the current signal and the voltage signal detected by the current sensor and the voltage sensor mounted on the power line, respectively.
A temperature detector for detecting a temperature of the AD conversion circuit,
A power-related amount calculating unit that calculates a frequency of the voltage signal and calculates a power-related amount of the power line based on an output of the AD conversion circuit;
A power-related quantity measuring device, comprising: a correction unit that corrects the power-related quantity using the temperature as an auxiliary variable.

The power-related amount measuring device according to claim 1, wherein the correction unit corrects an amplitude or a phase of the power-related amount.

4. The power-related amount according to claim 3, wherein the correction of the phase of the power-related amount by the correction unit is based on the following expression expressed by calculating a rotation matrix based on an amplitude and phase correction matrix at the time of initial adjustment. 5. Quantity measuring device.

A power-related amount calculating unit, an active power calculating unit that obtains active power from the current signal and the voltage signal output from the AD conversion circuit;
90-degree phase conversion means for performing a 90-degree phase conversion on the voltage signal with respect to the current signal, or the phase of the current signal with respect to the voltage signal by 90 degrees;
3. The power-related quantity measuring device according to claim 1, further comprising: a reactive power calculating unit that obtains a reactive power from the phase output converted by the 90-degree phase converting unit.

The power-related-amount calculating device according to claim 4, wherein the correction formula is determined for each predetermined range of the auxiliary variable.