JP2003333899A - Method for controlling induction motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To stabilize a variable speed control operation near the zero speed of an induction motor for vector-controlling by a speed sensor. <P>SOLUTION: A method for controlling the induction motor comprises steps of: inserting a lower limit setting means 31 to a path from a conventional primary angle frequency command arithmetic means 17 to an integrator 13 and a subtracter 20; and providing a lower limit value based on values of the rotation angle frequency command value ωR* and the slip-angle frequency ωS of the motor 2 to a primary angle frequency command value ω<SB>1</SB>* of the motor 2 by the means 31, and hence eliminating reverse drive near the zero of the primary angle frequency of the motor 2. <P>COPYRIGHT: (C)2004,JPO

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a method for controlling an induction motor driven by a PWM inverter, which is vector-controlled without a speed sensor.

[0002]

2. Description of the Related Art FIG. 2 is a block diagram showing a conventional example of a method for controlling an induction motor driven by a PWM inverter, which performs vector control without a speed sensor.

In FIG. 2, reference numeral 1 denotes a semiconductor power conversion circuit, and three-phase output voltage command values v _U ^* , v _V ^* , v _W ^{* which} will be described later ^.
A PWM inverter that outputs a sinusoidal three-phase voltage obtained by performing a PWM operation on each of them, 2 is an induction motor driven by the PWM inverter 1, and 3 is a primary current i _U , i _V , i of the induction motor 2 that flows from the PWM inverter 1. A current detector for detecting _W , 4 are primary voltages v _U , v _{V of} the induction motor 2,
A voltage detector for detecting v _W , 10 is a PWM inverter 1
Is a control device for vector-controlling the induction motor 2 via a speed sensorless via the.

This control device 10 has a three-phase / two-phase conversion means 1
1, coordinate conversion means 12, integrator 13, three-phase / two-phase conversion means 14, coordinate conversion means 15, induced voltage calculation means 16, primary angular frequency command calculation means 17, magnetic flux estimation means 18, slip angular frequency calculation means 19 , Subtractor 20, speed controller 21, magnetic flux controller 22, current controller 23, coordinate conversion means 24, 2.
The phase-to-phase conversion means 25 is included.

The operation of each block constituting the control device 10 shown in FIG. 1 will be described below.

1 of the induction motor 2 detected by the current detector 3
The secondary currents i _U , i _V , and i _W are converted into two-phase quantities iα and iβ on the fixed coordinates by the three-phase / two-phase conversion means 11, and further, the coordinate conversion means 12 described below has a primary angular frequency command value ω ₁ ^*. I _M (magnetization axis component current) as a two-phase quantity on the rotation coordinate based on the magnetization axis based on the phase angle command value θ ^* of the induction motor 2 obtained by time integration in the integrator 13, and Is converted to i _T (torque axis component current).

1 of the induction motor 2 detected by the voltage detector 4
The next voltages v _U , v _V , v _W are converted into two-phase quantities vα, vβ on the fixed coordinates by the three-phase / two-phase conversion means 14, and further, by the coordinate conversion means 15 on the rotation coordinates with the magnetization axis as a reference. v _T (torque axis component voltage).

In the induced voltage calculation means 16, the above-mentioned i _M , i
By inputting _T , v _T and ω ₁ ^* , the torque axis component e _T of the induced voltage shown in the following formula (1) is derived.

[0009]

[Number 1] _{e T = v T -R 1 ·} i T -Lσ (di / dt) i T -jω 1 * · Lσ · i M ... (1) However, R ₁ is the primary resistance of the induction motor 2, Lσ is the leakage inductance of the induction motor 2, and j is an imaginary unit.

The primary angular frequency command computing means 17 computes the primary angular frequency command value ω ₁ ^* according to the following equation (2).

[0011]

[Equation 2] ω ₁ ^* = e _T / φ ^# (2) Estimated magnetic flux value φ of the induction motor 2 in the above equation (2)
^# Is derived by the magnetic flux estimating means 18 based on the following equation (3).

[0012]

## EQU3 ## φ ^# = e _T / ω ₁ ^* (3) The slip angular frequency calculator 19 inputs the above-mentioned i _T , φ ^#, and the slip frequency ω of the induction motor 2 shown in the following formula (4) _S is derived.

[0013]

Ω _S = R ₂ · i _T / φ ^# (4) where R ₂ is the secondary resistance of the induction motor 2.

Rotational angular frequency estimated value ω of induction motor 2
_R ^# is derived by subtracting the value of the slip angular frequency ω _S from the primary angular frequency command value ω ₁ ^* in the subtractor 20.

The speed adjuster 21 performs an adjustment operation to make the deviation between the commanded rotational angular frequency command value ω _R ^{* of the} induction motor 2 and the rotational angular frequency estimated value ω _R ^# zero, and the calculated result is the torque. It is output as the axial component current command value i _T ^* .

The magnetic flux controller 22 performs an adjustment calculation to make the deviation between the commanded magnetic flux command value φ ^* of the induction motor 2 and the estimated magnetic flux value φ ^# zero, and the calculation result is used as the magnetization axis component current command value i. It is output as _M ^* .

In the current controller 23, the magnetization axis component is
Current command value i_M ^*And magnetization axis component current i _MDeviation from the value of
Adjustment calculation, and the calculation result is the magnetization axis voltage command.
Value v _M ^*As the torque axis component current finger
I i_T ^*And torque axis component current i_TThe deviation from the value of
Adjustment calculation, and the calculated result is the torque axis voltage command value.
v_T ^*Is output as.

The magnetizing axis voltage command value v _M ^* and the torque axis voltage command value v _T ^* on the rotating coordinates with the magnetization axis as a reference are converted by the coordinate converting means 24 into two-phase voltage command values v on the fixed coordinates.
was converted to α and v?, further it was transformed by passing through the two-to-three phase conversion unit 25 outputs the voltage command values of three phases to the PWM inverter ^{_{1 v U *, v V *}} , v W * , respectively .

[0019]

In the conventional control method based on the speed sensorless vector control shown in FIG.
The voltage output from the WM inverter 1, that is, the primary voltage of the induction motor 2 is fed back to set the induction motor 2 to 1
A secondary angular frequency command value ω ₁ ^* and a rotational angular frequency estimated value ω _R ^# are derived, and the induction motor 2 is controlled at a variable speed by using these, and at this time, a voltage detector that detects the primary voltage is used. 4 is very small due to the offset and noise.
A large amount of error is included in the detected value in the secondary voltage region, and due to this error, the derived primary angular frequency command value ω _{1 in} the low speed region where the rotational angular frequency of the induction motor 2 is near zero. ^*
Also, the error of the rotational angular frequency estimated value ω _R ^# becomes large, and as a result, the variable speed control operation of the induction motor 2 in the low speed region becomes unstable, and especially when the induction motor 2 is started, The primary angular frequency command value ω ₁ ^* may be a value in the direction opposite to the rotation direction based on the rotational angular frequency command value ω _R ^* of the induction motor 2, and this value causes the induction motor 2 to rotate.
There is a risk that the motor may be once driven in the opposite direction.

An object of the present invention is to provide a method of controlling an induction motor which solves the above problems when starting an induction motor driven by a PWM inverter.

[0021]

The first invention is a PW.
A coordinate axis parallel to the magnetic field of the electric motor (M
Axis) component current and a vector control based on a coordinate axis (T axis) component current orthogonal to the M axis, the method of controlling an induction motor for variable speed control of the electric motor, wherein the value of the T axis component current is the command value. When the primary angular frequency command value given to the induction motor is controlled so that
The next angular frequency command value is provided with a lower limit value based on the rotational angular frequency command value of the electric motor and the value of the slip angular frequency.

A second aspect of the present invention is the induction motor control method according to the first aspect, wherein the rotational angular frequency command value of the induction motor is positive and the primary angular frequency command value of the electric motor is the same as that of the electric motor. When it is smaller than the value of the slip angular frequency, the value of the slip angular frequency is set as a new primary angular frequency command value, the rotational angular frequency command value of the induction motor has a negative polarity, and the primary angular frequency command value of the electric motor. Is larger than the value of the slip angular frequency of the electric motor, the value of the slip angular frequency is set as a new primary angular frequency command value.

According to the present invention, the lower limit value based on the rotation angular frequency command value of the induction motor and the slip angular frequency value is provided to the primary angular frequency command value of the induction motor driven by the PWM inverter, so that it will be described later. As described above, it is possible to eliminate the unstable variable speed control operation in the low speed region where the rotational angular frequency of the induction motor is near zero.

[0024]

FIG. 1 is a block diagram showing an embodiment of a method for controlling an induction motor according to the present invention. Components having the same functions as those of the conventional example configuration shown in FIG. Therefore, the description is omitted here.

That is, the block configuration shown in FIG.
It is composed of an M inverter 1, an induction motor 2, a current detector 4, a voltage detector 5, and a control device 30. The control device 30 has a three-phase / two-phase conversion means 11 and a coordinate conversion means 12 having the same functions as those of the conventional configuration. , Integrator 13, three-phase / two-phase conversion unit 14, coordinate conversion unit 15, induced voltage calculation unit 16, primary angular frequency command calculation unit 17, magnetic flux estimation unit 18, slip angular frequency calculation unit 19, subtractor 20, speed Controller 21, magnetic flux controller 2
2, the current controller 23, the coordinate conversion means 24, the two-phase three-phase conversion means 25, the lower limit setting means 31 in the path from the primary angular frequency command calculation means 17 to the integrator 13 and the subtractor 20 as shown in the figure. Has been inserted.

Hereinafter, the control method of the induction motor of the present invention based on the control device 30 shown in FIG. 1 will be described focusing on the operation of the lower limit setting means 31.

First, the rotational angular frequency command value ω _R ^{* of the} induction motor 2 instructed via the control device 30 shown in FIG. 1 has a positive polarity (ω _R ^* > 0), and the primary angle command value computing means. When the primary angular frequency command value ω ₁ ^* of the induction motor 2 derived by 17 is smaller than the value of the slip angular frequency ω _S of the induction motor 2 derived by the slip angular frequency calculator 19, (ω ₁ ^* <ω _S )
In this case, the lower limit setting means 31 outputs the value of ω _S as a new primary angular frequency command value, so that the new 1
The next angular frequency command value ω ₁ ^* is in the “+” direction (> ω _S ) from the value of the slip frequency ω _{S to be} compensated, and as a result, especially when the induction motor 2 is started, the error in the voltage detector 4 described above is generated. Therefore, the primary angular frequency command value ω ₁ ^* derived by the lower limit setting means 31 is opposite to the rotation direction based on the rotational angular frequency command value ω _R ^* of the induction motor 2. It can be prevented from becoming a value.

At this time, the rotational angular frequency estimated value ω _R ^# of the induction motor 2 which is the output of the subtracter 20 also becomes positive or zero (ω _R ^# ≧ 0), and is erroneously on the opposite side (-). Side) is prevented from being performed.

Next, the rotational angular frequency command value ω _R ^{* of the} induction motor 2 instructed via the control device 30 shown in FIG. 1 has a negative polarity (ω _R ^* <0) and the primary angle command value calculation is performed. When the primary angular frequency command value ω ₁ ^{* of} the induction motor 2 derived by the means 17 is larger than the slip angular frequency ω _S of the induction motor 2 derived by the slip angular frequency calculator 19 (ω ₁ ^* > ω _S )
In this case, the lower limit setting means 31 outputs the value of ω _S as a new primary angular frequency command value, so that the new 1
The next angular frequency command value ω ₁ ^* is in the “−” direction (<ω _S ) from the value of the slip frequency ω _{S to be} compensated, and as a result, especially when the induction motor 2 is started, the error in the voltage detector 4 described above is generated. Therefore, the primary angular frequency command value ω ₁ ^* derived by the lower limit setting means 31 is opposite to the rotation direction based on the rotational angular frequency command value ω _R ^* of the induction motor 2. It can be prevented from becoming a value.

Further, at this time, the rotation angular frequency estimated value ω _R ^# of the induction motor 2 which is the calculation output of the subtractor 20 also becomes negative or zero (ω _R ^# ≤0), and is erroneously detected on the opposite side ( The estimation calculation operation to the (+ side) is prevented.

[0031]

According to the present invention, in the vector control without using the speed sensor of the induction motor using the primary voltage of the induction motor, by providing a lower limit limiter to the primary angular frequency command value of the motor, the motor is controlled. Primary voltage is small,
It is possible to stably perform the variable speed control operation in the low speed region where the detection error becomes a problem.

[Brief description of drawings]

FIG. 1 is a block configuration diagram showing an embodiment of the present invention.

FIG. 2 is a block diagram showing a conventional example.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 1 ... PWM inverter, 2 ... Induction motor, 3 ... Current detector, 4 ... Voltage detector, 10 ... Control device, 11 ... Three-phase / two-phase conversion means, 12 ... Coordinate conversion means, 13 ... Integrator, 14 ...
Three-phase / two-phase conversion means, 15 ... Coordinate conversion means, 16 ... Induced voltage calculation means, 17 ... Primary angular frequency command calculation means, 18 ...
Magnetic flux estimation means, 19 ... Slip angular frequency calculator, 20 ... Subtractor, 21 ... Velocity controller, 22 ... Flux controller, 23 ... Current controller, 24 ... Coordinate conversion means, 25 ... Two-phase / three-phase conversion means, 30 ... Control device, 31 ... Lower limit setting means.

Claims (2)

[Claims] 1. A coordinate axis (M-axis) component current parallel to a magnetic field of the electric motor obtained by coordinate-converting a primary current of an induction motor driven by a PWM inverter, and a coordinate axis (T-axis) orthogonal to the M-axis. In a method for controlling an induction motor in which a variable speed control of the electric motor is performed by vector control based on a component current, a primary angular frequency command value to be given to the induction motor so that the value of the T-axis component current matches the command value. A control method for an induction motor, wherein a lower limit value based on a rotation angular frequency command value of the electric motor and a value of a slip angular frequency is provided to the primary angular frequency command value during control. 2. The method for controlling an induction motor according to claim 1, wherein the rotational angular frequency command value of the induction motor is positive, and the primary angular frequency command value of the electric motor is a slip angular frequency of the electric motor. When the value is smaller than the value, the value of the slip angular frequency is set as a new primary angular frequency command value, the rotational angular frequency command value of the induction motor is negative, and the primary angular frequency command value of the motor is the negative primary angular frequency command value of the electric motor. A method for controlling an induction motor, wherein when the value of the slip angular frequency is larger than the value of the slip angular frequency, the value of the slip angular frequency is set as a new primary angular frequency command value.